AND8161/D Implementing a DC/DC Single−Ended Forward Converter with the NCP1216A http://onsemi.com Prepared by: Roman Stuler APPLICATION NOTE • Current−Mode Operation This document describes how the NCP 1216A controller can be used to design a DC/DC single−ended forward converter suitable for telecommunication applications. The requirements for the converter are as follows: − Input voltage range from 36 V to 72 VDC − Continuous output power greater than 30 W for a 12 V output voltage − Small PCB dimensions − Efficiency greater then 85% − Input to output isolation voltage of 1500 V The NCP1216A controller is an attractive solution for this application, due to the following features: • • • 50% Maximum Duty Cycle Operation • • • Forward converters usually limit the maximum duty cycle to 50%. Since the voltage reset is constrained to be equal to the input voltage (1:1 reset ratio), it is not desirable to exceed 50% DC to avoid saturating the transformer core. No Auxiliary Winding Operation The DSS (Dynamic Self−Supply) function allows the NCP1216A derive power directly from the HV line without having to supply VCC either from the secondary output inductance (creepage distance and isolation issues) or via an auxiliary winding delivering a variable voltage of N x Vin. 500 mA Peak Current Capability The NCP1216A can drive a MOSFET directly without any additional driver stage. If the selected MOSFET gate charge would overload the DSS capability, then an auxiliary winding could be used solely to supply the driver pulses. Semiconductor Components Industries, LLC, 2004 May, 2004 − Rev. 0 Cycle−by−cycle primary current monitoring eliminates any overcurrent situations, e.g. resulting from a secondary short−circuit. Direct Optocoupler Connection In applications where the input to output isolation is required, a direct connection eases the design stage, saving external components. Extremely Low No−Load Power Consumption Extremely low consumption in no−load operation is a great advantage of the NCP1216A controller. Today’s maximum stand−by consumption standards can be easily met if this function is used. Short−Circuit Protection By monitoring the activity on the feedback line, the NCP1216A simplifies the task of secondary side short−circuit protection. Coupling problems are eliminated thanks to this feature and the DSS implementation. The 35 W DC/DC Converter Board Specifications The schematic of the proposed converter is shown in Figure 1. This converter has the following specifications: Minimum Input Voltage 36 VDC Maximum Input Voltage 72 VDC Output Voltage 12 VDC Continued Output Current 3.0 A Operating Frequency 100 kHz No−load Consumption at 48 V 1.8 mA Maximum Ambient Temperature 70°C 1 Publication Order Number: AND8161/D D2 C7 R6 T1 100 R MURA240T3 2n2 L2 + C8 220 / 25 V L1 C1 22 / 100 V D4B 10 H C2 22 / 100 V + C3 22 / 100 V + C5 1.5 n R5 8k2 + C9 220 / 25 V + C10 220 / 25 V + +12 V 1.0 H C11 220 / 25 V MURB1620CT AND8161/D Figure 1. 2 http://onsemi.com R7 560 R D3 MURA240T3 IC1 ADJ HV Q1 FQD18N20 FB R1 12 k 100 H D4A 36− 72 V L3 CS VCC GND DRV IC2 R4 0R NCP1216A C4 22 R2 D1 1k8 1N4148 R8 C12 18 k 33 n PC817 T2 R9 39 k IC3 TLV431 R3 10 R C6 4n7/Y2 R10 4k3 AND8161/D Description of Converter Connection The primary magnetization current does not directly participate in the energy transfer and cause additive losses on the power switch and the primary winding. When the switch is off, the transformer core must be reset in order to let the internal flux return to zero. This is done via a dedicated reset circuit. Consequently the magnetizing current Imag must be kept smaller than the productive component of the primary current. The core flux density excursion B has to be chosen with respect to the characteristics of the core material: the saturation flux density Bmax or Bsat, the residual flux density Br, hysteretic losses and the core temperature behavior. With respect to these characteristics, the flux density excursion in high frequency converters should be between 0.15 T and 0.2 T. If a higher value is chosen, greater losses will be generated. The primary turn count Np can be calculated by rearranging equation 4: Capacitors C1, C2, C3 and inductor L1 form the input filter. Diode D3, capacitor C5 and resistor R5 provide the primary clamping network which combats leakage inductance between the reset winding and the primary winding. The link between both windings occurs via D2 when the switch is off. Transformer T2 with diode D1 and resistors R2, R3 serve as the primary current sensing circuit. Thanks to low insertion losses, the final efficiency of the converter benefits greatly from this configuration. IC1 is the main driving circuit of the power converter. The secondary circuitry has D4A as the forward diode and D4B as the freewheeling diode. Capacitor C6 offers a path for common−mode (CM) currents circulating via the various transformer stray capacitances during switching events. Resistors R7, R8, R9, and R10 together with capacitor C12, shunt regulator IC3, and optocoupler IC2 form an isolated feedback circuit for output voltage regulation. A snubber network (R6, C7) is connected across inductor L2 in order to damp high frequency oscillations. L2, C8, C9 and C10 form the basic LC output filter. L3 and C11 form an additional output filter to reduce high frequency noise. Design considerations for various sections of the converter are described below. Np In a forward converter, the core magnetization is ensured by applying a voltage Vin on the primary side. This action creates the core flux which links both primary and secondary windings. Using Faraday’s law, we can write that E = N.d / dt, where E is the voltage generated by a winding of N turns, energized by a flux . By integrating this formula, and rearranging it in terms of the input voltage Vin and the on time ton, we can see that the internal flux depends on the volt−second product: (eq. 1) where: Ae is the total core area B is the core flux density Thus, the maximum core flux density BMAX and the peak primary magnetization current IPKMAG of the transformer are given by the primary inductance value L1 and the maximum input voltage according to equations (2) and (3): V IPKMAG in max · 1 · max L1 fop (eq. 2) V · max BMAX in max Np · fop · Ae (eq. 3) (eq. 4) For an EFD25 core with a total core area of 58mm2 (Bmax = 0.2 T, Vin max = 80 V, fop = 100 kHz and maximum duty cycle max = 0.5) then the number of primary turns Np = 35. The number of reset winding turns depends on design tradeoffs. When the number of turns of the reset winding is lower than the that of the primary winding, the reflected voltage on the power switch drain will be lower than 2*Vin max. However, this limits the maximum duty cycle excursion to less than 50%. Conversely, if the reset turns are larger than the primary turns, the maximum allowed duty cycle will increase but the MOSFET voltage stress will exceed 2*Vin max. Due to these issues, the practical number of turns for the reset winding is usually chosen to be the same as the primary winding, or a 1:1 ratio. It is important to provide a very good coupling between these two windings. A high leakage inductance between these windings would require a hard voltage clamp that would hurt the converter efficiency. The number of turns on the secondary winding Ns can be obtained from equation 5: Transformer Design Vin · ton N · N · Ae · B Vin max · max BMAX · fop · Ae Ns Np · Vout max Vf Vin min (eq. 5) where: Vout is the desired output voltage Vf is the voltage drop of the output rectifier Vin min is the minimum input voltage In the example using the EFD 25, equation (5) gives Ns = 25 turns. where: Vin max is the maximum input voltage L1 is the primary winding inductance fop is the operating frequency is the maximum duty cycle max Np is the count of the primary turns http://onsemi.com 3 AND8161/D resistor configuration. If a classical current sense resistor were used in this application, the associated power loss would be about 3.0 W. When the current sense transformer is used, power losses are about 50 mW. The disadvantage of this solution lies in the current error brought by the magnetization current of current sense transformer. This error is additive so it should accounted for and reduced. A toroidal core with 38 turns of the secondary winding was used in NCP1216A demo board. The primary winding is created by one turn of isolated wire. The peak current I2pk of the current sense resistor can be obtained from equation 8: The primary and the secondary windings must be wound to limit the skin effect. This can be done by using several wires wound in parallel. The maximum diameter Dmax (in mm) of each single wire in the winding is given by equation 6: D max 2 · 75 fop (eq. 6) The total area of the selected wire for primary and secondary windings is a tradeoff between the desired output power, allowable conduction losses in the windings and thermal considerations. The current density in the transformer winding can generally range from 2 to 3.5 A/mm2. If a cooling fan is used, the current density can be increased. The reset winding can be made with a single wire technique, given the low magnetization current flowing into it. In some cases, a small air gap can be inserted into the magnetic circuit of the forward transformer. This solution brings the residual flux density Br to a lower value than without a gap. The main drawback lies in the primary inductance decrease which forces a higher magnetizing current. I2pk I1pk · 1 Imagpk Ns where: I1pk is the peak current of the power switch Ns is the count of secondary turns Imagpk is the peak value of the magnetization current Figure 2 shows the current sense transformer circuit. The peak value of the magnetization current is given by equation 9: V · max Imagpk csth max Ls · fop (eq. 9) Q1 Output Inductor Design The value of the output inductor selected depends on the acceptable level of ripple current. For a small ripple current, a large inductance is needed. On the other hand, when the current ripple is high, large output capacitors must be used to reduce the voltage ripple. In practice, it is usual to limit the current ripple to about 10−20% of the average current of the inductor. The maximum current ripple Imax in a forward converter occurs at 50% duty cycle. Its value can be found via equation (7): I max (eq. 8) V sec max 4 · fop · L2 I1/Ns D1 I2 T2 Imag I1 RSENSE Ns Np (eq. 7) Figure 2. Implementation of the Current Sense Transformer where: Vsec max is the maximum secondary voltage L2 is the inductance of inductor L2 In the NCP1216A demo board, where a 100 H inductor is used, the maximum output ripple will be Imax = 2.0 A. This is rather high, but the allowable dimensions of the inductor limit a higher inductance value selection. The values and types of output capacitors must be chosen with respect to the maximum allowable output voltage excursion as well as the RMS current that will flow in them. where: Vcsth max is the maximum threshold voltage of the current sense input Ls is the inductance of the secondary winding The value of the current sense resistor Rsense can be calculated by using equation 10: V Rsense csth max I2pk Current Sense Transformer Design (eq. 10) The NCP1216A Leading Edge Blanking circuit (LEB) allows the designer to avoid using a RC network to suppress voltage spikes during the switch turn−on event. The current sense transformer is used to reduce power losses traditionally found in the standard current sense http://onsemi.com 4 AND8161/D Primary RCD Clamp and Inductor Snubber Network Design Cclamp Because of manufacturing constraints, the leakage inductance between primary and secondary windings is never equal to zero. The energy stored in this leakage inductance during ton will cause large voltage spikes when the switch is turning off. To protect the power switch from a catastrophic voltage spike, a RCD clamping network must be used. The values of these components depend not only on the leakage inductance value but also on the reflected voltage, the parasitic influence of the layout, and the RCD capacitor. The power dissipation of the RCD clamp can be obtained from equation 11: is the ripple voltage level on the clamping capacitor; this ripple should be minimized. An RC snubber network is connected across the inductor L2 to dampen the parasitic oscillations caused when the freewheel and forward diodes are switched. Both the clamp and snubber networks dissipate heat and affect the converter efficiency. Regulation Loop Design A standard loop topology with a TLV431 shunt regulator is used. The optocoupler provides good isolation between input and output sides of the converter. The output voltage is set up by the R9 and R10 divider ratio according to equation 14: where: Lleak Vclamp Vrefl is value of the leakage inductance is value of the clamp voltage is value of the reflected voltage (Vrefl = Vin max for forward converters with max. DC = 50%) The optimal values of the clamping devices are given by equations 12 and 13: 2 · Vclamp · (Vclamp Vrefl) Lleak · I1pk2 · fop (eq. 13) where: Vripple Vclamp (eq. 11) Pclamp 1 · I1pk2 · Lleak · fop · 2 Vclamp Vrefl Rclamp Vclamp Vripple · fop · Rclamp Vout 1, 25 · 1 R9 R10 (eq. 14) The maximum current flowing through the optocoupler LED is determined by resistor R7. The internal consumption of the TLV431 is low, thus avoiding another biasing element, bypassing the LED. Resistor R8 and C12 constitute the feedback loop compensation circuit. The optimal values for these components are based on the feedback response measurements. (eq. 12) http://onsemi.com 5 AND8161/D Figure 3. PCB Layout (Top Side) Figure 4. PCB Layout (Bottom Side) Figure 5. Component Arrangement (Top Side) Figure 6. Component Arrangement (Bottom Side) http://onsemi.com 6 AND8161/D PCB Layout Design 86 A double−sided PCB is used to minimize the size of the converter. The board is designed with respect to the power dissipation created by the power devices, thus large cooling areas are used. Sound grounding techniques and appropriate isolation distances were incorporated into the layout. The PCB layout and component arrangement can be seen on Figures 3, 4, 5 and 6. EFFICIENCY (%) 85, 5 84, 5 BILL OF MATERIALS L1 10 H DS3316P−103−Coilcraft L2 100 H B0754−A−Coilcraft L3 1.0 H DS3316P−102−Coilcraft 85 84 36 C0972−A−Coilcraft T2 Toroid 6.0 mm, Material T30−Epcos Ns = 38 turns C1, C2, C3 22 /100 V Nippon Chemi−Con−KMF 90 C4 22 /25 V Nippon Chemi−Con−KMF 85 C5 1,5 nF/500 V Through Hole Ceramic Capacitor C6 4n7 Y2 Type Capacitor C7 2,2 nF/500 V Through Hole Ceramic Capacitor R1 12 k SMD 0805 R2 1,8 k SMD 0805 R3 10 SMD 0805 R4 0R SMD1206 R5 8,2 k1.0 W Through Hole R6 100 /1.0 W Through Hole R7 560 SMD1206 R8 18 k SMD 0805 R9 39 k SMD 0805 R10 4,3 k SMD 0805 D1 MMSD914T1−ON Semiconductor D2, D3 MURA2403T3−ON Semiconductor D4 MURB1620CT−ON Semiconductor Q1 FQD18N20V2TF−Fairchild IC1 NCP1216A−ON Semiconductor IC2 PC817−SHARP IC3 TLV431BSN1T1−ON Semiconductor 76 86 EFFICIENCY (%) 80 75 70 65 60 0 5 10 15 20 25 30 35 OUTPUT POWER (W) Figure 8. DC/DC Converter Efficiency vs. Output Power (Vin = 48 V) The no−load consumption as a function of input voltage is shown in Figure 9. 150 NO LOAD CONSUMPTION (mW) 33 nF SMD 1206 66 Figure 7. DC/DC Converter Efficiency vs. Input Voltage 220 /25 V Nippon Chemi−Con−LXZ C12 56 INPUT VOLTAGE (V) T1 C8, C9, C10, C11 46 140 130 120 110 100 90 80 70 36 Performance of the Converter 40 44 48 52 56 60 64 68 72 76 80 INPUT VOLTAGE (V) The power conversion efficiency of the DC/DC converter is shown in Figures 7 and 8. Figure 9. No Load Consumption vs. Input Voltage http://onsemi.com 7 AND8161/D The gate (trace 1) and drain (trace 2) waveforms of the power MOSFET Q1 are shown in Figures 10, 11, 12 and 13 for several converter conditions. GATE GATE DRAIN DRAIN Figure 10. Vinput = 48 V, Iout = 3.0 A Figure 13. Detailed Burst During Overload The load regulation for an output current step from 10% to 100% can be seen in Figure 14. GATE DRAIN Figure 11. No Load Operation Figure 14. Load Regulation (Iout changing from 10% to 100%−0.3 A to 3.0 A) GATE DRAIN Figure 12. Overload Operation http://onsemi.com 8 AND8161/D Notes http://onsemi.com 9 AND8161/D ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. 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