HB-LLC LED Driver Board

AND8479/D
HB-LLC LED Driver Board
Prepared by: Fabien Franc & Roman Stuler
ON Semiconductor
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APPLICATION NOTE
Introduction
The HB−LLC board controls up to 6 strings of LEDs and
regulates the anode voltage by automatically adjusting to the
highest LED string voltage and therefore providing
optimum efficiency. The board provides open LED and
shorted LED detection and protection.
The main supply consists of a high−voltage 400 V DC
supply coming from the PFC boost unit which is powered
directly from the AC lines as shown in Figure 1. The 400 V
is regulated down in the HB−LLC board with the
ON Semiconductor NCP1397 flyback resonant LLC
half−bridge converter to generate the anode supply voltage
for driving the LEDs. In addition to the 400 V, the PFC unit
provides a regulated 15 V supply needed to bias the
NCP1397 converter IC.
The LED load is connected externally to the HB−LLC
board via a connector.
This document describes the Half−Bridge LLC
(HB−LLC) evaluation board that includes an LLC resonant
converter and LED driver circuitry. The HB−LLC board is
targeting large LCD backlight applications where there is a
need to drive long strings of LEDs of about 100 V forward
voltage directly from a PFC converter output. The HB−LLC
provides a high efficiency power conversion (PLED / PIN
around 90%) and guarantees tight LED current matching
between 6 LED strings.
The LED driver function is achieved by the
ON Semiconductor CAT4026 6−channel LED controller.
The CAT4026 drives six long strings of LEDs with an
equivalent anode output voltage centered around 105 V
(90 V min and 121 V max). The complete power−supply
system includes an off−line PFC unit (or high voltage power
supply), the HB−LLC board and the LED load, as shown in
Figure 1. This note covers in detail the operation of the
HB−LLC board.
Figure 1. System Block Diagram
© Semiconductor Components Industries, LLC, 2011
January, 2011 − Rev. 0
1
Publication Order Number:
AND8479/D
AND8479/D
Board Description
The CAT4026 provides a current feedback (IFB) output
connected through a current amplifier (4x) and the
optocoupler back to the NCP1397 feedback (FB) pin to
automatically adjust the anode voltage to the minimum level
required to drive the LEDs.
LED current dimming can be achieved in two ways, PWM
dimming using the PWM input, or analog dimming with the
analog voltage ANLG input.
A 15 V DC supply from the PFC unit (hot side) powers the
NCP1397A controller. A 5 V DC external power supply is
required to power the CAT4026 LED controller and is also
connected to the optocoupler (cold side).
The HB−LLC evaluation board, shown on Figures 2 and
3, provides an isolated power supply with the NCP1397A
resonant mode controller circuitry (on the left side) and the
CAT4026 6−channel LED controller circuitry (on the right
side).
The NCP1397A circuitry converts the 400 V supply to a
dynamically adjustable anode voltage to power the LED
strings. The CAT4026 linear driver controls the constant
current in each of the 6 LED channels using external power
transistors to regulate the LED current.
Figure 2. HB−LLC Board − Top Side
Figure 3. HB−LLC Board − Bottom Side
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Detailed Operation
The board includes a half−bridge LLC power stage and an
LED driver section. Each section is described below.
Timer Based Fault Protection
The controller stops operation after a programmed delay
when the overload condition occurs. Under transient loading
conditions, the converter output will not be turned off, unless
the overload conditions exceed the user programmed
timeout.
LLC Power Stage Operation
The following sections describe the 130 V/1 A output
LLC power stage intended for use as a main power converter
in ON Semiconductor’s linear LED backlight reference
design. This power stage provides insulation and regulated
voltage to the linear LED current regulator driven by the
CAT4026 controller. Output voltage of the LLC stage is
controlled by the CAT4026 from the secondary side to
provide optimum voltage for the LED string with the highest
forward voltage.
Two Level Overcurrent Protection
The primary current of an LLC stage can grow to
excessive values when overload occurs on the output. The
NCP1397A features OCP comparator that speeds−up the
fault timer duration in case short circuit appears in the
application.
Adjustable Dead−Time (DT)
The NCP1397A/B provides the designer flexibility in
adjusting the appropriate dead time to protect switches
against cross−conduction. The length of the DT is chosen
based on the total output capacitance of MOSFETs used in
the application. If the DT is too short, there is not enough
time to re−charge this capacitance and the opposite
MOSFET is turned on under “hard switching conditions”.
The result is poor efficiency and EMI. On the other hand, it
is not good to choose a DT too long as the resonant tank
stores only a finite amount of energy to maintain ZVS
condition.
Topology Overview
The power stage uses a Half Bridge Resonant LLC
topology since it provides several beneficial features:
• Limited number of components (no secondary
regulation coil needed)
• Small size resonant inductance
• Zero Voltage Switching (ZVS) condition for primary
switches under all load conditions
• Zero Current Switching (ZCS) for secondary diodes
under all load conditions
• Higher power density than other topologies: better fit
for SLIM design
The selected LLC power stage topology improves
efficiency, reduces EMI signature and provides better
transformer utilization compared to conventional
topologies. The NCP1397A is used as a controller for this
power stage.
Consideration was also given to optimize the step load
transient response and transformer audible noise as the LCD
TV backlight application works permanently under on/off
mode as required in linear PWM dimming mode operation.
Table 1 shows the specifications of the main power stage.
Adjustable Minimum and Maximum Frequency
Excursion
Using external resistor, the designer can program its
lowest frequency point, obtained in lack of feedback
voltage. Internally trimmed capacitors offer a 3% precision
on the selection of the minimum switching frequency. The
maximum operating frequency clamp is less critical for
application thus the NCP1397 provides tolerance of 12%
maximally.
Open Feedback Loop (FB) Detection
Upon start−up or during operation, if the FB signal is
missing, the fault timer starts to charge timer capacitor. If the
loop is really broken, the FB level does not rise before the
timer ends charging. The controller then stops all pulses and
waits until the timer pin voltage collapses to 1 V typically
before a new attempt to re−start. A hiccup takes place if the
FB fault is permanently present in the application.
Table 1. LLC POWER STAGE SPECIFICATION
Requirement
Min
Max
Unit
Input Voltage (DC)
350
425
V
Output Voltage (DC)
80
125
V
Output Current
0
1
A
Total Output Power
0
150
W
High Side HV Driver
The NCP1397 enables direct connection of the high side
MOSFET Q1 thanks to built−in high side driver (HSD). This
“floating” driver accepts voltages up to 600 V and features
high dV/dt immunity. The HSD is powered from bootstrap
capacitor that is refilled through a bootstrap diode.
Under−voltage detection ensures the high side MOSFET
will be turned on only if there is enough voltage to properly
drive the MOSFET.
Please refer to the datasheet for more information and a
detailed description of the NCP1397A/B LLC controller.
The NCP1397A provides the following beneficial
features to an LLC power stage:
Brown−Out (BO) Protection Input
The BO input pin has two functions. First, the BO input
permanently monitors the bulk voltage and ensures the
SMPS works in the proper Vbulk range. The second
function is to latch−off the device when the BO input is
pulled above 4 V. The BO latch could be used to provide
OVP (overvoltage protection) or OCP (overcurrent
protection) if needed.
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Detailed Power Stage Description
A complete schematic of the power stage is shown in Figure 23. Partial schematics are used in the following descriptions
for better clarity.
Figure 4. The LLC Stage Connection – Primary Side
Primary Side Power Loop Connection
The LLC controller is powered from the independent
auxiliary power supply 15 V VCC connected to the
connector CON1 pin P2. Resistor R31 together with
capacitor C11 forms local filter to assure decoupling of the
controller supply voltage from other blocks.
The output voltage of the PFC stage is connected to
capacitor C7 via input connector CON1. Connector CON1
pins P7 and P8 are the 400 V + Vbulk terminal, and pins P3,
P4 are the GND terminals (refer to Figure 4). Capacitor C7
provides filtering of the high frequency LLC stage primary
current. Another purpose of the capacitor C7 is to deliver
energy to the LLC stage during transient loading (together
with bulk capacitors that are present in PFC stage). The LLC
stage power loop is closed through transistors Q1 and Q2,
transformer TR1, resonant coil L1 and resonant capacitor
C16. The NCP1397A LLC controller features 600 V
high−side gate driver and is capable of driving the HB power
stage directly without the use of a driver transformer.
Resistors R21 and R22 are used to suppress ringing and
control EMI noise on the power MOSFET gates. Bootstrap
capacitor C29 provides the energy required for controlling
the high side MOSFET. When Q2 is turned−on, the HB pin
voltage drops and bootstrap capacitor C29 is charged
through resistor R46 and high−voltage diode D13. At
turn−on and after any restart, the LLC controller turns on
MOSFET Q2 first to charge up the bootstrap capacitor.
Brown−out Protection and Fault Latch
The LLC converter provides excellent parameters over a
specific bulk voltage range only. Thus it is needed to monitor
the PFC output voltage and turn the LLC controller off in
case the bulk voltage drops. Resistor divider R2, R3, R4, R7,
R8, R10, R17 and R45 provides LLC controller with bulk
voltage information. The resistor divider ratio has been
calculated in such a way that the application starts operation
for Vbulk greater than 380 Vdc and turns−off when Vbulk
drops below 350 V. Capacitor C37 is used to filter
Brown−out pin voltage to overcome unwanted latch of the
device due to noise that is always present in switching
applications.
Some of the PFC stages feature PFC_OK output flag. The
Brown−out divider from Vbulk is then not needed. The
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period (depends on the PWM input dimming signal duty
cycle). As already mentioned above, the LLC controller
stops the power stage operation via skip/disable pin in such
case. The FB pin voltage increases when the load
diminishes. Once the load current is too low, the LLC stage
is not able to maintain regulation because the operating
frequency cannot increase further (Fmax clamp – resistor
R55). The FB voltage then increases above the Vfb_max
limit of 5.3 V. The resistor divider R53 and R60 provides the
FB pin voltage to the Skip/Fault input. The output drivers are
thus automatically turned−off and the device terminates
operation until the skip/disable input voltage does not drop
down again.
PFC_OK signal can be brought to the BO input from the
input connector CON1 (pin P1) via resistors R30, R130 and
R117.
In addition to the basic Brown−out function, the BO input
can also be used to latch−off the device in case of a
secondary fault. The optocoupler OK1, which is activated
by a secondary fault signal, pulls up the BO pin via resistors
R92, R131 and thus latches application fully off. The LLC
controller can be then restarted only via Vcc restart.
The latch−off state after fault detection is not acceptable
in some applications. This reference design provides also
another alternative – i.e. to stop the application only during
the time the fault flag is present. This is done using
NCP1397A skip/enable input. The OK1 pulls up the
skip/enable input via resistors R93, R132 when these are
implemented.
Overload and Short Circuit Protection
FB Loop and Skip Mode
The minimum operating frequency of the LLC stage is set
by resistor R12 connected to the Rt pin (refer to Figure 5).
The maximum operating frequency is set by resistor R55
connected to the FMAX pin. The LLC stage will reach
maximum operating frequency during no load conditions –
before entering skip mode.
Figure 5. The Primary FB Loop and
Skip Mode Circuitry Connection
Secondary regulator (CAT4026) drives optocoupler OK2
and thus provides feedback to the primary side. The
optocoupler current adjusts the FB pin voltage. The LLC
stage operating frequency is thus modulated to assure output
voltage regulation. Resistor R40 is implemented to pull up
the FB pin in case of light load conditions or when the
secondary regulator path is opened. Application then
naturally increases operating frequency and/or stops
operation – thus reducing output voltage. The internal FB
fault comparator present in the NCP1397 LLC controller
turns the application off after programmed time out in case
there is no voltage present on the FB pin. Application is thus
protected against overpower or short circuit on the FB pin of
the primary side.
The LCD TV backlight application works permanently in
dimming mode. This means that the LLC converter delivers
nearly zero output power for certain time within dimming
Figure 6. The Output Overload and/or Short
Circuit Protection
The Over Current Protection (OCP – Figure 6) is
implemented in this reference design to protect the
application against overload conditions. The primary
current is sensed indirectly by monitoring the resonant
inductor flux. Voltage from auxiliary windings is rectified
by dual diode D4, scaled down by resistor divider R54, R61,
filtered by capacitor C32 and connected to the fault input of
the LLC controller.
The Soft Start capacitor discharge switch on CSS pin 1 is
turned−on once the Fault/SF pin 9 voltage reaches the
VRef_fault threshold (1.04 V). The LLC stage operating
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Vref_OCP threshold (1.55 V). The controller then increases
timer capacitor charging current to shorten fault timer period
and protect the power stage components from thermal
damage.
Additional hardware cycle−by−cycle overcurrent
protection has been implemented in this design to optimize
power MOSFETs cost, peak power capability and transient
response in dimming mode. This hardware protection if
formed by two HV clamping diodes D3, D7. Resonant
capacitor voltage is clamped to Vbulk level and thus the
maximum peak power and primary current are limited.
In summary, the hardware OCP protection protects
primary switches against damage due to high peak current
that may occur during transformer secondary winding short.
The fault management of the LLC controller then protects
application against average overload induced by secondary
overcurrent. This technique allows the application to deliver
high peak power needed during startup and recovery from
disable mode while keeping precise average overcurrent
threshold.
frequency is thus automatically increased as the Soft Start
capacitor voltage drops and higher current flows out from
the RT pin. The frequency shift naturally reduces the
primary current and protects the primary MOSFETs against
excessive overcurrent. The Itimer1 current source is
activated on pin 3 simultaneously and external timing
capacitor C22 begins charging. If the overload condition
lasts longer than the time given by the timer pin components
(C22, R56), the controller enters protection mode and output
drivers are disabled. Once timer capacitor C22 is discharged
to 1 V by resistor R56, the application attempts to restart via
Soft Start.
The fault timer duration is too long to protect the
application against damage due to a short circuit on the
secondary side (output terminals short or secondary
transformer winding short). To protect against this
possibility, there is a second OCP comparator monitoring
the fault pin voltage. When the frequency shift (via Soft Stat
pin and resistors R9, R11) is no longer sufficient to keep the
primary current limited, the fault input voltage reaches the
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LLC Power Stage Secondary Side
Figure 7. The LLC Secondary Side Schematic
leaded rectifiers (DO201 or DO41) to allow for output
current increase if needed.
Capacitor C2 provides main filtering of the LLC stage
output current. A single 82 mF capacitor has been used to
achieve acceptable output voltage ripple and to minimize
transformer audible noise during dimming operation. Of
Full bridge rectification is used on the secondary side. The
efficiency degradation compared to the solution with center
taped windings is negligible as the nominal output current is
rather small. Advantage of this solution lies in the
transformer design and construction simplification which
impacts cost. PCB design features positions for two kinds of
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Cheap Primary Current Sensing
course one needs to check whether the selected capacitor is
able to handle corresponding RMS current (refer to [11] on
how to design secondary filtering capacitor in LLC
converter).
Capacitor C2 provides the bulk of filtering for the
secondary current, but it does not fully filter out narrow
glitches produced when the secondary winding reverses.
Thus an additional LC filter (L2, C4) has been implemented.
The resonant frequency of this filter should be as low as
possible, however, it can affect system loop gain if the
selected frequency is too close to the crossover frequency. A
resonant frequency of 23 kHz has been selected for this
design. The filter inductor of 2.2 mH features acceptable DC
resistance. A filtering capacitor C4 of 22 mF (low impedance
type) has been implemented. The filter provides higher
peaking around the resonant frequency when a low ESR
capacitor is used. On the other hand, if a capacitor with too
high ESR is used, the output voltage drop during fast
transient loading increases.
The output regulation is assured by the CAT4026 linear
LED driver. The IFB pin of the CAT4026 provides control
current from the optocoupler OK2 via bipolar transistors Q3
and Q19.
Flux of the external resonant coil can be easily sensed by
an auxiliary winding. A cost effective current sensor can be
implemented compared to traditional solutions which
require HV charge pump capacitors.
The resonant tank is designed in such a way that the LLC
stage is operated in, or very close to, the series resonant
frequency (fs) for full load conditions and nominal bulk
voltage. Efficiency is optimized for these operating
conditions. The LLC stage operating frequency is increased
up to 235 kHz to maintain output voltage regulation when
the load diminishes. When the output load drops further, the
maximum operating frequency clamp is reached and the
application enters skip mode operation to maintain output
voltage regulation and to reduce the LLC stage power losses.
On the other hand, when the bulk voltage drops, the
secondary regulator decreases the LLC stage operating
frequency down to 118 kHz to achieve the necessary gain for
output voltage regulation. Please refer to AND8460 for
detail design steps of an LLC resonant tank circuitry.
As mentioned above, the bridge rectifier has been used in
this design. This arrangement allows replacing center
tapped secondary winding configuration. Center tap
winding configuration is not suitable for applications with
high output voltage / low output current because it
complicates transformer design while providing only
negligible efficiency increase. The situation is even more
critical in LLC converter as both secondary windings needs
to be well matched to overcome circuitry imbalance.
Final specification of resonant tank we used in this
application is as follows:
• Resonant inductance Ls = 100 mH
• Resonant capacitor Cs = 15 nF
• Transformer magnetizing inductance Lm = 350 mH
• Transformer turns ratio n = 36/23
• Transformer leakage inductance Llk= 9.5 mH
The LLC stage gains needed for output voltage regulation
under full load conditions and selected bulk voltage range
(350 Vdc – 425 Vdc) can be calculated based on Equations
1 − 3.
Resonant Tank and Transformer Design
An LLC resonant tank with external resonant coil has
been selected in this reference design for several reasons:
Ultra low profile solution: The transformer with high
leakage inductance (or resonant tank with integrated
resonant inductance) cannot be easily used in low profile
solutions. This is because the application metal cover is
placed too close to the transformer. Transformers with high
stray flux induce eddy currents into the metal cower that
generates significant losses and affects resonant tank
parameters. A standard transformer with minimum leakage
inductance features only small stray flux and does not cause
any issues. External resonant inductance with shielded gap
is then needed to increase total resonant inductance value.
Transformer Winding Window Utilization
The winding window is usually purely used in transformer
designs with high leakage inductance. This is because the
primary and secondary windings have to be separated
enough to reach required leakage. Another disadvantage of
this solution is that special transformer bobbin is needed.
The standard transformer with low leakage is much easier to
design with lower manufacturing cost.
Design Flexibility
Specific LLC stage design requires specific magnetizing
and resonant inductance values to achieve required gain
over a given range of operating frequencies. The resonant
tank with external resonant coil provides much more design
flexibility because it allows simple adjustment of both
mentioned parameters.
G min +
2 @ (V out ) V f)
2 @ (80 ) 1.4)
+
+ 0.38
425
V bulk_max
G nom +
2 @ (V out ) V f)
2 @ (100 ) 1.4)
+
+ 0.51 (eq. 2)
396
V bulk_nom
G max +
2 @ (V out ) V f)
2 @ (125 ) 1.4)
+
+ 0.72 (eq. 3)
350
V bulk_min
(eq. 1)
Where:
Vf
– is the expected drop of the rectification
Vbulk_max
– is the maximum operating bulk voltage
Vbulk_nom
– is the nominal operating bulk voltage
Vbulk_min
– is the minimum operating bulk voltage
Vout
– is the required output voltage (including
worst case conditions i.e. temperature drift of the LED string).
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A simulation model can be built to verify the full load gain characteristic of the proposed LLC design with external resonant
inductor (Figure 8).
Figure 8. Simulation Model for the LLC Stage with External Resonant Inductance
Figure 9. Simulated Gain Characteristic for the LLC Stage Design with
External Resonant Inductance, Full Load Conditions
The simulated full load gain characteristic can be seen in Figure 9. Application features enough gain to cover whole output
voltage operating range including Vbulk and LED string forward voltage variations.
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LED Driver Operation
The CAT4026 controller regulates the current
independently in the 6 LED strings by using external NPN
power transistors and monitoring the voltage across the
sense resistors tied to ground. Accurate constant current is
guaranteed in each string so that the device is ideal for large
LCD backlight applications. The controller senses each
cathode string voltage and provides an output current
feedback (IFB pin) to be interfaced to a DC/DC converter for
automatically adjusting the anode voltage to the lowest level
and therefore maximizes the power supply efficiency. The
CAT4026 also detects shorted LEDs within a string or an
open LED string fault condition. Both PWM and analog
voltage inputs are available for dimming control.
Anode Voltage Range
The board is configured such that the anode voltage
(VOUT) cannot exceed 122 V in operation. This is not a
limitation of the CAT4026 but a limit set by an external
resistor divider (R42, R43) connected to the OCA pin. If the
anode voltage reaches a threshold of 122 V, the OCA fault
flag (FLT−OCA output) is triggered. The board is set−up to
accommodate LED strings with a total forward voltage of
around 105 V typical at nominal current (range between
95 V and 121 V).
In normal operation (PWM high), the anode voltage is
equal to the highest total LED forward voltage plus the
cathode voltage (3.6 V typical) of the LED string.
On the HB−LLC board, the CAT4026 IFB current
feedback in conjunction with the current amplifier (4x) via
the optocoupler controls the NCP1397 FB feedback which
sets the anode voltage. The IFB pin can sink up to a
maximum of 1 mA current.
LED Current Setting
The HB−LLC board is configured for LED current of
100 mA per channel. The LED current is set by external
resistors (10 W) located between each CAT4026 RSET[x]
pin and GND, as shown on Figure 10, and can be calculated
as follows:
LED current +
V RSET
1V
+
+ 100 mA
Rx
10 W
Fault Protection
The CAT4026 can detect two types of fault conditions:
open cathode−anode (OCA) caused by an open LED, and
short cathode−anode (SCA) caused by an LED string
voltage mismatch.
(eq. 4)
The LED current can be changed by replacing discrete
resistors. The following formula is used to set the LED
channel current:
Rx +
1V
LED current
(eq. 5)
Figure 10. Partial Schematic Around the CAT4026 (only 1 channel shown)
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Open LED
Dimming Control
To protect against an LED channel being open−circuit, the
maximum anode voltage is set on the HB−LLC board to
122 V. An open cathode−anode (OCA) fault is triggered
once the CAT4026 OCA pin voltage exceeds the 1 V
threshold. On the HB−LLC board, a resistor divider (R42,
R43) sets the open LED anode threshold (VOTH) to 121 V,
as follows:
LED dimming can be done by two methods: PWM
dimming using the CAT4026 PWM input, or analog
dimming using the ANLG input.
For PWM dimming, the duty cycle of the PWM input
signal sets the brightness. A function generator can be set
with a 100 to 300 Hz frequency and 5 V amplitude output
and connected directly to the PWM input on the connector
CON4 P8. For full LED brightness, the PWM input should
be connected to the 5 V supply.
The ANLG input (CON4 P7) is connected on the board to
a 5 kW pull−up to the 5 V supply. So there is no need to drive
the ANLG input externally (outside the board), ANLG is
pulled high in normal operation (no analog dimming). If
analog dimming is desired, a supply voltage can be
connected the ANLG input externally through a 500 W
resistor to limit the current in case the FLT−SCA fault turns
on. An ANLG voltage below 3 V will dim the LEDs linearly.
V OTH +
(eq. 6)
1 V @ (R42 ) R43)
1 V @ (1 k ) 121 k)
+
+ 122 V
R42
1k
The FLT−OCA open drain output is pulled low (active
low) and latched once the OCA fault is detected. Any
open−channel will automatically be disabled and removed
from the feedback loop when the OCA fault is triggered. The
remaining “good” channels can continue to operate
normally.
The FLT−OCA output is cleared only once the CAT4026
enters shutdown mode.
Feedback Circuit
The anode output voltage (VOUT) is controlled by the
CAT4026 IFB current feedback as shown in Figure 11. The
IFB pin sink current (1 mA max) is amplified by a current
mirror amplifier (4x) and drives the optocoupler which
controls the NCP1397 feedback FB pin voltage. The IFB pin
can sink up to a maximum of 1 mA current, with a typical of
0.5 mA in operation which translate in 4 mA max and 2 mA
typical in the optocoupler diode.
When PWM input is kept low, the CAT4026 is in
shutdown mode and the IFB pin does not sink current.
Therefore the optocoupler diode is not conducting. The
NCP1397A feedback FB pin voltage is around 7.7 V and the
SKIP pin is high at about 1 V (above the 0.66 V threshold)
which triggers the skip/disable mode. The LLC is not
switching. The anode voltage VOUT is zero volt. All LEDs
are off.
When PWM input is high (with 100 V LED string at
600 mA total load current), the CAT4026 is turned on and
the IFB pin current is around 0.3 mA. The optocoupler diode
is on. The NCP1397A feedback FB pin voltage is around
4.74 V and the SKIP pin voltage is about 0.61 V (below the
threshold). The LEDs are on.
Shorted LED
In case there is a large mismatch between LED string
forward voltage, such as the occurrence of a short between
several LEDs (anode to cathode), the power dissipated in the
external BJT transistors (Q6 to Q17) can become very large
causing the transistor package temperature to increase
excessively. The CAT4026 can detect a large voltage
mismatch of about 28 V or greater by sensing the cathode
voltages and detecting an SCA fault condition via the SCA
input pin. This SCA fault activates the FLT−SCA open−
drain output (active low). A derating circuitry on the board
decreases the LED current to 20% of nominal when the SCA
fault is on. The SCA pin is connected to each LED cathode
via a diode array and a voltage level translator. The threshold
voltage of the detector can be adjusted by using an external
Zener diode (D16) with 15 V breakdown in series with a
10 kW resistor. An unlatched signal will be produced by the
FLT−SCA pin. The fault FLT−SCA output is connected to
the ANLG pin through a diode (R160) and pulls the ANLG
pin lower to 0.6 V when the SCA fault is present (FLT−SCA
low), thereby limiting the current in each channel to
0.6 V/3 V x 100 mA or 20 mA.
Figure 22 shows the operation of the SCA fault
occurrence during power−up.
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Figure 11. Feedback Circuitry
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TYPICAL CHARACTERISTICS
Below are shown some typical transient waveforms during power−up, PWM dimming, and under open LED and shorted
LED fault conditions.
Normal Operation
Figure 12. Power Up with PWM High
Figure 13. Power Up with 300 Hz PWM
Figure 14. Power Up with PWM Transition High
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TYPICAL CHARACTERISTICS
PWM Dimming
Figure 15. PWM Dimming 300 Hz,
90% Duty Cycle
Figure 16. PWM Dimming 300 Hz,
50% Duty Cycle
Figure 17. PWM Dimming 300 Hz,
10% Duty Cycle
Figure 18. PWM Dimming 100 Hz,
1% Duty Cycle
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TYPICAL CHARACTERISTICS
Open LED and Short LED Detections
Figure 19 to Figure 21 show the open LED fault detection during power−up and in normal operation (LED disconnect).
Figure 22 shows the short LED detection, where the FLT−SCA output goes low when there is a voltage mismatch greater
than about 28 V between two cathode voltages of any LED strings. Once the SCA fault is triggered, all channels will run at
lower current (20% of the nominal current).
Figure 19. Single Channel Open LED, at
Power−up
Figure 20. Single Channel Open LED, at
Power−up
Figure 21. Single Channel Open LED in
Operation (LED Disconnect)
Figure 22. Shorted LED Channel at Power−up
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AND8479/D
Test Procedure
Before powering−up the HB−LLC board, an LED load
with a total forward voltage of about 100 V (for example 30
LEDs in series with 3.3 V Vf each) should be connected to
all the six LED channels on the connector CON2. The
connector CON2 includes 6 LED cathode pins and 6 anode
voltage pins shorted together.
Warning: It is important to power−up each supply and each
board in the correct sequence listed below in order to avoid
having the HB−LLC running in open loop; this could happen
for example when the PFC is powered prior to the 5 V supply
for the CAT4026.
Turn on the 5 V external supply.
Turn on the 15 V external supply.
Turn on the PFC supply unit which provides the 400 V rails.
With the PWM input set high (100% duty cycle), the 5 V
supply current should be around 20 mA.
The FLT−SCA and FLT−OCA should be both high.
Warnings:
Due to the very high−voltage present during operation of
the boards, power supplies and LED load, the set−up should
be handled with care. Make sure the hot side ground (for
400 V and 15 V supplies) is never connected to the cold side
ground (for 5 V supply).
When using a separate 400 VDC power supply, it is
recommended for safety reasons to set the power supply
current limit to 0.4 A.
The following steps are needed for the installation of the
set−up.
Connect a 5 V external supply to connector CON3 P1.
Connect the external supply Ground to connector CON3 P2.
Connect the PFC board ground to the connector CON1 P3
& P4.
Connect the PFC board 15 V rail to the connector CON1 P2.
Connect the PFC 400 V rail to the connector CON1 P7 & P8.
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AND8479/D
Board Schematic
Figure 23. HB−LLC Board Schematic (NCP1397 Section)
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AND8479/D
Figure 24. HB−LLC Board Schematic (CAT4026 Section)
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AND8479/D
BOARD LIST OF COMPONENTS
Table 2. HB−LLC BOARD LIST OF COMPONENTS
Name
Description
Value
Manufacturer
Part Number
Qty
C1
Electrolytic capacitor
NU
−
−
1
C2
Electrolytic capacitor
82 mF 250 V, 20%
Nippon Chemicon
EKXJ251BC820MJ40S
1
C3
Electrolytic capacitor
NU
−
−
1
C4
Electrolytic capacitor
22 mF 250 V, 20%
Nippon Chemicon
EKXG251BC220MJ25S
1
C6, C15, C24, C38
Ceramic capacitor
NU
−
−
4
C7
Electrolytic capacitor
39 mF 450 V, 20%
Nippon Chemicon
EKXJ451BC390MJ50S
1
C8
Electrolytic capacitor
C10, C29, C40, R40
Ceramic capacitor
C11, C36
Electrolytic capacitor
4u7/50 V, 20%
C12
Ceramic capacitor
C14
Ceramic capacitor
C16
Metalized Polypropylene Film
Capacitor
C17
C18
C21
C22
−
−
−
1
100 n, 10%
AVX Corporation
08055C104KAZ2A
4
Koshin
KLH−50V4, 7 mF
2
0.047 m, 10%
AVX Corporation
08053C473KAT2A
1
0.33 m, 10%
AVX Corporation
0805YC334KAT2A
1
15 nF/630 V, 5%
Xiamen Faratronic
CO., LTD.
C312J153K60C000
1
Ceramic capacitor
2n2, 10%
AVX Corporation
12062C222KAT2A
1
Ceramic capacitor
100 pF, 10%
AVX Corporation
08051A101KAT2A
1
Electrolytic capacitor
NU
−
−
1
Ceramic capacitor
NU
−
−
1
C23, C32
Ceramic capacitor
10 n, 10%
AVX Corporation
08055C103KAT2A
2
C35
Ceramic capacitor
220 n, 10%
AVX Corporation
08053C224KAT2A
1
C37
Ceramic capacitor
150 nF, 10%
AVX Corporation
08053C154KAT2A
1
C39
Ceramic capacitor
2n2, 15%
AVX Corporation
08055C222KAT2A
1
CY3
Ceramic capacitor
1 nF/Y1, 20%
Murata
DE1E3KX102MA5B
1
CON1, CON4
Connector
−
LEAMAX Enterprise
4324−08R
2
CON2
Connector
−
LEAMAX Enterprise
4324−12R
1
CON3
Connector
−
LEAMAX Enterprise
4324−03R
1
CON5
Connector
−
LEAMAX Enterprise
4324−07R
1
D1, D2, D5, D6
2.0 A, 200 V Ultrafast Rectifier
MUR220
ON Semiconductor
MUR220RLG
4
D3, D7
1 A 600 V Fast−Recovery
Rectifier
1N4937
ON Semiconductor
1N4937RLG
2
D4, D9, D10, D11, D12, D14,
D15, D20
Switching Diode, 250 V
BAS21
ON Semiconductor
BAS21LT1G
8
D8, D17, D18, D19
200 V Ultrafast Rectifier
NU
−
−
4
D13
Surface Mount Ultrafast Power
Rectifier
MURA160
ON Semiconductor
MURA160T3G
1
D16
Zener Diode 500 mW
15 V, 5%
ON Semiconductor
MMSZ15VT1G
1
D21
Switching Diode, 250 V
NU
−
−
1
D22
Zener Diode 500 mW
NU
−
−
1
D23
Zener Diode 500 mW
NU
−
−
1
D24, R160
Switching Diode, 100 V
MMSD4148
ON Semiconductor
MMSD4148T1G
2
HEATSINK_1
Aluminium Heatsink
−
Columbia−Staver
TP209ST,80.0,
7.0,NA,−−,02B
1
HEATSINK_2
Aluminium Heatsink
−
Columbia−Staver
TP209ST,120,
7.0,NA,−−,02B
1
HOLE1 − HOLE6
Ground Lugs
−
Kang Yang
GND−15
6
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AND8479/D
Table 2. HB−LLC BOARD LIST OF COMPONENTS
Name
Description
Value
Manufacturer
Part Number
Qty
IC1
High Performance Resonant
Mode Controller with Integrated
High Voltage Drivers
NCP1397A
ON Semiconductor
NCP1397ADR2G
1
IC2
General Purpose Transistor
PNP
BC856A
ON Semiconductor
BC856ALT1G
1
IC4
6−Channel LED Controller
CAT4026
ON Semiconductor
CAT4026V−T1
1
IC5
Low Input Bias Current, 1.8 V
Op Amp
NU
−
−
1
L1
Resonance Coil
L2
Inductor
TR1
Transformer, 5%
OK1, OK2
Optocoupler
Q1, Q2
N−MOSFET Power Transistor
Q4
100 mH, 7%
TDK
Y09750−01
1
2u2, 20%
Coilcraft
RFB0807−2R2L
1
−
TDK
X09738−01
1
PC817
Sharp
PC817X2J000F
2
STP5NK50ZFP
STMicroelectronics
STP5NK50ZFP
2
General Purpose Transistor
PNP
BC856A
ON Semiconductor
BC856ALT1G
1
Q5, Q18
General Purpose High Voltage
Transistor NPN
MSD42W
ON Semiconductor
MSD42WT1G
2
Q6, Q7, Q8, Q9, Q10, Q11
High Voltage Power Transistor
NPN
MJD340
ON Semiconductor
MJD340G
6
Q12, Q13, Q14, Q15, Q16,
Q17
Bipolar Power NPN
NU
ON Semiconductor
MJF47G
6
R1, R29, R168
Resistor SMD
27 k, 1%
Vishay Draloric
CRCW080527K0FKEA
3
R2, R3, R4, R6, R7, R10,
R17
Resistor SMD
180 k, 1%
Vishay Draloric
CRCW0805180KFKEA
7
R5, R13, R59, R147, R148
Resistor SMD
10 k, 1%
Vishay Draloric
CRCW080510K0FKEA
5
R8
Resistor SMD
160 k, 1%
Vishay Dale
CRCW0805160KFKEA
1
R9
Resistor SMD
11 k, 1%
Vishay Draloric
CRCW080511K0FKEA
1
R11
Resistor SMD
15 k, 1%
Vishay Draloric
CRCW080515K0FKEA
1
R12
Resistor SMD
16 k, 1%
Vishay Draloric
CRCW080516K0FKEA
1
R14, R15, R16, R18, R19,
R20, R23, R25, R26, R30,
R44, R47, R51, R52, R57,
R58, R62, R63, R64, R65,
R66, R67, R68, R69, R70,
R71, R72, R73, R74, R75,
R76, R77, R78, R79, R80,
R81, R83, R87, R91, R95
Resistor SMD
NU
−
−
40
R21, R22, R31
Resistor SMD
10 R, 1%
Vishay Draloric
CRCW080510R0FKEA
3
C9, R24, R27, R28, R34,
R36, R48, R49, R82, R84,
R85, R89, R101, R102,
R103, R104, R105, R106,
R107, R134, R156, R157,
R158, R159, R161, R162,
R163, R164, R165, R166,
R167
Resistor SMD
0 R, 1%
Vishay Draloric
CRCW08050000Z0EA
31
R43, R86, R99, R100, R135,
R136, R137, R138, R139,
R140, R141, R142, R143,
R144, R145, R146, R154
Resistor SMD
1 k, 1%
Vishay Draloric
CRCW08051K00FKEA
17
R32
Resistor SMD
200 R, 1%
Vishay Dale
CRCW0805200RFKEA
1
R33
Resistor trough hole, High
Voltage
4M7, 5%
Welwyn
VRW37−4M7JI
1
R35
Resistor SMD
6.04 k, 1%
Vishay Dale
CRCW08056K04FKEA
1
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AND8479/D
Table 2. HB−LLC BOARD LIST OF COMPONENTS
Name
Description
Value
Manufacturer
Part Number
Qty
R38, R151
Resistor SMD
47 k, 1%
Vishay Draloric
CRCW080547K0FKEA
2
R37, R39, R41, R50, R133,
R153
Resistor SMD
NU
−
−
6
R42
Resistor SMD
121 k, 1%
Vishay Draloric
CRCW1206121KFKEA
1
R45
Resistor SMD
3k9, 1%
Vishay Draloric
CRCW08053K90FKEA
1
R46
Resistor SMD
18 R, 1%
Vishay Dale
CRCW080518R0FKEA
1
R53
Resistor SMD
8k45, 1%
Vishay Dale
CRCW08058K45FKEA
1
R54, R55
Resistor SMD
9k1, 1%
Vishay Draloric
CRCW08059K10FKEA
2
R56
Resistor SMD
150 k, 1%
Vishay Draloric
CRCW0805150KFKEA
1
R60
Resistor SMD
1k24, 1%
Vishay Draloric
CRCW08051K24FKEA
1
R61
Resistor SMD
1k2, 1%
Vishay Draloric
CRCW08051K20FKEA
1
R88, Rx2
Resistor SMD
301 R, 1%
Vishay Dale
CRCW0805301RFKEA
2
R90, Rx3
Resistor SMD
100 R, 1%
Vishay Dale
CRCW0805100RFKEA
2
R149
Resistor SMD
1k8, 1%
Vishay Draloric
CRCW08051K80FKEA
1
R92, R93, R117, R130,
R131, R132
Resistor SMD
NU
−
−
6
R94
Ceramic capacitor
1 m, 10%
AVX Corporation
08053C105KAT2A
1
R96, R97, R98, R108, R109,
R110, R111, R112, R113,
R114, R115, R116, R118,
R119, R120, R121, R122,
R123, R124, R125, R126,
R127, R128, R129
Resistor SMD
0R, 5%
Vishay Draloric
CRCW12060000Z0EA
24
R150
Resistor SMD
100 k, 1%
Vishay Draloric
CRCW0805100KFKEA
1
R152
Resistor SMD
18 k, 1%
Vishay Draloric
CRCW080518K0FKEA
1
R155
Resistor SMD
4.99 k, 1%
Vishay Draloric
CRCW08054K99FKEA
1
Rx1
Resistor SMD
49R9, 1%
Vishay Dale
CRCW080549R9FKEA
1
Rx4
Resistor SMD
499 k, 1%
Vishay Dale
CRCW0805499KFKEA
1
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AND8479/D
References
1. CAT4026 data sheet
2. NCP1397 data sheet
3. Application note AND8281/D
4. Application note AND8255/D
5. Application note AND8257/D
6. Application note AND8344/D
7. Application note AND8241/D
8. Application note AND8327/D
9. Application note AND8460/D
10. Bo Yang − Topology Investigation for Front−End DC−DC Power Conversion for Distributed Power System
11. M. B. Borage, S. R. Tiwari and S. Kotaiah − Design Optimization for an LCL − Type Series Resonant Converter
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