LT8302 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/3.6A Switch Features Description 2.8V to 42V Input Voltage Range n3.6A, 65V Internal DMOS Power Switch n Low Quiescent Current: 106µA in Sleep Mode 380µA in Active Mode n Quasi-Resonant Boundary Mode Operation at Heavy Load n Low Ripple Burst Mode® Operation at Light Load n Minimum Load < 0.5% (Typ) of Full Output n No Transformer Third Winding or Opto-Isolator Required for Output Voltage Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n Temperature Compensation for Output Diode n Output Short-Circuit Protection n Thermally Enhanced 8-Lead SO Package The LT®8302 is a monolithic micropower isolated flyback converter. By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with two external resistors and a third optional temperature compensation resistor. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 3.6A, 65V DMOS power switch is integrated along with all the high voltage circuitry and control logic into a thermally enhanced 8-lead SO package. n Applications Isolated Automotive, Industrial, Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies n The LT8302 operates from an input voltage range of 2.8V to 42V and delivers up to 18W of isolated output power. The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522. Typical Application 2.8V to 32VIN/5VOUT Isolated Flyback Converter 3:1 470pF 10µF 39Ω VIN LT8302 1µF INTVCC • SW EN/UVLO GND 9µH • 1µH VOUT– 10mA TO 1.1A (VIN = 5V) 10mA TO 2.0A (VIN = 12V) 10mA TO 2.9A (VIN = 24V) RREF 115k TC 10k FRONT PAGE APPLICATION 85 220µF 154k RFB 90 VOUT+ 5V EFFICIENCY (%) VIN 2.8V TO 32V Efficiency vs Load Current 80 75 70 VIN = 5V VIN = 12V VIN = 24V 65 8302 TA01a 60 0 0.5 2.0 1.5 1.0 LOAD CURRENT (A) 2.5 3.0 8302 TA01b 8302fb For more information www.linear.com/LT8302 1 LT8302 Absolute Maximum Ratings (Note 1) SW (Note 2)...............................................................65V VIN.............................................................................42V EN/UVLO.....................................................................VIN RFB.........................................................VIN – 0.5V to VIN Current Into RFB.....................................................200µA INTVCC, RREF, TC..........................................................4V Operating Junction Temperature Range (Notes 3, 4) LT8302E, LT8302I.............................. –40°C to 125°C LT8302H............................................. –40°C to 150°C LT8302MP.......................................... –55°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec).................... 300°C Pin Configuration TOP VIEW EN/UVLO 1 INTVCC 2 VIN 3 GND 4 8 9 GND TC 7 RREF 6 RFB 5 SW S8E PACKAGE 8-LEAD PLASTIC SO θJA = 33°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8302ES8E#PBF LT8302ES8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C LT8302IS8E#PBF LT8302IS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C LT8302HS8E#PBF LT8302HS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C LT8302MPS8E#PBF LT8302MPS8E#TRPBF 8302 8-Lead Plastic SO –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 8302fb 2 For more information www.linear.com/LT8302 LT8302 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted. SYMBOL PARAMETER VIN VIN Voltage Range IQ VIN Quiescent Current CONDITIONS MIN l TYP 2.8 VEN/UVLO = 0.3V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 0.5 53 106 380 EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.3 0.75 EN/UVLO Enable Threshold Falling l 1.178 1.214 EN/UVLO Enable Hysteresis MAX UNIT 42 V 2 µA µA µA µA V 1.250 14 V mV IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.3 –0.1 0 2.5 0 0.1 2.7 0.1 µA µA µA VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.85 3 3.1 V IINTVCC INTVCC Current Limit VINTVCC = 2.8V 10 13 16 mA INTVCC UVLO Threshold Falling 2.39 2.47 2.55 INTVCC UVLO Hysteresis (RFB – VIN) Voltage 105 IRFB = 75µA to 125µA –50 RREF Regulation Voltage RREF Regulation Voltage Line Regulation l 2.8V ≤ VIN ≤ 42V 50 1.00 1.02 V –0.01 0 0.01 %/V TC Pin Voltage ITC TC Pin Current 1.00 fMIN Minimum Switching Frequency tON(MIN) Minimum Switch-On Time tOFF(MAX) Maximum Switch-Off Time ISW(MAX) Maximum Switch Current Limit ISW(MIN) Minimum Switch Current Limit RDS(ON) Switch On-Resistance ISW = 1.5A 80 ILKG Switch Leakage Current VSW = 65V 0.1 tSS Soft-Start Timer V 12 15 –200 18 µA µA 11.3 12 12.7 kHz 160 Backup Timer ns 170 µs 3.6 4.5 5.4 0.78 0.87 0.96 11 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 65V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 65V as shown in Figure 5. Note 3: The LT8302E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The mV 0.98 VTC VTC = 1.2V VTC = 0.8V V mV A A mΩ 0.5 µA ms LT8302I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8302H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8302MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8302 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 8302fb For more information www.linear.com/LT8302 3 LT8302 Typical Performance Characteristics Output Load and Line Regulation 5.3 5.00 4.95 VIN = 5V VIN = 12V VIN = 24V 4.80 0.5 0 1.0 2.0 1.5 LOAD CURRENT (A) 2.5 5.1 RTC = 115k 5.0 RTC = OPEN 4.9 4.7 –50 –25 0 VSW 20V/DIV VOUT 50mV/DIV VOUT 50mV/DIV 8302 G04 8302 G05 VIN Quiescent Current, Active Mode TJ = 150°C IQ (µA) 110 TJ = –55°C 30 VIN (V) 40 50 8302 G07 80 TJ = 25°C 380 TJ = –55°C 360 340 90 20 TJ = 150°C 400 TJ = 25°C 100 10 8302 G06 20µs/DIV FRONT PAGE APPLICATION VIN = 12V IOUT = 10mA 120 0 3.0 420 130 IQ (µA) IQ (µA) 0 2.5 VOUT 50mV/DIV 140 TJ = 150°C TJ = 25°C TJ = –55°C 2 2.0 1.0 1.5 LOAD CURRENT (A) Burst Mode Waveforms VIN Quiescent Current, Sleep Mode 4 0.5 VSW 20V/DIV 2µs/DIV FRONT PAGE APPLICATION VIN = 12V IOUT = 0.5A VIN Shutdown Current 6 0 8302 G03 Discontinuous Mode Waveforms VSW 20V/DIV 8 VIN = 5V VIN = 12V VIN = 24V 8302 G02 Boundary Mode Waveforms 10 200 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G01 2µs/DIV FRONT PAGE APPLICATION VIN = 12V IOUT = 2A 300 100 4.8 3.0 FRONT PAGE APPLICATION 400 FREQUENCY (kHz) 5.05 500 FRONT PAGE APPLICATION VIN = 12V IOUT = 1A 5.2 5.10 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 5.15 4.85 Switching Frequency vs Load Current Output Temperature Variation 5.20 4.90 TA = 25°C, unless otherwise noted. 0 10 20 30 VIN (V) 50 40 8302 G08 320 0 10 20 30 VIN (V) 40 50 8302 G09 8302fb 4 For more information www.linear.com/LT8302 LT8302 Typical Performance Characteristics EN/UVLO Enable Threshold TA = 25°C, unless otherwise noted. EN/UVLO Hysteresis Current 1.240 INTVCC Voltage vs Temperature 5 RISING 1.220 FALLING 3.00 3 IHYST (µA) VEN/UVLO (V) 1.225 1.215 3.05 4 1.230 VINTVCC (V) 1.235 3.10 2 1.210 1 0 IINTVCC = 10mA 2.85 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 2.95 2.90 1.205 1.200 –50 –25 IINTVCC = 0mA 0 2.80 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G10 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G12 8302 G11 INTVCC Voltage vs VIN INTVCC UVLO Threshold 3.10 2.8 3.05 2.7 (RFB-VIN) Voltage 40 30 IRFB = 125µA 2.6 IINTVCC = 0mA 2.95 VINTVCC (V) VINTVCC (V) 3.00 IINTVCC = 10mA RISING 2.5 2.90 2.4 2.85 2.3 VOLTAGE (mV) 20 FALLING 10 IRFB = 100µA 0 –10 –20 2.80 5 10 15 20 25 30 VIN (V) 35 40 –30 2.2 –50 –25 45 0 RREF Line Regulation TC Pin Voltage 1.008 1.006 1.006 1.004 1.004 1.002 1.002 1.5 1.4 1.3 1.2 VTC (V) VRREF (V) VRREF (V) 1.008 1.000 0.998 0.998 0.996 0.996 0.994 0.994 0.992 0.992 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G16 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G15 1.010 1.000 0 8302 G14 RREF Regulation Voltage 0.990 –50 –25 –40 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G13 1.010 IRFB = 75µA 0.990 1.1 1.0 0.9 0.8 0 10 20 30 VIN (V) 40 50 8302 G17 0.7 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G18 8302fb For more information www.linear.com/LT8302 5 LT8302 Typical Performance Characteristics Switch Current Limit 5 160 4 120 3 80 40 MAXIMUM CURRENT LIMIT 0 MINIMUM CURRENT LIMIT 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 Minimum Switching Frequency 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G22 25 50 75 100 125 150 TEMPERATURE (°C) Minimum Switch-Off Time 400 400 300 300 200 100 0 0 8302 G21 TIME (ns) TIME (ns) FREQUENCY (kHz) 0 –50 –25 0 –50 –25 Minimum Switch-On Time 16 4 200 8302 G20 20 8 300 100 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G19 12 400 2 1 0 –50 –25 Maximum Switching Frequency 500 FREQUENCY (kHz) 200 ISW (A) RESISTANCE (mΩ) RDS(ON) TA = 25°C, unless otherwise noted. 0 –50 –25 200 100 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G23 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 8302 G24 8302fb 6 For more information www.linear.com/LT8302 LT8302 Pin Functions EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8302. Pull the pin below 0.3V to shut down the LT8302. This pin has an accurate 1.214V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from VIN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. INTVCC (Pin 2): Internal 3V Linear Regulator Output. The INTVCC pin is supplied from VIN and powers the internal control circuitry and gate driver. Do not overdrive the INTVCC pin with any external supply, such as a third winding supply. Locally bypass this pin to ground with a minimum 1µF ceramic capacitor. VIN (Pin 3): Input Supply. The VIN pin supplies current to the internal circuitry and serves as a reference voltage for the feedback circuitry connected to the RFB pin. Locally bypass this pin to ground with a capacitor. GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed pad provides both electrical contact to ground and good thermal contact to the printed circuit board. Solder the exposed pad directly to the ground plane. SW (Pin 5): Drain of the Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. RFB (Pin 6): Input Pin for External Feedback Resistor. Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the RREF resistor, times the internal voltage reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. RREF (Pin 7): Input Pin for External Ground Referred Reference Resistor. The resistor at this pin should be in the range of 10k, but for convenience in selecting a resistor divider ratio, the value may range from 9.09k to 11.0k. TC (Pin 8): Output Voltage Temperature Compensation. The voltage at this pin is proportional to absolute temperature (PTAT) with temperature coefficient equal to 3.35mV/°K, i.e., equal to 1V at room temperature 25°C. The TC pin voltage can be used to estimate the LT8302 junction temperature. Connect a resistor from this pin to the RREF pin to compensate the output diode temperature coefficient. 8302fb For more information www.linear.com/LT8302 7 LT8302 Block Diagram T1 N:1 VIN CIN L1A RFB 3 2 6 1:4 M3 M2 REN2 OSCILLATOR – 1.214V + 1V – A1 VOUT– START-UP, REFERENCE, CONTROL BOUNDARY DETECTOR + 1 EN/UVLO COUT SW 25µA REN1 L1B VIN LDO CINTVCC • VOUT+ 5 RFB VIN INTVCC • DOUT INTVCC – gm + S A3 R Q M1 DRIVER 2.5µA PTAT VOLTAGE M4 + A2 RSENSE – RREF 7 TC GND 4, EXPOSED PAD PIN 9 8 RTC 8302 BD RREF 8302fb 8 For more information www.linear.com/LT8302 LT8302 Operation The LT8302 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8302 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8302 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. The LT8302 is a simple to use micropower isolated flyback converter housed in a thermally enhanced 8-lead SO package. The output voltage is programmed with two external resistors. An optional TC resistor provides easy output diode temperature compensation. By integrating the loop compensation and soft-start inside, the part reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sampleand-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Quasi-Resonant Boundary Mode Operation The LT8302 features quasi-resonant boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero and the SW rings to its valley. Boundary conduction mode is a variable frequency, variable peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on at its valley. 8302fb For more information www.linear.com/LT8302 9 LT8302 Operation Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit subharmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8302 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 380kHz. Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8302 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8302 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-andhold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effective quiescent current to improve light load efficiency. In this condition, the LT8302 runs in low ripple Burst Mode operation. The typical 12kHz minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement. 8302fb 10 For more information www.linear.com/LT8302 LT8302 Applications Information Output Voltage The RFB and RREF resistors as depicted in the Block Diagram are external resistors used to program the output voltage. The LT8302 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the VIN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = Output diode forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current, IRFB, by the RFB resistor and the flyback pulse sense circuit (M2 and M3). This current, IRFB, also flows through the RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sampleand-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The internal reference voltage, VREF, 1.00V, feeds to the noninverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes the voltage at the RREF pin to be nearly equal to the internal reference voltage VREF. The resulting relationship between VFLBK and VREF can be expressed as: ⎛ VFLBK ⎞ ⎜⎝ R ⎟⎠ •RREF = VREF or FB ⎛ R ⎞ VFLBK = VREF • ⎜ FB ⎟ ⎝ RREF ⎠ VREF = Internal reference voltage 1.00V Combination with the previous VFLBK equation yields an equation for VOUT, in terms of the RFB and RREF resistors, transformer turns ratio, and diode forward voltage: ⎛ R ⎞ ⎛ 1 ⎞ VOUT = VREF • ⎜ FB ⎟ • ⎜ – VF ⎝ RREF ⎠ ⎝ NPS ⎟⎠ Output Temperature Compensation The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage, VF, has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. The LT8302 junction temperature usually tracks the output diode junction temperature to the first order. To compensate the negative temperature coefficient of the output diode, a resistor, RTC, connected between the TC and RREF pins generates a proportional-to-absolute-temperature (PTAT) current. The PTAT current is zero at 25°C, flows into the RREF pin at hot temperature, and flows out of the RREF pin at cold temperature. With the RTC resistor in place, the output voltage equation is revised as follows: VOUT = VREF • RFB 1 • – VF (TO) – ( VTC / T ) • RREF NPS ( T –TO) • RFB 1 • – ( VF / T ) • ( T–TO) R TC NPS TO=Room temperature 25°C ( ( VF / T ) = Output diode forward voltage temperature coefficient VTC / T ) = 3.35mV/ °C 8302fb For more information www.linear.com/LT8302 11 LT8302 Applications Information To cancel the output diode temperature coefficient, the following two equations should be satisfied: VOUT = VREF • ( RFB 1 • – VF (TO) RREF NPS R 1 VTC / T) • FB • = – ( VF / T ) R TC NPS Selecting Actual RREF, RFB, RTC Resistor Values The LT8302 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the RFB and RTC resistor values. Therefore, a simple 2-step sequential process is recommended for selecting resistor values. Rearrangement of the expression for VOUT in the previous sections yields the starting value for RFB: RFB = ( RREF •NPS • VOUT + VF (TO) VREF ) VOUT = Output voltage VF (TO) = Output diode forward voltage at 25°C = ~0.3V NPS = Transformer effective primary-to-secondary turns ratio The equation shows that the RFB resistor value is independent of the RTC resistor value. Any RTC resistor connected between the TC and RREF pins has no effect on the output voltage setting at 25°C because the TC pin voltage is equal to the RREF regulation voltage at 25°C. The RREF resistor value should be approximately 10k because the LT8302 is trimmed and specified using this value. If the RREF resistor value varies considerably from 10k, additional errors will result. However, a variation in RREF up to 10% is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios. First, build and power up the application with the starting RREF, RFB values (no RTC resistor yet) and other components connected, and measure the regulated output voltage, VOUT(MEAS). The new RFB value can be adjusted to: RFB(NEW) = VOUT VOUT(MEAS) •RFB Second, with a new RFB resistor value selected, the output diode temperature coefficient in the application can be tested to determine the RTC value. Still without the RTC resistor, the VOUT should be measured over temperature at a desired target output load. It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT8302. If freeze spray or a heat gun is used, there can be a significant mismatch in temperature between the two devices that causes significant error. Attempting to extrapolate the data from a diode data sheet is another option if there is no method to apply uniform heating or cooling such as an oven. With at least two data points spreading across the operating temperature range, the output diode temperature coefficient can be determined by: – ( δVF /δT ) = VOUT ( T1) – VOUT ( T2) T1– T2 Using the measured output diode temperature coefficient, an exact RTC value can be selected with the following equation: R TC = (δVTC /δT ) • ⎛ RFB ⎞ – ( δVF /δT ) ⎜⎝ NPS ⎟⎠ Once the RREF, RFB, and RTC values are selected, the regulation accuracy from board to board for a given application will be very consistent, typically under ±5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. 8302fb 12 For more information www.linear.com/LT8302 LT8302 Applications Information Output Power A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter. A boost converter has a relatively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinuous input and output currents which make it similar to a nonisolated buck-boost converter. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 50V during the switch-off time. 15V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 50V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 8V and a maximum input voltage of 32V. A three-to-one winding ratio fits this design example perfectly and outputs equal to 15.3W at 32V but lowers to 7.7W at 8V. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 20 20 MAXIMUM OUTPUT POWER 15 N = 6:1 OUTPUT POWER (W) OUTPUT POWER (W) MAXIMUM OUTPUT POWER N = 4:1 N = 3:1 10 N = 2:1 5 0 0 20 10 30 INPUT VOLTAGE (V) OUTPUT POWER (W) OUTPUT POWER (W) N = 3:1 N = 2:1 N = 1:1 5 10 N = 1:2 5 0 20 INPUT VOLTAGE (V) 30 20 10 30 40 MAXIMUM OUTPUT POWER N = 1:1 15 N = 2:3 N = 1:2 10 N = 1:3 5 0 0 10 20 INPUT VOLTAGE (V) 8302 F02 Figure 2. Output Power for 5V Output 40 8302 F03 Figure 3. Output Power for 12V Output N = 4:1 0 10 20 10 N = 1:1 INPUT VOLTAGE (V) MAXIMUM OUTPUT POWER 15 0 N = 3:2 8302 F01 Figure 1. Output Power for 3.3V Output 20 15 0 40 N = 2:1 30 40 8302 F04 Figure 4. Output Power for 24V Output 8302fb For more information www.linear.com/LT8302 13 LT8302 Applications Information The equations below calculate output power: the power switch shorter than approximately 160ns. This minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor current exceeds the desired current limit during that time, oscillation may occur at the output as the current control loop will lose its ability to regulate. Therefore, the following equation relating to maximum input voltage must also be followed in selecting primary-side magnetizing inductance: POUT = η • VIN • D • ISW(MAX) • 0.5 η = Efficiency = ~85% D = Duty Cycle = ( VOUT + VF ) •NPS ( VOUT + VF ) •NPS + VIN ISW(MAX) = Maximum switch current limit = 3.6A (MIN) LPRI ≥ Primary Inductance Requirement ISW(MIN) tON(MIN) = Minimum switch-on time = 160ns (TYP) The LT8302 obtains output voltage information from the reflected output voltage on the SW pin. The conduction of secondary current reflects the output voltage on the primary SW pin. The sample-and-hold error amplifier needs a minimum 350ns to settle and sample the reflected output voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of 350ns. The following equation gives the minimum value for primary-side magnetizing inductance: LPRI ≥ tON(MIN) • VIN(MAX) In general, choose a transformer with its primary magnetizing inductance about 40% to 60% larger than the minimum values calculated above. A transformer with much larger inductance will have a bigger physical size and may cause instability at light load. Selecting a Transformer tOFF(MIN) •NPS • ( VOUT + VF ) Transformer specification and design is perhaps the most critical part of successfully applying the LT8302. In addition to the usual list of guidelines dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. ISW(MIN) tOFF(MIN) = Minimum switch-off time = 350ns (TYP) ISW(MIN) = Minimum switch current limit = 0.87A (TYP) Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT8302. Table 1 shows the details of these transformers. In addition to the primary inductance requirement for the minimum switch-off time, the LT8302 has minimum switch-on time that prevents the chip from turning on Table 1. Predesigned Transformers–Typical Specifications TRANSFORMER PART NUMBER RSEC (mΩ) VENDOR TARGET APPLICATION VIN (V) VOUT (V) IOUT (A) DIMENSIONS (W × L × H) (mm) LPRI (µH) LLKG (µH) NP:NS RPRI (mΩ) 750311625 17.75 × 13.46 × 12.70 9 0.35 4:1 43 6 Würth Elektronik 8 to 32 3.3 2.1 750311564 17.75 × 13.46 × 12.70 9 0.12 3:1 36 7 Würth Elektronik 8 to 32 5 1.5 750313441 15.24 × 13.34 x 11.43 9 0.6 2:1 75 18 Würth Elektronik 8 to 32 5 1.3 750311624 17.75 × 13.46 × 12.70 9 0.18 3:2 34 21 Würth Elektronik 8 to 32 8 0.9 750313443 15.24 × 13.34 × 11.43 9 0.3 1:1:1 85 100 Würth Elektronik 8 to 36 ±12 0.3 750313445 15.24 × 13.34 × 11.43 9 0.25 1:2 85 190 Würth Elektronik 8 to 36 24 0.3 750313457 15.24 × 13.34 × 11.43 9 0.25 1:4 85 770 Würth Elektronik 8 to 36 48 0.15 750313460 15.24 × 13.34 × 11.43 12 0.7 4:1 85 11 Würth Elektronik 4 to 18 5 0.9 750311342 15.24 × 13.34 × 11.43 15 0.44 2:1 85 22 Würth Elektronik 4 to 18 12 0.4 750313439 15.24 × 13.34 × 11.43 12 0.6 2:1 115 28 Würth Elektronik 18 to 42 3.3 2.1 750313442 15.24 × 13.34 × 11.43 12 0.75 3:2 150 53 Würth Elektronik 18 to 42 5 1.6 8302fb 14 For more information www.linear.com/LT8302 LT8302 Applications Information Turns Ratio Note that when choosing an RFB/RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, the use of simple ratios of small integers, e.g., 3:1, 2:1, 1:1, etc., provides more freedom in settling total turns and mutual inductance. Typically, choose the transformer turns ratio to maximize available output power. For low output voltages (3.3V or 5V), a N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 65V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure NPS < 65V – VIN(MAX) – VLEAKAGE VOUT + VF For larger N:1 values, choose a transformer with a larger physical size to deliver additional current. In addition, choose a large enough inductance value to ensure that the switch-off time is long enough to accurately sample the output voltage. For lower output power levels, choose a 1:1 or 1:N transformer for the absolute smallest transformer size. A 1:N transformer will minimize the magnetizing inductance (and minimize size), but will also limit the available output power. A higher 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within ±1%. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8302, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8302. Leakage Inductance and Snubbers Transformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 50V. This leaves at least 15V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. 8302fb For more information www.linear.com/LT8302 15 LT8302 Applications Information <65V VLEAKAGE <50V VSW tOFF > 350ns tSP < 250ns TIME 8302 F05 then add capacitance until the period of the ringing is 1.5 to 2 times longer. The change in period determines the value of the parasitic capacitance, from which the parasitic inductance can be also determined from the initial period. Once the value of the SW node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically damp the ringing. The equation for deriving the optimal series resistance using the observed periods ( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is: Figure 5. Maximum Voltages for SW Pin Flyback Waveform In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trigger boundary mode detector, the LT8302 internally blanks the boundary mode detector for approximately 250ns. Any remaining voltage ringing after 250ns may turn the power switch back on again before the secondary current falls to zero. In this case, the LT8302 enters continuous conduction mode. So the leakage inductance spike ringing should be limited to less than 250ns. To clamp and damp the leakage voltage spikes, a (RC + DZ) snubber circuit in Figure 6 is recommended. The RC (resistor-capacitor) snubber quickly damps the voltage spike ringing and provides great load regulation and EMI performance. And the DZ (diode-Zener) ensures well defined and consistent clamping voltage to protect SW pin from exceeding its 65V absolute maximum rating. Lℓ Z D • C R • CPAR = CSNUBBER 2 ⎛ tPERIOD(SNUBBED) ⎞ ⎜⎝ ⎟⎠ – 1 t PERIOD LPAR = tPERIOD2 CPAR • 4π 2 RSNUBBER = LPAR CPAR Note that energy absorbed by the RC snubber will be converted to heat and will not be delivered to the load. In high voltage or high current applications, the snubber needs to be sized for thermal dissipation. A 470pF capacitor in series with a 39Ω resistor is a good starting point. For the DZ snubber, proper care should be taken when choosing both the diode and the Zener diode. Schottky diodes are typically the best choice, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch voltage protection. The best compromise is to choose the largest voltage breakdown with 5V margin. Use the following equation to make the proper choice: VZENNER(MAX) ≤ 60V – VIN(MAX) 8302 F06 Figure 6. (RC + DZ) Snubber Circuit The recommended approach for designing an RC snubber is to measure the period of the ringing on the SW pin when the power switch turns off without the snubber and For an application with a maximum input voltage of 32V, choose a 24V Zener diode, the VZENER(MAX) of which is around 26V and below the 28V maximum. The power loss in the DZ snubber determines the power rating of the Zener diode. A 1.5W Zener diode is typically recommended. 8302fb 16 For more information www.linear.com/LT8302 LT8302 Applications Information Undervoltage Lockout (UVLO) A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO enable falling threshold is set at 1.214V with 14mV hysteresis. In addition, the EN/UVLO pin sinks 2.5µA when the voltage on the pin is below 1.214V. This current provides user programmable hysteresis based on the value of R1. The programmable UVLO thresholds are: 1.228V • (R1+R2) + 2.5µA •R1 R2 1.214V • (R1+R2) VIN(UVLO– ) = R2 VIN(UVLO+ ) = Figure 7 shows the implementation of external shutdown control while still using the UVLO function. The NMOS grounds the EN/UVLO pin when turned on, and puts the LT8302 in shutdown with quiescent current less than 2µA. during light load conditions to ensure accurate output voltage information. The minimum energy delivery creates a minimum load requirement, which can be approximately estimated as: ILOAD(MIN) = LPRI •ISW(MIN)2 • fMIN 2 • VOUT LPRI = Transformer primary inductance ISW(MIN) = Minimum switch current limit = 0.96A (MAX) fMIN = Minimum switching frequency = 12.7kHz (MAX) The LT8302 typically needs less than 0.5% of its full output power as minimum load. Alternatively, a Zener diode with its breakdown of 10% higher than the output voltage can serve as a minimum load if pre-loading is not acceptable. For a 5V output, use a 5.6V Zener with cathode connected to the output. Output Short Protection VIN R1 EN/UVLO RUN/STOP CONTROL (OPTIONAL) R2 LT8302 GND 8302 F07 Figure 7. Undervoltage Lockout (UVLO) Minimum Load Requirement The LT8302 samples the isolated output voltage from the primary-side flyback pulse waveform. The flyback pulse occurs once the primary switch turns off and the secondary winding conducts current. In order to sample the output voltage, the LT8302 has to turn on and off for a minimum amount of time and with a minimum frequency. The LT8302 delivers a minimum amount of energy even When the output is heavily overloaded or shorted to ground, the reflected SW pin waveform rings longer than the internal blanking time. After the 350ns minimum switch-off time, the excessive ringing falsely triggers the boundary mode detector and turns the power switch back on again before the secondary current falls to zero. Under this condition, the LT8302 runs into continuous conduction mode at 380kHz maximum switching frequency. If the sampled RREF voltage is still less than 0.6V after 11ms (typ) soft-start timer, the LT8302 initiates a new soft-start cycle. If the sampled RREF voltage is larger than 0.6V after 11ms, the switch current may run away and exceed the 4.5A maximum current limit. Once the switch current hits 7.2A over current limit, the LT8302 also initiates a new soft-start cycle. Under either condition, the new soft-start cycle throttles back both the switch current limit and switch frequency. The output short-circuit protection prevents the switch current from running away and limits the average output diode current. 8302fb For more information www.linear.com/LT8302 17 LT8302 Applications Information Design Example Step 2: Determine the primary inductance. Use the following design example as a guide to designing applications for the LT8302. The design example involves designing a 5V output with a 1.5A load current and an input range from 8V to 32V. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ VIN(MIN) = 8V, VIN(NOM) = 12V, VIN(MAX) = 32V, VOUT = 5V, IOUT = 1.5A Step 1: Select the transformer turns ratio. NPS < 65V – VIN(MAX) – VLEAKAGE VOUT + VF VLEAKAGE = Margin for transformer leakage spike = 15V VF = Output diode forward voltage = ~0.3V Example: NPS < 65V – 32V – 15V = 3.4 5V + 0.3V LPRI ≥ tOFF(MIN) • NPS • ( VOUT + VF ) ISW(MIN) tON(MIN) • VIN(MAX) ISW(MIN) tOFF(MIN) = 350ns tON(MIN) = 160ns ISW(MIN) = 0.87A Example: 350ns • 3 • ( 5V + 0.3V ) = 6.4µH 0.87A 160ns • 32V = 5.9µH LPRI ≥ 0.87A LPRI ≥ The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 2 shows the switch voltage stress and output current capability at different transformer turns ratio. Table 2. Switch Voltage Stress and Output Current Capability vs Turns Ratio NPS VSW(MAX) at VIN(MAX) (V) IOUT(MAX) at VIN(MIN) (A) DUTY CYCLE (%) 1:1 37.3 0.92 14-40 2:1 42.6 1.31 25-57 3:1 47.9 1.53 33-67 Most transformers specify primary inductance with a tolerance of ±20%. With other component tolerance considered, choose a transformer with its primary inductance 40% to 60% larger than the minimum values calculated above. LPRI = 9µH is then chosen in this example. Once the primary inductance has been determined, the maximum load switching frequency can be calculated as: fSW = Clearly, only NPS = 3 can meet the 1.5A output current requirement, so NPS = 3 is chosen as the turns ratio in this example. ISW = 1 1 = LPRI •ISW tON + tOFF LPRI •ISW + VIN NPS • ( VOUT + VF ) VOUT •IOUT • 2 η • VIN • D 8302fb 18 For more information www.linear.com/LT8302 LT8302 Applications Information Example: D= to calculate the output capacitance: (5V + 0.3V ) • 3 = 0.57 (5V + 0.3V ) • 3 + 12V 5V • 1.5A • 2 ISW = 0.8 • 12V • 0.57 fSW = 277kHz Design for output voltage ripple less than ±1% of VOUT, i.e., 100mV. Step 3: Choose the output diode. Two main criteria for choosing the output diode include forward current rating and reverse-voltage rating. The maximum load requirement is a good first-order guess at the average current requirement for the output diode. Under output short-circuit condition, the output diode needs to conduct much higher current. Therefore, a conservative metric is 60% of the maximum switch current limit multiplied by the turns ratio: IDIODE(MAX) = 0.6 • ISW(MAX) • NPS COUT = 2 9µH • ( 4.5A ) = 182µF 2 • 5V • 0.1V Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 220µF, 6.3V rating ceramic capacitor is chosen. Step 5: Design snubber circuit. The snubber circuit protects the power switch from leakage inductance voltage spike. A (RC + DZ) snubber is recommended for this application. A 470pF capacitor in series with a 39Ω resistor is chosen as the RC snubber. The maximum Zener breakdown voltage is set according to the maximum VIN: VZENNER(MAX) ≤ 60V – VIN(MAX) Example: Example: IDIODE(MAX) = 8.1A Next calculate reverse voltage requirement using maximum VIN: LPRI •ISW2 2 • VOUT • ΔVOUT Example: The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 7A is necessary to work with the LT8302. The 750311564 from Würth is chosen as the flyback transformer. VREVERSE = VOUT + COUT = VIN(MAX) NPS VZENNER(MAX) ≤ 60V – 32V = 28V A 24V Zener with a maximum of 26V will provide optimal protection and minimize power loss. So a 24V, 1.5W Zener from Central Semiconductor (CMZ5934B) is chosen. Choose a diode that is fast and has sufficient reverse voltage breakdown: Example: VREVERSE > VSW(MAX) 32V VREVERSE = 5V + = 15.7V 3 VSW(MAX) = VIN(MAX) + VZENNER(MAX) The PDS835L (8A, 35V diode) from Diodes Inc. is chosen. Step 4: Choose the output capacitor. The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the following equation Example: VREVERSE > 60V A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen. 8302fb For more information www.linear.com/LT8302 19 LT8302 Applications Information Step 6: Select the RREF and RFB resistors. Example: Use the following equation to calculate the starting values for RREF and RFB: RFB = ( – ( δVF /δT ) = ) RREF • NPS • VOUT + VF ( TO) VREF RTC = 5.189V – 5.041V = 1.48mV / °C 100°C – ( 0°C) 3.35mV/°C ⎛ 154 ⎞ •⎜ ⎟ = 115k 1.48mV/°C ⎝ 3 ⎠ RREF = 10k Step 9: Select the EN/UVLO resistors. Example: Determine the amount of hysteresis required and calculate R1 resistor value: RFB = 10k • 3 • ( 5V + 0.3V ) = 159k 1.00V VIN(HYS) = 2.5µA • R1 For 1% standard values, a 158k resistor is chosen. Step 7: Adjust RFB resistor based on output voltage. Build and power up the application with application components and measure the regulated output voltage. Adjust RFB resistor based on the measured output voltage: RFB(NEW) = VOUT VOUT(MEASURED) Choose 2V of hysteresis, R1 = 806k Determine the UVLO thresholds and calculate R2 resistor value: • RFB VIN(UVLO+) = 1.228V • (R1+ R2) + 2.5µA • R1 R2 Example: Set VIN UVLO rising threshold to 7.5V: Example: RFB = Example: 5V • 158k = 154k 5.14V Step 8: Select RTC resistor based on output voltage temperature variation. R2 = 232k VIN(UVLO+) = 7.5V VIN(UNLO–) = 5.5V Step 10: Ensure minimum load. Measure output voltage in a controlled temperature environment like an oven to determine the output temperature coefficient. Measure output voltage at a consistent load current and input voltage, across the operating temperature range. The theoretical minimum load can be approximately estimated as: Calculate the temperature coefficient of VF: Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the converter delivers more energy than what is consumed at the output. The real minimum load for this application is about 10mA. In this example, a 500Ω resistor is selected as the minimum load. VOUT ( T1) – VOUT ( T2) T1– T2 3.35mV/°C ⎛ RFB ⎞ RTC = • – ( δVF /δT ) ⎜⎝ NPS ⎟⎠ – ( δVF /δT ) = 2 9µH • ( 0.96A ) • 12.7kHz ILOAD(MIN) = =10.5mA 2 • 5V 8302fb 20 For more information www.linear.com/LT8302 LT8302 Typical Applications 8V to 32VIN/12VOUT Isolated Flyback Converter VIN 8V TO 32V Z1 C1 10µF R1 806k D1 VIN EN/UVLO R2 232k SW RFB RREF INTVCC C2 1µF D2 • C4 47µF 9µH R6 OPEN TC D1: DIODES DFLS1100 D2: DIODES PDS540 T1: WURTH 750313443 Z1: CENTRAL CMZ5934B R5 10k 8302 TA02a Load and Line Regulation 95 12.4 90 12.2 OUTPUT VOLTAGE (V) EFFICIENCY (%) Efficiency vs Load Current 85 80 75 70 65 200 400 600 800 1000 LOAD CURRENT (mA) 12.0 11.8 11.6 11.4 VIN = 12V VIN = 24V 0 VOUT+ 12V 5mA TO 0.8A (VIN = 12V) 5mA TO 1.1A (VIN = 24V) VOUT– R4 121k LT8302 GND C3 470pF R3 9µH 39Ω • T1 1:1 11.2 1200 VIN = 12V VIN = 24V 0 200 400 600 800 1000 LOAD CURRENT (mA) 8302 TA02c 8302 TA02b 8V to 32VIN/3.3VOUT Isolated Flyback Converter C1 10µF Z1 R1 806k R2 232k C2 1µF D1 VIN SW EN/UVLO R4 140k LT8302 GND INTVCC RFB RREF TC C3 470pF R3 9µH 39Ω • R6 105k R5 10k T1 4:1 D2 • 0.56µH Output Temperature Variation VOUT+ 3.3V 20mA TO 2.7A (VIN = 12V) 20mA TO 3.8A (VIN = 24V) C4 470µF – VOUT D1: DIODES DFLS1100 D2: DIODES PDS1040L T1: WURTH 750311625 Z1: CENTRAL CMZ5934B 8302 TA03 3.50 3.45 OUTPUT VOLTAGE (V) VIN 8V TO 32V 1200 VIN = 12V IOUT = 1A 3.40 3.35 3.30 RTC = 105k 3.25 RTC = OPEN 3.20 3.15 3.10 –50 –25 0 25 50 75 100 125 150 AMBIENT TEMPERATURE (°C) 8302 TA03b 8302fb For more information www.linear.com/LT8302 21 LT8302 Typical Applications 8V to 36VIN/±12VOUT Isolated Flyback Converter T1 D2 1:1:1 VIN 8V TO 36V Z1 C1 10µF R1 806k R2 232k C2 1µF D1 VIN EN/UVLO SW RFB RREF INTVCC TC • R4 121k LT8302 GND C3 470pF R3 9µH 39Ω • C4 22µF D3 • R6 OPEN 9µH R5 10k 9µH C5 22µF VOUT1+ 12V 5mA TO 0.4A (VIN = 12V) 5mA TO 0.55A (VIN = 24V) VOUT2– VOUT2+ 12V 5mA TO 0.4A (VIN = 12V) 5mA TO 0.55A (VIN = 24V) VOUT2– 8302 TA04 D1: DIODES DFLS1100 D2, D3: DIODES PDS360 T1: WURTH 750313443 Z1: CENTRAL CMZ5934B 8V to 36VIN/24VOUT Isolated Flyback Converter VIN 8V TO 36V Z1 C1 10µF R1 806k R2 232k C2 1µF D1 VIN EN/UVLO SW RFB INTVCC RREF TC D2 • 36µH R6 OPEN R5 10k VOUT+ 24V 2.5mA TO 0.4A (VIN = 12V) 2.5mA TO 0.55A (VIN = 24V) C4 10µF VOUT– R4 121k LT8302 GND C3 470pF R3 9µH 39Ω • T1 1:2 D1: DIODES DFLS1100 D2: DIODES SBR2U150SA T1: WURTH 750313445 Z1: CENTRAL CMZ5934B 8302 TA05 8V to 36VIN/48VOUT Isolated Flyback Converter VIN 8V TO 36V Z1 C1 10µF R1 806k R2 232k C2 1µF D1 VIN EN/UVLO SW RFB INTVCC RREF TC D2 • 144µH R4 121k LT8302 GND C3 470pF R3 9µH 39Ω • T1 1:4 R6 OPEN R5 10k VOUT+ 48V 1.2mA TO 0.2A (VIN = 12V) 1.2mA TO 0.27A (VIN = 24V) C4 2.2µF VOUT– D1: DIODES DFLS1100 D2: DIODES SBR1U200P1 T1: WURTH 750313457 Z1: CENTRAL CMZ5934B 8302 TA06 8302fb 22 For more information www.linear.com/LT8302 LT8302 Typical Applications 8V to 32VIN/5VOUT Isolated Flyback Converter with LT8309 C3 470pF R3 9µH 39Ω • Z1 C1 10µF R1 806k R2 232k C2 1µF D1 VIN EN/UVLO SW RFB RREF INTVCC • R6 OPEN 1µH TC D2 R8 2.1k VCC DRAIN LT8309 GATE INTVCC GND M1 D1: DIODES DFLS1100 D2: CENTRAL CMMSH1-60 M1: INFINEON BSC059N04LS T1: WURTH 750311564 Z1: CENTRAL CMZ5934B 95 90 C4 220µF R7 5Ω C4 10µF R5 10k Efficiency vs Load Current VOUT+ 5V/1.1A (VIN = 5V) 5V/2.0A (VIN = 12V) 5V/2.9A (VIN = 24V) R4 154k LT8302 GND T1 3:1 EFFICIENCY (%) VIN 8V TO 32V 85 80 75 70 C5 4.7µF 8302 TA07 VOUT– 65 0 0.5 2.0 1.5 1.0 LOAD CURRENT (A) 2.5 3.0 8302 TA07b –4V to –42VIN/12VOUT Buck-Boost Converter VIN RFB EN/UVLO C1 10µF R4 Z1 118k SW LT8302 RREF INTVCC C2 1µF D1: DIODES PMEG6030EP L1: WÜRTH 744770112 Z1: CENTRAL CMHZ5243B R5 10k GND VIN –4V TO –42V 95 VOUT 12V/0.45A (VIN = –5V) 12V/0.8A (VIN = –12V) 12V/1.1A (VIN = –24V) C3 47µF 12V/1.3A (VIN = –42V) D1 90 EFFICIENCY (%) L1 12µH Efficiency vs Load Current 8302 TA08a 85 80 75 VIN = –5V VIN = –12V VIN = –24V VIN = –42V 70 65 0 200 400 600 800 1000 1200 1400 LOAD CURRENT (mA) 8302 TA08b –18V to –42VIN/–12VOUT Negative Buck Converter Efficiency vs Load Current 100 C3 47µF R1 806k C1 10µF R2 232k EN/UVLO INTVCC C2 1µF SW LT8302 EN/UVLO VIN –18V TO –42V L1 12µH VIN VOUT –12V 1.8A R4 118k D1: DIODES PMEG6030EP L1: WÜRTH 744770112 Z1: CENTRAL CMHZ5243B RFB RREF R5 10k 95 EFFICIENCY (%) Z1 D1 90 85 80 70 8302 TA09a VIN = –18V VIN = –24V VIN = –42V 75 0 500 1000 1500 LOAD CURRENT (mA) 2000 8302 TA09b 8302fb For more information www.linear.com/LT8302 23 LT8302 Package Description Please refer to http://www.linear.com/product/LT8302#packaging for the most recent package drawings. S8E Package 8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad (Reference LTC DWG # 05-08-1857 Rev C) .050 (1.27) BSC .189 – .197 (4.801 – 5.004) NOTE 3 .045 ±.005 (1.143 ±0.127) 8 .089 .160 ±.005 (2.26) (4.06 ±0.127) REF .245 (6.22) MIN .150 – .157 .080 – .099 (2.032 – 2.530) (3.810 – 3.988) NOTE 3 .228 – .244 (5.791 – 6.197) 1 .030 ±.005 (0.76 ±0.127) TYP .005 (0.13) MAX 7 5 6 .118 (2.99) REF 3 2 .118 – .139 (2.997 – 3.550) 4 RECOMMENDED SOLDER PAD LAYOUT .010 – .020 × 45° (0.254 – 0.508) .008 – .010 (0.203 – 0.254) .053 – .069 (1.346 – 1.752) 0°– 8° TYP .016 – .050 (0.406 – 1.270) .014 – .019 (0.355 – 0.483) TYP NOTE: 1. DIMENSIONS IN INCHES (MILLIMETERS) 2. DRAWING NOT TO SCALE 3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm) 4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542) 5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542) 4 5 .004 – .010 0.0 – 0.005 (0.101 – 0.254) (0.0 – 0.130) .050 (1.270) BSC S8E 1015 REV C 8302fb 24 For more information www.linear.com/LT8302 LT8302 Revision History REV DATE DESCRIPTION A 11/14 Modified IQ and IHYS Conditions 3 B 11/15 PAGE NUMBER Modified LPRI Equation 14 Modified Schematic 23 Updated Related Parts 26 Revised Package Drawing 24 8302fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. For more information www.linear.com/LT8302 25 LT8302 Typical Application 4V to 42VIN/48VOUT Boost Converter L1 22µH VIN 4V TO 42V VIN SW RFB EN/UVLO C1 10µF VOUT 48V/1.4A (VIN = 42V) 48V/0.8A (VIN = 24V) 48V/0.4A (VIN = 12V) 48V/0.15A (VIN = 5V) D1 LT8302 C3 10µF R4 Z1 464k RREF INTVCC C2 1µF R3 1M D1: DIODES PDS560 L1: WÜRTH 7443551221 Z1: CENTRAL CMHZ5262B R5 10k GND 8302 TA10a Efficiency vs Load Current 100 EFFICIENCY (%) 95 90 85 80 VIN = 5V VIN = 12V VIN = 24V VIN = 42V 75 70 0 250 500 750 1000 LOAD CURRENT (mA) 1250 1500 8302 TA10b Related Parts PART NUMBER DESCRIPTION COMMENTS LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback 5-Lead TSOT-23 LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573/LT3574 LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12) LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3757A/LT3759 LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch 8302fb 26 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT8302 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8302 LT 1115 REV B • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2013