PHILIPS SA5212AD

INTEGRATED CIRCUITS
SA5212A
Transimpedance amplifier (140MHz)
Product specification
Replaces datasheet NE/SA/SE5212A of 1995 Apr 26
IC19 Data Handbook
1998 Oct 07
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
DESCRIPTION
PIN CONFIGURATION
The SA5212A is a 14kΩ transimpedance, wideband, low noise
differential output amplifier, particularly suitable for signal recovery in
fiber optic receivers and in any other applications where very low
signal levels obtained from high-impedance sources need to be
amplified.
N, FE, D Packages
FEATURES
• Extremely low noise: 2.5pA/√Hz
• Single 5V supply
• Large bandwidth: 140MHz
• Differential outputs
• Low input/output impedances
• 14kΩ differential transresistance
• ESD hardened
IIN
1
8
GND2
VCC
2
7
OUT (–)
GND1
3
6
GND2
GND1
4
5
OUT (+)
SD00336
Figure 1. Pin Configuration
• Wideband gain block
• Medical and scientific instrumentation
• Sensor preamplifiers
• Single-ended to differential conversion
• Low noise RF amplifiers
• RF signal processing
APPLICATIONS
• Fiber-optic receivers, analog and digital
• Current-to-voltage converters
ORDERING INFORMATION
TEMPERATURE RANGE
ORDER CODE
DWG #
8-Pin Plastic Small Outline (SO) Package
DESCRIPTION
-40°C to +85°C
SA5212AD
SOT96-1
8-Pin Plastic Dual In-Line Package (DIP)
-40°C to +85°C
SA5212AN
SOT97-1
8-Pin Ceramic Dual In-Line Package (DIP)
-40°C to +85°C
SA5212AFE
0580A
ABSOLUTE MAXIMUM RATINGS
SYMBOL
VCC
PARAMETER
SA5212A
UNIT
6
V
8-Pin Plastic DIP
1100
mW
8-Pin Plastic SO
750
mW
8-Pin Cerdip
750
mw
5
mA
Power Supply
Power dissipation, TA=25°C (still air)1
PD MAX
IIN MAX
Maximum input current2
TA
Operating ambient temperature range
-40 to 85
°C
TJ
Operating junction
-55 to 150
°C
Storage temperature range
-65 to 150
°C
TSTG
NOTES:
1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance:
8-Pin Plastic DIP: 110°C/W
8-Pin Plastic SO: 160°C/W
8-Pin Cerdip: 165°C/W
2. The use of a pull-up resistor to VCC, for the PIN diode, is recommended
1998 Oct 07
2
853-1266 20142
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
RECOMMENDED OPERATING CONDITIONS
SYMBOL
VCC
PARAMETER
RATING
UNIT
Supply voltage range
4.5 to 5.5
V
TA
Ambient temperature ranges
-40 to +85
°C
TJ
Junction temperature ranges
-40 to +105
°C
DC ELECTRICAL CHARACTERISTICS
Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V
and TA=25°C1.
SYMBOL
Min
Typ
Max
UNIT
VIN
Input bias voltage
0.55
0.8
1.05
V
VO±
Output bias voltage
2.5
3.3
3.8
V
VOS
Output offset voltage
120
mV
ICC
Supply current
20
26
33
mA
Output sink/source current
3
4
mA
IOMAX
IIN
IN MAX
PARAMETER
TEST CONDITIONS
Maximum input current (2% linearity)
Test Circuit 6, Procedure 2
±40
±80
µA
Maximum input current overload threshold
Test Circuit 6, Procedure 4
±60
±120
µA
NOTES:
1. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.
1998 Oct 07
3
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
AC ELECTRICAL CHARACTERISTICS
Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V
and TA=25°C5.
PARAMETER
SYMBOL
TEST CONDITIONS
Min
Typ
Max
UNIT
DC tested, RL = ∞
Test Circuit 6, Procedure 1
9.0
14
19
kΩ
RT
Transresistance (differential output)
RO
Output resistance (differential output)
DC tested
14
30
46
Ω
RT
Transresistance (single-ended output)
DC tested, RL = ∞
4.5
7
9.5
kΩ
RO
Output resistance (single-ended output)
DC tested
7
15
23
Ω
100
140
100
120
70
110
150
Ω
10
18
pF
Test Circuit 1
D package,
f3dB
Bandwidth (-3dB)
TA = 25°C
MHz
N, FE packages,
TA = 25°C
RIN
Input resistance
CIN
Input capacitance
∆R/∆V
Transresistance power supply sensitivity
VCC = 5 ±0.5V
9.6
%/V
∆R/∆T
Transresistance ambient
temperature sensitivity
D package
∆TA = TA MAX-TA MIN
0.05
%/°C
IN
RMS noise current spectral density
(referred to input)
Test Circuit 2
f = 10MHz TA = 25°C
2.5
pA/√Hz
TA = 25°C Test Circuit 2
∆f = 50MHz
20
∆f = 100MHz
27
∆f = 200MHz
40
∆f = 50MHz
22
∆f = 100MHz
32
∆f = 200MHz
52
Integrated
g
RMS noise current over the bandwidth (referred to input) CS = 01
IT
CS = 1pF
PSRR
Power supply rejection ratio2
Any package
DC tested
∆VCC = 0.1V
Equivalent AC
Test Circuit 3
PSRR
Power supply rejection ratio2
(ECL configuration)
Any package
f = 0.1MHz1
Test Circuit 4
VO MAX
Maximum differential output voltage swing
RL = ∞
Test Circuit 6, Procedure 3
VIN MAX
Maximum input amplitude for output duty
cycle of 50 ±5%3
tR
Rise time for 50mV output signal4
20
nA
33
dB
23
dB
3.2
VP-P
Test Circuit 5
325
mVP-P
Test Circuit 5
2.0
ns
1.7
NOTES:
1. Package parasitic capacitance amounts to about 0.2pF.
2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in VCC line.
3. Guaranteed by linearity and over load tests.
4. tR defined as 20-80% rise time. It is guaranteed by -3dB bandwidth test.
5. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.
1998 Oct 07
4
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS
SINGLE-ENDED
Rt V OUT
V IN
DIFFERENTIAL
2 S21 R
Rt V OUT
4 S21 R
V IN
1 S22
RO ZO
33
1 S22
1 S22
R O 2Z O
66
1 S22
SPECTRUM ANALYZER
NETWORK ANALYZER
VCC
AV = 60DB
OUT
33
1µF
S-PARAMETER TEST SET
PORT 1
NC
PORT 2
IN
DUT
33
1µF
OUT
RL = 50
VCC
GND1
ZO = 50Ω
0.1µF
OUT
GND2
1µF
33
R = 1k
IN
DUT
1µF
33
OUT
RL = 50Ω
50
GND1
GND2
Test Circuit 1
Test Circuit 2
SD00337
Figure 2. Test Circuits 1 and 2
NETWORK ANALYZER
5V + ∆V
10µF
S-PARAMETER TEST SET
10µF
PORT 1
10µF
PORT 2
CURRENT PROBE
1mV/mA
0.1µF
16
CAL
VCC
33
1µF
OUT
NC
50
100
BAL.
IN DUT
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
1µF
GND1
GND2
Test Circuit 3
Figure 3. Test Circuit 3
1998 Oct 07
5
SD00338
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS (Continued)
NETWORK ANALYZER
–5.2V + ∆V
10µF
S-PARAMETER TEST SET
0.1µF
PORT 1
10µF
PORT 2
CURRENT PROBE
1mV/mA
0.1µF
16
GND1
CAL
GND2
1µF
33
OUT
NC
50
100
BAL.
IN
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
1µF
VCC
Test Circuit 4
SD00339
Figure 4. Test Circuit 4
PULSE GEN.
5V
33
1µF
OUT
0.1µF 1k IN
A
DUT
33
OUT
ZO = 50Ω
OSCILLOSCOPE
B
1µF
ZO = 50Ω
50
GND1
GND2
Test Circuit 5
Figure 5. Test Circuit 5
1998 Oct 07
6
Measurement done using
differential wave forms
SD00545
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS (Continued)
Typical Differential Output Voltage
vs Current Input
5V
+
OUT +
IN
VOUT (V)
DUT
–
OUT –
IIN (µA)
GND1
GND2
2.00
DIFFERENTIAL OUTPUT VOLTAGE (V)
1.60
1.20
0.80
0.40
0.00
–0.40
–0.80
–1.20
–1.60
–2.00
–200
–160
–120
–80
–40
0
40
80
120
160
200
CURRENT INPUT (µA)
NE5212A TEST CONDITIONS
Procedure 1
RT measured at 30µA
RT = (VO1 – VO2)/(+30µA – (–30µA))
Where: VO1 Measured at IIN = +30µA
VO2 Measured at IIN = –30µA
Procedure 2
Linearity = 1 – ABS((VOA – VOB) / (VO3 – VO4))
Where: VO3 Measured at IIN = +60µA
VO4 Measured at IIN = –60µA
R T ( 60A) V
OA
OB
V
R T ( 60A) V
OB
OB
V
Procedure 3
VOMAX = VO7 – VO8
Where: VO7 Measured at IIN = +130µA
VO8 Measured at IIN = –130µA
Procedure 4
IIN Test Pass Conditions:
VO7 – VO5 > 20mV and V06 – VO5 > 20mV
Where: VO5 Measured at IIN = +800µA
VO6 Measured at IIN = –80µA
VO7 Measured at IIN = +130µA
VO8 Measured at IIN = –130µA
Test Circuit 8
Figure 6. Test Circuit 8
1998 Oct 07
7
SD00340
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS
NE5212A Supply Current
vs Temperature
NE5212A Input Bias Voltage
vs Temperature
3.50
950
29
28
27
26
VCC = 5.0V
900
OUTPUT BIAS VOLTAGE (V)
VCC = 5.0V
INPUT BIAS VOLTAGE (mV)
30
SUPPLY CURRENT (mA)
NE5212A Output Bias Voltage
vs Temperature
850
800
750
700
VCC = 5.0V
3.45
PIN 5
3.40
PIN 7
3.35
3.30
650
25
–60 –40 –20
0
600
–60 –40 –20
20 40 60 80 100 120 140
AMBIENT TEMPERATURE (°C)
3.2
3.0
2.8
2.6
2.4
–60 –40 –20
0
40
20
0
–20
–40
AMBIENT TEMPERATURE (°C)
0
20 40 60 80 100 120 140
17.0
16.5
16.0
14.5
14.0
–60 –40 –20
36
35
34
33
–60 –40 –20 0 20 40 60 80 100 120 140
AMBIENT TEMPERATURE (°C)
50 PIN 5
SINGLE-ENDED
40 RL = 50Ω
VCC = 5.0V
16
DC TESTED
15
14
13
PIN 7
12
PIN 5
20
10
10
0
112.5
0
20 40 60 80 100 120 140
N, F PKG
VCC = 5.0V
TA = 25°C
30
11
9
–60 –40 –20
20 40 60 80 100 120 140
NE5212A Typical
Bandwidth Distribution
(75 Parts from 3 Wafer Lots)
POPULATION (%)
37
OUTPUT RESISTANCE ( Ω )
∆VCC = ±0.1V
DC TESTED
OUTPUT REFERRED
0
AMBIENT TEMPERATURE (°C)
17
VCC = 5.0V
RL = ∞
15.0
NE5212A Output Resistance
vs Temperature
40
VCC = 5.0V
DC TESTED
15.5
AMBIENT TEMPERATURE (°C)
NE5212A Power Supply Rejection Ratio
vs Temperature
POWER SUPPLY REJECTION RATIO (dB)
VCC = 5.0V
VOS = VOUT5 – VOUT7
–60
–60 –40 –20
20 40 60 80 100 120 140
DIFFERENTIAL TRANSRESISTANCE (kΩ )
3.4
RL = ∞
OUTPUT OFFSET VOLTAGE (mV)
DIFFERENTIAL OUTPUT SWING (V)
3.6
60
20 40 60 80 100 120 140
NE5212A Differential Transresistance
vs Temperature
80
VCC = 5.0V
DC TESTED
0
AMBIENT TEMPERATURE (°C)
NE5212A Output Offset Voltage
vs Temperature
3.8
38
3.25
–60 –40 –20
20 40 60 80 100 120 140
AMBIENT TEMPERATURE (°C)
NE5212A Differential Output Swing
vs Temperature
39
0
122.5 132.5 142.5 152.5
FREQUENCY (MHz)
162.5
AMBIENT TEMPERATURE (°C)
SD00341
Figure 7. Typical Performance Characteristics
1998 Oct 07
8
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
Gain vs Frequency
11
10
9
8
7
6
5
4
3
N PKG
PIN 5
TA = 25°C
100
4.5V
N PKG
PIN 7
TA = 25°C
0.1
1
10
FREQUENCY (MHz)
Output Resistance
vs Frequency
11
6
GAIN (dB)
N PKG
PIN 5
VCC = 5V
TA = 25°C
–135
5
–225
4
3
0.1
1
10
FREQUENCY (MHz)
100
PHASE (o)
7
–45
GAIN (dB)
GAIN (dB)
8
11
7
125°C
N PKG
PIN 7
VCC = 5V
5
11
10
9
8
7
6
5
4
3
0.1
125°C
25°C
9
–55°C
6
100
85°C
10
–55°C
8
7
–55°C
6
5
N PKG
PIN 5
VCC = 5V
4
125°C
3
1
10
FREQUENCY (MHz)
0.1
100
Gain and Phase Shift
vs Frequency
φ
11
PIN 7
1
10
FREQUENCY (MHz)
Gain vs Frequency
8
0.1
100
Gain and Phase Shift
vs Frequency
10
9
PIN 5
0.1
–55°C
125°C
10
9
4
3
1
10
FREQUENCY (MHz)
N PKG
VCC = 5V
TA = 25°C
Gain vs Frequency
D PKG
TA = 25°C
VCC = 5V
0.1
80
70
60
50
40
30
20
10
GAIN (dB)
OUTPUT RESISTANCE (Ω )
100
90
80
70
60
50
40
30
20
10
100
–180
φ
D PKG
PIN 7
VCC = 5V
TA = 25°C
–270
–360
1
10
FREQUENCY (MHz)
100
1
10
FREQUENCY (MHz)
100
Gain and Phase Shift
vs Frequency
GAIN (dB)
1
10
FREQUENCY (MHz)
5.0V
11
10
9
8
7
6
5
4
3
0.1
–180
φ
N PKG
PIN 7
VCC = 5V
TA = 25°C
1
10
FREQUENCY (MHz)
–270
–360
100
SD00342
Figure 8. Typical Performance Characteristics (cont.)
1998 Oct 07
9
PHASE (o)
GAIN (dB)
GAIN (dB)
4.5V
0.1
5.5V
OUTPUT RESISTANCE (Ω )
5.5V
5.0V
Output Resistance
vs Frequency
PHASE (o)
12
11
10
9
8
7
6
5
4
3
Gain vs Frequency
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
DIFFERENTIAL OUTPUT VOLTAGE (V)
11
10
0
8
7
–90
D PKG
PIN 5
VCC = 5V
TA = 25°C
6
5
PHASE (o )
GAIN (dB)
9
–180
4
3
0.1
1
10
FREQUENCY (MHz)
4.5
Differential Output Voltage
vs Input Current
DIFFERENTIAL OUTPUT VOLTAGE (V)
Output Voltage
vs Input Current
Gain and Phase Shift
vs Frequency
125°C
85°C
25°C
–55°C
125°C
85°C
–55°C25°C
2.0
–150.0
100
0
INPUT CURRENT (µA)
2.0
5.5V
5.0V
4.5V
0
5.5V
5.0V
–2.0
–150.0
150.0
4.5V
0
INPUT CURRENT (µA)
150.0
Group Delay
vs Frequency
Differential Output Voltage
vs Input Current
10
2.000
25°C
8
85°C
6
125°C
4
DELAY (ns)
OUTPUT VOLTAGE (V)
–55°C
0
2
0
–55°C
25°C
85°C
125°C
–2.000
–150.0
INPUT CURRENT (µA)
0.1
150.0
20
40
60
80 100 120 140
FREQUENCY (MHz)
160
Output Step Response
VCC = 5V
TA = 25°C
20mV/Div
0
2
4
6
8
10
(ns)
12
14
16
18
20
SD00343
Figure 9. Typical Performance Characteristics (cont.)
1998 Oct 07
10
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
THEORY OF OPERATION
R IN Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The SA5212A is a wide
bandwidth (typically 140MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The SA5212A is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 10. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
source to the emitter of Q3 is approximately the value of the
feedback resistor, RF=7kΩ. The gain from the second stage (A2)
and emitter followers (A3 and A4) is about two. Therefore, the
differential transresistance of the entire amplifier, RT is
RT V IN
RF
7.2K 103W
70
I IN
1 A VOL
More exact calculations would yield a higher value of 110Ω.
Thus CIN and RIN will form the dominant pole of the entire amplifier;
f 3dB 1
2p R IN C IN
Assuming typical values for RF = 7.2kΩ, RIN = 110Ω, CIN = 10pF
f 3dB 1
145MHz
2p (110) 10 10 12
The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascade input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
a source capacitance of 1pF, input stage voltage gain of 70, RIN =
60Ω then the total input capacitance, CIN = (1+7.5) pF which will
lead to only a 12% bandwidth reduction.
V OUT(diff)
2R F 2(7.2K) 14.4kW
I IN
OUTPUT +
The single-ended transresistance of the amplifier is typically 7.2kΩ.
A3
The simplified schematic in Figure 11 shows how an input current is
converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
the feedback resistor RF. The transistor Q1 provides most of the
open loop gain of the circuit, AVOL≈70. The emitter follower Q2
minimizes loading on Q1. The transistor Q4, resistor R7, and VB1
provide level shifting and interface with the Q15 – Q16 differential
pair of the second stage which is biased with an internal reference,
VB2. The differential outputs are derived from emitter followers Q11 –
Q12 which are biased by constant current sources. The collectors of
Q11 – Q12 are bonded to an external pin, VCC2, in order to reduce
the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.
INPUT
A1
RF
A4
OUTPUT –
SD00327
Figure 10. SA5212A – Block Diagram
NOISE
Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of
3.5pA/√Hz. The transresistance configuration assures that the
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
equivalent input RMS noise current is strongly determined by the
quiescent current of Q1, the feedback resistor RF, and the
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 52nA RMS
in a 200MHz bandwidth.
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 12, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
capacitance, CIN, in parallel with the source, IS, is approximately
7.5pF, assuming that CS=0 where CS is the external source
capacitance.
Since the input is driven by a current source the input must have a
low input resistance. The input resistance, RIN, is the ratio of the
incremental input voltage, VIN, to the corresponding input current, IIN
and can be calculated as:
1998 Oct 07
A2
11
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
VCC1
VCC2
R3
R1
Q2
INPUT
R13
Q4
Q11
+
Q3
Q1
R12
Q12
Q15
R2
Q16
R14
GND1
R7
PHOTODIODE
OUT–
R15
+
OUT+
VB2
R5
R4
GND2
SD00328
Figure 11. Transimpedance Amplifier
No. of incident photons/sec= where P=optical incident power
VCC
IC1
R1
INPUT
Q2
IB
IIN
P
No. of incident photons/sec = hc
l
R3
where P = optical incident power
Q3
Q1
P
No. of generated electrons/sec = h @ hc
l
R2
IF
VIN
VEQ3
where η = quantum efficiency
RF
no. of generated electron hole paris
no. of incident photons
P
NI + h @ hc @ e Amps (Coulombsńsec.)
l
where e = electron charge = 1.6 × 10-19 Coulombs
h@e
Responsivity R = hc Amp/watt
l
+
R4
SD00329
Figure 12. Shunt-Series Input Stage
I + P@R
DYNAMIC RANGE
The electrical dynamic range can be defined as the ratio of
maximum input current to the peak noise current:
Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the
noise parameter Z may be calculated as:1
Electrical dynamic range, DE, in a 200MHz bandwidth assuming
IINMAX = 120µA and a wideband noise of IEQ=52nARMS for an
external source capacitance of CS = 1pF.
Z+
DE
where Z is the ratio of RMS noise output to the peak response to a
single hole-electron pair. Assuming 100% photodetector quantum
efficiency, half mark/half space digital transmission, 850nm
lightwave and using Gaussian approximation, the minimum required
optical power to achieve 10-9 BER is:
(Max. input current)
+
(Peak noise current)
D E(dB) + 20 log
(120 @ 10 *6)
(Ǹ2 52nA)
D E(dB) + 20 log
(120mA)
+ 64dB
(73nA)
P avMIN + 12 hc B Z + 12 (2.3 @ 10 *19)
l
200 @ 10 6 1625 + 897nW + * 30.5dBm,
In order to calculate the optical dynamic range the incident optical
power must be considered.
where h is Planck’s Constant, c is the speed of light, λ is the
wavelength. The minimum input current to the SA5212A, at this
input power is:
For a given wavelength λ;
I avMIN + qP avMIN l
hc
Energy of one Photon = hc watt sec (Joule)
l
Where h=Planck’s Constant = 6.6 × 10-34 Joule sec.
*9
@ 10 *19
+ 897 @ 10 @ 1.6
2.3 @ 10 *19
= 624nA
c = speed of light = 3 × 108 m/sec
c / λ = optical frequency
1998 Oct 07
ǒ Ǔ
I EQ
Amp
52 @ 10 *9
+
+ 1625
Amp
qB
(1.6 @ 10 *19)(200 @ 10 6)
12
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
As with any high-frequency device, some precautions must be
observed in order to enjoy reliable performance. The first of these is
the use of a well-regulated power supply. The supply must be
capable of providing varying amounts of current without significantly
changing the voltage level. Proper supply bypassing requires that a
good quality 0.1µF high-frequency capacitor be inserted between
VCC1 and VCC2, preferably a chip capacitor, as close to the package
pins as possible. Also, the parallel combination of 0.1µF capacitors
with 10µF tantalum capacitors from each supply, VCC1 and VCC2, to
the ground plane should provide adequate decoupling. Some
applications may require an RF choke in series with the power
supply line. Separate analog and digital ground leads must be
maintained and printed circuit board ground plane should be
employed whenever possible.
Choosing the maximum peak overload current of IavMAX=120µA, the
maximum mean optical power is:
VIN
OUT–
R = 560
IN
NE5212A OUT+
a. Non-inverting 20dB Amplifier
VIN
OUT+
R = 560
IN
NE5212A OUT–
BASIC CONFIGURATION
b. Inverting 20dB Amplifier
VIN
OUT+
R = 560
IN
A trans resistance amplifier is a current-to-voltage converter. The
forward transfer function then is defined as voltage out divided by
current in, and is stated in ohms. The lower the source resistance,
the higher the gain. The SA5212A has a differential transresistance
of 14kΩ typically and a single-ended transresistance of 7kΩ
typically. The device has two outputs: inverting and non-inverting.
The output
voltage in the differential output mode is twice that of the output
voltage in the single-ended mode. Although the device can be used
without coupling capacitors, more care is required to avoid upsetting
the internal bias nodes of the device. Figure 13 shows some basic
configurations.
NE5212A OUT–
c. Differential 20dB Amplifier
SD00344
Figure 13. Variable Gain Circuit
P avMAX hcI avMAX
2.3 10 19(120 10 6)
lq
1.6 10 19
VARIABLE GAIN
= 172µW or –7.6dBm
Figure 14 shows a variable gain circuit using the SA5212A and the
SA5230 low voltage op amp. This op amp is configured in a
non-inverting gain of five. The output drives the gate of the SD210
DMOS FET. The series resistance of the FET changes with this
output voltage which in turn changes the gain of the SA5212A. This
circuit has a distortion of less than 1% and a 25dB range, from
-42.2dBm to -15.9dBm at 50MHz, and a 45dB range, from -60dBm
to -14.9dBm at 10MHz with 0 to 1V of control voltage at VCC.
Thus the optical dynamic range, DO is:
DO = PavMAX - PavMIN = -30.5 -(-7.6) = 22.8dB.
This represents the maximum limit attainable with the SA5212A
operating at 200MHz bandwidth, with a half mark/half space digital
transmission at 820nm wavelength.
APPLICATION INFORMATION
Package parasitics, particularly ground lead inductances and
parasitic capacitances, can significantly degrade the frequency
response. Since the SA5212A has differential outputs which can
feed back signals to the input by parasitic package or board layout
capacitances, both peaking and attenuating type frequency
response shaping is possible. Constructing the board layout so that
Ground 1 and Ground 2 have very low impedance paths has
produced the best results. This was accomplished by adding a
ground-plane stripe underneath the device connecting Ground 1,
Pins 8–11, and Ground 2, Pins 1 and 2 on opposite ends of the
SO14 package. This ground-plane stripe also provides isolation
between the output return currents flowing to either VCC2 or Ground
2 and the input photodiode currents to flowing to Ground 1. Without
this ground-plane stripe and with large lead inductances on the
board, the part may be unstable and oscillate near 800MHz. The
easiest way to realize that the part is not functioning normally is to
measure the DC voltages at the outputs. If they are not close to their
quiescent values of 3.3V (for a 5V supply), then the circuit may be
oscillating. Input pin layout necessitates that the photodiode be
physically very close to the input and Ground 1. Connecting Pins 3
and 5 to Ground 1 will tend to shield the input but it will also tend to
increase the capacitance on the input and slightly reduce the
bandwidth.
1998 Oct 07
OUT+
0.1µF
RFIN
SD210
IN
51
NE5212A
RFOUT
OUT–
+5V
VCC
0–5V
0–1V
10k
2.4k
SD00345
Figure 14. Variable Gain Circuit
13
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
16MHZ CRYSTAL OSCILLATOR
DIGITAL FIBER OPTIC RECEIVER
Figure 15 shows a 16MHz crystal oscillator operating in the series
resonant mode using the SA5212A. The non-inverting input is fed
back to the input of the SA5212A in series with a 2pF capacitor. The
output is taken from the inverting output.
Figures 16 and 17 show a fiber optic receiver using off-the-shelf
components.
The receiver shown in Figure 16 uses the SA5212A, the Philips
Semiconductors 10116 ECL line receiver, and Philips/Amperex
BPF31 PIN diode. The circuit is a capacitor-coupled receiver and
utilizes positive feedback in the last stage to provide the hysteresis.
The amount of hysteresis can be tailored to the individual application
by changing the values of the feedback resistors to maintain the
desired balance between noise immunity and sensitivity. At room
temperature, the circuit operates at 50Mbaud with a BER of 10E-10
and over the automotive temperature range at 40Mbaud with a BER
of 10E-9. Higher speed experimental diodes have been used to
operate this circuit at 220Mbaud with a BER of 10E-10.
+5V
OUT+
NE5212A
OUT–
IN
Figure 17 depicts a TTL receiver using the SA5212A and the
SA5214 fast amplifier system along with the Philips/Amperex PIN
diode. The system shown is optimized for 50 Mb/s Non Return to
Zero (NRZ) data. A link status indication is provided along with a
jamming function when the input level is below a
user-programmable threshold level.
SD00346
Figure 15. 16MHz Crystal Oscillator
VEE
VBB1
VBB1
1.0µF
VCC
+5.0
4.7
0.01µF
510
510
NE5212A
9
7
5
10
6 8
3 4
BPF31
1k
0.1µF
2
1
510
1k
1k
100pF
1
16 7
1/3
10116
2
1/3
4 10116
100pF
15
1/3
12 10116
3
11 6
0.1µF
13
5
8
ECL
ECL
14
0.01µF
1k
1k
0.01µF
510
–15V
510
1k
510
0.1µF
0.01µF
VBB1
VEE
VBB1
2.7µH
–5.2V
4.7µF
0.1µF
4.7µF
VEE
NOTE:
1. Tie all VBB points together.
SD00347
Figure 16. ECL Fiber Optic Receiver
1998 Oct 07
14
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
+VCC
GND
47µF
C1
C2
2
CPKDET
3
THRESH
4
GNDA
5
FLAG
100pF
R3
47k
L2
10µH
6
C11
.01µF
C13
.01µF
7
VCCD
8
VCCA
9
GNDD
10
C12
10µF
JAM
TTLOUT
IN1B
20
IN1A
19
CAZP 18
CAZN
C7
100pF
C8
0.1µF
17
5
OUT+
6
GND2
7
OUT–
8
NE5212A
LED
GND2
IN8B
15
OUT2A
14
IN8A
13
RHYST
12
GND1
4
GND1
3
VCC
2
IIN
C5
1.0µF
.01µF
C4
.01µF
C6
1
BPF31
OPTICAL
INPUT
OUT2B 16
NE5214
1
C9
L3
10µH
R1
100
C3
10µF
D1
LED
R2
220
C10
10µF
L1
10µH
.01µF
RPKDET 11
R4
5.1k
VOUT (TTL)
SD00348
Figure 17. A 50Mb/s TTL Digital Fiber Optic Receiver
1998 Oct 07
15
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
GND2
1
8
2
7
GND1
3
6
GND2
GND1
4
5
OUT+
IIN
VCC
OUT–
ECN No.: 99918
1990 Jul 5
SD00489
Figure 18. SA5212A Bonding Diagram
carriers, it is impossible to guarantee 100% functionality through this
process. There is no post waffle pack testing performed on
individual die.
Die Sales Disclaimer
Due to the limitations in testing high frequency and other parameters
at the die level, and the fact that die electrical characteristics may
shift after packaging, die electrical parameters are not specified and
die are not guaranteed to meet electrical characteristics (including
temperature range) as noted in this data sheet which is intended
only to specify electrical characteristics for a packaged device.
Since Philips Semiconductors has no control of third party
procedures in the handling or packaging of die, Philips
Semiconductors assumes no liability for device functionality or
performance of the die or systems on any die sales.
All die are 100% functional with various parametrics tested at the
wafer level, at room temperature only (25°C), and are guaranteed to
be 100% functional as a result of electrical testing to the point of
wafer sawing only. Although the most modern processes are
utilized for wafer sawing and die pick and place into waffle pack
1998 Oct 07
Although Philips Semiconductors typically realizes a yield of 85%
after assembling die into their respective packages, with care
customers should achieve a similar yield. However, for the reasons
stated above, Philips Semiconductors cannot guarantee this or any
other yield on any die sales.
16
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
DIP8: plastic dual in-line package; 8 leads (300 mil)
1998 Oct 07
SOT97-1
17
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
SO8: plastic small outline package; 8 leads; body width 3.9mm
1998 Oct 07
18
SOT96-1
2. Dimension and tolerancing per ANSI Y14. 5M-1982.
3. “T”, “D”, and “E” are reference datums on the body
and include allowance for glass overrun and meniscus
on the seal line, and lid to base mismatch.
4. These dimensions measured with the leads
constrained to be perpendicular to plane T.
0.303 (7.70)
0.245 (6.22)
–E–
PIN # 1
0.100 (2.54) BSC
–D–
5. Pin numbers start with Pin #1 and continue
counterclockwise to Pin #8 when viewed
from the top.
0.408 (10.36)
0.376 (9.55)
0.320 (8.13)
0.290 (7.37)
(NOTE 4)
19
0.070 (1.78)
0.050 (1.27)
0.200 (5.08)
0.165 (4.19)
–T–
SEATING
PLANE
0.165 (4.19)
0.125 (3.18)
0.023 (0.58)
0.015 (0.38)
T
E D
0.175 (4.45)
0.145 (3.68)
0.035 (0.89)
0.020 (0.51)
Philips Semiconductors
NOTES:
1. Controlling dimension: Inches. Millimeters are
shown in parentheses.
8-PIN (300 mils wide) CERAMIC DUAL IN-LINE (F) PACKAGE
0.055 (1.40)
0.030 (0.76)
Transimpedance amplifier (140MHz)
0580A
853–0580A 006688
1998 Oct 07
0.055 (1.40)
0.030 (0.76)
BSC
0.300 (7.62)
(NOTE 4)
0.010 (0.254)
0.015 (0.38)
0.010 (0.25)
0.395 (10.03)
0.300 (7.62)
Product specification
SA5212A
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
Data sheet status
Data sheet
status
Product
status
Definition [1]
Objective
specification
Development
This data sheet contains the design target or goal specifications for product development.
Specification may change in any manner without notice.
Preliminary
specification
Qualification
This data sheet contains preliminary data, and supplementary data will be published at a later date.
Philips Semiconductors reserves the right to make chages at any time without notice in order to
improve design and supply the best possible product.
Product
specification
Production
This data sheet contains final specifications. Philips Semiconductors reserves the right to make
changes at any time without notice in order to improve design and supply the best possible product.
[1] Please consult the most recently issued datasheet before initiating or completing a design.
Definitions
Short-form specification — The data in a short-form specification is extracted from a full data sheet with the same type number and title. For
detailed information see the relevant data sheet or data handbook.
Limiting values definition — Limiting values given are in accordance with the Absolute Maximum Rating System (IEC 134). Stress above one
or more of the limiting values may cause permanent damage to the device. These are stress ratings only and operation of the device at these or
at any other conditions above those given in the Characteristics sections of the specification is not implied. Exposure to limiting values for extended
periods may affect device reliability.
Application information — Applications that are described herein for any of these products are for illustrative purposes only. Philips
Semiconductors make no representation or warranty that such applications will be suitable for the specified use without further testing or
modification.
Disclaimers
Life support — These products are not designed for use in life support appliances, devices or systems where malfunction of these products can
reasonably be expected to result in personal injury. Philips Semiconductors customers using or selling these products for use in such applications
do so at their own risk and agree to fully indemnify Philips Semiconductors for any damages resulting from such application.
Right to make changes — Philips Semiconductors reserves the right to make changes, without notice, in the products, including circuits, standard
cells, and/or software, described or contained herein in order to improve design and/or performance. Philips Semiconductors assumes no
responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright, or mask work right to these
products, and makes no representations or warranties that these products are free from patent, copyright, or mask work right infringement, unless
otherwise specified.
 Copyright Philips Electronics North America Corporation 1998
All rights reserved. Printed in U.S.A.
Philips Semiconductors
811 East Arques Avenue
P.O. Box 3409
Sunnyvale, California 94088–3409
Telephone 800-234-7381
Date of release: 10-98
Document order number:
1998 Oct 07
20
9397 750 04625