INTERSIL ISL6741

Flexible Double Ended Voltage and Current Mode PWM
Controllers
ISL6740, ISL6741
Features
The ISL6740, ISL6741 family of adjustable frequency, low
power, pulse width modulating (PWM) voltage mode (ISL6740)
and current mode (ISL6741) controllers is designed for a wide
range of power conversion applications using half-bridge, full
bridge, and push-pull configurations. These controllers provide
an extremely flexible oscillator that allows precise control of
frequency, duty cycle, and deadtime.
• Precision Duty Cycle and Deadtime Control
This advanced BiCMOS design features low operating current,
adjustable switching frequency up to 1MHz, adjustable softstart, internal and external over-temperature protection, fault
annunciation, and a bidirectional SYNC signal that allows the
oscillator to be locked to paralleled units or to an external
clock for noise sensitive applications.
• Bidirectional Synchronization
Ordering Information
• Adjustable Soft-start
PART
NUMBER
(Notes 2, 3)
PART
MARKING
TEMP. RANGE
(°C)
PACKAGE
PKG.
DWG. #
-40 to +105
16 Ld SOIC
(Pb-free)
M16.15
ISL6740IVZ
(Note 1)
ISL67 40IVZ
-40 to +105
16 Ld TSSOP
(Pb-free)
M16.173
ISL6741IB
ISL6741IB
-40 to +105
16 Ld SOIC
M16.15
ISL6741IBZ 6741IBZ
(Note 1)
-40 to +105
16 Ld SOIC
(Pb-free)
M16.15
ISL6741IVZ
(Note 1)
-40 to +105
16 Ld TSSOP
(Pb-free)
M16.173
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details
on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pbfree material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is
RoHS compliant and compatible with both SnPb and Pb-free
soldering operations). Intersil Pb-free products are MSL classified
at Pb-free peak reflow temperatures that meet or exceed the Pbfree requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information
page for ISL6740, ISL6741. For more information on MSL please
see techbrief TB363.
December 2, 2011
FN9111.6
ISL674x
x=
CONTROL MODE
0
Voltage Mode
1
Current Mode
1
• Adjustable Delayed Overcurrent Shutdown and Re-start
(ISL6740)
• Adjustable Short Circuit Shutdown and Re-start
• Adjustable Oscillator Frequency Up to 2MHz
• Inhibit Signal
• Internal Over-Temperature Protection
• System Over-Temperature Protection Using a Thermistor or
Sensor
• Adjustable Input Undervoltage Lockout
ISL6740IBZ 6740IBZ
(Note 1)
ISL67 41IVZ
• 95µA Start-up Current
• Fault Signal
• Tight Tolerance Voltage Reference Over Line, Load, and
Temperature
• Pb-Free Available (RoHS Compliant)
Applications
• Telecom and Datacom Power
• Wireless Base Station Power
• File Server Power
• Industrial Power Systems
• DC Transformers and Buss Regulators
Pin Configuration
ISL6740, ISL6741
(16 LD SOIC, 16 LD TSSOP)
TOP VIEW
OUTA 1
16 OUTB
GND 2
15 VREF
SCSET 3
14 VDD
CT 4
13 RTD
SYNC 5
12 RTC
CS 6
11 OTS
VERROR 7
UV 8
10 FAULT
9 SS
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2003, 2004, 2007, 2011. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
Functional Block Diagram
ISL6740
VDD
SYNC
VREF
VREF
5.00 V
1%
2
+
-
FL
100
Q
OUTA
Q
PWM TOGGLE
OUTB
ENABLE
T
BG +-
4.5k
GND
OC S/D
SC S/D
UV
VREF
70µA
S Q
BI-DIRECTIONAL
SYNCHRONIZATION SYNC IN
R Q
SC LATCH
SS LOW
EXT. SYNC
ON
SS
IRTC
RTC
RTD
IRTD
SCSET
SHORT CIRCUIT
DETECTION
CT
SS HI
0.6V
0.4
+
--
+
-
Q
50µs
RETRIGGERABLE
ONE SHOT
FN9111.6
December 2, 2011
OTS
SS LOW
S Q
R Q
PWM LATCH
RESET
DOMINANT
PWM
COMPARATOR
0.27V
R Q
SC S/D
FAULT
VREF
OC S/D
SS
0.5
+
-
4.25V
FAULT LATCH
SET DOMINANT
S Q FL
INHIBIT
OC DETECT
VERROR
0.4
+
-
Q
SS DONE
CS
15µA
R Q
300k
4.5V
SS CLAMP
CLK
OC LATCH
S Q
SS DONE
+
OSCILLATOR
VREF UV 4.65V
VREF/2
+
+
BG +-
ISL6740, ISL6741
INTERNAL
OT SHUTDOWN
+130°C TO +150°C
INHIBIT/VIN UV
1.00V
+
INHIBIT
-
N_SYNC OUT
Functional Block Diagram (Continued)
ISL6741
SYNC
VREF
VDD
VREF
5.00 V
1%
3
+
-
FL
100
Q
OUTA
Q
PWM TOGGLE
OUTB
ENABLE
T
BG +-
4.5k
GND
SC S/D
UV
VREF
70µA
SC LATCH
S Q
ON
R Q
SS
RTC
RTD
IRTC
+
OSCILLATOR
IRTD
4.5V
SS CLAMP
CLK
SCSET
15µA
SS DONE
300k
SHORT CIRCUIT
DETECTION
CT
SS DONE
SS LOW
CS
0.6V
80mV
+-
+
--
+
-
S Q
VERROR
0.2
FAULT
SC S/D
VREF
SS
0.25
OTS
R Q
PWM LATCH
RESET
DOMINANT
PWM
COMPARATOR
0.27V
FAULT LATCH
SET DOMINANT
S Q
FL
R Q
INHIBIT
OC DETECT
+
-
VREF UV 4.65V
VREF/2
+
+
BG +-
ISL6740, ISL6741
1.00 V
N_SYNC OUT
INTERNAL
OT SHUTDOWN
BI-DIRECTIONAL
+130°C TO +150°C
SYNCHRONIZATION SYNC IN
INHIBIT/VIN UV
+
EXT. SYNC
INHIBIT
-
FN9111.6
December 2, 2011
Typical Application (ISL6740) - 48V Input DC Transformer, 12V @ 8A Output (ISL6740EVAL1)
SP1
VIN+
+12V
QR1
L1
C11
QH
C2
QR3
T1
L3
R8
C9
C13
R10
TP1
4
C1
QR2
C14
QR4
R11
C12
CR2
C3
CR1
C7
U1
HIP2101
VDD LO
HB VSS
HO LI
HS HI
C4
R14
R5
TP4
TP2
R6
VREF
C10
RT1
C5
OTS
OUTA
CS
CT
RTC
VREF
RTD
TP6
UV
VDD
R3
SCSET
R17
ISL6740
OUTB
R7
R19
VERROR
GND
VIN-
SYNC
FAULT
U3
SS
C18
TP5
Q5
R13
C15
C17
D1
FN9111.6
December 2, 2011
R18
C6
R12
C16
R15
ISL6740, ISL6741
R1
R9
QL
CR3
RTN
L2
T2
R2
C8
Typical Application (ISL6740) - 36V to 75V Input, Regulated 12V @ 8A Output (ISL6740EVAL2Z)
SP1
VIN+
CR5
L1
QH
C11
QR1
T1
L3
C2
+12V
R26
QR3
R8
+
C13
C8
RTN
L2
R10
TP1
C21
C9
CR4
5
C1
T2
CR6
R27
QL
R2
C14
CR3
QR2
R11
36V TO 75V
R9
QR4
R1
C3
CR1
C7
U1
HIP2101
VDD LO
HB VSS
HO LI
HS HI
C4
R14
R5
TP4
TP2
R6
VREF
C10
C5
GND
R4
R20
R19
CT
SCSET
RTC
UV
VREF
RTD
R3
TP6
CS
ISL6740
OUTA
VDD
R19
VERROR
OUTB
R17
SYNC
OTS
VIN-
FAULT
U3
SS
C18
TP5
R7
+ 12V
RT1
Q5
R23
C22
C20
R25
R13
C15
R21
C19
U2
C17
D1
R18
FN9111.6
December 2, 2011
C6
R12
C16
R15
U4
D2
R24
ISL6740, ISL6741
C12
CR2
Typical Application (ISL6741) - 48V to 5V Push-Pull DC/DC Converter
+ 5V
RTN
+48V
QR1
R18
R19
R20
T1
+
C9
CR1
EL7242
+5V
L1
6
C1
CR2
QR2
Q1
U5
RT1
Q2
T3
OUTB
R11
SYNC
VERROR
OUTA
CS
CT
+ 5V
SCSET
VREF
RTD
R4
OUTA
CR4
RTC
UV
VDD
CR3
U3
SS
GND
OUTB
ISL6741
VIN-
SYNC
R3
OTS
R1
FAULT
R2
R21
R14
R13
R15
C8
R6
R5
C3
Q3
R7
R8
C4
C5
R16
R10
C7
U2
C6
FN9111.6
December 2, 2011
VR1
U4
R17
C2
R9
ISL6740, ISL6741
R12
ISL6740, ISL6741
Absolute Maximum Ratings (Note 6)
Thermal Information
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +20.0V
OUTA, OUTB, Signal Pins . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VREF
VREF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .GND - 0.3V to 6.0V
Peak GATE Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0.5A
ESD Classification
Human Body Model (Per MIL-STD-883 Method 3015.7) . . . . . . . . 1500V
Charged Device Model (Per EOS/ESD DS5.3, 4/14/93). . . . . . . . 1000V
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
16 Lead SOIC (Notes 4, 5) . . . . . . . . . . . . . .
74
33
16 Lead TSSOP (Notes 4, 5) . . . . . . . . . . . .
98
30
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . .-55°C to +150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Temperature Range
ISL6740Ix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +105°C
ISL6741Ix. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +105°C
Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . . . . 9VDC - 16VDC
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379.
5. For θJC, the “case temp” location is taken at the package top center.
6. All voltages are with respect to GND.
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on
page 2 and page 3 and Typical Application Schematics on page 4 to page 6. 9V < VDD < 20 V, RTD = 51.1kΩ, RTC = 10kΩ, CT = 470pF, TA = -40°C
to +105°C, Typical values are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +105°C
PARAMETER
TEST CONDITIONS
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
SUPPLY VOLTAGE
Start-Up Current, IDD
VDD < START Threshold
-
95
140
µA
Operating Current, IDD
RLOAD, COUTA,B = 0
-
5.0
8.0
mA
COUTA,B = 1nF
-
7.0
12.0
mA
UVLO START Threshold
6.50
7.25
8.00
V
UVLO STOP Threshold
6.00
6.75
7.50
V
Hysteresis
0.25
0.50
0.75
V
4.900
5.000
5.050
V
-
3
-
mV
Fault Voltage
4.10
4.55
4.75
V
VREF Good Voltage
4.25
4.75
VREF - 0.05
V
Hysteresis
75
165
250
mV
Operational Current (Source)
-20
-
-
mA
5
-
-
mA
-25
-
-100
mA
0.55
0.6
0.65
V
CS to OUT Delay
-
35
50
ns
CS Sink Current
-
10
-
mA
-1.00
-
1.00
µA
-
80
-
mV
REFERENCE VOLTAGE
Overall Accuracy
IVREF = 0, -20mA
Long Term Stability
TA = +125°C, 1000 hours
Operational Current (Sink)
Current Limit
CURRENT SENSE
Current Limit Threshold
VERROR = VREF
Input Bias Current
CS to PWM Comparator Input Offset (ISL6741)
7
(Note 7)
FN9111.6
December 2, 2011
ISL6740, ISL6741
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on
page 2 and page 3 and Typical Application Schematics on page 4 to page 6. 9V < VDD < 20 V, RTD = 51.1kΩ, RTC = 10kΩ, CT = 470pF, TA = -40°C
to +105°C, Typical values are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +105°C (Continued)
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
-
4
-
V/V
SCSET Input Impedance
1
-
-
MΩ
SC Setpoint Accuracy
-
10
-
%
400
-
-
kΩ
VERROR < CS Offset (ISL6741)
-
-
0
%
VERROR < CT Valley Voltage (ISL6740)
-
-
0
%
VERROR > 4.75V (Note 9)
-
83
-
%
VERROR to PWM Comparator Input Offset (ISL6741)
0.4
1.0
1.25
V
VERROR to PWM Comparator Input Gain (ISL6741)
-
0.25
-
VERROR to PWM Comparator Input Gain (ISL6740)
-
0.4
-
V/V
CT to PWM Comparator Input Gain (ISL6740)
-
0.4
-
V/V
SS to PWM Comparator Input Gain (ISL6740)
-
0.5
-
V/V
SS to PWM Comparator Input Gain (ISL6741)
-
0.2
-
V/V
333
351
369
kHz
PARAMETER
TEST CONDITIONS
ACS = ΔVERROR/ΔVCS
Gain (ISL6741)
PULSE WIDTH MODULATOR
VERROR Input Impedance
Minimum Duty Cycle
Maximum Duty Cycle
OSCILLATOR
Frequency Accuracy
TA = +25°C
Frequency Variation with VDD
T = +105°C (f20V- - f9V)/f9V
-
2
3
%
T = -40°C (f20V- - f9V)/f9V
-
2
3
%
Temperature Stability
-
8
-
%
Charge Current Gain
1.88
2.0
2.12
µA/µA
45
55
65
µA/µA
CT Valley Voltage
0.75
0.80
0.85
V
CT Peak Voltage
2.70
2.80
2.90
V
-
2.000
-
V
Input High Threshold (VIH), Minimum
4.0
-
-
V
Input Low Threshold (VIL), Maximum
-
-
0.8
V
4.5
-
kΩ
Free
Running
-
1.67 x
Free Running
Hz
Discharge Current Gain
RTD, RTC Voltage
RLOAD = 0
SYNCHRONIZATION
Input Impedance
Input Frequency Range
High Level Output Voltage (VOH)
ILOAD = -1mA
-
4.5
-
V
Low Level Output Voltage (VOL)
ILOAD = 10µA
-
-
100
mV
SYNC Output Current
VOH > 2.0V
-10
-
-
mA
SYNC Output Pulse Duration (Minimum)
(Note 8)
250
-
532
ns
SYNC Advance
SYNC rising edge to GATE falling edge,
CGATE = CSYNC = 100pF
-
5
-
ns
-45
-55
-75
µA
SOFT-START
Charging Current
SS = 2V
8
FN9111.6
December 2, 2011
ISL6740, ISL6741
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on
page 2 and page 3 and Typical Application Schematics on page 4 to page 6. 9V < VDD < 20 V, RTD = 51.1kΩ, RTC = 10kΩ, CT = 470pF, TA = -40°C
to +105°C, Typical values are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +105°C (Continued)
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
SS Clamp Voltage
4.35
4.5
4.65
V
Sustained Overcurrent Threshold Voltage (ISL6740) Charged Threshold minus:
0.20
0.25
0.30
V
PARAMETER
TEST CONDITIONS
Overcurrent/Short Circuit Discharge Current
SS = 2V
13
18
23
µA
Fault SS Discharge Current
SS = 2V
-
10.0
-
mA
0.25
0.27
0.33
V
Reset Threshold Voltage
FAULT
Fault High Level Output Voltage (VOH)
ILOAD = -10mA
2.85
3.5
-
V
Fault Low Level Output Voltage (VOL)
ILOAD = 10mA
-
0.4
0.9
V
Fault Rise Time
CLOAD = 100pF
-
15
-
ns
Fault Fall Time
CLOAD = 100pF
-
15
-
ns
High Level Output Voltage (VOH)
VREF - OUTA or OUTB,
IOUT = -50mA
-
0.5
1.0
V
Low Level Output Voltage (VOL)
OUTA or OUTB - GND, IOUT = 50mA
-
0.5
1.0
V
Rise Time
CGATE = 1nF, VDD = 15V
-
50
100
ns
Fall Time
CGATE = 1nF, VDD = 15V
-
40
80
ns
Thermal Shutdown
135
145
155
°C
Thermal Shutdown Clear
120
130
140
°C
Hysteresis, Internal Protection
-
15
-
°C
Reference, External Protection
2.375
2.50
2.625
V
Hysteresis, External Protection
18
25
30
µA
0.97
1.00
1.03
V
7
10
15
µA
4.8
-
-
V
1
-
-
MΩ
OUTPUT
THERMAL PROTECTION
SUPPLY UVLO/INHIBIT
Input Voltage Low/Inhibit Threshold
Hysteresis, Switched Current Amplitude
Input High Clamp Voltage
Input Impedance
NOTES:
7. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.
8. SYNC pulse width is the greater of this value or the CT discharge time.
9. This is the maximum duty cycle achievable using the specified values of RTC, RTD, and CT. Larger or smaller maximum duty cycles may be obtained
using other values for these components. See Equations 2 through 4.
9
FN9111.6
December 2, 2011
ISL6740, ISL6741
Typical Performance Curves
65
CT DISCHARGE CURRENT GAIN
NORMALIZED VREF
1.001
1.000
0.999
0.998
0.997
-40
-25 -10
5
20
35
50
65
80
95 110
60
55
50
45
40
0
50
100 150 200 250 300 350 400 450 500
TEMPERATURE (°C)
RTD CURRENT (µA)
FIGURE 1. REFERENCE VOLTAGE vs TEMPERATURE
FIGURE 2. OSCILLATOR CT DISCHARGE CURRENT GAIN
1•106
CT (pF) =
1000
680
470
1•103 330
220
100
FREQUENCY (Hz)
DEADTIME - TD (ns)
1•104
100
10
10
20
30
40
50
60
70
80
90
100
RTD (kΩ)
1•105
RTD = 10k
CT (pF) =
100
220
330
470
1•104
10
20
30
680
1000
40
50
60
70
80
90
100
RTC (kΩ)
FIGURE 3. DEADTIME (TD) vs CAPACITANCE
FIGURE 4. CAPACITANCE vs FREQUENCY
Pin Descriptions
VDD - VDD is the power connection for the IC. To optimize noise
immunity, bypass VDD to GND with a ceramic capacitor as close
to the VDD and GND pins as possible.
The total supply current, IDD, will be dependent on the load
applied to outputs OUTA and OUTB. Total IDD current is the sum
of the quiescent current and the average output current. Knowing
the operating frequency, fSW, and the output loading capacitance
charge, Q, per output, the average output current can be
calculated from:
I OUT = 2 • Q • f SW
A
(EQ. 1)
SYNC - A bidirectional synchronization signal used to coordinate
the switching frequency of multiple units. Synchronization may
be achieved by connecting the SYNC signal of each unit together
or by using an external master clock signal. The oscillator timing
capacitor, CT, is always required regardless of the
synchronization method used. The paralleled unit with the
highest oscillator frequency assumes control. Selfsynchronization is not recommended for oscillator frequencies
above 900kHz. For higher switching frequencies, an external
10
clock with a pulse width less than one-half of the oscillator period
must be used.
RTC - This is the oscillator timing capacitor charge current control
pin. A resistor is connected between this pin and GND. The
current flowing through the resistor determines the magnitude of
the charge current. The charge current is nominally twice this
current. The PWM maximum ON time is determined by the
timing capacitor charge duration.
RTD - This is the oscillator timing capacitor discharge current
control pin. A resistor is connected between this pin and GND.
The current flowing through the resistor determines the
magnitude of the discharge current. The discharge current is
nominally 50x this current. The PWM deadtime is determined by
the timing capacitor discharge duration.
CT - The oscillator timing capacitor is connected between this pin
and GND.
VERROR - The inverting input of the PWM comparator. The error
voltage is applied to this pin to control the duty cycle. Increasing
the signal level increases the duty cycle. The node may be driven
with an external error amplifier or opto-coupler.
FN9111.6
December 2, 2011
ISL6740, ISL6741
The ISL6740, ISL6741 features a built-in soft-start. Soft-start is
implemented as a clamp on the error voltage input.
current limit characteristic of peak current mode control limits
the output current to acceptable levels.
OTS - The non-inverting input to the over-temperature shutdown
comparator. The signal input at this pin is compared to an
internal threshold of VREF/2. If the voltage at this pin exceeds the
threshold, the Fault signal is asserted and the outputs are
disabled until the condition clears. There is a nominal 25μA
switched current source used for hysteresis. The amount of
hysteresis is adjustable by varying the source impedance of the
signal into this pin.
GND - Reference and power ground for all functions on this
device. Due to high peak currents and high frequency operation,
a low impedance layout is necessary. Ground planes and short
traces are highly recommended.
OTS may be used to monitor parameters other than temperature,
such as voltage. Any signal for which a high out-of-bounds
monitor is desired may utilize the OTS comparator.
FAULT - The Fault signal is asserted high whenever the outputs,
OUTA and OUTB, are disabled. This occurs during an overtemperature fault, an input UV fault, a VREF UV fault, or during an
overcurrent (ISL6740) or short circuit shutdown fault. Fault can
be used to disable synchronous rectifiers whenever the outputs
are disabled.
Fault is a three-state output and is high impedance during the
soft-start cycle. Adding a pull-up resistor to VREF or a pull-down
resistor to ground determines the state of Fault during soft-start.
This feature allows the designer to use the Fault signal to enable
or disable output synchronous rectifiers during soft-start.
UV - Undervoltage monitor input pin. A resistor divider between
the input source voltage and GND sets the undervoltage lock out
threshold. The signal is compared to an internal 1.00V reference
to detect an undervoltage or inhibit condition.
CS - This is the input to the current sense comparator(s). The IC
has the PWM comparator for peak current mode control
(ISL6741) and an overcurrent protection comparator. The
overcurrent comparator threshold is set at 0.600V nominal.
The CS pin is shorted to GND at the end of each switching cycle.
Depending on the current sensing source impedance, a series
input resistor may be required due to the delay between the
internal clock and the external power switch. This delay may
allow an overlap such that the CS signal may be discharged while
the current signal is still active. If the current sense source is low
impedance, it will cause increased power dissipation.
ISL6740 - Exceeding the overcurrent threshold will start a
delayed shutdown sequence. Once an overcurrent condition is
detected, the soft-start charge current source is disabled. The
soft-start capacitor begins discharging through a 25μA current
source, and if it discharges to less than 4.25V (Sustained
Overcurrent Threshold), a shutdown condition occurs and the
OUTA and OUTB outputs are forced low. When the soft-start
voltage reaches 0.27V (Reset Threshold) a soft-start cycle begins.
An overcurrent condition must be absent for 50μs before the
delayed shutdown control resets. If the overcurrent condition
ceases, and an additional 50μs period elapses before the
shutdown threshold is reached, no shutdown occurs. The SS
charging current is re-enabled and the soft-start voltage is
allowed to recover.
ISL6741 - The ISL6741 current mode controller does not
shutdown due to an overcurrent condition. The pulse-by-pulse
11
OUTA and OUTB - Alternate half cycle output stages. Each output
is capable of 0.5A peak currents for driving logic level power
MOSFETs or MOSFET drivers. Each output provides very low
impedance to overshoot and undershoot.
VREF - The 5.00V reference voltage output. +1%/-2% tolerance
over line, load and operating temperature. Bypass to GND with a
0.047μF to 2.2μF ceramic capacitor. Capacitors outside of this
range may cause oscillation.
SS - Connect the soft-start timing capacitor between this pin and
GND to control the duration of soft-start. The value of the
capacitor determines the rate of increase of the duty cycle during
start up, controls the overcurrent shutdown delay (ISL6740), and
the overcurrent and short circuit hiccup restart period.
SCSET - Sets the duty cycle threshold that corresponds to a short
circuit condition. A resistive divider between RTC and GND or RTD
and GND, or a voltage between 0V and 2V may be used to adjust
the SCSET threshold. If using a resistor divider from either RTC or
RTD, the impedance to GND affects the oscillator timing and
should be considered when determining the oscillator timing
components. Connecting SCSET to GND disables short circuit
shutdown and hiccup.
Functional Description
Features
The ISL6740, ISL6741 PWMs are an excellent choice for low cost
bridge and push-pull topologies for applications requiring
accurate duty cycle and deadtime control. With its many
protection and control features, a highly flexible design with
minimal external components is possible. Among its many
features are current mode control (ISL6741), adjustable softstart, overcurrent protection, thermal protection, bidirectional
synchronization, fault indication, and adjustable frequency.
Oscillator
The ISL6740, ISL6741 have an oscillator with a programmable
frequency range to 2MHz, which can be programmed with two
resistors and capacitor. The use of three timing elements, RTC,
RTD, and CT allow great flexibility and precision when setting the
oscillator frequency.
The switching period may be considered the sum of the timing
capacitor charge and discharge durations. The charge duration is
determined by RTC and CT. The discharge duration is determined
by RTD and CT.
t C ≈ 0.5 • R TC • C T
t D ≈ 0.02 • R TD • C T
1
t SW = t C + t D = ----------F SW
(EQ. 2)
S
(EQ. 3)
S
S
(EQ. 4)
FN9111.6
December 2, 2011
ISL6740, ISL6741
where tC and tD are the charge and discharge times, respectively,
tSW is the oscillator free running period, and f is the oscillator
frequency. One output switching cycle requires two oscillator
cycles. The actual times will be slightly longer than calculated
due to internal propagation delays of approximately
10ns/transition. This delay ads directly to the switching duration,
but also causes overshoot of the timing capacitor peak and valley
voltage thresholds, effectively increasing the peak-to-peak
voltage on the timing capacitor. Additionally, if very low charge
and discharge currents are used, there will be increased error
due to the input impedance at the CT pin.
The maximum duty cycle, D, and percent deadtime, DT, can be
calculated from:
tC
D = ---------t SW
(EQ. 5)
DT = 1 – D
(EQ. 6)
Implementing Synchronization
The oscillator can be synchronized to an external clock applied to
the SYNC pin or by connecting the SYNC pins of multiple ICs
together. If an external master clock signal is used, the free
running frequency of the oscillator should be ~10% slower than
the desired synchronous frequency. The external master clock
signal should have a pulse width greater than 20ns. The SYNC
circuitry will not respond to an external signal during the first
60% of the oscillator switching cycle. Self-synchronization is not
recommended for oscillator frequencies above 900 kHz. For
higher switching frequencies, an external clock with a pulse
width less than one-half of the oscillator period must be used.
The SYNC input is edge triggered and its duration does not affect
oscillator operation. However, the deadtime is affected by the
SYNC frequency. A higher frequency signal applied to the SYNC
input will shorten the deadtime. The shortened deadtime is the
result of the timing capacitor charge cycle being prematurely
terminated by the external SYNC pulse. Consequently, the timing
capacitor is not fully charged when the discharge cycle begins.
This effect is only a concern when an external master clock is
used, or if units with different operating frequencies are
paralleled.
recovery from a Fault condition or overcurrent/short circuit
shutdown. The soft-start voltage is clamped to 4.5V.
The Fault signal output is high impedance during the soft-start
cycle. A pull-up resistor to VREF or a pull-down resistor to ground
should be added to achieve the desired state of Fault during softstart.
Gate Drive
The ISL6740, ISL6741 are capable of sourcing and sinking 0.5A
peak current, but are primarily intended to be used in
conjunction with a MOSFET driver due to the 5V drive level. To
limit the peak current through the IC, an external resistor may be
placed between the totem-pole output of the IC (OUTA or OUTB
pin) and the gate of the MOSFET. This small series resistor also
damps any oscillations caused by the resonant tank of the
parasitic inductances in the traces of the board and the FET’s
input capacitance.
Undervoltage Monitor and Inhibit
The UV input is used for input source undervoltage lockout and
inhibit functions. If the node voltage falls below 1.00V a UV
shutdown fault occurs. This may be caused by low source voltage
or by intentional grounding of the pin to disable the outputs.
There is a nominal 10μA switched current source used to create
hysteresis. The current source is active only during an UV/Inhibit
fault; otherwise, it is inactive and does not affect the node
voltage. The magnitude of the hysteresis is a function of the
external resistor divider impedance. If the resistor divider
impedance results in too little hysteresis, a series resistor
between the UV pin and the divider may be used to increase the
hysteresis. A soft-start cycle begins when the UV/Inhibit fault
clears.
The voltage hysteresis created by the switched current source
and the external impedance is generally small due to the large
resistor divider ratio required to scale the input voltage down to
the UV threshold level. A small capacitor placed between the UV
input and ground may be required to filter noise out.
VIN
R1
Soft-start Operation
1.00V
The ISL6740, ISL6741 feature a soft-start using an external
capacitor in conjunction with an internal current source. soft-start
reduces stresses and surge currents during start up.
Upon start up, the soft-start circuitry clamps the error voltage
input (VERROR pin) indirectly to a value equal to the soft-start
voltage. The soft-start clamp does not actually clamp the error
voltage input as is done in many implementations. Rather the
PWM comparator has two inverting inputs such that the lower
voltage is in control.
The output pulse width increases as the soft-start capacitor
voltage increases. This has the effect of increasing the duty cycle
from zero to the regulation pulse width during the soft-start
period. When the soft-start voltage exceeds the error voltage,
soft-start is completed. soft-start occurs during start-up, after
12
+
-
R3
10μA
R2
ON
FIGURE 5. UV HYSTERESIS
As VIN decreases to a UV condition, the threshold level is:
R1 + R2
V IN ( DOWN ) = ---------------------R2
V
(EQ. 7)
FN9111.6
December 2, 2011
ISL6740, ISL6741
The hysteresis voltage, ΔV, is:
ΔV = 10
–5
R1 + R2
• 〈 R1 + R3 • ⎛ ----------------------⎞ 〉
⎝ R2 ⎠
V
(EQ. 8)
Setting R3 equal to zero results in the minimum hysteresis, and
yields:
ΔV = 10
–5
• R1
V
(EQ. 9)
As VIN increases from a UV condition, the threshold level is:
V IN ( UP ) = V IN ( DOWN ) + ΔV
V
(EQ. 10)
Overcurrent Operation
ISL6740 - Overcurrent delayed shutdown is enabled once the
soft-start cycle is complete. If an overcurrent condition is
detected, the soft-start charging current source is disabled and
the soft-start capacitor is allowed to discharge through a 15μA
source. At the same time a 50μs re-triggerable one-shot timer is
activated. It remains active for 50μs after the overcurrent
condition ceases. If the soft-start capacitor discharges by more
then 0.25V to 4.25V, the output is disabled and the Fault signal
asserted. This state continues until the soft-start voltage reaches
270mV, at which time a new soft-start cycle is initiated. If the
overcurrent condition stops at least 50μs prior to the soft-start
voltage reaching 4.25V, the soft-start charging currents revert to
normal operation and the soft-start voltage is allowed to recover.
sets a threshold that is compared to the voltage on the timing
capacitor, CT. The resistor divider percentage corresponds to the
fraction of the maximum duty cycle below which a short circuit
may exist. If the timing capacitor voltage fails to exceed the
threshold before an overcurrent pulse is detected, a short circuit
condition exists. A shutdown and soft-start cycle will begin if 8
short circuit events occur within 32 oscillator cycles. Connecting
SCSET to GND disables this feature.
Since the current sourced from both RTC and RTD determine the
charge and discharge currents for the timing capacitor, the effect
of the SCSET divider must be included in the timing calculations.
Typically the resistor between RTC and GND is formed by two
series resistors with the center node connected to SCSET.
Alternatively, SCSET may be set using a voltage between 0V and
2V. This voltage divided by 2 determines the percentage of the
maximum duty cycle that corresponds to a short circuit when
current limit is active. For example, if the maximum duty cycle is
95% and 1V is applied to SCSET, then the short circuit duty cycle
is 50% of 95% or 47.5%.
Fault Conditions
A fault condition occurs if VREF falls below 4.65V, the UV input
falls below 1.00V, the thermal protection is triggered, or if OTS
faults. When a fault is detected, OUTA and OUTB outputs are
disabled, the Fault signal is asserted, and the soft-start capacitor
is quickly discharged. When the fault condition clears and the
soft-start voltage is below the reset threshold, a soft-start cycle
begins. The Fault signal is high impedance during the soft-start
cycle.
The duration of the OC shutdown period can be increased by
adding a resistor between VREF and SS. The value of the resistor
must be large enough so that the minimum specified SS
discharge current is not exceeded. Using a 422kΩ resistor, for
example, will result in a small current being injected into SS,
effectively reducing the discharge current. This will increase the
OFF time by about 60%, nominally. The external pull-up resistor
will also decrease the SS duration, so its effect should be
considered when selecting the value of the SS capacitor.
An overcurrent condition that results in shutdown (ISL6740), or a
short circuit shutdown also cause assertion of the Fault signal.
The difference between a current fault and the faults described
earlier is that the soft-start capacitor is not quickly discharged.
The initiation of a new soft-start cycle is delayed while the softstart capacitor is discharged at a 15μA rate. This keeps the
average output current to a minimum.
Latching OC shutdown is also possible by using a lower valued
resistor between VREF and SS. If the SS node is not allowed to
discharge below the SS reset threshold, the IC will not recover
from an overcurrent fault. The value of the resistor must be low
enough so that the maximum specified discharge current is not
sufficient to pull SS below 0.33V. A 200kΩ resistor, for example,
prevents SS from discharging below ~0.4V. Again, the external
pull-up resistor will decrease the SS duration, so its effect should
be considered when selecting the value of the SS capacitor.
Two methods of over-temperature protection are provided. The
first method is an on board temperature sensor that protects the
device should the junction temperature exceed 145°C. There is
approximately 15°C of hysteresis.
ISL6741 - Overcurrent results in pulse-by-pulse duty cycle
reduction as occurs in any peak current mode controller. This
results in a well controlled decrease in output voltage with
increasing current beyond the overcurrent threshold. An
overcurrent condition in the ISL6741 will not cause a shutdown.
Thermal Protection
The second method uses an internal comparator with a 2.5V
reference (VREF/2). The non-inverting input to the comparator is
accessible through the OTS pin. A thermistor or thermal sensor
located at or near the area of interest may be connected to this
input. There is a nominal 25μA switched current source used to
create hysteresis. The current source is active only during an OT
fault; otherwise, it is inactive and does not affect the node
voltage. The magnitude of the hysteresis is a function of the
external resistor divider impedance. Either a positive
temperature coefficient (PTC) or a negative temperature
Short Circuit Operation
A short circuit condition is defined as the simultaneous
occurrence of current limit and a reduced duty cycle. The degree
of reduced duty cycle is user adjustable using the SCSET input. A
resistor divider between either RTD or RTC and GND to RCSET
13
FN9111.6
December 2, 2011
ISL6740, ISL6741
coefficient (NTC) thermistor may be used. If a NTC is desired,
position R1 may be substituted.
VREF
VREF
ON
Circuit Element Descriptions
R1
The converter design may be broken down into the following
functional blocks:
25μA
VREF/2
R3
The input voltage range is 48 ±10%VDC. The output is a nominal
12V when the input voltage is at 48V. Since this is an
unregulated topology, the output voltage will vary proportionately
with input voltage. The load regulation is a function of resistance
between the source and the converter output. The output is rated
at 8A.
+
-
Input Filtering: L1, C1, R1
R2
Half-Bridge Capacitors: C2, C3
Isolation Transformer: T1
Primary Snubber: C13, R10
Start Bias Regulator: CR3, R2, R7, C6, Q5, D1
FIGURE 6. OTS HYSTERESIS
Supply Bypass Components: R3, C15, C4, C5
If a PTC is desired, then position R2 may be substituted. The
threshold with increasing temperature is set by making the fixed
resistance equal in value to the thermistor resistance at the
desired trip temperature.
V TH↑ = 2.5V and R1 = R2 (HOT)
5
Ω
(EQ. 11)
If the hysteresis resistor, R3, is not desired, the value of the
thermistor resistance at the reset temperature can be
determined from:
2.5 • R2
R1 = ----------------------------------------–5
2.5 – 10 • R2
Ω
2.5 • R1
R2 = ----------------------------------------–5
2.5 + 10 • R1
Ω
Current Sense Network: T2, CR1, CR2, R5, R6, R11, C10, C14
Control Circuit: U3, RT1, R14, R19, R13, R15, R17, R18, C16, C18,
C17
Output Rectification and Filtering: QR1, QR2, QR3, QR4, L2, C9, C8
To determine the value of the hysteresis resistor, R3, select the
value of thermistor resistance that corresponds to the desired
reset temperature.
10 • ( R1 – R2 ) – R1 • R2
R3 = ---------------------------------------------------------------------R1 + R2
Main MOSFET Power Switch: QH, QL
( NTC )
(EQ. 12)
Secondary Snubber: R8, R9, C11, C12
FET Driver: U1
ZVS Resonant Delay (Optional): L3, C7
Design Criteria
The following design requirements were selected:
Switching Frequency, Fsw: 235kHz
VIN: 48 ±10%V
VOUT: 12V (nominal) @ IOUT = 8A
( PTC )
(EQ. 13)
POUT: 100W
Efficiency: 95%
The OTS comparator may also be used to monitor signals other
than suggested above. It may also be used to monitor any
voltage signal for which an excess requires a response as
described above. Input or output voltage monitoring are
examples of this.
Ground Plane Requirements
Careful layout is essential for satisfactory operation of the device.
A good ground plane must be employed. VDD should be bypassed
directly to GND with good high frequency capacitance.
Ripple: 1%
Transformer Design
The design of a transformer for a half-bridge application is a
straight forward affair, although iterative. It is a process of many
compromises, and even experienced designers will produce
different designs when presented with identical requirements.
The iterative design process is not presented here for clarity.
Typical Application
The Typical Application Schematic features the ISL6740 in an
unregulated half-bridge DC/DC converter configuration, often
referred to as a DC Transformer or Bus Regulator. The
ISL6740EVAL1 demonstration unit implements this design and is
available for evaluation.
14
FN9111.6
December 2, 2011
ISL6740, ISL6741
The abbreviated design process follows:
Ae = 0.62cm2 or 6.2e -5m2
• Select a core geometry suitable for the application.
Constraints of height, footprint, mounting preference, and
operating environment will affect the choice.
Using Faraday’s Law, V = N dΦ/dt, the number of primary turns
can be determined once the maximum flux density is set. An
acceptable Bmax is ultimately determined by the allowable
power dissipation in the ferrite material and is influenced by the
lossiness of the core, core geometry, operating ambient
temperature, and air flow. The TDK datasheet for PC44 material
indicates a core loss factor of ~400mW/cm3 with a ±2000
gauss 100kHz sinusoidal excitation. The application uses a
235kHz square wave excitation, so no direct comparison
between the application and the data can be made. Interpolation
of the data is required. The core volume is approximately
1.6cm3, so the estimated core loss is
• Determine the turns ratio.
• Select suitable core material(s).
• Select maximum flux density desired for operation.
• Select core size. Core size will be dictated by the capability of
the core structure to store the required energy, the number of
turns that have to be wound, and the wire gauge needed. Often
the window area (the space used for the windings) and power
loss determine the final core size.
• Determine maximum desired flux density. Depending on the
frequency of operation, the core material selected, and the
operating environment, the allowed flux density must be
determined. The decision of what flux density to allow is often
difficult to determine initially. Usually the highest flux density
that produces an acceptable design is used, but often the
winding geometry dictates a larger core than is indicated
based on flux density alone.
• Determine the number of primary turns.
f act
3
mW
200kHz
P loss ≈ ----------- • cm • --------------- = 0.4 • 1.6 • --------------------- = 1.28
3
f meas
100kHz
cm
W
(EQ. 15)
1.28W of dissipation is significant for a core of this size.
Reducing the flux density to 1200 gauss will reduce the
dissipation by about the same percentage, or 40%. Ultimately,
evaluation of the transformer’s performance in the application
will determine what is acceptable.
From Faraday’s Law and using 1200 gauss peak flux density (ΔB
= 2400 gauss or 0.24 tesla)
• Select the wire gauge for each winding.
• Determine winding order and insulation requirements.
–6
V IN • T ON
53 • 2 • 10
N = ----------------------------- = ------------------------------------------------------ = 3.56
–5
2 • A e • ΔB
2 • 6.2 • 10 • 0.24
• Verify the design.
nSR
(EQ. 16)
Rounding up yields 4 turns for the primary winding. The peak flux
density using 4 turns is ~1100 gauss. From Equation 1, the
number of secondary turns is 2.
nS
nP
nS
The volts/turn for this design ranges from 5.4V at VIN = 43V to
6.6V at VIN = 53V. Therefore, the synchronous rectifier (SR)
windings may be set at 1 turn each with proper FET selection.
Selecting 2 turns for the synchronous rectifier windings would
also be acceptable, but the gate drive losses would increase.
nSR
FIGURE 7. TRANSFORMER SCHEMATIC
For this application we have selected a planar structure to
achieve a low profile design. A PQ style core was selected
because of its round center leg cross section, but there are many
suitable core styles available.
Since the converter is operating open loop at nearly 100% duty
cycle, the turns ratio, N, is simply the ratio of the input voltage to
the output voltage divided by 2.
V IN
48
N = ---------------------- = ---------------- = 2
V OUT • 2
12 • 2
turns
(EQ. 14)
The factor of 2 divisor is due to the half-bridge topology. Only half
of the input voltage is applied to the primary of the transformer.
A PC44HPQ20/6 “E-Core” plus a PC44PQ20/3 “I-Core” from TDK
were selected for the transformer core. The ferrite material is
PC44.
The next step is to determine the equivalent wire gauge for the
planar structure. Since each secondary winding conducts for only
50% of the period, the RMS current is
I RMS = I OUT • D = 10 • 0.5 = 7.07
A
(EQ. 17)
where D is the duty cycle. Since an FR-4 PWB planar winding
structure was selected, the width of the copper traces is limited
by the window area width, and the number of layers is limited by
the window area height. The PQ core selected has a usable
window area width of 0.165 inches. Allowing one turn per layer
and 0.020 inches clearance at the edges allows a maximum
trace width of 0.125 inches. Using 100 circular mils(c.m.)/A as a
guideline for current density, and from Equation 17, 707c.m. are
required for each of the secondary windings (a circular mil is the
area of a circle 0.001 inches in diameter). Converting c.m. to
The core parameter of concern for flux density is the effective
core cross sectional area, Ae. For the PQ core pieces selected:
15
FN9111.6
December 2, 2011
ISL6740, ISL6741
square mils yields 555mils2 (0.785 sq. mils/c.m.). Dividing by
the trace width results in a copper thickness of 4.44 mils
(0.112mm). Using 1.3 mils/oz. of copper requires a copper
weight of 3.4oz. For reasons of cost, 3oz. copper was selected.
between the two secondary windings. The winding layout
appears below.
One layer of each secondary winding also contains the
synchronous rectifier winding. For this layer the secondary trace
width is reduced by 0.025 inches to 0.100 inches(0.015 inches
for the SR winding trace width and 0.010 inches spacing
between the SR winding and the secondary winding).
The choice of copper weight may be validated by calculating the
DC copper losses of the secondary winding as follows. Ignoring
the terminal and lead-in resistance, the resistance of each layer
of the secondary may be approximated using Equation 18.
2πρ
R = -----------------------⎛ r 2⎞
t • ln ⎜ -----⎟
⎝ r 1⎠
FIGURE 7A. TOP LAYER: 1 TURN SECONDARY AND SR
WINDINGS
Ω
(EQ. 18)
where
R = Winding resistance
ρ = Resistivity of copper = 669e-9Ω-inches at 20°C
t = Thickness of the copper (3 oz.) = 3.9e-3 inches
r2 = Outside radius of the copper trace = 0.324 or 0.299 inches
r1 = Inside radius of the copper trace = 0.199 inches
The winding without the SR winding on the same layer has a DC
resistance 2.21mΩ. The winding that shares the layer with the
SR winding has a DC resistance of 2.65mΩ. With the secondary
configured as a 4 turn center tapped winding (2 turns each side
of the tap), the total DC power loss for the secondary at +20°C is
486mW.
FIGURE 7B. INT. LAYER 1: 1 TURN SECONDARY WINDING
The primary windings have an RMS current of approximately 5A
(IOUT x NS/NP at ~ 100% duty cycle). The primary is configured as
2 layers, 2 turns per layer to minimize the winding stack height.
Allowing 0.020 inches edge clearance and 0.010 inches between
turns yields a trace width of 0.0575 inches. Ignoring the terminal
and lead-in resistance, and using Equation 18, the inner trace
has a resistance of 4.25mΩ, and the outer trace has a resistance
of 5.52mΩ. The resistance of the primary then is 19.5mΩ at
+20°C. The total DC power loss for the secondary at +20°C is
489mW.
Improved efficiency and thermal performance could be achieved
by selecting heavier copper weight for the windings. Evaluation in
the application will determine its need.
FIGURE 7C. INT. LAYER 2: 2 TURNS PRIMARY WINDING
The order and geometry of the windings affects the AC
resistance, winding capacitance, and leakage inductance of the
finished transformer. To mitigate these effects, interleaving the
windings is necessary. The primary winding is sandwiched
FIGURE 7D. INT. LAYER 3: 2 TURNS PRIMARY WINDING
16
FN9111.6
December 2, 2011
ISL6740, ISL6741
transition voltage amplitude. If the leakage inductance energy is
too low, ZVS operation is not possible and near or partial ZVS
operation occurs. As the leakage energy increases, the voltage
amplitude increases until it is clamped by the FET body diode to
ground or VIN, depending on which FET conducts. When the
leakage energy exceeds the minimum required for ZVS
operation, the voltage is clamped until the energy is transferred.
This behavior increases the time window for ZVS operation. This
behavior is not without consequences, however. The transition
time and the period of time during which the voltage is clamped
reduces the effective duty cycle.
FIGURE 7E. INT. LAYER 4: 1 TURN SECONDARY WINDING
The gate charge affects the switching speed of the FETs. Higher
gate charge translates into higher drive requirements and/or
slower switching speeds. The energy required to drive the gates is
dissipated as heat.
The maximum input voltage, VIN, plus transient voltage,
determines the voltage rating required. With a maximum input
voltage of 53V for this application, and if we allow a 10% adder
for transients, a voltage rating of 60V or higher will suffice.
The RMS current through the each primary side FET can be
determined from Equation 17, substituting 5A of primary current
for IOUT. The result is 3.5A RMS. Fairchild FDS3672 FETs, rated at
100V and 7.5A (rDS(ON) = 22mΩ), were selected for the halfbridge switches.
FIGURE 7F. BOTTOM LAYER: 1 TURN SECONDARY AND SR
WINDINGS
∅0.689
∅0.358
0.807
0.639
0.403
The synchronous rectifier FETs must withstand approximately
one half of the input voltage assuming no switching transients
are present. This suggests a device capable of withstanding at
least 30V is required. Empirical testing in the circuit revealed
switching transients of 20V were present across the device
indicating a rating of at least 60V is required.
The RMS current rating of 7.07A for each SR FET requires a low
rDS(ON) to minimize conduction losses, which is difficult to find in a
60V device. It was decided to use two devices in parallel to simplify
the thermal design. Two Fairchild FDS5670 devices are used in
parallel for a total of four SR FETs. The FDS5670 is rated at 60V
and 10A (rDS(ON) = 14mΩ).
Oscillator Component Selection
0.169
0.000
0.000 0.184
0.479
0.774
1.054
FIGURE 7G. PWB DIMENSIONS
MOSFET Selection
The criteria for selection of the primary side half-bridge FETs and
the secondary side synchronous rectifier FETs is largely based on
the current and voltage rating of the device. However, the FET
drain-source capacitance and gate charge cannot be ignored.
The zero voltage switch (ZVS) transition timing is dependent on
the transformer’s leakage inductance and the capacitance at the
node between the upper FET source and the lower FET drain. The
node capacitance is comprised of the drain-source capacitance
of the FETs and the transformer parasitic capacitance. The
leakage inductance and capacitance form an LC resonant tank
circuit which determines the duration of the transition. The
amount of energy stored in the LC tank circuit determines the
17
The desired operating frequency of 235kHz for the converter was
established in “Design Criteria” on page 14. The oscillator
frequency operates at twice the frequency of the converter
because two clock cycles are required for a complete converter
period.
During each oscillator cycle the timing capacitor, CT, must be
charged and discharged. Determining the required discharge
time to achieve zero voltage switching (ZVS) is the critical design
goal in selecting the timing components. The discharge time sets
the deadtime between the two outputs, and is the same as ZVS
transition time. Once the discharge time is determined, the
remainder of the period becomes the charge time.
The ZVS transition duration is determined by the transformer’s
primary leakage inductance, Llk, by the FET Coss, by the
transformer’s parasitic winding capacitance, and by any other
parasitic elements on the node. The parameters may be
FN9111.6
December 2, 2011
ISL6740, ISL6741
determined by measurement, calculation, estimate, or by some
combination of these methods.
π L lk • ( 2C oss + C xfrmr )
t zvs ≈ -----------------------------------------------------------------2
(EQ. 19)
S
Device output capacitance, Coss, is non-linear with applied
voltage. To find the equivalent discrete capacitance, Cfet, a
charge model is used. Using a known current source, the time
required to charge the MOSFET drain to the desired operating
voltage is determined and the equivalent capacitance is
calculated.
Ichg • t
Cfet = ------------------V
(EQ. 20)
F
Once the estimated transition time is determined, it must be
verified directly in the application. The transformer leakage
inductance was measured at 125nH and the combined
capacitance was estimated at 2000pF. Calculations indicate a
transition period of ~ 25ns. Verification of the performance
yielded a value of tD closer to 45ns.
The remainder of the switching half-period is the charge time, tC,
and can be found from
–9
1
1
= 2.08
t C = --------------- – t D = ----------------------------------- – 45 • 10
3
2 • FS
2 • 235 • 10
μs
(EQ. 21)
where FS is the converter switching frequency.
Using Figure 4, the capacitor value appropriate to the desired
oscillator operating frequency of 470kHz can be selected. A CT
value of 100pF, 220pF, or 330pF is appropriate for this
frequency. A value of 220pF was selected.
To obtain the proper value for RTD, Equation 3 is used. Since
there is a 10ns propagation delay in the oscillator circuit, it must
be included in the calculation. The value of RTD selected is
8.06kΩ.
A similar procedure is used to determine the value of RTC using
Equation 2. The value of RTC selected is the series combination of
17.4kΩ and 1.27kΩ. See section “Overcurrent Component
Selection” on page 18 for further explanation.
VOUT is the output voltage
If we allow a current ramp, ΔI, of 5% of the rated output current,
the minimum inductance required is:
V L • T ON
0.25 • 2.08
L ≥ ---------------------- = ----------------------------- = 1.04
ΔI
0.5
The output filter inductor and capacitor selection is simple and
straightforward. Under steady state operating conditions the
voltage across the inductor is very small due to the large duty
cycle. Voltage is applied across the inductor only during the
switch transition time, about 45ns in this application. Ignoring
the voltage drop across the SR FETs, the voltage across the
inductor during the ON time with VIN = 48V is:
V IN • N S • ( 1 – D )
V L = V S – V OUT = --------------------------------------------- ≈ 250
2N P
mV
(EQ. 23)
An inductor value of 1.4μH, rated for 18A was selected.
With a maximum input voltage of 53V, the maximum output
voltage is about 13V. The closest higher voltage rated capacitor
is 16V. Under steady state operating conditions the ripple current
in the capacitor is small, so it would seem appropriate to have a
low ripple current rated capacitor. However, a high rated ripple
current capacitor was selected based on the nature of the
intended load, multiple buck regulators. To minimize the output
impedance of the filter, a Sanyo OSCON 16SH150M capacitor in
parallel with a 22μF ceramic capacitor were selected.
Overcurrent Component Selection
There are two circuit areas to consider when selecting the
components for overcurrent protection, current limit and short
circuit shutdown. The current limit threshold is fixed at 0.6V while
the short circuit threshold is set to a fraction of the duty cycle the
designer wishes to define as a short circuit.
The current level that corresponds to the overcurrent threshold
must be chosen to allow for the dynamic behavior of an open
loop converter. In particular, the low inductor ripple current under
steady state operation increases significantly as the duty cycle
decreases.
14
V (L1:1)
I (L1)
13
12
11
10
9
8
0.9950
Output Filter Design
μH
0.9960
0.9970
0.9980
0.9990
1.000
TIME (ms)
FIGURE 8. STEADY STATE SECONDARY WINDING VOLTAGE
AND INDUCTOR CURRENT
(EQ. 22)
where
VL is the inductor voltage
VS is the voltage across the secondary winding
18
FN9111.6
December 2, 2011
ISL6740, ISL6741
and line regulation performance are demonstrated in the
following figures.
15
V (L1:1)
I (L1)
EFFICIENCY (%)
100
10
95
90
85
80
75
70
0
5
0.986
0.988
0.990
0.992
0.994
0.996
0.998
1
2
3
4
5
6
LOAD CURRENT (A)
7
8
9
FIGURE 10. EFFICIENCY vs LOAD VIN = 48Vt
1.000
TIME (ms)
The short circuit protection involves setting a voltage between 0
and 2V on the SCSET pin. The applied voltage divided by 2 is the
percent of maximum duty cycle that corresponds to a short
circuit when the peak current limit is active. A divider from RTC to
ground provides an easy method to achieve this. The divider
between RTC and GND formed by R13 and R15 determines the
percent of maximum duty cycle that corresponds to a short
circuit. The divider ratio formed by R13 and R15 is:
R15
1.27k
----------------------------- = ------------------------------------- = 0.068
R13 + R15
1.27k + 17.4k
(EQ. 24)
Therefore, the duty cycle that corresponds to a short circuit is
6.8% of D max (97.9%), or ~6.6%.
Performance
The major performance criteria for the converter are efficiency,
and to a lesser extent, load regulation. Efficiency, load regulation
19
OUTPUT VOLTAGE (V)
Figures 8 and 9 show the behavior of the inductor ripple under
steady state and overcurrent conditions. In this example, the
peak current limit is set at 11A. The peak current limit causes
the duty cycle to decrease resulting in a reduction of the average
current through the inductor. The implication is that the converter
can not supply the same output current in current limit that it can
supply under steady state conditions. The peak current limit
setpoint must take this behavior into consideration. A 3.32Ω
current sense resistor was selected for the rectified secondary of
current transformer T2, corresponding to a peak current limit
setpoint of 16.5A.
12.5
12.25
12.00
11.75
11.50
11.25
11
0
1
2
3
4
5
6
7
LOAD CURRENT (A)
8
9
FIGURE 11. LOAD REGULATION AT VIN = 48V
14.0
OUPUT VOLTAGE (V)
FIGURE 9. SECONDARY WINDING VOLTAGE AND INDUCTOR
CURRENT DURING CURRENT LIMIT OPERATION
13.5
13.0
12.5
12.0
11.5
11.0
45
46
47
48 49 50 51 52
INPUT VOLTAGE (V)
53
54
FIGURE 12. LINE REGULATION AT I OUT = 1A
As expected, the output voltage varies considerably with line and
load when compared to an equivalent converter with closed loop
feedback. However, for applications where tight regulation is not
required, such as those application that use downstream DC/DC
converters, this design approach is viable.
FN9111.6
December 2, 2011
ISL6740, ISL6741
Waveforms
Typical waveforms can be found in the following Figures. Figure
13 shows the output voltage during start up.
FIGURE 15. FET DRAIN-SOURCE VOLTAGE
FIGURE 13. OUTPUT SOFT-START
Figure 14 shows the output voltage ripple and noise at a 5A load.
FIGURE 16. FET D-S VOLTAGE NEAR-ZVS TRANSITION
FIGURE 14. OUTPUT RIPPLE AND NOISE (20MHz BW)
Figures 15 and 16 show the voltage waveforms at the switching
node shared by the upper FET source and the lower FET drain. In
particular, Figure 16 shows near ZVS operation at 8A of load when
the upper FET is turning off and the lower FET turning on. There is
insufficient energy stored in the leakage inductance to allow
complete ZVS operation. However, since the energy stored in the
node capacitance is proportional to V2, a significant portion of the
energy is still recovered. Figure 17 shows the switching transition
between outputs, OUTA and OUTB during steady state operation.
The deadtime duration of 48.6ns is clearly shown.
FIGURE 17. OUTA TO OUTB TRANSITION
20
FN9111.6
December 2, 2011
ISL6740, ISL6741
Adding Line Only Regulation - Feed Forward
Component List
REFERENCE
DESIGNATOR
VALUE
DESCRIPTION
C1
1.0μF
Capacitor, 1812, X7R, 100V, 20%
TDK C4532X7R2A105M
C2, C3
3.3μF
Capacitor, 1812, X5R, 50V, 20%
TDK C4532X5R1H335M
C4, C6
1.0μF
Capacitor, 0805, X5R, 16V, 10%
TDK C2012X5R1C105K
C5, C15, C16
0.1μF
Capacitor, 0603, X7R, 50V, 10%
TDK C1608X7R1H104K
C7
Open
Capacitor, 0603, Open
C8
22μF
Capacitor, 1812, X5R, 16V, 20%
TDK C4532X5R1C226M
C9
150μF
Output voltage variation caused by changes in the supply voltage
may be virtually removed through a technique known as feed
forward compensation. Using feed forward, the duty cycle is
directly controlled based on changes in the input voltage only. No
closed loop feedback system is required. Voltage feed forward
may be implemented as shown in Figure 18.
1.5V
Capacitor, 0603, COG, 16V, 5%
TDK C1608COG1C221J
C18
0.047μF Capacitor, 0603, X7R, 16V, 10%
TDK C1608X7R1C473K
CR1, CR2
Diode, Schottky, BAT54S
CR3
Diode, Schottky, BAT54
D1
Zener, 10V, Philips BZX84-C10
L1
190nH
Pulse, P2004T
L2
1.5μH
Pulse, PG0077.142
L3
Short
Jumper or Optional Discrete Leakage
Inductance
Q5
Transistor, ON MJD31C
Q L, Q H
FET, Fairchild FDS3672
QR1, QR2, QR3,
QR4
FET, Fairchild FDS5670
0.8V
+VIN
R106
100K
R103
49.9k
R100
69.8k
+
-
Capacitor, Radial, Sanyo 16SH150M
220pF
R111
806
VREF
U100A
U100B
+
-
R105
100k
R102
100k
C10, C11, C12, 1000pF Capacitor, 0603, X7R, 50V, 10%
C13, C14
TDK C1608X7R1H102K
C17
R110
698
R109
3.48k
to VERROR
R104
100k
R101
2k
R107
100k
C100
1nF
R108
100k
FIGURE 18. VOLTAGE FEED FORWARD CIRCUIT
The circuit provides feed forward compensation for a 2:1 input
voltage range. Resistors R100 and R101 set the input voltage
divider to generate a 1V signal at the input voltage that
corresponds to maximum duty cycle (VIN minimum). Resistors
R109, R110, and R111 form a voltage divider from VREF to create
reference voltages for the amplifiers. The first stage uses U100A,
R102, R103, R104, and C100 to form a unity gain inverting
amplifier. Its output varies inversely with input voltage and
ranges from 1V to 2V. The bandwidth of the circuit may be
controlled by varying the value of C100. The gain of the first
amplifier stage is:
R1, R10
3.3
Resistor, 2512, 5%
R2
3.01k
Resistor, 2512, 1%
R3, R6
10.0
Resistor, 0603, 1%
R5
3.32
Resistor, 0603, 1%
R7
75.0k
Resistor, 0805, 1%
R8, R9
20.0
Resistor, 0805, 1%
VA = Output voltage of U100A
R11
100
Resistor, 0603, 1%
VD = The input divider voltage
R12
8.06k
Resistor, 0603, 1%
R13
17.4k
Resistor, 0603, 1%
R14
Open
Resistor, 0603, Open
R15
1.27k
Resistor, 0603, 1%
R17
97.6k
Resistor, 0603, 1%
The second stage uses U100B, R105, R106, R107, and R108 to
form a summing amplifier which offsets the first stage output by
0.8V (the value of CT valley voltage). The signal applied to the
VERROR input now matches the offset and amplitude of the
oscillator sawtooth so that the duty cycle varies linearly from
100% to 50% of maximum with a 2:1 input voltage variation.
R18
3.01k
Resistor, 0603, 1%
R19, RT1
10.0k
Resistor, 0603, 1%
T1
Midcom 31718
T2
Pulse P8205T
U1
Intersil HIP2101IB
U3
ISL6740IB
21
V A = – V D + 3.00
V
(EQ. 25)
where:
FN9111.6
December 2, 2011
ISL6740, ISL6741
Other duty ranges are possible, but are still limited to a 2:1 ratio.
The voltage applied to VERROR must be scaled to the peak-topeak voltage on CT, and offset by the valley voltage. Since the
peak-to-peak CT voltage is 2.00V nominal, the voltage at the
output of U100A must be divided by 2.0V to obtain the desired
duty cycle. For example, if an 80% duty cycle was required at the
minimum operating voltage, the output of U100A must be 1.60V
(80% of 2.00V). From (Equation 25), the divider voltage must be
set to 1.4V for the input voltage that corresponds to the 80% duty
cycle.
It should be noted that the synchronous rectifiers (SRs), being
driven from the transformer secondary, are only gated on during
the ON time of the primary FETs. Conduction continues through
the body diodes during the OFF time when operating in
continuous inductor current mode. This mode of operation
usually results in significant conduction and switching losses in
the SR FETs. These losses may be reduced considerably by either
adding schottky diodes in parallel to the SR FETs or by driving the
SR FETs directly with a control signal.
A block diagram of the feedback control loop follows in
Figure 19.
POWER
STAGE
PWM
VOUT
ERROR AMPLIFIER
Z2
ISOLATION
+
Z1
REF
FIGURE 19. CONTROL LOOP BLOCK DIAGRAM
The loop compensation is placed around the Error Amplifier (EA)
on the secondary side of the converter. A Type 3 error amplifier
configuration was selected.
Adding Regulation - Closed Loop Feedback
The second Typical Application schematic adds closed loop
feedback with isolation. The ISL6740EVAL2Z demonstration
platform implements this design and is available for evaluation.
The input voltage range was increased to 36V to 75V, which
necessitates a few modifications to the open loop design. The
output inductor value was increased to 4.0μH, schottky rectifier
CR4 was added to minimize SR FET body diode conduction, the
turns ratio of the main transformer was changed to 4:3, and the
synchronous rectifier gate drives were modified. The design
process is essentially the same as it was for the unregulated
version, so only the feedback control loop design will be
discussed.
The major components of the feedback control loop are a
programmable shunt regulator and an opto-coupler. The optocoupler is used to transfer the error signal across the isolation
barrier. The opto-coupler offers a convenient means to cross the
isolation barrier, but it adds complexity to the feedback control
loop. It adds a pole at about 10kHz and a significant amount of
gain variation due the current transfer ratio (CTR). The CTR of the
opto-coupler varies with initial tolerance, temperature, forward
current, and age.
VOUT
-
VERR
+
REF
FIGURE 20. TYPE 3 ERROR AMPLIFIER
The control to output transfer function may be represented as [1]
s
1 + -----V IN
NS
vo
ωz
----- = --------------- • ------- • -----------------------------------------------vc
VS • 2 NP
s 2
s
1 + ---------------- + ⎛ -------⎞
⎝ω ⎠
( Q )ω
o
(EQ. 26)
o
where:
Ro
Q = --------------ωo • L
1
ω o = ----------LC
or
1
f o = -----------------2π LC
1
ω z = ---------Rc C
or
1
f z = -----------------2πR c C
Ro = Output Load Resistance
L = Output Inductance
C = Output Capacitance
Rc = Output Capacitance ESR
VS = Sawtooth Ramp Amplitude
Gain and phase plots of (Equation 26) appear below using
L = 4.0μH, C = 150μF, Rc = 28mΩ, Ro = 1.2Ω, and VIN = 75V.
22
FN9111.6
December 2, 2011
ISL6740, ISL6741
The higher the desired bandwidth of the converter, the more
difficult it is to create a solution that is stable over the entire
operating range. A good rule of thumb is to limit the bandwidth to
about fSW/4, where fSW is the switching frequency of the
converter. However, due to the bandwidth constraints of the optocoupler and the LM431 shunt regulator, the bandwidth was
reduced to about 25kHz.
40
30
GAIN (dB)
20
10
0
-10
-20
10
100
1•103
1•104
FREQUENCY (Hz)
1•105
1•106
FIGURE 21A. CONTROL-TO-OUTPUT GAIN
50
Some liberties where taken with the generally accepted
compensation procedure described above due to the transfer
characteristics of the opto coupler. The effects of the optocoupler tend to dominate over those of the LM431 so the GBWP
effects of the LM431 are not included here.
0
PHASE (DEGREES)
The first pole is placed at the origin by default (C20 is an
integrating capacitor). If the two zeroes are placed at the same
frequency, they should be placed at fLC/2, where fLC is the
resonant frequency of the output L-C filter. To reduce the gain
peaking at the L-C resonant frequency, the two zeroes are often
separated. When they are separated, the first zero may be placed
at fLC/5, and the second at just above fLC. The second pole is
placed at the lowest expected zero cause by the output capacitor
ESR. The third, and last pole is placed at about 1.5 times the
cross over frequency.
-50
The gain and phase characteristics of the opto coupler are shown
in Figure 22A.
-100
10
-150
5
10
100
1•103
1•104
FREQUENCY (Hz)
1•105
1•106
FIGURE 21B. CONTROL-TO-OUTPUT PHASE
The Type 3 compensation configuration has three poles and two
zeros. The first pole is at the origin, and provides the integration
characteristic which results in excellent DC regulation. Referring
to the Typical Application Schematic for the regulated output, the
remaining poles and zeros for the compensator are located at:
-5
-10
-15
-20
10
1
f p2 = ----------------------------------------2π • R21 • C20
1
f p3 ≈ ------------------------------------2π • R4 • C22
0
GAIN (dB)
-200
100
(EQ. 27)
1•103
1•104
1•105
1•106
FREQUENCY (Hz)
FIGURE 22A. OPTO COUPLER GAIN
C19 » C20
(EQ. 28)
90
1
f z1 = ----------------------------------------2π • R21 • C19
R23 » R4
(EQ. 30)
From (Equation 26), it can be seen that the control to output
transfer function frequency dependence is a function of the
output load resistance, the value of output capacitor and
inductor, and the output capacitance ESR. These variations must
be considered when compensating the control loop. The worst
case small signal operating point for a voltage mode converter
tends to be at maximum Vin, maximum load, maximum COUT,
and minimum ESR.
23
PHASE (DEGREES)
1
f z2 ≈ ----------------------------------------2π • R23 • C22
(EQ. 29)
45
0
-45
-90
10
100
1•103
1•104
1•105
1•106
FREQUENCY (Hz)
FIGURE 22B. OPTO COUPLER
FN9111.6
December 2, 2011
ISL6740, ISL6741
The following compensation components were selected
R23 = 9.53kΩ
The gain and phase plots combined with the opto coupler’s
transfer characteristics appear in Figures 24A and 24B:
30
R24 = 2.49kΩ
R4 = 499Ω
20
GAIN (dB)
R21 = 4.22kΩ
C22 = 1nF
C20 = 82pF
C19 = 0.22μF
10
0
From (Equations 27, 28, 29 and 30), the poles and zeroes are:
fz1 = 171Hz
-10
10
100
fz2 = 16.7kHz
1•103
1•104
1•105
1•106
FREQUENCY (Hz)
fp2 = 460kHz
FIGURE 24A. EA PLUS OPTO COUPLER GAIN
fp3 = 319kHz
90
The calculated gain and phase plots of the error amplifier appear
below using an ideal op amp.
PHASE (DEGREES)
45
20
GAIN (dB)
10
0
-45
-90
-135
0
-180
10
100
1•103
1•104
FREQUENCY (Hz)
1•105
1•106
FIGURE 24B. EA PLUS OPTO COUPLER GAIN
-10
10
100
1•103
1•104
1•105
1•106
Using the control-to-output transfer function combined with the
EA transfer function, the loop gain and phase may be predicted.
The predicted loop gain and phase margin of the converter
appear in Figures 25A and 25B:
FREQUENCY (Hz)
FIGURE 23A. IDEAL ERROR AMPLIFIER GAIN
90
50
40
45
20
0
GAIN (dB)
PHASE (°)
30
-45
10
0
-10
-20
-30
-90
10
100
1•103
1•104
1•105
1•106
FREQUENCY (Hz)
FIGURE 23B. IDEAL ERROR AMPLIFIER PHASE
-40
-50
100
1•103
1•104
1•105
FREQUENCY (Hz)
FIGURE 25A. PREDICTED LOOP GAIN
24
FN9111.6
December 2, 2011
ISL6740, ISL6741
as the FET on resistance, copper losses, and inductor resistance.
The phase discrepancy above 60kHz is not particularly relevant
to the loop performance since it occurs well above the cross over
frequency. The predicted behavior indicates a much gentler drop
off of phase than was observed in the measured performance.
The discrepancy was not investigated.
225
PHASE MARGIN (°)
180
135
90
45
Performance
0
The major performance criteria for the converter are efficiency
and load regulation. These quantities are detailed in Figures 27
and 28.
-45
-90
-135
100
1•103
1•104
1•105
95
FREQUENCY (Hz)
93
The actual loop gain and phase margin measured on the
ISL6740EVAL2Z demonstration board appear in Figures 26A and
26B:
EFFICIENCY (%)
FIGURE 25B. PREDICTED LOOP PHASE MARGIN
91
89
87
50
40
85
30
2
3
4
GAIN (dB)
20
5
6
7
8
9
10
LOAD CURRENT (A)
10
FIGURE 27. EFFICIENCY vs LOAD VIN = 48Vt
0
-10
-20
12.015
-30
-50
0.1k
1k
10k
100k
FREQUENCY (Hz)
FIGURE 26A. MEASURED LOOP GAIN
225
PHASE MARGIN (°)
180
12.010
12.005
12.000
11.995
135
0
1
2
3
4
5
6
7
8
9
10
LOAD CURRENT (A)
90
FIGURE 28. LOAD REGULATION AT VIN = 48V
45
0
-45
-90
-135
0.1k
OUTPUT VOLTAGE (V)
-40
1k
10k
100k
FREQUENCY (Hz)
FIGURE 26B. MEASURE LOOP PHASE MARGIN
The efficiency, although very good, could be further improved
using a controlled SR method instead of using a self-driven
method with an auxiliary schottky diode. The schottky diode
conducts when the main switching FETs are off. Its forward
voltage drop is considerably larger than that of the SR FETs and
causes a measurable reduction in efficiency. The effect becomes
more significant as the input voltage is increased due to the
reduction of duty cycle (and consequent increase in the OFF
time).
The only major discrepancies between the predicted behavior
and the measured results are the Q of the L-C filter and the phase
behavior above 60kHz. The actual Q appears to be significantly
less than predicted resulting in less gain peaking and a less rapid
phase shift near the resonant frequency. This is most likely the
result of neglecting other losses in the converter’s output, such
25
FN9111.6
December 2, 2011
ISL6740, ISL6741
Component List
REFERENCE
DESIGNATOR
VALUE
Component List (Continued)
DESCRIPTION
C1
1.0μF
Capacitor, 1812, X7R, 100V, 20%
TDK C4532X7R2A105M
C2, C3
3.3μF
Capacitor, 1812, X5R, 50V, 20%
TDK C4532X5R1H335M
C4, C6
1.0μF
Capacitor, 0805, X5R, 16V, 10%
TDK C2012X5R1C105K
REFERENCE
DESIGNATOR
VALUE
DESCRIPTION
R8, R9, R10
18
Resistor, 2512, 5%
R11
205
Resistor, 0603, 1%
R12
8.06k
Resistor, 0603, 1%
R13
18.2k
Resistor, 0603, 1%
R14
Open
Resistor, 0603, Open
R15
1.27k
Resistor, 0603, 1%
R16, R19
1.00k
Resistor, 0603, 1%
R17
97.6k
Resistor, 0603, 1%
R18
3.01k
Resistor, 0603, 1%
C5, C15, C16
0.1μF
Capacitor, 0603, X7R, 50V, 10%
TDK C1608X7R1H104K
C7
Open
Capacitor, 0603, Open
C8, C21
22μF
Capacitor, 1812, X5R, 16V, 20%
TDK C4532X5R1C226M
C9
150μF
Capacitor, Radial, Sanyo 16SH150M
R20
2.00k
Resistor, 0603, 1%
C10, C14, C22
1000pF
Capacitor, 0603, X7R, 50V, 10%
TDK C1608X7R1H102K
R21
4.22k
Resistor, 0603, 1%
R23
9.53k
Resistor, 0603, 1%
C11, C12
560 pF
Capacitor, 0603, X7R, 100V, 10%
TDK C1608X7R2A561K
R24
2.49k
Resistor, 0603, 1%
C13
220pF
Capacitor, 0603, X7R, 100V, 10%
TDK C1608X7R2A221K
R26, R27
5.11
Resistor, 0805, 1%
RT1
10.0k
Resistor, 0603, 1%
C17
220pF
C18
0.047μF Capacitor, 0603, X7R, 16V, 10%
TDK C1608X7R1C473K
C19
0.22μF
C20
82pF
Capacitor, 0603, COG, 16V, 5%
TDK C1608COG1C221J
Capacitor, 0603, X7R, 16V, 10%
TDK C1608X7R1C224K
Capacitor, 0603, X7R, 16V, 10%
TDK C1608X7R1C820K
CR1, CR2
Diode, Schottky, BAT54S
CR3, CR5, CR6
Diode, Schottky, BAT54
CR4
Diode, Schottky, IR 12CWQ06FNPBF
D1
Zener, 10V, Philips BZX84-C10
D2
Zener, 6.8V, Philips BZX84-C6V8
L1
190nH
Pulse, P2004NL
L2
4.0μH
BI Technologies, HM65-H4R0LF
L3
Short
0 Ohm Jumper
Q5
Transistor, ONSemi MJD31CG
QL, QH, QR1,
QR2, QR3, QR4
FET, Fairchild FDS3672
R1
3.3
Resistor, 2512, 5%
R2
3.01k
Resistor, 2512, 2%
R3
10.0
Resistor, 0603, 1%
R4, R25
499
Resistor, 0603, 1%
R5
2.20
Resistor, 0805, 1%
R6
200
Resistor, 0603, 1%
R7
75.0k
Resistor, 0805, 1%
26
T1
Midcom 31660-LF1
T2
Pulse P8205NL
U1
Intersil HIP2101IBZ
U2
NEC PS2801-1-A
U3
ISL6740IBZ
U4
National LM431BIM3/NOPB
References
[1] Dixon, Lloyd H., “Closing the Feedback Loop”, Unitrode Power
Supply Design Seminar, SEM-700, 1990.
FN9111.6
December 2, 2011
ISL6740, ISL6741
Package Outline Drawing
M16.173
16 LEAD THIN SHRINK SMALL OUTLINE PACKAGE (TSSOP)
Rev 2, 5/10
A
1
3
5.00 ±0.10
SEE DETAIL "X"
9
16
6.40
PIN #1
I.D. MARK
4.40 ±0.10
2
3
0.20 C B A
1
8
B
0.65
0.09-0.20
END VIEW
TOP VIEW
1.00 REF
- 0.05
H
C
1.20 MAX
SEATING
PLANE
0.90 +0.15/-0.10
GAUGE
PLANE
0.25 +0.05/-0.06 5
0.10 M C B A
0.10 C
0°-8°
0.05 MIN
0.15 MAX
SIDE VIEW
0.25
0.60 ±0.15
DETAIL "X"
(1.45)
NOTES:
1. Dimension does not include mold flash, protrusions or gate burrs.
(5.65)
Mold flash, protrusions or gate burrs shall not exceed 0.15 per side.
2. Dimension does not include interlead flash or protrusion. Interlead
flash or protrusion shall not exceed 0.25 per side.
3. Dimensions are measured at datum plane H.
4. Dimensioning and tolerancing per ASME Y14.5M-1994.
5. Dimension does not include dambar protrusion. Allowable protrusion
shall be 0.08mm total in excess of dimension at maximum material
condition. Minimum space between protrusion and adjacent lead
(0.65 TYP)
(0.35 TYP)
TYPICAL RECOMMENDED LAND PATTERN
is 0.07mm.
6. Dimension in ( ) are for reference only.
7. Conforms to JEDEC MO-153.
27
FN9111.6
December 2, 2011
ISL6740, ISL6741
Small Outline Plastic Packages (SOIC)
M16.15 (JEDEC MS-012-AC ISSUE C)
N
INDEX
AREA
H
0.25(0.010) M
16 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE
B M
INCHES
E
-B1
2
3
L
SEATING PLANE
-A-
A
D
h x 45°
-C-
e
A1
B
0.25(0.010) M
C
0.10(0.004)
C A M
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.0532
0.0688
1.35
1.75
-
A1
0.0040
0.0098
0.10
0.25
-
B
0.013
0.020
0.33
0.51
9
C
0.0075
0.0098
0.19
0.25
-
D
0.3859
0.3937
9.80
10.00
3
E
0.1497
0.1574
3.80
4.00
4
e
α
B S
0.050 BSC
1.27 BSC
-
H
0.2284
0.2440
5.80
6.20
-
h
0.0099
0.0196
0.25
0.50
5
L
0.016
0.050
0.40
N
α
NOTES:
MILLIMETERS
16
0°
1.27
16
8°
0°
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95.
6
7
8°
Rev. 1 6/05
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above
the seating plane, shall not exceed a maximum value of 0.61mm (0.024
inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions are not
necessarily exact.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
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28
FN9111.6
December 2, 2011