LT1936 1.4A, 500kHz Step-Down Switching Regulator U FEATURES DESCRIPTIO ■ The LT®1936 is a current mode PWM step-down DC/DC converter with an internal 1.9A power switch, packaged in a tiny, thermally enhanced 8-lead MSOP. The wide input range of 3.6V to 36V makes the LT1936 suitable for regulating power from a wide variety of sources, including automotive batteries, 24V industrial supplies and unregulated wall adapters. Its high operating frequency allows the use of small, low cost inductors and ceramic capacitors, resulting in low, predictable output ripple. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Wide Input Range: 3.6V to 36V Short-Circuit Protected Over Full Input Range 1.9A Guaranteed Minimum Switch Current 5V at 1.4A from 10V to 36V Input 3.3V at 1.4A from 7V to 36V Input 5V at 1.2A from 6.3V to 36V Input 3.3V at 1.2A from 4.5V to 36V Input Output Adjustable Down to 1.20V 500kHz Fixed Frequency Operation Soft-Start Uses Small Ceramic Capacitors Internal or External Compensation Low Shutdown Current: <2µA Thermally Enhanced 8-Lead MSOP Package Cycle-by-cycle current limit, frequency foldback and thermal shutdown provide protection against shorted outputs, and soft-start eliminates input current surge during startup. Transient response can be optimized by using external compensation components, or board space can be minimized by using internal compensation. The low current (<2µA) shutdown mode enables easy power management in battery-powered systems. U APPLICATIO S ■ ■ Automotive Battery Regulation Industrial Control Supplies Unregulated Wall Adapters , LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. U ■ TYPICAL APPLICATIO 3.3V Step-Down Converter Efficiency 95 VIN 4.5V TO 36V 4.7µF BOOST SHDN SW 10µH LT1936 17.4k COMP VC VOUT 3.3V 1.2A FB GND VOUT = 5V 90 0.22µF 10k 22µF EFFICIENCY (%) ON OFF VIN VIN = 12V 85 VOUT = 3.3V 80 75 70 1936 TA01a 65 0 0.5 1 LOAD CURRENT (A) 1.5 1936 TA01b 1936fa 1 LT1936 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) VIN Voltage ............................................... – 0.4V to 36V BOOST Voltage ........................................................ 43V BOOST Above SW Voltage ....................................... 20V SHDN Voltage ........................................... – 0.4V to 36V FB, VC, COMP Voltage ............................................... 6V Operating Temperature Range (Note 2) LT1936E ............................................. – 40°C to 85°C LT1936I ............................................ – 40°C to 125°C LT1936H .......................................... – 40°C to 150°C Maximum Junction Temperature LT1936E, LT1936I ............................................ 125°C LT1936H ......................................................... 150°C Storage Temperature Range .................. –65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW BOOST VIN SW GND 1 2 3 4 9 8 7 6 5 LT1936EMS8E LT1936IMS8E LT1936HMS8E COMP VC FB SHDN MS8E PACKAGE 8-LEAD PLASTIC MSOP MS8E PART MARKING θJA = 40°C/W, θJC = 10°C/W EXPOSED PAD (PIN 9) IS GND MUST BE CONNECTED TO PCB LTBMT LTBRV LTBWB Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult factory for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2) PARAMETER CONDITIONS MIN Undervoltage Lockout TYP MAX UNITS 3.45 3.6 V Quiescent Current VFB = 1.5V 1.8 2.5 mA Quiescent Current in Shutdown VSHDN = 0V 0.1 2 µA 1.200 1.215 V 50 50 200 300 nA nA ● FB Voltage 1.175 ● ● FB Pin Bias Current (Note 4) VFB = 1.20V, E and I Grades H Grade FB Voltage Line Regulation VIN = 5V to 36V 0.01 %/V Error Amp gm VC = 0.5V, IVC = ±5µA 250 µS Error Amp Voltage Gain VC = 0.8V, 1.2V 150 VC Clamp 1.8 VC Switch Threshold 0.7 V Internal Compensation R 50 kΩ 150 pF Internal Compensation C VCOMP = 1V COMP Pin Leakage VCOMP = 1.8V, E and I Grades H Grade Switching Frequency VFB = 1.1V VFB = 0V 400 ● Maximum Duty Cycle Switch Current Limit Switch VCESAT ● ● ISW = 1.2A 500 40 ISW = 1.2A 1 2 µA µA 600 kHz kHz 87 92 1.9 2.2 2.6 A 410 520 mV 2 µA 2.2 V Switch Leakage Current Minimum BOOST Voltage Above SW V 2 % 1936fa 2 LT1936 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2) PARAMETER CONDITIONS TYP MAX BOOST Pin Current ISW = 1.2A MIN 28 50 mA BOOST Pin Leakage VSW = 0V 0.1 1 µA SHDN Input Voltage High UNITS 2.3 V SHDN Input Voltage Low SHDN Pin Current VSHDN = 2.3V (Note 5) VSHDN = 12V VSHDN = 0V Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: The LT1936E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT1936I specifications are guaranteed over the –40°C to 125°C temperature range. The LT1936H 34 140 0.01 0.3 V 50 240 0.1 µA µA µA specifications are guaranteed over the –40°C to 150°C temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime at junction temperatures greater than 125°C is derated to 1000 hours. Note 3: Current limit guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycle. Note 4: Current flows out of pin. Note 5: Current flows into pin. U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency, VOUT = 5V Switch Current Limit Efficiency, VOUT = 3.3V 3.0 100 100 2.5 EFFICIENCY (%) EFFICIENCY (%) VIN = 24V 80 70 VOUT = 5V TA = 25°C D1 = DFLS140L L1 = 15µH, TOKO D63CB 60 0 0.5 1.0 LOAD CURRENT (A) VIN = 5V 90 1.5 1936 G01 CURRENT LIMIT (A) VIN = 12V 90 VIN = 12V 80 VIN = 24V 70 VOUT = 3.3V TA = 25°C D1 = DFLS140L L1 = 10µH, TOKO D63CB TYP 2.0 MIN 1.5 1.0 0.5 0 60 0 0.5 1.0 LOAD CURRENT (A) 1.5 1936 G02 0 20 60 40 DUTY CYCLE (%) 80 100 1936 G03 1936fa 3 LT1936 U W TYPICAL PERFOR A CE CHARACTERISTICS Maximum Load Current 1.8 VOUT = 5V VOUT = 3.3V 1.6 LOAD CURRENT (A) LOAD CURRENT (A) 1.6 Switch Voltage Drop 600 SWITCH VOLTAGE DROP (mV) 1.8 Maximum Load Current L = 15µH 1.4 L = 10µH 1.2 L = 10µH 1.4 L = 6.8µH 1.2 1.0 0 5 10 15 20 INPUT VOLTAGE (V) 25 0 5 10 15 20 INPUT VOLTAGE (V) 1936 G04 25 3.2 1.185 3.0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 500 450 TA = 25°C 200 SHDN Pin Current 200 TA = 25°C DC = 30% TA = 25°C 2.5 SHDN PIN CURRENT (µA) 600 300 25 50 75 100 125 150 TEMPERATURE (°C) 1936 G09 Soft-Start 3.0 400 0 1936 G08 SWITCH CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) 550 400 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 1936 G07 Frequency Foldback 1.5 1936 G06 SWITCHING FREQUENCY (kHz) UVLO (V) FEEDBACK VOLTAGE (V) 3.4 1.190 500 0.5 1.0 SWITCH CURRENT (A) Switching Frequency 3.6 1.195 0 600 1.205 700 100 30 3.8 0 200 Undervoltage Lockout Feedback Voltage –50 –25 TA = –45°C 300 1936 G05 1.210 1.200 TA = 85°C TA = 25°C 400 0 1.0 30 500 2.0 1.5 1.0 150 100 50 0.5 100 0 0 0 1.0 0.5 FB PIN VOLTAGE (V) 1.5 1936 G10 0 0 1 2 3 SHDN PIN VOLTAGE (V) 4 1936 G11 0 12 4 8 SHDN PIN VOLTAGE (V) 16 1936 G12 1936fa 4 LT1936 U W TYPICAL PERFOR A CE CHARACTERISTICS Minimum Input Voltage 6.0 VOUT = 5V TA = 25°C L = 15µH VOUT = 3.3V TA = 25°C L = 10µH 5.5 INPUT VOLTAGE (V) 6 TO RUN TO START 5.0 4.5 4.0 5 TO RUN 3.5 3.0 1 100 10 LOAD CURRENT (mA) 1000 0 10 100 LOAD CURRENT (mA) 1000 1936 G13 VSW 10V/DIV IL 500mA/DIV IL 500mA/DIV VOUT 20mV/DIV VOUT 20mV/DIV 1936 G16 VIN = 12V VOUT = 3.3V IOUT = 50mA L = 10µH COUT = 22µF VC Voltages 1.0 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1936 G15 1936 G17 1µs/DIV Error Amp Output Current 60 2.5 TA = 25°C VC = 0.5V 40 2.0 CURRENT LIMIT CLAMP 1.5 1.0 SWITCHING THRESHOLD 0.5 0 –50 –25 1.5 Switching Waveforms, Discontinuous Mode VSW 10V/DIV 1µs/DIV 2.0 1936 G14 Switching Waveforms VIN = 12V VOUT = 3.3V IOUT = 1A L = 10µH COUT = 22µF 2.5 0.5 VC PIN CURRENT (µA) 4 VC VOLTAGE (V) INPUT VOLTAGE (V) 7 TO START Switch Current Limit 3.0 SWITCH CURRENT LIMIT (A) Minimum Input Voltage 8 20 0 –20 –40 –60 0 25 50 75 100 125 150 TEMPERATURE (°C) 1936 G18 0 1 FB PIN VOLTAGE (V) 2 1936 G19 1936fa 5 LT1936 U U U PI FU CTIO S BOOST (Pin 1): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. FB (Pin 6): The LT1936 regulates its feedback pin to 1.200V. Connect the feedback resistor divider tap to this pin. Set the output voltage according to VOUT = 1.200V (1 + R1/R2). A good value for R2 is 10k. VIN (Pin 2): The VIN pin supplies current to the LT1936’s internal regulator and to the internal power switch. This pin must be locally bypassed. SW (Pin 3): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. VC (Pin 7): The VC pin is used to compensate the LT1936 control loop by tying an external RC network from this pin to ground. The COMP pin provides access to an internal RC network that can be used instead of the external components. GND (Pin 4): Tie the GND pin to a local ground plane below the LT1936 and the circuit components. Return the feedback divider to this pin. COMP (Pin 8): To use the internal compensation network, tie the COMP pin to the VC pin. Otherwise, tie COMP to ground or leave it floating. Exposed Pad (Pin 9): The Exposed Pad must be soldered to the PCB and electrically connected to ground. Use a large ground plane and thermal vias to optimize thermal performance. SHDN (Pin 5): The SHDN pin is used to put the LT1936 in shutdown mode. Tie to ground to shut down the LT1936. Tie to 2.3V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. SHDN also provides a soft-start function; see the Applications Information. Do not drive SHDN more than 5V above VIN. W BLOCK DIAGRA 2 VIN VIN C2 INT REG AND UVLO ON OFF SLOPE COMP R3 5 Σ BOOST R Q S Q D2 1 SHDN C3 C4 DRIVER Q1 SW OSC L1 VOUT 3 D1 FREQUENCY FOLDBACK C1 R1 FB VC RC 50k 7 VC 8 COMP 6 gm R2 CC 150pF 1.200V 4 GND 1936 BD R4 C5 1936fa 6 LT1936 U OPERATIO (Refer to Block Diagram) The LT1936 is a constant frequency, current mode stepdown regulator. A 500kHz oscillator enables an RS flipflop, turning on the internal 1.9A power switch Q1. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC pin. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp (not shown) on the VC pin provides current limit. The VC pin is also clamped to the voltage on the SHDN pin; soft-start is implemented by generating a voltage ramp at the SHDN pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. This regulator includes an undervoltage lockout to prevent switching when VIN is less than ~3.45V. The SHDN pin is used to place the LT1936 in shutdown, disconnecting the output and reducing the input current to less than 2µA. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. The oscillator reduces the LT1936’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. 1936fa 7 LT1936 U W U U APPLICATIO S I FOR ATIO FB Resistor Network Inductor Selection and Maximum Output Current The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: A good first choice for the inductor value is ⎛V ⎞ R1 = R2⎜ OUT – 1⎟ ⎝ 1.200 ⎠ R2 should be 20k or less to avoid bias current errors. Reference designators refer to the Block Diagram. Input Voltage Range The input voltage range for LT1936 applications depends on the output voltage and the Absolute Maximum Ratings of the VIN and BOOST pins. The minimum input voltage is determined by either the LT1936’s minimum operating voltage of ~3.45V or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: VOUT + VD DC = VIN – VSW + VD where VD is the forward voltage drop of the catch diode (~0.5V) and VSW is the voltage drop of the internal switch (~0.5V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VD – VD + VSW DCMAX with DCMAX = 0.87. The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins and by the minimum duty cycle DCMIN = 0.08: VIN(MAX) = VOUT + VD – VD + VSW DCMIN Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the absolute maximum ratings of the VIN and BOOST pins. L = 2.2 (VOUT + VD) where VD is the voltage drop of the catch diode (~0.4V) and L is in µH. With this value the maximum output current will be above 1.2A at all duty cycles and greater than 1.4A for duty cycles less than 50% (VIN > 2 VOUT). The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be about 30% higher. For robust operation in fault conditions (start-up or short circuit) and high input voltage (>30V), the saturation current should be above 2.6A. To keep the efficiency high, the series resistance (DCR) should be less than 0.1Ω, and the core material should be intended for high frequency applications. Table 1 lists several vendors and suitable types. Table 1. Inductor Vendors VENDOR URL PART SERIES TYPE Murata www.murata.com LQH55D Open TDK www.component.tdk.com SLF7045 SLF10145 Shielded Shielded Toko www.toko.com D62CB D63CB D75C D75F Shielded Shielded Shielded Open Sumida www.sumida.com CR54 CDRH74 CDRH6D38 CR75 Open Shielded Shielded Open Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than 1.2A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. There are several graphs in the Typical Performance Characteristics section 1936fa 8 LT1936 U W U U APPLICATIO S I FOR ATIO of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode operation, which is okay but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), there is a minimum inductance required to avoid subharmonic oscillations. Choosing L greater than 1.6 (VOUT + VD) µH prevents subharmonic oscillations at all duty cycles. Catch Diode A 1A Schottky diode is recommended for the catch diode, D1. The diode must have a reverse voltage rating equal to or greater than the maximum input voltage. The ON Semiconductor MBRM140 is a good choice. It is rated for 1A DC at a case temperature of 110°C and 1.5A at a case temperature of 95°C. Diode Incorporated’s DFLS140L is rated for 1.1A average current; the DFLS240L is rated for 2A average current. The average diode current in an LT1936 application is approximately IOUT (1 – DC). Input Capacitor Bypass the input of the LT1936 circuit with a 4.7µF or higher value ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7µF ceramic is adequate to bypass the LT1936 and will easily handle the ripple current. However, if the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1936 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 4.7µF capacitor is capable of this task, but only if it is placed close to the LT1936 and the catch diode; see the PCB Layout section. A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT1936. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1936 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1936’s voltage rating. This situation is easily avoided; see the Hot Plugging Safety section. For space sensitive applications, a 2.2µF ceramic capacitor can be used for local bypassing of the LT1936 input. However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and may couple noise into other circuitry. Also, the increased voltage ripple will raise the minimum operating voltage of the LT1936 to ~3.7V. Output Capacitor The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT1936 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT1936’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good value is: COUT = 150 VOUT where COUT is in µF. Use X5R or X7R types. This choice will provide low output ripple and good transient response. Transient performance can be improved with a high value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used, but transient performance will suffer. With an external compensation network, the loop gain can be lowered to compensate for the lower capacitor value. When using the internal compensation network, the lowest value for stable operation is: COUT > 66 VOUT 1936fa 9 LT1936 U W U U APPLICATIO S I FOR ATIO Table 2. Capacitor Vendors Vendor Phone URL Part Series Comments Panasonic (714) 373-7366 www.panasonic.com Ceramic, Polymer, Tantalum EEF Series Kemet Sanyo Murata (864) 963-6300 (408) 749-9714 (404) 436-1300 AVX Taiyo Yuden (864) 963-6300 www.kemet.com Ceramic, Tantalum www.sanyovideo.com Ceramic, Polymer, Tantalum www.murata.com Ceramic www.avxcorp.com Ceramic, Tantalum www.taiyo-yuden.com This is the minimum output capacitance required, not the nominal capacitor value. For example, a 3.3V output requires 20µF of output capacitance. If a small 22µF, 6.3V ceramic capacitor is used, the circuit may be unstable because the effective capacitance is lower than the nominal capacitance when biased at 3.3V. Look carefully at the capacitor’s data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage rating, may be required. High performance electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. T494, T495 POSCAP TPS Series Ceramic parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR. An alternative to using external compensation components is to use the internal RC network by tying the COMP pin to the VC pin. This reduces component count but does not provide the optimum transient response when the output capacitor value is high, and the circuit may not be stable when the output capacitor value is low. If the internal compensation network is not used, tie COMP to ground or leave it floating. Loop compensation determines the stability and transient performance. Designing the compensation network is a LT1936 CURRENT MODE POWER STAGE gm = 2mho SW ERROR AMPLIFIER R1 – + Frequency compensation is provided by the components tied to the VC pin, as shown in Figure 1. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in CPL FB gm = 250µmho Frequency Compensation The LT1936 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT1936 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. OUTPUT ESR 1.25V C1 + 600k C1 50k VC CF RC 150pF COMP POLYMER OR TANTALUM GND CERAMIC R2 CC 1936 F01 Figure 1. Model for Loop Response 1936fa 10 LT1936 U W U U APPLICATIO S I FOR ATIO bit complicated and the best values depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 1 shows an equivalent circuit for the LT1936 control loop. The error amplifier is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that the output capacitor integrates this current, and that the capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 2 compares the transient response across several output capacitor choices and compensation schemes. In each case the load current is stepped from 200mA to 800mA and back to 200mA. COUT = 22µF (AVX 1210ZD226MAT) (2a) VOUT 100mV/DIV COMP VC COUT = 22µF ×2 (2b) VOUT 100mV/DIV COMP VC COUT = 150µF (4TPC150M) (2c) VOUT 100mV/DIV COMP VC COUT = 150µF (4TPC150M) (2d) VOUT 100mV/DIV COMP VC 220k 100pF IOUT 500mA/DIV 800mA 200mA 50µs/DIV 1936 F02 Figure 2. Transient Load Response of the LT1936 with Different Output Capacitors as the Load Current is Stepped from 200mA to 800mA. VOUT = 3.3V 1936fa 11 LT1936 U W U U APPLICATIO S I FOR ATIO BOOST Pin Considerations Capacitor C3 and diode D2 are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.22µF capacitor and fast switching diode (such as the 1N4148 or 1N914) will work well. Figure 3 shows two ways to arrange the boost circuit. The BOOST pin must be at least 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 3a) is best. For outputs between 2.8V and 3V, use a 0.47µF capacitor and a Schottky diode. For lower output voltages the boost diode can be tied to the input (Figure 3b), or to another supply greater than 2.8V. The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower voltage. You must also be sure that the maximum voltage rating of the BOOST pin is not exceeded. A 2.5V output presents a special case. This is a popular output voltage, and the advantage of connecting the boost circuit to the output is that the circuit will accept a 36V maximum input voltage rather than 20V (due to the BOOST pin rating). However, 2.5V is marginally adequate to support the boosted drive stage at low ambient temperatures. Therefore, special care and some restrictions on operation are necessary when powering the BOOST pin from a 2.5V output. Minimize the voltage loss in the boost circuit by using a 1µF boost capacitor and a good, low drop D2 C3 BOOST LT1936 VIN VIN VOUT SW GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT (3a) D2 C3 BOOST Schottky diode (such as the ON Semi MBR0540). Because the required boost voltage increases at low temperatures, the circuit will supply only 1A of output current when the ambient temperature is –45°C, increasing to 1.2A at 0°C. Also, the minimum input voltage to start the boost circuit is higher at low temperature. See the Typical Applications section for a 2.5V schematic and performance curves. The minimum operating voltage of an LT1936 application is limited by the undervoltage lockout (~3.45V) and by the maximum duty cycle as outlined above. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT1936 is turned on with its SHDN pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 4 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT1936, requiring a higher input voltage to maintain regulation. Soft-Start LT1936 VIN VIN SW VOUT GND 1933 F03 VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN (3b) Figure 3. Two Circuits for Generating the Boost Voltage The SHDN pin can be used to soft-start the LT1936, reducing the maximum input current during start-up. The SHDN pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 5 shows the start-up waveforms with and without the soft-start circuit. By choosing a large RC time constant, the peak start-up 1936fa 12 LT1936 U U W U APPLICATIO S I FOR ATIO Minimum Input Voltage VOUT = 5V Minimum Input Voltage VOUT = 3.3V 6.0 5.5 5.0 4.5 4.0 TO START 6 TO RUN 5 TO RUN 3.5 3.0 4 0 VOUT = 5V TA = 25°C L = 15µH 7 TO START INPUT VOLTAGE (V) INPUT VOLTAGE (V) 8 VOUT = 3.3V TA = 25°C L = 10µH 10 100 LOAD CURRENT (mA) 1000 1 100 10 LOAD CURRENT (mA) 1000 1936 G13 1936 G14 Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit RUN 5V/DIV RUN SHDN IIN 500mA/DIV GND VOUT 5V/DIV RUN 1936 F05a 0.5ms/DIV 1936 F05b RUN 5V/DIV 15k SHDN 0.22µF 50µs/DIV IIN 500mA/DIV GND VOUT 5V/DIV Figure 5. To Soft-Start the LT1936, Add a Resistor and Capacitor to the SHDN Pin. VIN = 12V, VOUT = 3.3V, COUT = 2 × 22µF, RLOAD = 3.3Ω current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 60µA when the SHDN pin reaches 2.3V. Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1936 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1936 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1936’s output. If the VIN pin is allowed to float and the SHDN pin is held high (either by a logic signal or because it is tied to VIN), then the LT1936’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground 1936fa 13 LT1936 U W U U APPLICATIO S I FOR ATIO the SHDN pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT1936 can pull large currents from the output through the SW pin and the VIN pin. Figure 6 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. IN MINIMIZE LT1936 C2, D1 LOOP C2 R4 D2 C3 R2 D1 D4 MBRS140 VIN GND R1 VIN BOOST L1 LT1936 SHDN C1 VOUT SW GND VC OUT COMP GND FB VIAS 1936 F07 BACKUP Figure 7. A Good PCB Layout Ensures Low EMI Operation 1936 F06 High Temperature Considerations Figure 6. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output; It Also Protects the Circuit from a Reversed Input. The LT1936 Runs Only When the Input is Present PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 7 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT1936’s VIN and SW pins, the catch diode (D1) and the input capacitor (C2). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT1936 to additional ground planes within the circuit board and on the bottom side. The die temperature of the LT1936 must be lower than the maximum rating of 125°C (150°C for the H grade). This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT1936. The maximum load current should be derated as the ambient temperature approaches 125°C (150°C for the H grade). The die temperature is calculated by multiplying the LT1936 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT1936 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The resulting temperature rise at full load is nearly independent of input voltage. Thermal resistance depends on the layout of the circuit board, but values from 40°C/W to 60°C/W are typical. Die temperature rise was measured on a 4-layer, 5cm × 6.5cm circuit board in still air at a load current of 1.4A. For 12V input to 3.3V output the die temperature elevation above ambient was 26°C; for 24V in to 3.3V out the rise was 31°C; for 12V in to 5V the rise was 31°C and for 24V in to 5V the rise was 34°C. 1936fa 14 LT1936 U W U U APPLICATIO S I FOR ATIO Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1936 circuits. However, these capacitors can cause problems if the LT1936 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor combined with stray inductance in series with the power source forms an under damped tank circuit, and the voltage at the VIN pin of the LT1936 can ring to twice the CLOSING SWITCH SIMULATES HOT PLUG IIN VIN nominal input voltage, possibly exceeding the LT1936’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1936 into an energized supply, the input network should be designed to prevent this overshoot. Figure 8 shows the waveforms that result when an LT1936 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The first plot is the response with a 4.7µF ceramic capacitor at the input. The input voltage rings as high as 50V and the input current peaks at 26A. One DANGER VIN 20V/DIV RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING OF THE LT1936 LT1936 + 4.7µF LOW IMPEDANCE ENERGIZED 24V SUPPLY IIN 10A/DIV STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR 20µs/DIV (8a) LT1936 + 22µF 35V AI.EI. VIN 20V/DIV + 4.7µF IIN 10A/DIV (8b) 0.7Ω LT1936 20µs/DIV VIN 20V/DIV + 0.1µF 4.7µF IIN 10A/DIV (8c) 20µs/DIV 1936 F08 Figure 8. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation When the LT1936 is Connected to a Live Supply 1936fa 15 LT1936 U W U U APPLICATIO S I FOR ATIO method of damping the tank circuit is to add another capacitor with a series resistor to the circuit. In Figure 8b an aluminum electrolytic capacitor has been added. This capacitor’s high equivalent series resistance damps the circuit and eliminates the voltage overshoot. The extra capacitor improves low frequency ripple filtering and can slightly improve the efficiency of the circuit, though it is likely to be the largest component in the circuit. An alternative solution is shown in Figure 8c. A 0.7Ω resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1µF capacitor improves high frequency filtering. This solution is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is minor, reducing efficiency by one percent for a 5V output at full load operating from 24V. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 100 shows how to generate a bipolar output supply using a buck regulator. U TYPICAL APPLICATIO S 3.3V Step-Down Converter D2 VIN 4.5V TO 36V ON OFF C1 4.7µF VIN BOOST SHDN SW C3 0.22µF D1 LT1936 COMP VC L1 10µH R1 17.4k VOUT 3.3V 1.2A FB GND R2 10k C2 47µF 1936 TA03 5V Step-Down Converter D2 VIN 6.3V TO 36V ON OFF C1 4.7µF VIN BOOST SHDN SW C3 0.22µF D1 LT1936 COMP VC L1 15µH R1 31.6k VOUT 5V 1.2A FB GND R2 10k C2 22µF 1936 TA04 1936fa 16 LT1936 U TYPICAL APPLICATIO S 1.8V Step-Down Converter Efficiency, 1.8V Output 90 D2 BOOST SHDN SW C1 4.7µF C3 0.22µF D1 LT1936 COMP L1 4.7µH R1 10k VOUT 1.8V 1.3A FB VC GND C2 47µF ×2 R2 20k 2.0 VOUT = 1.8V TA = 25°C VIN = 5V 80 1.5 VIN = 12V 70 1.0 60 POWER LOSS (W) ON OFF VIN EFFICIENCY (%) VIN 3.6V TO 20V 0.5 POWER LOSS D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 1936 TA05a 50 0 0 1.5 0.5 1 LOAD CURRENT (A) 1936 TA05b 1.2V Step-Down Converter Efficiency, 1.2V Output 80 D2 BOOST SHDN SW C1 4.7µF C3 0.22µF D1 LT1936 COMP VC L1 3.3µH 75 VOUT 1.2V 1.3A FB GND 100k C2 47µF ×2 VIN = 5V 1.5 VIN = 12V 1.0 70 65 60 0.5 55 D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 2.0 VOUT = 1.2V TA = 25°C POWER LOSS (W) ON OFF VIN EFFICIENCY (%) VIN 3.6V TO 20V POWER LOSS 1936 TA06a 50 0 0.5 1 LOAD CURRENT (A) 0 1.5 1936 TA06b 1936fa 17 LT1936 U TYPICAL APPLICATIO S 2.5V Step-Down Converter D2 VIN 3.6V TO 36V ON OFF VIN BOOST SHDN SW C1 4.7µF C3 1µF D1 LT1936 COMP VC L1 6.2µH FB GND R2 10k C2 47µF D1: DFLS140L D2: MBRO540 L1: TOKO D63CB 1936 TA07a Efficiency, 2.5V Output Minimum Input Voltage 5.5 100 VOUT 2.5V 1.2A TA > 0°C R1 11k VOUT = 2.5V VOUT = 2.5V TA = 25°C 5.0 INPUT VOLTAGE (V) EFFICIENCY (%) 90 VIN = 5V 80 VIN = 12V 70 TO START TA = –45°C 4.5 TO START TA = 25°C 4.0 TO RUN TA = –45°C 3.5 TO RUN TA = 25°C 3.0 60 0 0.5 1.0 LOAD CURRENT (A) 1.5 1936 TA07b 1 100 10 LOAD CURRENT (mA) 1000 1936 TA07c 1936fa 18 LT1936 U PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP (Reference LTC DWG # 05-08-1662) BOTTOM VIEW OF EXPOSED PAD OPTION 2.06 ± 0.102 (.081 ± .004) 1 5.23 (.206) MIN 1.83 ± 0.102 (.072 ± .004) 0.889 ± 0.127 (.035 ± .005) 2.794 ± 0.102 (.110 ± .004) 2.083 ± 0.102 3.20 – 3.45 (.082 ± .004) (.126 – .136) 8 0.42 ± 0.038 (.0165 ± .0015) TYP 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 0.65 (.0256) BSC 8 7 6 5 0.52 (.0205) REF RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 4.90 ± 0.152 (.193 ± .006) DETAIL “A” 0° – 6° TYP GAUGE PLANE 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 1 2 3 4 1.10 (.043) MAX 0.86 (.034) REF 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 0.65 (.0256) BSC 0.127 ± 0.076 (.005 ± .003) MSOP (MS8E) 0603 NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 1936fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT1936 U TYPICAL APPLICATIO 2.5V Step-Down Converter Minimum Input Voltage D2 5.5 ON OFF VIN BOOST SHDN SW C1 4.7µF C3 0.22µF D1 LT1936 COMP VC L1 8.2µH R1 11k FB GND R2 10k VOUT = 2.5V 5.0 VOUT 2.5V 1.3A C2 47µF INPUT VOLTAGE (V) VIN 3.6V TO 20V CONNECTING THE BOOST CIRCUIT TO THE INPUT LOWERS THE MINIMUM INPUT VOLTAGE TO RUN AND TO START TO LESS THAN 3.7V AT ALL LOADS 4.5 4.0 3.5 D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 1936 TA08a 3.0 1 100 10 LOAD CURRENT (mA) 1000 1936 TA08b RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1676 60V, 440mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 2.5µA, SO-8 Package LT1765 25V, 2.75A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD = 15µA, SO-8 and 16-Lead TSSOPE Packages LT1766 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 25µA, 16-Lead TSSOP/TSSOPE Packages LT1767 25V, 1.2A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD = 6µA, MS8/MS8E Packages LT1776 40V, 550mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.4V to 40V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 30µA, N8/SO-8 Packages LT1933 600mA, 500kHz, Step-Down Switching Regulator in SOT-23 VIN: 3.6V to 36V, VOUT(MIN) = 1.25V, IQ = 1.6mA, ISD < 1µA, ThinSOTTM Package LT1940 25V, Dual 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, ISD < 1µA, 16-Lead TSSOPE Package LT1956 60V, 1.2A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 25µA, 16-Lead TSSOP/TSSOPE Packages LT1976 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode® Operation VIN: 3.3V to 60V, VOUT(MIN) = 1.20V, IQ = 100µA, ISD < 1µA, 16-Lead TSSOPE Package LT3010 80V, 50mA, Low Noise Linear Regulator VIN: 1.5V to 80V, VOUT(MIN) = 1.28V, IQ = 30µA, ISD < 1µA, MS8E Package LTC®3407 Dual 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, ISD < 1µA, 10-Lead MSE Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, 16-Lead TSSOPE Package LTC3414 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter VIN: 2.3V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, ISD < 1µA, 20-Lead TSSOPE Package LT3430/LT3431 60V, 2.75A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 30µA, 16-Lead TSSOPE Package Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. 1936fa 20 Linear Technology Corporation LT/LT 0705 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005