L6911D 5 BIT PROGRAMMABLE STEP DOWN CONTROLLER WITH SYNCHRONOUS RECTIFICATION ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ OPERATING SUPPLY IC VOLTAGE FROM 5V TO 12V BUSES UP TO 1.3A GATE CURRENT CAPABILITY TTL-COMPATIBLE 5 BIT PROGRAMMABLE OUTPUT COMPLIANT WITH VRM 9.0 : 1.100V TO 1.850V WITH 0.025V BINARY STEPS VOLTAGE MODE PWM CONTROL EXCELLENT OUTPUT ACCURACY: ±1% OVER LINE AND TEMPERATURE VARIATIONS VERY FAST LOAD TRANSIENT RESPONSE: FROM 0% TO 100% DUTY CYCLE POWER GOOD OUTPUT VOLTAGE OVERVOLTAGE PROTECTION AND MONITOR OVERCURRENT PROTECTION REALIZED USING THE UPPER MOSFET'S RdsON 200KHz INTERNAL OSCILLATOR OSCILLATOR EXTERNALLY ADJUSTABLE FROM 50KHz TO 1MHz SOFT START AND INHIBIT FUNCTIONS APPLICATIONS ■ POWER SUPPLY FOR ADVANCED MICROPROCESSOR CORE ■ DISTRIBUTED POWER SUPPLY ■ HIGH POWER DC-DC REGULATORS SO-20 ORDERING NUMBERS: L6911D L6911DTR (Tape and Reel) DESCRIPTION The device is a power supply controller specifically designed to provide a high performance DC/DC conversion for high current microprocessors. A precise 5-bit digital to analog converter (DAC) allows adjusting the output voltage from 1.30V to 2.05V with 50mV binary steps and from 2.10V to 3.50V with 100mV binary steps. The high precision internal reference assures the selected output voltage to be within ±1%. The high peak current gate drive affords to have fast switching to the external power mos providing low switching losses. The device assures a fast protection against load overcurrent and load overvoltage. An external SCR is triggered to crowbar the input supply in case of hard over-voltage. An internal crowbar is also provided turning on the low side mosfet as long as the overvoltage is detected. In case of over-current detection, the soft start capacitor is discharged and the system works in HICCUP mode. BLOCK DIAGRAM Vcc 5 to 12V Vin 5V to12V VCC PGOOD OCSET BOOT SS MONITOR and PROTECTION UGATE OVP RT PHASE OSC LGATE VD0 VD1 VD2 VD3 VD4 PGND - D/A + + - D98IN957_2 November 2001 Vo 1.100V to 1.850V E/A GND PWM VSEN VFB COMP 1/17 L6911D ABSOLUTE MAXIMUM RATINGS Symbol Value Unit VCC to GND, PGND 15 V Boot Voltage 15 V 15 V -0.3 to Vcc+0.3 V 7 V 6.5 V Value Unit Thermal Resistance Junction to Ambient 110 °C/W Maximum junction temperature 150 °C VCC VBOOT-VPHASE Parameter VHGATE-VPHASE OCSET, LGATE, PHASE RT, SS, FB, PGOOD, VSEN, VID0-4 OVP, COMP THERMAL DATA Symbol Rth j-amb Tj Parameter Tstg Storage temperature range -40 to 150 °C TJ Junction temperature range 0 to 125 °C PIN CONNECTION (Top view) VSEN 1 20 RT OCSET 2 19 OVP SS/INH 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD 10 11 GND FB D98IN958 2/17 L6911D PIN FUNCTION Pin Num. Name 1 VSEN 2 OCSET A resistor connected from this pin and the upper Mos Drain sets the current limit protection. The internal 200µA current generator sinks a current from the drain through the external resistor. The Over-Current threshold is due to the following equation: I OCSE T ⋅ R O CS ET IP = ---------------------------------------------R DS on 3 SS/INH The soft start time is programmed connecting an external capacitor from this pin and GND. The internal current generator forces through the capacitor 10µA. This pin can be used to disable the device forcing a voltage lower than 0.4V 4-8 VID0 - 4 Voltage Identification Code pins. These input are internally pulled-up and TTL compatible. They are used to program the output voltage as specified in Table 1 and to set the overvoltage and power good thresholds. Connect to GND to program a ‘0’ while leave floating to program a ‘1’. 9 COMP This pin is connected to the error amplifier output and is used to compensate the voltage control feedback loop. 10 FB This pin is connected to the error amplifier inverting input and is used to compensate the voltage control feedback loop. 11 GND 12 PGOOD This pin is an open collector output and is pulled low if the output voltage is not within the above specified thresholds. If not used may be left floating. 13 PHASE This pin is connected to the source of the upper mosfet and provides the return path for the high side driver. This pin monitors the drop across the upper mosfet for the current limit 14 UGATE High side gate driver output. 15 BOOT Bootstrap capacitor pin. Through this pin is supplied the high side driver and the upper mosfet. Connect through a capacitor to the PHASE pin and through a diode to Vcc (cathode vs. boot). 16 PGND Power ground pin. This pin has to be connected closely to the low side mosfet source in order to reduce the noise injection into the device 17 LGATE This pin is the lower mosfet gate driver output 18 VCC Device supply voltage. The operative nominal supply voltage ranges from 5 to 12V. DO NOT CONNECT VIN TO A VOLTAGE GREATER THAN VCC. 19 OVP Over voltage protection. If the output voltage reaches the 17% above the programmed voltage this pin is driven high and can be used to drive an external SCR that crowbar the supply voltage. If not used, it may be left floating. 20 RT Description Connected to the output voltage is able to manage over-voltage conditions and the PGOOD signal. All the internal references are referred to this pin. Connect it to the PCB signal ground. Oscillator switching frequency pin. Connecting an external resistor from this pin to GND, the external frequency is increased according to the equation: 6 4.94 ⋅ 10 f S = 200kHz + ------------------------R T ( kΩ ) Connecting a resistor from this pin to Vcc (12V), the switching frequency is reduced according to the equation: 7 4.306 ⋅ 10 f S = 200kHz – ----------------------------RT ( kΩ ) If the pin is not connected, the switching frequency is 200KHz. The voltage at this pin is fixed at 1.23V (typ). Forcing a 50µA current into this pin, the built in oscillator stops to switch. 3/17 L6911D ELECTRICAL CHARACTERISTCS (VCC = 12V, Tamb = 25°C unless otherwise specified) Symbol Parameter Test Condition Min. Typ. Max. Unit VCC SUPPLY CURRENT Icc Vcc Supply current UGATE and LGATE open 5 mA POWER-ON Turn-On Vcc threshold VOCSET=4.5V Turn-Off Vcc threshold VOCSET=4.5V 4.6 3.6 Rising VOCSET threshold ISS Soft start Current V V 1.24 V 10 µA OSCILLATOR ∆Vosc Free running frequency RT = OPEN 180 Total Variation 6 KΩ < RT to GND < 200 KΩ -15 Ramp amplitude RT = OPEN 200 220 KHz 15 % 1.9 Vp-p REFERENCE AND DAC DACOUT Voltage Accuracy VID0, VID1, VID2, VID3, VID4 see Table1; Tamb = 0 to 70°C -1 VID Pull-Up voltage 1 % 4 V 88 dB ERROR AMPLIFIER DC Gain GBWP SR Gain-Bandwidth Product Slew-Rate COMP=10pF 10 MHz 10 V/µS 1.3 A GATE DRIVERS 1 IUGATE High Side Source Current VBOOT - VPHASE=12V, VUGATE - VPHASE= 6V RUGATE High Side Sink Resistance VBOOT-VPHASE=12V, IUGATE = 300mA ILGATE Low Side Source Current Vcc=12V, VLGATE = 6V RLGATE Low Side Sink Resistance Vcc=12V, ILGATE = 300mA 1.5 Output Driver Dead Time PHASE connected to GND 120 Over Voltage Trip (VSEN/DACOUT) VSEN Rising 117 120 % OCSET Current Source VOCSET = 4.5V 170 200 230 µA OVP Sourcing Current VSEN > OVP Trip, VOVP=0V 60 Upper Threshold (VSEN/DACOUT) VSEN Rising 110 112 114 % Lower Threshold (VSEN/DACOUT) VSEN Falling 86 88 90 % Hysteresis (VSEN/DACOUT) Upper and Lower threshold PGOOD Voltage Low IPGOOD = -5mA 2 0.9 4 1.1 Ω A 3 Ω ns PROTECTIONS IOCSET IOVP mA POWER GOOD VPGOOD 4/17 2 % 0.5 V L6911D Table 1. VID Settings VID4 VID3 VID2 VID1 VID0 Output Voltage (V) VID4 VID3 VID2 VID1 VID0 Output Voltage (V) 1 1 1 1 1 Output OFF 0 1 1 1 1 1.475 1 1 1 1 0 1.100 0 1 1 1 0 1.500 1 1 1 0 1 1.125 0 1 1 0 1 1.525 0 1 1 0 0 1.550 1 1 1 0 0 1.150 1 1 0 1 1 1.175 0 1 0 1 1 1.575 1 1 0 1 0 1.200 0 1 0 1 0 1.600 1 1 0 0 1 1.225 0 1 0 0 1 1.625 1 1 0 0 0 1.250 0 1 0 0 0 1.650 1 0 1 1 1 1.275 0 0 1 1 1 1.675 1 0 1 1 0 1.300 0 0 1 1 0 1.700 1 0 1 0 1 1.325 0 0 1 0 1 1.725 1 0 1 0 0 1.350 0 0 1 0 0 1.750 1 0 0 1 1 1.375 0 0 0 1 1 1.775 1 0 0 1 0 1.400 0 0 0 1 0 1.800 1 0 0 0 1 1.425 0 0 0 0 1 1.825 1 0 0 0 0 1.450 0 0 0 0 0 1.850 Device Description The device is an integrated circuit realized in BCD technology. It provides complete control logic and protections for a high performance step-down DC-DC converter optimized for microprocessor power supply. It is designed to drive N-Channel Mosfets in a synchronous-rectified buck topology. The device works properly with Vcc ranging from 5V to 12V and regulates the output voltage starting from a 1.26V power stage supply voltage (Vin). The output voltage of the converter can be precisely regulated, programming the VID pins, from 1.100V to 1.850V with 25mV binary steps, with a maximum tolerance of ±1% over temperature and line voltage variations. The device provides voltage-mode control with fast transient response. It includes a 200kHz free-running oscillator that is adjustable from 50kHz to 1MHz. The error amplifier features a 15MHz gain-bandwidth product and 10V/µs slew rate which permits high converter bandwidth for fast transient performance. The resulting PWM duty cycle ranges from 0% to 100%. The device protects against over-current conditions entering in HICCUP mode. The device monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. The device is available in SO20 package Oscillator The switching frequency is internally fixed to 200kHz. The internal oscillator generates the triangular waveform for the PWM charging and discharging with a constant current an internal capacitor. The current delivered to the oscillator is typically 50µA (Fsw=200KHz) and may be varied using an external resistor (R T) connected between RT pin and GND or VCC. Since the RT pin is maintained at fixed voltage (typ. 1.235V), the frequency is varied proportionally to the current sunk (forced) from (into) the pin. In particular connecting it to GND the frequency is increased (current is sunk from the pin), according to the following relationship: 6 4.94 ⋅ 10 f S = 200kH z + ------------------------R T ( kΩ ) Connecting RT to VCC=12V or to VCC=5V the frequency is reduced (current is forced into the pin), according to the following relationships: 5/17 L6911D 7 4.306 ⋅ 10 f S = 200kH z + ----------------------------R T ( kΩ ) VCC = 12V 7 15 ⋅ 10 f S = 200kH z + -------------------R T ( kΩ ) VCC = 5V Switching frequency variations vs. RT are reported in Fig.1. Note that forcing a 50µA current into this pin, the device stops switching because no current is delivered to the oscillator. Figure 1. 1 0 00 0 R e s is ta n ce [kO h m ] 1 00 0 10 0 10 R T to G N D R T to V C C =1 2 V R T to V C C =5 V 10 100 1000 F re q u e n cy [kH z] Digital to Analog Converter The built-in digital to analog converter allows the adjustment of the output voltage from 1.30V to 2.05V with 50mV binary steps and from 2.10V to 3.50V with 100mV binary steps as shown in the previous table 1. The internal reference is trimmed to ensure the precision of 1%. The internal reference voltage for the regulation is programmed by the voltage identification (VID) pins. These are TTL compatible inputs of an internal DAC that is realized by means of a series of resistors providing a partition of the internal voltage reference. The VID code drives a multiplexer that selects a voltage on a precise point of the divider. The DAC output is delivered to an amplifier obtaining the VPROG voltage reference (i.e. the set-point of the error amplifier). Internal pull-ups are provided (realized with a 5µA current generator); in this way, to program a logic "1" it is enough to leave the pin floating, while to program a logic "0" it is enough to short the pin to GND. The voltage identification (VID) pin configuration also sets the power-good thresholds (PGOOD) and the overvoltage protection (OVP) thresholds. The VID code "11111" disable the device (as a short on the SS pin) and no output voltage is regulated. Soft Start and Inhibit At start-up a ramp is generated charging the external capacitor CSS by means of a 10µA constant current, as shown in figure 1. When the voltage across the soft start capacitor (VSS) reaches 0.5V the lower power MOS is turned on to dis- 6/17 L6911D charge the output capacitor. As V SS reaches 1V (i.e. the oscillator triangular wave inferior limit) also the upper MOS begins to switch and the output voltage starts to increase. The VSS growing voltage initially clamps the output of the error amplifier, and consequently VOUT linearly increases, as shown in figure 2. In this phase the system works in open loop. When V SS is equal to VCOMP the clamp on the output of the error amplifier is released. In any case another clamp on the input of the error amplifier remains active, allowing to VOUT to grow with a lower slope (i.e. the slope of the VSS voltage, see figure 2). In this second phase the system works in closed loop with a growing reference. As the output voltage reaches the desired value VPROG, also the clamp on the error amplifier input is removed, and the soft start finishes. Vss increases until a maximum value of about 4V. The Soft-Start will not take place, and the relative pin is internally shorted to GND, if both VCC and OCSET pins are not above their own turn-on thresholds. During normal operation, if any under-voltage is detected on one of the two supplies, the SS pin is internally shorted to GND and so the SS capacitor is rapidly discharged. The device goes in INHIBIT state forcing SS pin below 0.4V. In this condition both external MOSFETS are kept off. Figure 2. Soft Start V cc Turn-on threshold V cc Vin V in Turn-on threshold 1V V ss to G ND 0.5V LGATE Vout Timing Diagram Aquisition: CH1 = PHASE; CH2 = V OUT; CH3 = PGOOD; CH4 = VSS Driver Section The driver capability on the high and low side drivers allows using different types of power MOS (also multiple MOS to reduce the RDSON), maintaining fast switching transition. The low-side mos driver is supplied directly by Vcc while the high-side driver is supplied by the BOOT pin. Adaptative dead time control is implemented to prevent cross-conduction and allow to use several kinds of mosfets. The upper mos turn-on is avoided if the lower gate is over about 200mV while the lower mos turn-on is avoided if the PHASE pin is over about 500mV. The upper mos is in any case turned-on after 200nS from the low side turn-off. The peak current is shown for both the upper (fig. 3) and the lower (fig. 4) driver at 5V and 12V. A 4nF capacitive load has been used in these measurements. For the lower driver, the source peak current is 1.1A @ Vcc=12V and 500mA @ Vcc=5V, and the sink peak current is 1.3A @ Vcc=12V and 500mA @ Vcc=5V. Similarly, for the upper driver, the source peak current is 1.3A @ Vboot-Vphase=12V and 600mA @ VbootVphase =5V, and the sink peak current is 1.3A @ Vboot-Vphase =12V and 550mA @ Vboot-Vphase = 5V. 7/17 L6911D Figure 3. High Side driver peak current. Vboot-Vphase=12V (left) Vboot-Vphase=5V (right) CH1 = High Side Gate CH4 = Gate Current Figure 4. Low Side driver peak current. Vcc=12V (left) Vcc=5V (right) CH1 = Low Side Gate CH4 = Gate Current Monitoring and Protections The output voltage is monitored by means of pin 1 (VSEN). If it is not within ±12% (typ.) of the programmed value, the powergood output is forced low. The device provides overvoltage protection, when the output voltage reaches a value 17% (typ.) grater than the nominal one. If the output voltage exceeds this threshold, the OVP pin is forced high, triggering an external SCR to shuts the supply (VIN) down, and also the lower driver is turned on as long as the over-voltage is detected. To perform the overcurrent protection the device compares the drop across the high side MOS, due to the RDSON, with the voltage across the external resistor (ROCS) connected between the OCSET pin and drain of the upper MOS. Thus the overcurrent threshold (I P) can be calculated with the following relationship: I OCS ⋅ R OCS I P = --------------------------------R DSON Where the typical value of IOCS is 200µA. To calculate the ROCS value it must be considered the maximum RDSON (also the variation with temperature) and the minimum value of IOCS. To avoid undesirable trigger of 8/17 L6911D overcurrent protection this relationship must be satisfied: ∆l I P ≥ I OUT MAX + ----- = I PEA K 2 Where ∆I is the inductance ripple current and IOUTMAX is the maximum output current. In case of output short circuit the soft start capacitor is discharged with constant current (10µA typ.) and when the SS pin reaches 0.5V the soft start phase is restarted. During the soft start the over-current protection is always active and if such kind of event occurs, the device turns off both mosfets, and the SS capacitor is discharged again (after reaching the upper threshold of about 4V). The system is now working in HICCUP mode, as shown in figure 5a. After removing the cause of the over-current, the device restart working normally without power supplies turn off and on. Figure 5. 9 L=1.5µH, Vin=12V Inductor Ripple [A] 8 7 L=2µH, Vin=12V 6 L=3µH, Vin=12V 5 4 L=1.5µH, Vin=5V 3 L=2µH, Vin=5V 2 L=3µH, Vin=5V 1 0 0.5 1.5 2.5 3.5 Output Voltage [V ] a: Hiccup Mode b: Inductor Ripple Current vs. Vout Inductor design The inductance value is defined by a compromise between the transient response time, the efficiency, the cost and the size. The inductor has to be calculated to sustain the output and the input voltage variation to maintain the ripple current ∆IL between 20% and 30% of the maximum output current. The inductance value can be calculated with this relationship: V IN – V OUT V OUT L = ------------------------------ ⋅ --------------V IN f S ⋅ ∆I L Where fSW is the switching frequency, VIN is the input voltage and VOUT is the output voltage. Figure 5b shows the ripple current vs. the output voltage for different values of the inductor, with VIN = 5V and VIN = 12V. Increasing the value of the inductance reduces the ripple current but, at the same time, reduces the converter response time to a load transient. If the compensation network is well designed, the device is able to open or close the duty cycle up to 100% or down to 0%. The response time is now the time required by the inductor to change its current from initial to final value. Since the inductor has not finished its charging time, the output current is supplied by the output capacitors. Minimizing the response time can minimize the output capacitance required. The response time to a load transient is different for the application or the removal of the load: if during the application of the load the inductor is charged by a voltage equal to the difference between the input and the output voltage, during the removal it is discharged only by the output voltage. The following expressions give approximate response time for ∆I load transient in case of enough fast compensation network response: 9/17 L6911D L ⋅ ∆I t a pplic atio n = -----------------------------V IN – V OUT L ⋅ ∆I t rem ov al = --------------V OUT The worst condition depends on the input voltage available and the output voltage selected. Anyway the worst case is the response time after removal of the load with the minimum output voltage programmed and the maximum input voltage available. Output Capacitor Since the microprocessors require a current variation beyond 10A doing load transients, with a slope in the range of tenth A/µsec, the output capacitor is a basic component for the fast response of the power supply. In fact for first few microseconds they supply the current to the load. The controller recognizes immediately the load transient and sets the duty cycle at 100%, but the current slope is limited by the inductor value. The output voltage has a first drop due to the current variation inside the capacitor (neglecting the effect of the ESL): ∆VOUT = ∆IOUT · ESR A minimum capacitor value is required to sustain the current during the load transient without discharge it. The voltage drop due to the output capacitor discharge is given by the following equation: 2 ∆I OUT L ∆V OUT = --------------------------------------------------------------------------------------------2 ⋅ C OUT ⋅ ( V INM IN ⋅ D M AX – V OUT ) Where DMAX is the maximum duty cycle value that is 100%. The lower is the ESR, the lower is the output drop during load transient and the lower is the output voltage static ripple. Input Capacitor The input capacitor has to sustain the ripple current produced during the on time of the upper MOS, so it must have a low ESR to minimize the losses. The rms value of this ripple is: I rm s = I OUT D ⋅ ( 1 – D ) Where D is the duty cycle. The equation reaches its maximum value with D=0.5. The losses in worst case are: 2 P = ESR ⋅ I rm s Compensation network design The control loop is a voltage mode (figure 7) that uses a droop function to satisfy the requirements for a VRM module, reducing the size and the cost of the output capacitor. This method "recovers" part of the drop due to the output capacitor ESR in the load transient, introducing a dependence of the output voltage on the load current: at light load the output voltage will be higher than the nominal level, while at high load the output voltage will be lower than the nominal value. 10/17 L6911D Figure 6. Output transient response without (a) and with (b) the droop function ESR DROP ESR DROP VMAX VDROOP VNOM VMIN (a) (b) As shown in figure 6, the ESR drop is present in any case, but using the droop function the total deviation of the output voltage is minimized. In practice the droop function introduces a static error (Vdroop in figure 6) proportional to the output current. Since a sense resistor is not present, the output DC current is measured by using the intrinsic resistance of the inductance (a few mΩ). So the low-pass filtered inductor voltage (that is the inductor current) is added to the feedback signal, implementing the droop function in a simple way. Referring to the schematic in figure 7, the static characteristic of the closed loop system is: R3 + R 8 // R9 R L ⋅ R8 // R9 V OUT = V PRO G + V PRO G ⋅ ------------------------------------- – ----------------------------------- ⋅ I OUT R8 R2 Where VPROG is the output voltage of the digital to analog converter (i.e. the set point) and R L is the inductance resistance. The second term of the equation allows a positive offset at zero load (∆V+); the third term introduces the droop effect (∆VDROOP). Note that the droop effect is equal the ESR drop if: R L ⋅ R 8 // R9 ----------------------------------- = ESR R8 Figure 7. Compensation network V V IN V COM P R L2 PHASE L V O UT PW M ESR R8 C18 Z C 6-15 F C 20 R4 R9 C 25 R3 V ZI PROG R2 Considering the previous relationships R2, R3, R8 and R9 may be determined in order to obtain the desired droop effect as follow: ■ Choose a value for R2 in the range of hundreds of KΩ to obtain realistic values for the other components. 11/17 L6911D ■ From the above equations, it results: + ∆V ⋅ R 2 R L ⋅ I M AX R8 = ----------------------- ⋅ --------------------------- ; V PRO G ∆V DROOP ∆V DROOP 1 R9 = R8 ⋅ --------------------------- ⋅ ------------------------------------- ; ∆V DROOP R L ⋅ I M AX 1 + --------------------------R L ⋅ I M AX Where IMAX is the maximum output current. The component R3 must be chosen in order to obtain R3<<R8//R9 to permit these and successive simplifications. ■ Therefore, with the droop function the output voltage decreases as the load current increases, so the DC output impedance is equal to a resistance ROUT. It is easy to verify that the output voltage deviation under load transient is minimum when the output impedance is constant with frequency. To choose the other components of the compensation network, the transfer function of the voltage loop is considered. To simplify the analysis is supposed that R3 << Rd, where Rd = (R8//R9). Figure 8. Compensation network definition |A v | 2 fLC fC E f2 f1 fEC fC C f |R | R0 fD f3 f |G lo o p | G0 fc f ConverterS ingularity fLC = 1 / 2π ⋅ LC fCE = 1 / 2π ⋅ ESR ⋅ C OUT = 1 / 2π ⋅ ESR ⋅ Cceramic f EC = 1 / 2π ⋅ Rceramic ⋅ Cceramic f CC ESRzero f1 = 1 / 2π ⋅ R 4 ⋅ C 20 f2 = 1 / 2π ⋅ ( R 3 + R 4 ) ⋅ C 20 Introduced by f3 = 1 / 2π ⋅ R 3 ⋅ C 25 CeramicCapacitor fd = 1 / 2π ⋅ Rd ⋅ C 25 doublepole Compensati onNetworkS ingularity The transfer function may be evaluated neglecting the connection of R8 to PHASE because, as will see later, this connection is important only at low frequencies. So R4 is considered connected to VOUT. Under this assumption, the voltage loop has the following transfer function: 12/17 L6911D ZC ( s ) Zf ( s ) Vin G loo p ( s ) = Av ( s ) ⋅ R ( s ) = Av ( s ) ⋅ -------------- Where Av ( s ) = ---------------- ⋅ ------------------------------------∆V o sc Z C ( s ) + Z L ( s ) Zi ( s ) Where ZC(s) and ZL(s) are the output capacitor and inductor impedance respectively. The expression of ZI(s) may be simplified as follow: 2 R3 1 1 Rd 1 + s ⋅ ( τ1 + τ d ) + s ⋅ -------- ⋅ τ 1 ⋅ τ d Rd ⋅ --- ⋅ C 25 R4 + --s- ⋅ C20 ⋅ R 3 Rd s Z I ( s ) = ---------------------------------- + ------------------------------------------------------ = --------------------------------------------------------------------------------------------------- = (1 + s ⋅ τ2 ) ⋅ (1 + s ⋅ τd ) 1 Rd + --- ⋅ C25 R 4 + 1 --- ⋅ C20 + R3 s s 1 + s R3 -------- ⋅ τ d ⋅ ( 1 + s ⋅ τ1 ) Rd = R d --------------------------------------------------------------------( 1 + s ⋅ τ2) ⋅ ( 1 + s ⋅ τd ) Where: τ1 = R4×C20, τ2 = (R4+R3)×C20 and τd = Rd×C25. The regulator transfer function became now: ( 1 + s ⋅ τ2 ) ⋅ ( 1 + s ⋅ τd ) R ( s ) ≈ -------------------------------------------------------------------------------------------------------R3 s ⋅ C 18 ⋅ R d ⋅ 1 + s -------- ⋅ τ d ⋅ ( 1 + s ⋅ τ1 ) Rd Figure 8 shows a method to select the regulator components (please note that the frequencies f EC and fCC corresponds to the singularities introduced by additional ceramic capacitors in parallel to the output main electrolytic capacitor). ■ To obtain a flat frequency response of the output impedance, the droop time constant τd has to be equal to the inductor time constant (see the note at the end of the section): L τ d = R d ⋅ C 25 = ------- = τ L RL ■ To obtain a constant -20dB/dec Gloop(s) shape the singularity f1 and f2 are placed in proximity of fCE and fLC respectively. This implies that: f LC f2 ---- = --------f CE f1 f1 = f CE ■ L ⇒ C25 = ----------------------( R L ⋅ Rd ) ⇒ ⇒ f LC R4 = R3 ⋅ --------- – 1 f CE C20 1 = --- ⋅ π ⋅ R 4 ⋅ f CE 2 To obtain a Gloop bandwidth of fC, results: G0 ⋅ f LC = 1 ⋅ fC ⇒ fC V IN C20 // C25 G 0 = A 0 ⋅ R 0 = ------------------ ⋅ ----------------------------- = -------C18 ∆Vosc fLC C20 ⋅ C25 f LC VIN ⇒ C 18 = ------------------ ⋅ ----------------------------- ⋅ -------∆Vosc C20 + C25 f C Note. To understand the reason of the previous assumption, the scheme in figure 9 must be considered. In this scheme, the inductor current has been substituted by the load current, because in the frequencies range of interest for the Droop function these current are substantially the same and it was supposed that the droop network don't represent a charge for the inductor. 13/17 L6911D Figure 9. Voltage regulation with droop function block scheme V com p V out A v(s) R (s ) It results: R OUT ⋅ Iout 1 + s ⋅τL 1 + s ⋅τd 1 + sτ L 1 + sτL G LO O P Vo Z OUT = ---------------- = R d ⋅ ------------------ ⋅ ----------------------------- = R OUT ⋅ -----------------I LO AD 1 + sτ d 1 + G L O O P 1 + sτd Because in the interested range |Gloop|>>1. To obtain a flat shape, the relationship considered will naturally follow. Application Idea: 1.100V to 1.850V / 25A Figure 10 shows an application schematic for a 1.100V to 1.850V conversion with 25A of current capability. Since the device's high gate drive, more than one mosfet for both high side and low side can be used: three STS11NF30L (30V, 9mW typ @ Vgs=10V) mosfet are suggested for high side while four of them are suggested as low side switch. Figure 10. Schematic Circuit L1 F1 +5 VIN C1-3 D1 C24 OVP 19 15 R10 VCC +12Vcc C21-22 C23 BOOT OCSET 18 R7 2 GND C17 11 UGATE R13 14 VID0 VID0 4 Q1,Q2, Q3 L2 PHASE VID1 VID1 13 VOUTCORE 5 U1 L6911D VID2 VID2 6 VID3 VID3 LGATE R14 17 Q4,Q5,Q6 7 D2 C4-9 R15 R6 PGND 16 Vss VID4 VID25mV 8 PGOOD R1 OSC 12 PWRGD 20 SS VSEN 3 1 9 C16 10 R8 VFB COMP C18 C19 R3 C20 R5 R2 14/17 R9 R4 C17 L6911D Part List Resistor R1 Not Mounted R2 470 SMD 0805 R3 1k SMD 0805 R4 82 SMD 0805 R5 Not Mounted SMD 0805 R6 1k SMD 0805 R7 1k SMD 0805 R8 13k SMD 0805 R9 100k SMD 0805 R10 Not Mounted SMD 0805 R12 20k SMD 0805 R13, R14 Short Circuit SMD 0805 1% SMD 0805 Capacitor C1-C3 680µF - 6.3V OSCON 6SP680M Radial 10x10.5 C4-C9 820µF - 4V 680µF - 6.3V OSCON 6SP680M OSCON 4SP820M Radial 10x10.5 Radial 10x10.5 C16 100n SMD 0805 C17 100n SMD 0805 C18 2.2n SMD 0805 C19 Not Mounted SMD 0805 C20 100n SMD 0805 C23 1n SMD 0805 C24 100n SMD 0805 C25 47n SMD 0805 Magnetics L1 1.5µH T44-52 Core, 7T-18AWG L2 1.8µH T50-52B Core, 7T-16AWG STS12NF30L STMicroelectronics SO8 D1 1N4148 STMicroelectronics SOT23 D2 STPS3L25U STMicroelectronics SMB L6911D STMicroelectronics SO20 251015A-15A Littlefuse AXIAL Transistors Q1-Q5 Diodes Ics U1 Fuse F1 15/17 L6911D mm inch OUTLINE AND MECHANICAL DATA DIM. MIN. TYP. MAX. MIN. TYP. MAX. A 2.35 2.65 0.093 0.104 A1 0.1 0.3 0.004 0.012 B 0.33 0.51 0.013 0.020 C 0.23 0.32 0.009 0.013 D 12.6 13 0.496 0.512 E 7.4 7.6 0.291 0.299 e 1.27 0.050 H 10 10.65 0.394 0.419 h 0.25 0.75 0.010 0.030 L 0.4 1.27 0.016 0.050 SO20 K 0˚ (min.)8˚ (max.) L h x 45˚ A B e A1 K H D 20 11 E 1 0 1 SO20MEC 16/17 C L6911D Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics 2001 STMicroelectronics - All Rights Reserved STMicroelectronics GROUP OF COMPANIES Australia - Brazil - China - Finland - France - Germany - Hong Kong - India - Italy - Japan - Malaysia - Malta - Morocco - Singapore - Spain - Sweden - Switzerland - United Kingdom - U.S.A. http://www.st.com 17/17