RT8810 - Richtek

RT8810
Dual-Phase Synchronous Buck PWM Controller
General Description
Features
The RT8810 is a dual phase synchronous buck controller
which can provide users with a compact, high efficient,
well protected and cost effective solution. The RT8810's
integrated high driving capability MOSFET drivers makes
it more attractive for high current application. The built-in
bootstrap diode simplifies the circuit design and reduces
external part count and PCB space. For output voltage
control, the RT8810 can precisely regulate feedback
voltage according to the internal reference voltage 0.6V or
external reference voltage from 0.4V to 2.5V.
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Single IC Supply Voltage : 4.5V to 13.2V
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Supports Manual / Auto Dynamic Phase Number
Control
Integrated Bootstrap Diode
Lossless RDS(ON) Current Sensing for Current Balance
Adjustable Operation Frequency : 100kHz to 1MHz
Adjustable Over Current Protection
Capacitor Programmable Soft-Start
Support 0% to 80% Duty Cycle
Selectable Internal/External VREF
Voltage Mode PWM Control with External
Feedback Loop Compensation
Phase Crosstalk Jitter Suspend (CJSTM)
Programmable Quick Response
Driver Shoot Through Protection
Supports Current Reporting
16-Lead WQFN and 24-Lead WQFN Packages
RoHS Compliant and Halogen Free
The MODE pin programs single phase or dual phase
operation, making the RT8810 suitable for dual power input
applications such as PCI-Express interface graphic cards.
To set RT8810 at automatic mode, the RT8810 operates
in single phase at light load condition and maintains high
efficiency over a wide range of output currents. In addition,
the RT8810 features adjustable gate driving voltage for
maximum efficiency and optimum performance.
The RT8810 adopts lossless RDS(ON) current sensing
technique for channel current balance and over current
protection. Other features include adjustable soft-start,
adjustable operation phase, and adjustable over current
threshold.
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Applications
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GPU Core Power
Desktop PC Memory, VTT Power
Low Output Voltage, High Power Density DC/DC
Converters
Voltage Regulator Modules
Ordering Information
RT8810
Package Type
QW : WQFN-16L 3x3 (W-Type)
QW : WQFN-24L 4x4 (W-Type)
Lead Plating System
G : Green (Halogen Free and Pb Free)
Z : ECO (Ecological Element with
Halogen Free and Pb free)
Product Classification
A : Only for WQFN-24L 4x4
B : Only for WQFN-16L 3x3
(With MODE Pin)
C : Only for WQFN-16L 3x3
(With REFIN Pin)
D : Only for WQFN-24L 4x4
DS8810-01
June 2011
Note :
Richtek products are :
`
RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.
`
Suitable for use in SnPb or Pb-free soldering processes.
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1
RT8810
Marking Information
RT8810AGQW
RT8810BGQW
RT8810CGQW
RT8810DGQW
EL=YM
DNN
JU=YM
DNN
JV=YM
DNN
02=YM
DNN
EL= : Product Code
JU= : Product Code
JV= : Product Code
02= : Product Code
YMDNN : Date Code
YMDNN : Date Code
YMDNN : Date Code
YMDNN : Date Code
RT8810AZQW
RT8810BZQW
RT8810CZQW
RT8810DZQW
EL YM
DNN
JU YM
DNN
JV YM
DNN
02 YM
DNN
EL : Product Code
JU : Product Code
JV : Product Code
02 : Product Code
YMDNN : Date Code
YMDNN : Date Code
YMDNN : Date Code
YMDNN : Date Code
Pin Configurations
LGATE1
PVCC9
VCC
LGATE2
PHASE1
LGATE1
PVCC9
PVCC
VCC
LGATE2
(TOP VIEW)
24 23 22 21 20 19
1
18
2
17
3
16
PGND
4
15
25
5
14
6
13
8
16 15 14 13
PHASE1
UGATE1
BOOT1
MODE
1
12
2
11
PGND
3
5
9 10 11 12
7
9
RT8810B
LGATE1
PVCC9
VCC
LGATE2
PHASE1
LGATE1
PVCC
PCVV9
VCC
LGATE2
RT8810A
24 23 22 21 20 19
12
2
11
PGND
3
10
17
4
7
8
IMAX
RT
COMP
FB
6
9
PHASE2
UGATE2
BOOT2
SS/EN
NC
PGND
UGATE1
BOOT1
AGND
REFIN
1
18
2
17
3
16
PGND
4
15
25
5
14
6
13
7
8
PHASE2
PGND
UGATE2
BOOT2
SS/EN
QR1
9 10 11 12
MODE
IMAX
RT
COMP
FB
QR2
1
PHASE2
UGATE2
BOOT2
SS/EN
8
WQFN-16L 3x3
5
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2
6
WQFN-24L 4x4
16 15 14 13
PHASE1
UGATE1
BOOT1
REFIN
10
17
4
MODE
IMAX
RT
COMP
FB
QR2
7
PHASE2
PGND
UGATE2
BOOT2
SS/EN
QR1
IMAX
RT
COMP
FB
NC
PGND
UGATE1
BOOT1
AGND
REFIN
WQFN-16L 3x3
WQFN-24L 4x4
RT8810C
RT8810D
DS8810-01
June 2011
RT8810
Typical Application Circuit
VIN
12V
R1
10
VCC
RBOOT1
0
CBOOT1
0.1µF
C1
1µF
PVCC9
C6
1µF
R2
0
C14
0.1µF
UGATE1
PHASE1
LGATE1
REFIN
RT
RRT
18k
CSS
0.1µF
SS/EN
C4
33pF
R6
20k
PHASE2
RUG2
0
LGATE2
C9
10µF x 4
/16V
C10
10µF x 5
Q3
L2
1µH
R12*
Q4
C18*
PGND
C11
820µF x 2
/2.5V
C12
10µF x 4
/16V
C13
NC
R7
1.5k
R9
NC
FB
COMP
C5
4.7nF
C17*
C8
820µF x 2
/2.5V
+
RIMAX
100k
R11*
Q2
RBOOT2
0
CBOOT2
0.1µF
UGATE2
IMAX
VOUT
1.1V
L1
1µH
BOOT2
MODE
RMODE
33k
RUG1
0
Q1
+
PVCC
C2
1µF
VREFIN
C7
10µF x 5
RT8810
BOOT1
AGND
QR2
QR1
R10 24k
C15
100pF
C16
NC
R8
1.8k
* : Option
Figure 1. RT8810A/D
DS8810-01
June 2011
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3
RT8810
VIN
12V
R1
10
VCC
RBOOT1
0
CBOOT1
0.1µF
C1
1µF
C6
1µF
R2
0
PVCC9
UGATE1
PHASE1
LGATE1
MODE
UGATE2
RT
CSS
0.1µF
C4
33pF
RBOOT2
0
CBOOT2
0.1µF
RUG2
0
R11*
C17*
C8
820µF x 2
/2.5V
C9
10µF x 4
/16V
C10
10µF x 5
Q3
Q4
L2
1µH
R12*
C18*
PGND
COMP
C5
4.7µF
Q2
SS/EN
LGATE2
VOUT
1.1V
L1
1µH
+
PHASE2
RRT
18k
Q1
BOOT2
IMAX
RIMAX
100k
RUG1
0
+
PVCC
C2
1µF
RMODE
33k
C7
10µF x 5
RT8810
BOOT1
C11
820µF x 2
/2.5V
C12
10µF x 4
/16V
C13
NC
R7
1.5k
R9
NC
FB
R8
1.8k
AGND
R6
20k
* : Option
Figure 2. RT8810B
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4
DS8810-01
June 2011
RT8810
VIN
12V
R1
10
VCC
RBOOT1
0
CBOOT1
0.1µF
C1
1µF
PVCC9
C6
1µF
R2
0
UGATE1
C14
0.1µF
LGATE1
REFIN
RT
CSS
0.1µF
SS/EN
C5
4.7nF
PHASE2
LGATE2
COMP
Q2
R11*
C17*
RBOOT2
0
CBOOT2
0.1µF
UGATE2
VOUT
1.1V
L1
1µH
RUG2
0
C8
820µF x 2
/2.5V
C9
10µF x 4
/16V
C10
10µF x 5
Q3
L2
1µH
+
RRT
18k
Q1
BOOT2
IMAX
RIMAX
100k
C4
33pF
PHASE1
RUG1
0
+
PVCC
C2
1µF
VREFIN
C7
10µF x 5
RT8810
BOOT1
Q4
R12*
C18*
PGND
C11
820µF x 2
/2.5V
C12
10µF x 4
/16V
C13
NC
R7
1.5k
R9
NC
FB
R8
1.8k
AGND
R6
20k
* : Option
Figure 3. RT8810C
DS8810-01
June 2011
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5
RT8810
Functional Pin Description
Pin No.
WQFN-16L
3x3
--
WQFN-24L
4x4
1
Pin Name
NC
2, 17,
17
25
PGND
(Exposed Pad)
(Exposed Pad)
2
3
UGATE1
3
4
BOOT1
--
5
AGND
4
(RT8810C)
6
REFIN
Pin Function
No Internal Connection.
Power Ground for the IC. These pins are ground returns for the
gate drivers. Tie these pins to the ground island/plane through the
lowest impedance connection available. The exposed pad must be
soldered to a large PCB and connected to PGND for maximum
power dissipation.
Upper Gate Driver Output for Channel 1. Connect this pin to the
gate of upper MOSFET. T his pin is monitored by the adaptive shoot
through protection circuitry to determine when the upper MOSFET
has turned off.
Bootstrap Supply for the Floating Upper Gate Driver of Channel 1.
Connect the bootstrap capacitor CBOOT 1 between BOOT1 pin and
the PHASE1 pin to form a bootstrap circuit. The bootstrap capacitor
provides the charge to turn on the upper MOSFET.
All voltages levels are measured with respect to this pin. Tie this pin
to the ground island/plane through the lowest impedance
connection available.
External Reference Input. This is the input pin for the external
reference voltage. If external reference voltage is not available,
leave this pin open for default internal 0.6V reference.
Operation Phase Control Input. Connect a resistor RMODE from this
pin to GND to set the threshold current level for single and dual
phase operations. The RT8810 operates in dual phase if the output
current is higher than the threshold current level; in single phase if
the output current is lower than the threshold current level; see the
related sections for detail. Tie this pin to GND for continuous single
phase operation. Leave this pin open for continuous dual phase
operation. Both upper and lower switches of PHASE2 are turned off
when operating in single phase.
4
(RT8810B)
7
MODE
5
8
IMAX
6
9
RT
7
10
COMP
8
11
FB
Feedback Voltage. This pin is the inverting input to the error
amplifier. A resistor divider from the output to GND is used to set the
regulation voltage.
--
12
QR2
Quick Response Setting Pin for Load Transition.
--
13
QR1
Quick Response Setting Pin for Load Transition.
9
14
SS/EN
Soft-Start Output. Connect a capacitor from this pin to GND to set
the soft-start interval. Pulling this pin low to 0.4V will shut down the
RT8810.
Output Current Indication. Connect this pin to ground with a resistor
to set the output over current protection level.
Operation Frequency Setting. Connect a resistor between this pin
and AGND to set the operation frequency.
Error Amplifier Output. This is the output of the Error Amplifier (EA)
and the non-inverting input of the PWM comparators. Use this pin in
combination with the FB pin to compensate the voltage-control
feedback loop of the converter
To be continued
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6
DS8810-01
June 2011
RT8810
Pin No.
WQFN-16L WQFN-24L
3x3
4x4
Pin Name
Pin Function
10
15
BOOT2
Bootstrap Supply for the Floating Upper Gate Driver of Channel 2.
Connect the bootstrap capacitor between BOOT2 pin and the PHASE2
pin to form a bootstrap circuit. The bootstrap capacitor provides the
charge to turn on the upper MOSFET.
11
16
UGATE2
Upper Gate Driver Output for Channel 2. Connect this pin to the gate of
upper MOSFET. This pin is monitored by the adaptive shoot through
protection circuitry to determine when the upper MOSFET has turned off.
12
18
PHASE2
Switch Node for Channel 2. Connect this pin to the source of the upper
MOSFET and the drain of the lower MOSFET. This pin is used as the sink
for the UGATE2 driver. This pin is also monitored by the adaptive shoot
through protection circuitry to determine when the upper MOSFET has
turned off.
13
19
LGATE2
Lower Gate Driver Output for Channel 2. Connect this pin to the gate of
lower MOSFET. This pin is monitored by the adaptive shoot through
protection circuitry to determine when the lower MOSFET has turn off.
VCC
Supply Voltage. This pin is the input pin of the internal 9V LDO, which
provides current for PVCC9 and PVCC pins. Place a minimum 1μF
ceramic capacitor physically near the pin to locally bypass the supply
voltage.
PVCC
Supply Input. This pin receives a supply voltage from 4.5V to 13.2V and
provides bias current for the internal control circuit. Physically place a
minimum 1μF ceramic capacitor near it. This pin to bypass it.
PVCC9
Supply Input. This pin is the output of the internal 9V LDO regulator. It
provides current for lower gate drivers and bootstrap current for upper
drivers.
LGATE1
Lower Gate Driver Output for Channel 1. Connect this pin to the gate of
lower MOSFET. This pin is monitored by the adaptive shoot through
protection circuitry to determine when the lower MOSFET has turn off.
PHASE1
Switch Node for Channel 1. Connect this pin to the source of the upper
MOSFET and the drain of the lower MOSFET. This pin is used as the sink
for the UGATE driver. This pin is also monitored by the adaptive shoot
through protection circuitry to determine when the upper MOSFET has
turned off.
14
--
15
16
1
DS8810-01
20
21
(RT8810A)
22
(RT8810D)
22
(RT8810A)
21
(RT8810D)
23
24
June 2011
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7
RT8810
Function Block Diagram
REFIN
REF
SEL
PVCC
VCC
Bias
LV
Regulator
HV
Regulator
-
SS/EN
POR
Soft-Start
+
+
FB
COMP
BOOT2
-
PVCC9
SD
-
UGATE2
Fault
Logic
+
Gate
Control
PHASE2
+
+
-
-
BOOT1
OC
UGATE1
Gate
Control
PHASE1
LGATE1
LGATE2
+
-
S/H
Current
Balance
S/H
PGND
-
VB
AGND
-
+
VB
Phase
Control
+
Oscillator
+
Transient
Response
Enhancement
MODE
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8
RT
QR1
QR2
IMAX
DS8810-01
June 2011
RT8810
Absolute Maximum Ratings
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(Note 1)
VCC, PVCC, PVCC9 to AGND ---------------------------------------------------------------BOOTx to PHASEx -----------------------------------------------------------------------------PHASEx to PGNDx
DC ---------------------------------------------------------------------------------------------------<20ns ----------------------------------------------------------------------------------------------UGATEx to PHASEx
DC ---------------------------------------------------------------------------------------------------<20ns ----------------------------------------------------------------------------------------------LGATEx to PGNDx
DC ---------------------------------------------------------------------------------------------------<20ns ----------------------------------------------------------------------------------------------Input, Output or I/O Voltage -------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C
WQFN-16L 3x3 ----------------------------------------------------------------------------------WQFN-24L 4x4 ----------------------------------------------------------------------------------Package Thermal Resistance (Note 2)
WQFN-16L 3x3, θJA -----------------------------------------------------------------------------WQFN-16L 3x3, θJC ----------------------------------------------------------------------------WQFN-24L 4x4, θJA -----------------------------------------------------------------------------WQFN-24L 4x4, θJC ----------------------------------------------------------------------------Junction Temperature ---------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) -----------------------------------------------------Storage Temperature Range ------------------------------------------------------------------ESD Susceptibility (Note 3)
HBM (Human Body Mode) --------------------------------------------------------------------MM (Machine Mode) -----------------------------------------------------------------------------
Recommended Operating Conditions
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15V
15V
−0.5V to 15V
−5V to 25V
−0.3V to (BOOTx − PHASEx + 0.3V)
−5V to (BOOTx − PHASEx + 5V)
−0.3V to (PVCC9 + 0.3V)
−5V to (PVCC9 + 5V)
(AGND − 0.3V) to 6V
1.471W
1.923W
68°C/W
7.5°C/W
52°C/W
7°C/W
150°C
260°C
−65°C to 150°C
2kV
200V
(Note 4)
Supply Voltage, VCC ----------------------------------------------------------------------------- 4.5V to 13.2V
Junction Temperature Range ------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range ------------------------------------------------------------------- −40°C to 85°C
DS8810-01
June 2011
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9
RT8810
Electrical Characteristics
(VCC = 12V, VPVCC9 = 9V, TA = 25°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Input
Bias Voltage
VPVCC
4.5
--
13.2
V
Regulated Bias Voltage
VPVCC9
8
9
10
V
Supply Current
I CC
UGATE, LGATE Open
--
6.5
--
mA
Shutdown Current
I SHDN
UGATE, LGATE Open
--
4
--
mA
3.8
4.1
4.4
V
--
0.3
--
V
175
200
225
kHz
100
--
1000
kHz
--
2
--
VP-P
Minimum Duty Cycle
0
--
--
%
Minimum LGATE Pulse
--
300
--
ns
0.59
0.6
0.61
V
Power-On Reset
VCC POR Threshold
VPVCC9R_th VCC9 Rising
Power On Reset Hysteresis
VPVCC9_hys
Oscillator
Frequency
f OSC
RRT = 30kΩ
Frequency Range
Ramp Amplitude
ΔVOSC
Reference
Nominal Feedback Voltage
VFB
Error Amplifier
Open Loop DC Gain
ADC
Guaranteed by Design
--
70
--
dB
Gain Bandwidth
GBW
Guaranteed by Design
--
10
--
MHz
Slew Rate
SR
Guaranteed by Design, CL = 10pF
--
6
--
V/μs
Transconductance
Maximum Current (Source &
Sink)
Soft-Start
gm
--
1.8
--
mA/V
I COMPsk
--
360
--
μA
SS Source Current
I SS
7
10
13
μA
--
0.5
--
V
145
165
185
μA/V
Re-Soft-Start Threshold
Level
VSS/EN = 0V
Current Sense
Current Sense Gain
Mode Pin Voltage
VMODE
--
0.6
--
V
Forced Single Phase
Operation
I MODE
250
--
--
μA
Forced Dual Phase
Operation
I MODE
--
--
1
μA
To be continued
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10
DS8810-01
June 2011
RT8810
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
PWM Controller Gate Driver
Upper Gate Sourcing Ability
IUGATEsr
VBOOTx − VPHASEx = 12V, Max.
Source Current
--
1.5
--
A
Upper Gate RDS(ON) Sinking
RUGATEsk
VUGATEx − VPHASEx = 0.1V
--
2
--
Ω
Lower Gate Sourcing Ability
ILGATEsr
VCC= 12V, Max. Source Current
--
1.5
--
A
Lower Gate RDS(ON) Sinking
RLGATEsk
VLGATEx = 0.1V
--
2
--
Ω
VUGATEx − VPHASEx = 1.2V to
VLGATEx = 1.2V
--
30
--
ns
Deadtime
Protection
Over Current Threshold
VIMAX
2.75
3
3.25
V
SS Enable Threshold
VEN
0.3
0.4
0.5
V
Note 1. Stresses listed as the above "Absolute Maximum Ratings" may cause permanent damage to the device. These are for
stress ratings. Functional operation of the device at these or any other conditions beyond those indicated in the
operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended
periods may remain possibility to affect device reliability.
Note 2. θJA is measured in natural convection at TA = 25°C on a high effective thermal conductivity four-layer test board of
JEDEC 51-7 thermal measurement standard. The measurement case position of θJC is on the exposed pad of the
packages.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
DS8810-01
June 2011
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11
RT8810
Typical Operating Characteristics
VREF vs. Temperature
Efficiency vs. Load Current
100
0.610
Phase 2 Active
95
0.608
90
0.606
0.604
VREF (V)
Efficiency (%)
85
80
75
70
0.602
0.600
0.598
0.596
65
0.594
60
VIN = VCC = 12V, VOUT = 1.1V
55
0.592
VIN = VCC = 12V, No Load
0.590
0
10
20
30
40
50
60
-50
-25
0
25
Frequency vs. Temperature
100
125
RRT vs. Frequency
315
650
VIN = VCC = 12V, No Load
600
310
550
Frequency (kHz)1
Frequency (kHz)1
75
Temperature (°C)
Load Current (A)
305
300
295
500
450
400
350
300
250
290
VIN = VCC = 12V, No Load
285
200
150
-50
Inductor Current (A)
50
32
30
28
26
24
22
20
18
16
14
12
10
8
6
4
2
0
-25
0
25
50
75
100
125
5
10
15
20
25
30
Temperature (°C)
RRT (k
)
(kΩ)
Inductor Current vs. Output Current
Power On from EN
35
40
SS/EN
(1V/Div)
Phase1
VOUT
(1V/Div)
UGATE1
(20V/Div)
Phase2
UGATE2
(20V/Div)
VIN = VCC = 12V
5
10
15
20
25
30
35
40
Output Current (A)
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12
45
50
55
60
VIN = VCC = 12V, IOUT = 40A
Time (4ms/Div)
DS8810-01
June 2011
RT8810
Power Off from EN
Power On from VCC
SS/EN
(1V/Div)
VCC
(10V/Div)
VOUT
(1V/Div)
UGATE1
(20V/Div)
VOUT
(1V/Div)
UGATE1
(20V/Div)
UGATE2
(20V/Div)
VIN = VCC = 12V, IOUT = 40A
UGATE2
(20V/Div)
VIN = VCC = 12V, IOUT = 40A
Time (200μs/Div)
Time (4ms/Div)
Power Off from VCC
Dynamic Output Voltage Control
VCC
(10V/Div)
V REFIN
(1V/Div)
VOUT
(1V/Div)
UGATE1
(20V/Div)
VOUT
(1V/Div)
UGATE1
(20V/Div)
UGATE2
(20V/Div)
VIN = VCC = 12V, IOUT = 40A
UGATE2
(20V/Div)
VIN = VCC = 12V, IOUT = 20A, VREFIN = 0V to 1.1V
Time (20ms/Div)
Time (400μs/Div)
Dynamic Output Voltage Control
Load Transient Response
VIN = VCC = 12V, IOUT = 0A to 40A
V REFIN
(1V/Div)
UGATE1
(20V/Div)
VOUT
(1V/Div)
UGATE1
(20V/Div)
UGATE2
(20V/Div)
UGATE2
(20V/Div)
VIN = VCC = 12V, IOUT = 20A, VREFIN = 1.1V to 0V
Time (400μs/Div)
DS8810-01
June 2011
IOUT
(50A/Div)
VOUT
(50mV/Div)
Time (10μs/Div)
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13
RT8810
Mode Transition
Load Transient Response
VIN = VCC = 12V, IOUT = 40A to 0A
UGATE1
(20V/Div)
UGATE1
(20V/Div)
UGATE2
(20V/Div)
UGATE2
(20V/Div)
IOUT
(50A/Div)
VOUT
(50mV/Div)
VOUT
(20mV/Div)
VIN = VCC = 12V, single to dual phase
Time (10μs/Div)
Time (10μs/Div)
Mode Transition
Over Current Protection
UGATE1
(20V/Div)
VOUT
(500mV/Div)
UGATE2
(20V/Div)
IL1
(10A/Div)
VOUT
(20mV/Div)
VIN = VCC = 12V, dual to single phase
Time (10μs/Div)
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14
IL2
(10A/Div)
VIN = VCC = 12V
Time (10ms/Div)
DS8810-01
June 2011
RT8810
Application Information
Frequency vs. RRT
Dual Supply Voltage (VCC, PVCC) with Internal
Regulator
The Power-On-Reset (POR) circuit monitors the supply
voltage at the PVCC pin. If PVCC exceeds the POR rising
threshold voltage, the controller is reset and prepares the
PWM for operation. If PVCC falls below the POR falling
threshold during normal operation, all MOSFETs stop
switching. The POR rising and falling threshold has a
hysteresis to prevent noise caused reset.
Soft-Start
The RT8810 provides external soft-start function to prevent
large inrush current and output voltage overshoot when
the converter starts up. The soft-start begins when OCP
programming is complete.
During soft-start, an internal current source (10μA) is used
to charge the external soft-start capacitor at the SS/EN
pin. VSS/EN rises up, and the PWM logic and gate drives
become enabled. When the feedback voltage crosses
0.6V, the internal 0.6V reference takes over the behavior
of the error operational transconductance amplifier and softstart is complete. The RT8810 turns off the internal 10μA
current source when soft-start is complete.
600
Frequency (kHz)1
The RT8810 requires an external bias supply for PVCC
and VCC. PVCC receives a supply voltage from 4.5V to
13.2V and provides bias current for internal control circuit.
VCC is the input pin of the internal 9V LDO which provides
current for the PVCC9 pin. PVCC9 is the output pin of the
internal 9V LDO regulator. It provides current for lower
gate drivers and bootstrap current for upper drivers.
Physically place a minimum 1μF ceramic capacitor near
PVCC and VCC to locally bypass the supply voltage.
700
500
400
300
200
100
5
10
15
20
25
30
35
RRT (kΩ)
Figure 4. RRT vs. Switching Frequency
A resistor of 8.6kΩ to 18kΩ corresponds to a switching
frequency of 500kHz to 300kHz, respectively.
External Reference Input
The RT8810 supports external reference input to provide
more flexible applications. The REFIN pin is implemented
to be the external reference input. The mode selection is
determined and latched after POR. If REFIN pin is floating,
a 10μA current source will pull high the REFIN pin and if
the pin voltage exceeds 2.8V, the FB pin will follow the
internal reference voltage 0.6V. On the other hand, if an
external voltage is applied to the REFIN pin, the RT8810
enters tracking mode and regulates FB to be close to this
voltage. The applied voltage must be within the tracking
range (typically between 0.4V to 2.5V).
If the applied voltage is less than 0.3V, the controller will
be shut down.
Switching Frequency
Current Sensing and Reporting
High frequency operation optimizes the application by
allowing smaller component size, but trades off efficiency
due to higher switching losses. Low frequency operation
offers the best overall efficiency, but at the expense of
component size and board space.
The RT8810 monitors per phase current for current balance
and over current protection. Per phase current is sensed
by the on-resistance of low side MOSFET when turned
on. The GM amplifier senses the voltage drop across the
lower switch and converts it into a current signal each
time it turns on. The sensed current is expressed as :
Connect a resistor (RRT) between RT and ground to set
the switching frequency (fSW) per phase. Users can refer
to Figure 4 for switching frequency setting.
DS8810-01
June 2011
ICS = 3.3 x IL x RDS(ON) x 10 −4 + 5.5μA
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15
RT8810
where IL is the per phase current in Ampere, RDS(ON) is the
on-resistance of low side MOSFET in mΩ, and 5.5μA is a
constant to compensate the offset of the current sensing
circuit. Note that the valley inductor current is sampled
and held. The sampled and hold current is the averaged
inductor current minus half of inductor ripple current :
IL_SH = IL_AVG − ⎛⎜ 1 x ΔIL ⎞⎟
⎝2
⎠
where ΔIL is the inductor ripple current
One half of the summation of the sampled and hold current
signal (ICS1 + ICS2) / 2 is injected to the IMAX pin, that
results in a voltage VIMAX across the resistor RIMAX
connecting IMAX and AGND for over current protection.
And VIMAX is equal to
VIMAX =
ICS1 + ICS2
x RIMAX
2
⎡ 3.3 x (IL1_SH + IL2_SH ) xRDS(ON) x10 −4 + 11μA ⎤
⎥
= ⎢
2
⎢
⎥
⎣
⎦
Therefore, IMAX pin could be used for current reporting.
Over Current Protection
The RT8810 features over current protection. The voltage
at the IMAX pin (VIMAX) is compared with a 3.0V reference
voltage. If VIMAX is higher than 3.0V, OCP is triggered.
The over current setting resistor (RIMAX) value for dual phase
threshold can be calculated according to
3V
x RDS(ON) x 10−4 + 5.5μA
And the RIMAX value for single phase threshold will be
RIMAX
=
3V
1.5 x ⎡1.65x (IO_MAX − ΔIL ) x RDS(ON) x10−4 + 2.75μA ⎤
⎣
⎦
The RT8810 features hiccup and shutdown mode OCP. If
OCP is triggered after soft-start ends, the RT8810 turns
off both upper and lower MOSFETs and discharges CSS
with a constant current of 10μA. When VSS exceeds 0.5V,
the RT8810 initiates another soft-start cycle. The RT8810
shuts down after 3 hiccups. If the OCP is triggered during
soft-start cycle, the RT8810 turns off both upper and lower
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16
Current Balance
The RT8810 senses each phase current from low side
MOSFET RDS(ON), and fine tunes the duty cycle of each
phase for current balance as shown in Figure 5. If the
current of PHASE1 is smaller than the current of PHASE2,
the RT8810 increases the duty cycle of the corresponding
phase to increase its phase current accordingly.
PWM1
+
-
PWM2
VCOMP
+
Ramp1
+
+
+
ICS1
ICS2
+
-
Ramp2
+
+
-
Figure 5. Current Balance Control Circuit
Dynamic Phase Number Control
The RT8810 adaptively controls the operation phase
number according to the load current. Figure 6 shows the
dynamic phase number control circuit. The phase adding
and dropping threshold can be set by a resistor, RMODE,
which is connected from the MODE pin to AGND. A
current, IMODE, flows through the resistor, RMODE, as
IMODE =
RIMAX =
1.65 x (IO_MAX − ΔIL )
MOSFETs but continues to charge CSS with a constant
current of 10μA until soft-start ends. The shutdown status
can only be reset by the POR function.
0.6
RMODE
Once IIMAX is higher than 3 / 5 of IMODE, the controller will
transit to 2-phase operation. When IIMAX is lower than 2 / 5
of IMODE, the active phase number will return to one phase.
For example, if RMODE = 30kΩ, RDS(ON) = 3mΩ,ΔIL = 5A.
The load current threshold for adding phase can be
calculated as
3 x IMODE
5
⎡ 3.3 x10 −4 x (IOUT_2P − 2.5 ) A x 3m Ω + 5.5μA ⎤
⎥
= ⎢
2
⎢
⎥
⎣
⎦
IOUT_2P = 21.2A
And the load current threshold for dropping phase can be
calculated as
DS8810-01
June 2011
RT8810
2 x IMODE
5
⎡ 3.3 x 10−4 x (IOUT_2P − 5 ) A x 3mΩ + 11μA ⎤
⎥
= ⎢
2
⎢
⎥
⎣
⎦
IOUT_2P = 10A
2/5 x IMODE
2/5
+
VFB
1µA
QR2
QR comp.
+
RQR2
EAP
Min. on
EAP
VFB
Drop Phase
VQR2
RT8810
FB
CQR1
QR
QR1
TQR
3/5 x IMODE
3/5
0.6V
+
Add Phase
Figure 7. Quick Response Active
-
+
IIMAX
-
ICS1
Feedback and Compensation
IMODE
ICS2
RMODE
Figure 6. Dynamic Phase Number Control Circuit
The RT8810 allows the output voltage of the DC/DC
converter to be adjusted from 0.6V to 85% of VIN supply
via an external resistor divider. It will try to maintain the
feedback pin at internal reference voltage (0.6V).
VOUT
Manual Phase Number Control
R1
FB
The RT8810 supports manual selecting of single phase or
dual phase operation. If IMODE is higher than 150μA, the
RT8810 operates in forced single phase mode. If IMODE is
smaller than 4μA, the RT8810 operates in forced dual
phase mode.
According to the resistor divider network above, the output
voltage is set as :
Note that, the MODE pin is not available for the RT8810C.
It supports only two phase operation.
⎛
⎞
VREF
R2 = R1 x ⎜
⎟
V
V
−
REF ⎠
⎝ OUT
Load Transient Quick Response
The RT8810 utilizes a new quick response feature to supply
heavy load current demand during instantaneous load
application transient. The RT8810 detects load transient
and reacts via VOUT pin. When VOUT drops during load
application transient, the quick response comparator will
send asserted signals to turn on high side MOSFETs and
turn off low side MOSFETs. The QR signal will turn on all
phase' high side MOSFETs while turning off low side
MOSFETs. Therefore, the influence of total quick response
function of the RT8810 is adjustable. The quick response
threshold can be set by RQR2. QR is triggered if VEAP >
1μA x RQR2 + VFB. The QR width can be set according
to :
C
x 0.8V
TQR = QR1
300μA
DS8810-01
June 2011
R2
The RT8810 is a voltage mode controller and requires
external compensation to have an accurate output voltage
regulation with fast transient response.
The RT8810 uses a high gain Operational
Transconductance Amplifier (OTA) as the error amplifier.
As Figure 8 shows, the OTA works as the voltage
controlled current source. The characteristic of OTA is as
below :
gm =
ΔIOUT
,
ΔVM
where ΔVM = ( VIN+ ) − ( VIN− ) and ΔVCOMP = ΔIOUT x ZOUT
VIN+
VIN-
+ GM
-
IOUT
VCOMP
ZOUT
Figure 8. Operational Transconductance Amplifier, OTA
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17
RT8810
Figure 9 shows a typical buck control loop using Type II
compensator. The control loop consists of the power stage,
PWM comparator and a compensator. The PWM
comparator compares V COMP with oscillator (OSC)
sawtooth wave to provide a Pulse-Width Modulated (PWM)
with an amplitude of VIN at the PHASE node. The PWM
wave is smoothen by the output filter LOUT and COUT. The
output voltage (VOUT) is sensed and fed to the inverting
input of the error amplifier.
VIN
UGATE
PWM
Comparator
Driver
Logic
+
-
PHASE
LOUT
COUT
LGATE
VOUT
VOSC
+
GM
-
VREF
FB
RFB1
The output LC filter introduces a double pole, 40dB/decade
gain slope above its corner resonant frequency, and a total
phase lag of 180 degrees. The resonant frequency of the
LC filter is expressed as :
fLC =
VIN
2π LOUT x COUT
The ESR zero is contributed by the ESR associated with
the output capacitance. Note that this requires the output
capacitor to have enough ESR to satisfy stability
requirements. The ESR zero of the output capacitor is
expressed as follows :
fESR =
RFB2
COMP
VCOMP
CC
RC
The DC gain of the modulator is the input voltage (VIN)
divided by the peak-to-peak oscillator voltage VOSC.
VIN
Gainmodulator =
ΔVOSC
CP
Figure 9. Typical Voltage Mode Buck Converter Control
Loop
The modulator transfer function is the small signal transfer
function of VOUT / VCOMP (output voltage over the error
amplifier output). This transfer function is dominated by a
DC gain, a double pole, and an ESR zero as shown in
Figure 10.
1
2π x COUT x ESR
The goal of the compensation network is to provide
adequate phase margin (usually greater than 45 degrees)
and the highest bandwidth (0dB crossing frequency). It is
also recommended to manipulate loop frequency response
so that its gain crosses over 0dB at a slope of −20dB/
dec. According to Figure 8, the compensation network
frequency is as below :
FP1 = 0
1
⎛ C x CP ⎞
2 π x RC x ⎜ C
⎟
⎝ CC + CP ⎠
1
=
2π x RC x CC
FP2 =
FZ1
Determining the 0dB crossing frequency (FC, control loop
bandwidth) is the first step of compensator design. Usually,
FC is set to 0.1 to 0.3 times the switching frequency. The
second step is to calculate the open loop modulator gain
and find out the gain loss at FC. The third step is to design
a compensator gain that can compensate the modulator
gain loss at FC. The final step is to design FZ1 and FZ2 to
allow the loop sufficient phase margin.
Figure 10. Typical Bode plot of a Voltage Mode Buck
Converter
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18
FZ1 is designed to cancel one of the double poles of
modulator. Usually, FZ1 is placed before fLC. FP2 is usually
placed below the switching frequency (typically, 0.5 to
1.0 times switching frequency) to eliminate high frequency
noise.
DS8810-01
June 2011
RT8810
The inductor plays an important role in the buck converter
because energy from the input power rail is stored in it
and then released to the load. From the viewpoint of
efficiency, the inductor's DC Resistance (DCR) should be
as small as possible since the inductor constantly carries
current. In addition, the inductor takes up most of the
board space, so its size is also important. Low profile
inductors can save board space, especially when there is
a height limitation.
Additionally, larger inductance results in lower ripple
current, and therefore lower power loss. However, the
inductor current rising time increases with inductance value.
This means the inductor will have a longer charging time
before its current reaches the required output current.
Since the response time is increased, the transient
response performance will be decreased. Therefore, the
inductor design is a trade-off between performance, size
and cost.
In general, inductance is designed such that the ripple
current ranges between 20% to 30% of full load current.
The inductance can be calculated using the following
equation.
L(MIN) =
fSW
VIN − VOUT
V
x OUT
x k x IOUT(MAX)
VIN
where k is 0.2 to 0.3.
Output Capacitor Selection
Output capacitors are used to maintain high performance
for the output beyond the bandwidth of the converter itself.
Two different settings of output capacitors can be found,
bulk capacitors closely located to the inductors and
ceramic output capacitors in close proximity to the load.
Latter ones are for mid frequency decoupling with
especially small ESR and ESL values, while the bulk
capacitors have to provide enough stored energy to
overcome the low frequency bandwidth gap between the
regulator and the GPU.
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
DS8810-01
June 2011
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :
PD(MAX) = (TJ(MAX) − TA) / θJA
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications of
the RT8810, the maximum junction temperature is 125°C
and TA is the ambient temperature. The junction to ambient
thermal resistance, θJA, is layout dependent. For WQFN16L 3x3 packages, the thermal resistance, θJA, is 68°C/
W on a standard JEDEC 51-7 four-layer thermal test board.
For WQFN-24L 4x4 packages, the thermal resistance,
θJA, is 52°C/W on a standard JEDEC 51-7 four-layer
thermal test board. The maximum power dissipation at TA
= 25°C can be calculated by the following formula :
PD(MAX) = (125°C − 25°C) / (68°C/W) = 1.471W for
WQFN-16L 3x3 package
PD(MAX) = (125°C − 25°C) / (52°C/W) = 1.923W for
WQFN-24L 4x4 package
The maximum power dissipation depends on the operating
ambient temperature for fixed T J(MAX) and thermal
resistance, θJA. For the RT8810 package, the derating
curves in Figure 11 allow the designer to see the effect of
rising ambient temperature on the maximum power
dissipation.
Maximum Power Dissipation (W)1
Inductor Selection
2.0
1.9
1.8
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0.0
Four-Layer PCB
WQFN-24L 4x4
WQFN-16L 3x3
0
25
50
75
100
125
Ambient Temperature (°C)
Figure 11. Derating Curves for the RT8810 Packages
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19
RT8810
Layout Considerations
Careful PC board layout is critical to achieve low switching
losses and clean, stable operation. The switching power
stage requires particular attention. If possible, mount all
of the power components on the top side of the board
with their ground terminals flush against one another.
Follow these guidelines for optimum PC board layout :
`
Power components should be placed first. Place the
input capacitors close to the power MOSFETs, then
locate the filter inductors and output capacitors between
the power MOSFETs and the load.
`
Place both the ceramic and bulk input capacitor close to
the drain pin of the high side MOSFET. This can reduce
the impedance presented by the input bulk capacitance
at high switching frequency. If there is more than one
high side MOSFET in parallel, each should have its own
individual ceramic capacitor.
`
Keep the power loops as short as possible. For low
voltage high current applications, power components
are the most critical part in the layout because they
switch a large amount of current. The current transition
from one device to another at high speed causes voltage
spikes due to the parasitic components on the circuit
board. Therefore, all of the high current switching loops
should be kept as short as possible with large and thick
copper traces to minimize the radiation of
electromagnetic interference.
`
Minimize the trace length between the power MOSFETs
and its drivers. Since the drivers use short, high current
pulses to drive the power MOSFETs, the driving traces
should be sized as short and wide as possible to reduce
the trace inductance. This is especially true for the low
side MOSFET, since this can reduce the possibility of
shoot through.
`
Provide enough copper area around the power MOSFETs
and the inductors to aid in heat sinking. Use thick
copper PCB to reduce the resistance and inductance
for improved efficiency.
`
The bank of output capacitor should be placed physically
close to the load. This can minimize the impedance
seen by the load, and then improve the transient
response.
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20
`
Place all of the high frequency decoupling ceramic
capacitors close to their decoupling targets.
`
Small signal components should be located as close to
the IC as possible. The small signal components include
the feedback components, current sensing components,
the compensation components, function setting
components and any bypass capacitors. These
components belong to the high impedance circuit loop
and are inherently sensitive to noise pick-up. Therefore,
they must be located close to their respective controller
pins and away from the noisy switching nodes.
`
A multi layer PCB design is recommended. Make use
of one single layer as the power ground and have a
separate control signal ground as the reference of all
signals.
DS8810-01
June 2011
RT8810
Outline Dimension
D
SEE DETAIL A
D2
L
1
E
E2
e
b
A
A1
1
1
2
2
DETAIL A
Pin #1 ID and Tie Bar Mark Options
A3
Note : The configuration of the Pin #1 identifier is optional,
but must be located within the zone indicated.
Dimensions In Millimeters
Dimensions In Inches
Symbol
Min
Max
Min
Max
A
0.700
0.800
0.028
0.031
A1
0.000
0.050
0.000
0.002
A3
0.175
0.250
0.007
0.010
b
0.180
0.300
0.007
0.012
D
2.950
3.050
0.116
0.120
D2
1.300
1.750
0.051
0.069
E
2.950
3.050
0.116
0.120
E2
1.300
1.750
0.051
0.069
e
L
0.500
0.350
0.020
0.450
0.014
0.018
W-Type 16L QFN 3x3 Package
DS8810-01
June 2011
www.richtek.com
21
RT8810
D2
D
SEE DETAIL A
L
1
E
E2
e
b
1
1
2
2
DETAIL A
Pin #1 ID and Tie Bar Mark Options
A
A3
A1
Note : The configuration of the Pin #1 identifier is optional,
but must be located within the zone indicated.
Dimensions In Millimeters
Dimensions In Inches
Symbol
Min
Max
Min
Max
A
0.700
0.800
0.028
0.031
A1
0.000
0.050
0.000
0.002
A3
0.175
0.250
0.007
0.010
b
0.180
0.300
0.007
0.012
D
3.950
4.050
0.156
0.159
D2
2.300
2.750
0.091
0.108
E
3.950
4.050
0.156
0.159
E2
2.300
2.750
0.091
0.108
e
L
0.500
0.350
0.020
0.450
0.014
0.018
W-Type 24L QFN 4x4 Package
Richtek Technology Corporation
Richtek Technology Corporation
Headquarter
Taipei Office (Marketing)
5F, No. 20, Taiyuen Street, Chupei City
5F, No. 95, Minchiuan Road, Hsintien City
Hsinchu, Taiwan, R.O.C.
Taipei County, Taiwan, R.O.C.
Tel: (8863)5526789 Fax: (8863)5526611
Tel: (8862)86672399 Fax: (8862)86672377
Email: [email protected]
Information that is provided by Richtek Technology Corporation is believed to be accurate and reliable. Richtek reserves the right to make any change in circuit design,
specification or other related things if necessary without notice at any time. No third party intellectual property infringement of the applications should be guaranteed
by users when integrating Richtek products into any application. No legal responsibility for any said applications is assumed by Richtek.
www.richtek.com
22
DS8810-01
June 2011