MCP6V06/7/8 300 µA, Auto-Zeroed Op Amps Features Description • High DC Precision: - VOS Drift: ±50 nV/°C (maximum) - VOS: ±3 µV (maximum) - AOL: 125 dB (minimum) - PSRR: 125 dB (minimum) - CMRR: 120 dB (minimum) - Eni: 1.7 µVP-P (typical), f = 0.1 Hz to 10 Hz - Eni: 0.54 µVp-p (typical), f = 0.01 Hz to 1 Hz • Low Power and Supply Voltages: - IQ: 300 µA/amplifier (typical) - Wide Supply Voltage Range: 1.8V to 5.5V • Easy to Use: - Rail-to-Rail Input/Output - Gain Bandwidth Product: 1.3 MHz (typical) - Unity Gain Stable - Available in Single and Dual - Single with Chip Select (CS): MCP6V08 • Extended Temperature Range: -40°C to +125°C The Microchip Technology Inc. MCP6V06/7/8 family of operational amplifiers has input offset voltage correction for very low offset and offset drift. These devices have a wide gain bandwidth product (1.3 MHz, typical) and strongly reject switching noise. They are unity gain stable, have no 1/f noise, and have good PSRR and CMRR. These products operate with a single supply voltage as low as 1.8V, while drawing 300 µA/amplifier (typical) of quiescent current. Typical Applications • • • • • Portable Instrumentation Sensor Conditioning Temperature Measurement DC Offset Correction Medical Instrumentation Design Aids • • • • • • SPICE Macro Models FilterLab® Software Mindi™ Circuit Designer & Simulator Microchip Advanced Part Selector (MAPS) Analog Demonstration and Evaluation Boards Application Notes Related Parts The Microchip Technology Inc. MCP6V06/7/8 op amps are offered in single (MCP6V06), single with Chip Select (CS) (MCP6V08), and dual (MCP6V07). They are designed in an advanced CMOS process. Package Types (top view) MCP6V06 2x3 TDFN * MCP6V06 SOIC NC 1 8 NC NC 1 VIN– 2 7 VDD VIN– 2 VIN+ 3 VSS 4 6 VOUT 5 NC VIN+ 3 VINA– 2 VINA+ 3 VSS 4 8 VDD 7 VDD 6 VOUT 5 NC MCP6V07 4x4 DFN * VOUTA 7 VOUTB V – INA 6 VINB– V + INA 5 VINB+ V SS MCP6V08 SOIC EP 9 VSS 4 MCP6V07 SOIC VOUTA 1 8 NC 8 VDD 1 2 3 EP 9 7 VOUTB 6 VINB– 5 VINB+ 4 MCP6V08 2x3 TDFN * NC 1 NC 1 8 CS VIN– 2 7 VDD VIN– 2 VIN+ 3 VSS 4 6 VOUT 5 NC VIN+ 3 VSS 4 8 CS EP 9 7 VDD 6 VOUT 5 NC * Includes Exposed Thermal Pad (EP); see Table 3-1. • MCP6V01/2/3: Spread clock, lower offset © 2008 Microchip Technology Inc. DS22093B-page 1 MCP6V06/7/8 Typical Application Circuit VIN R1 R2 R2 VDD/2 R3 VOUT C2 3 kΩ MCP6XXX MCP6V06 Offset Voltage Correction for Power Driver DS22093B-page 2 © 2008 Microchip Technology Inc. MCP6V06/7/8 1.0 ELECTRICAL CHARACTERISTICS 1.1 Absolute Maximum Ratings † † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. VDD – VSS .......................................................................6.5V Current at Input Pins ....................................................±2 mA Analog Inputs (VIN+ and VIN–) †† ... VSS – 1.0V to VDD+1.0V All other Inputs and Outputs ............ VSS – 0.3V to VDD+0.3V Difference Input voltage ...................................... |VDD – VSS| Output Short Circuit Current ................................ Continuous Current at Output and Supply Pins ............................±30 mA Storage Temperature ...................................-65°C to +150°C Max. Junction Temperature ........................................ +150°C ESD protection on all pins (HBM, MM) ................≥ 4 kV, 300V 1.2 †† See Section 4.2.1 “Rail-to-Rail Inputs”. Specifications TABLE 1-1: DC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to +5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, and CS = GND (refer to Figure 1-5 and Figure 1-6). Parameters Sym Min Typ Max Units Conditions Input Offset Input Offset Voltage VOS -3 — +3 µV Input Offset Voltage Drift with Temperature (linear Temp. Co.) TC1 -50 — +50 nV/°C Input Offset Voltage Quadratic Temp. Co. TC2 — ±0.15 — PSRR 125 142 — Input Bias Current IB — +6 — pA Input Bias Current across Temperature IB — +140 — pA TA = +85°C IB — +1500 +5000 pA TA = +125°C IOS — -85 — pA Power Supply Rejection TA = +25°C (Note 1) TA = -40 to +125°C (Note 1) nV/°C2 TA = -40 to +125°C dB (Note 1) Input Bias Current and Impedance Input Offset Current IOS — -85 — pA TA = +85°C IOS -1000 -190 1000 pA TA = +125°C Common Mode Input Impedance ZCM — 1013||6 — Ω||pF Differential Input Impedance ZDIFF — 1013||6 — Ω||pF Common-Mode Input Voltage Range VCMR VSS − 0.20 — VDD + 0.20 V (Note 2) Common-Mode Rejection CMRR 120 136 — dB VDD = 1.8V, VCM = -0.2V to 2.0V (Note 1, Note 2) CMRR 130 147 — dB VDD = 5.5V, VCM = -0.2V to 5.7V (Note 1, Note 2) AOL 125 147 — dB VDD = 1.8V, VOUT = 0.2V to 1.6V (Note 1) AOL 135 158 — dB VDD = 5.5V, VOUT = 0.2V to 5.3V (Note 1) Input Offset Current across Temperature Common Mode Open-Loop Gain DC Open-Loop Gain (large signal) Note 1: 2: Set by design and characterization. Due to thermal junction and other effects in the production environment, these parts can only be screened in production (except TC1; see Appendix B: “Offset Related Test Screens”). Figure 2-18 shows how VCMR changed across temperature for the first three production lots. © 2008 Microchip Technology Inc. DS22093B-page 3 MCP6V06/7/8 TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to +5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, and CS = GND (refer to Figure 1-5 and Figure 1-6). Parameters Sym Min Typ Max Units Conditions VOL, VOH ISC VSS + 15 — VDD − 15 mV G = +2, 0.5V input overdrive — ±7 — mA VDD = 1.8V ISC — ±22 — mA VDD = 5.5V Output Maximum Output Voltage Swing Output Short Circuit Current Power Supply VDD 1.8 — 5.5 V IQ 200 300 400 µA VPOR 1.15 — 1.65 V Supply Voltage Quiescent Current per amplifier POR Trip Voltage Note 1: 2: IO = 0 Set by design and characterization. Due to thermal junction and other effects in the production environment, these parts can only be screened in production (except TC1; see Appendix B: “Offset Related Test Screens”). Figure 2-18 shows how VCMR changed across temperature for the first three production lots. TABLE 1-2: AC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to +5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND (refer to Figure 1-5 and Figure 1-6). Parameters Sym Min Typ Max Units Conditions Amplifier AC Response Gain Bandwidth Product GBWP — 1.3 — MHz Slew Rate SR — 0.5 — V/µs Phase Margin PM — 65 — ° Eni — 0.54 — µVP-P f = 0.01 Hz to 1 Hz Eni — 1.7 — µVP-P f = 0.1 Hz to 10 Hz eni — 82 — nV/√Hz f < 2.5 kHz eni — 52 — nV/√Hz f = 100 kHz ini — 0.6 — fA/√Hz IMD — 32 — µVPK VCM tone = 50 mVPK at 1 kHz, GN = 1, VDD = 1.8V IMD — 25 — µVPK VCM tone = 50 mVPK at 1 kHz, GN = 1, VDD = 5.5V Start Up Time tSTR — 500 — µs VOS within 50 µV of its final value Offset Correction Settling Time tSTL — 300 — µs G = +1, VIN step of 2V, VOS within 50 µV of its final value Output Overdrive Recovery Time tODR — 100 — µs G = -100, ±0.5V input overdrive to VDD/2, VIN 50% point to VOUT 90% point (Note 2) G = +1 Amplifier Noise Response Input Noise Voltage Input Noise Voltage Density Input Noise Current Density Amplifier Distortion (Note 1) Intermodulation Distortion (AC) Amplifier Step Response Note 1: 2: These parameters were characterized using the circuit in Figure 1-7. Figure 2-37 and Figure 2-38 show both an IMD tone at DC and a residual tone at1 kHz; all other IMD and clock tones are spread by the randomization circuitry. tODR includes some uncertainty due to clock edge timing. DS22093B-page 4 © 2008 Microchip Technology Inc. MCP6V06/7/8 TABLE 1-3: DIGITAL ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to +5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND (refer to Figure 1-5 and Figure 1-6). Parameters Sym Min Typ Max Units Conditions RPD 3 5 — MΩ CS Logic Threshold, Low VIL VSS — 0.3VDD V CS Input Current, Low ICSL — 5 — pA CS Logic Threshold, High VIH 0.7VDD — VDD V CS Input Current, High ICSH — VDD/RPD — pA CS = VDD ISS — -0.7 — µA CS = VDD, VDD = 1.8V ISS — -2.3 — µA CS = VDD, VDD = 5.5V — 20 — pA CS = VDD CS Pull-Down Resistor (MCP6V08) CS Pull-Down Resistor CS Low Specifications (MCP6V08) CS = VSS CS High Specifications (MCP6V08) CS Input High, GND Current per amplifier Amplifier Output Leakage, CS High IO_LEAK CS Dynamic Specifications (MCP6V08) CS Low to Amplifier Output On Turn-on Time tON — 11 100 µs CS Low = VSS+0.3 V, G = +1 V/V, VOUT = 0.9 VDD/2 CS High to Amplifier Output High-Z tOFF — 10 — µs CS High = VDD – 0.3 V, G = +1 V/V, VOUT = 0.1 VDD/2 VHYST — 0.25 — V Internal Hysteresis TABLE 1-4: TEMPERATURE SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = +1.8V to +5.5V, VSS = GND. Parameters Sym Min Typ Max Units Specified Temperature Range TA -40 — +125 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C Thermal Resistance, 8L-2x3 TDFN θJA — 41 — °C/W Thermal Resistance, 8L-4x4 DFN θJA — 44 — °C/W Thermal Resistance, 8L-SOIC θJA — 150 — °C/W Conditions Temperature Ranges (Note 1) Thermal Package Resistances Note 1: 2: (Note 2) Operation must not cause TJ to exceed Maximum Junction Temperature specification (150°C). Measured on a standard JC51-7, four layer printed circuit board with ground plane and vias. © 2008 Microchip Technology Inc. DS22093B-page 5 MCP6V06/7/8 1.3 Timing Diagrams 1.4 1.8V to 5.5V 1.8V VDD 0V tSTR VOS + 50 µV VOS Test Circuits The circuits used for the DC and AC tests are shown in Figure 1-5 and Figure 1-6. Lay the bypass capacitors out as discussed in Section 4.3.8 “Supply Bypassing and Filtering”. RN is equal to the parallel combination of RF and RG to minimize bias current effects. VDD VOS – 50 µV VIN FIGURE 1-1: Amplifier Start Up. RISO VIN 100 nF RG VOS + 50 µV VOS VOS + 50 µV Offset Correction Settling CL RL VL RF FIGURE 1-5: AC and DC Test Circuit for Most Non-Inverting Gain Conditions. VDD FIGURE 1-2: Time. VOUT MCP6V0X VDD/3 tSTL 1 µF RN 1 µF VDD/3 RN RISO VOUT MCP6V0X VIN tODR RG VDD VDD/2 VSS FIGURE 1-3: CS Output Overdrive Recovery. VIL VIH tON IDD tOFF 1 µA (typical) ISS -2 µA (typical) ICS V /5 MΩ DD (typical) FIGURE 1-4: DS22093B-page 6 VL RF 20.0 kΩ 20.0 kΩ 50Ω 0.1% 0.1% 25 turn VREF High-Z High-Z 300 µA (typical) 300 µA (typical) 5 pA (typical) RL The circuit in Figure 1-7 tests the op amp input’s dynamic behavior (i.e., IMD, tSTR, tSTL and tODR). The potentiometer balances the resistor network (VOUT should equal VREF at DC). The op amp’s common mode input voltage is VCM = VIN/2. The error at the input (VERR) appears at VOUT with a noise gain of 10 V/V. 1 µA (typical) -2 µA (typical) VDD/5 MΩ (typical) Chip Select (MCP6V08). VIN 2.49 kΩ 2.49 kΩ VOUT CL FIGURE 1-6: AC and DC Test Circuit for Most Inverting Gain Conditions. tODR VOUT 100 nF VIN VDD 1 µF RISO 100 nF MCP6V0X VOUT CL RL VL 20.0 kΩ 20.0 kΩ 24.9 Ω 0.1% 0.1% FIGURE 1-7: Input Behavior. Test Circuit for Dynamic © 2008 Microchip Technology Inc. MCP6V06/7/8 2.0 TYPICAL PERFORMANCE CURVES Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. DC Input Precision 4 80 Samples TA = +25°C VDD = 1.8V and 5.5V Soldered on PCB 12% 10% 8% 6% 4% 2% 2 1 0 -1 -3 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 Power Supply Voltage (V) 4 Input Offset Voltage (µV) 20% FIGURE 2-4: Input Offset Voltage vs. Power Supply Voltage with VCM = VCMR_L. 80 Samples VDD = 1.8V and 5.5V Soldered on PCB 15% 10% 5% 2 1 0 -1 -3 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 Power Supply Voltage (V) 20% 15% 10% 5% Input Offset Voltage's Quadratic Temp Co; 2 TC2 (nV/°C ) FIGURE 2-3: Input Offset Voltage Quadratic Temp Co. © 2008 Microchip Technology Inc. 0.4 0.2 0.0 -0.2 0% Input Offset Voltage (µV) 25% FIGURE 2-5: Input Offset Voltage vs. Power Supply Voltage with VCM = VCMR_H. 4 80 Samples VDD = 1.8V and 5.5V Soldered on PCB -0.4 Percentage of Occurrences 30% Input Offset Voltage Drift. +125°C +85°C +25°C -40°C -2 Input Offset Voltage Drift; TC1 (nV/°C) FIGURE 2-2: VCM = VCMR_H Representative Part 3 -4 50 40 30 20 10 0 -10 -20 -30 -40 0% -50 Percentage of Occurrences 25% Input Offset Voltage. +125°C +85°C +25°C -40°C -2 Input Offset Voltage (µV) FIGURE 2-1: VCM = VCMR_L Representative Part 3 -4 2.0 1.5 1.0 0.5 0.0 -0.5 -1.0 -1.5 0% -2.0 Percentage of Occurrences 14% Input Offset Voltage (µV) 2.1 Representative Part 3 2 VDD = 5.5V 1 0 -1 VDD = 1.8V -2 -3 -4 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) FIGURE 2-6: Output Voltage. Input Offset Voltage vs. DS22093B-page 7 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. 6% 4% 2% -3 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 -4 FIGURE 2-11: CMRR, PSRR (dB) 155 25% VDD = 5.5V 20% 15% 10% DC Open-Loop Gain. 160 39 Samples TA = +25°C Soldered on PCB VDD = 1.8V 5% VDD = 5.5V VDD = 1.8V 150 145 140 135 CMRR 130 PSRR 125 1/CMRR (µV/V) FIGURE 2-9: DS22093B-page 8 CMRR. 0.4 0.3 0.2 0.2 0.1 0.0 -0.1 -0.2 -0.2 -0.3 0% -0.4 Percentage of Occurrences FIGURE 2-8: Input Offset Voltage vs. Common Mode Voltage with VDD = 5.5V. 30% 0.4 1/AOL (µV/V) Input Common Mode Voltage (V) 35% 0.3 VDD = 1.8V 0.3 -2 0.2 0 -1 VDD = 5.5V 0.1 1 40 Samples TA = +25°C Soldered on PCB -0.1 2 50% 45% 40% 35% 30% 25% 20% 15% 10% 5% 0% PSRR. -0.2 +125°C +85°C +25°C -40°C -0.3 Percentage of Occurrences Input Offset Voltage (µV) VDD = 5.5V Representative Part FIGURE 2-10: 0.0 FIGURE 2-7: Input Offset Voltage vs. Common Mode Voltage with VDD = 1.8V. 3 0.2 1/PSRR (µV/V) Input Common Mode Voltage (V) 4 0.1 0% 2.4 2.2 2.0 1.0 0.8 0.6 0.4 0.2 0.0 -0.2 -0.4 -0.6 -4 1.8 -3 1.6 -2 1.4 -40°C +25°C +85°C +125°C 8% 0.0 -1 10% -0.1 0 40 Samples TA = +25°C Soldered on PCB -0.2 1 12% -0.3 2 14% -0.4 Percentage of Occurrences VDD = 1.8V Representative Part 3 1.2 Input Offset Voltage (µV) 4 120 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-12: CMRR and PSRR vs. Ambient Temperature. © 2008 Microchip Technology Inc. MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. 10,000 155 Input Bias, Offset Currents (pA) DC Open-Loop Gain (dB) 160 VDD = 5.5V VDD = 1.8V 150 145 140 135 130 125 120 -25 0 25 50 75 Ambient Temperature (°C) 100 -IOS 100 10 125 IB 200 100 IB 50 0 -50 IOS -100 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 -150 Common Mode Input Voltage (V) FIGURE 2-14: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +85°C. 1600 1400 1200 1000 800 600 400 200 0 -200 -400 1.E-02 10m 1m 1.E-03 100µ 1.E-04 10µ 1.E-05 1µ 1.E-06 100n 1.E-07 10n 1.E-08 1n 1.E-09 100p 1.E-10 10p 1.E-11 1p 1.E-12 Input Current Magnitude (A) TA = +85°C VDD = 5.5V 150 25 35 45 55 65 75 85 95 105 115 125 Ambient Temperature (°C) FIGURE 2-16: Input Bias and Offset Currents vs. Ambient Temperature with VDD = +5.5V. FIGURE 2-13: DC Open-Loop Gain vs. Ambient Temperature. Input Bias, Offset Currents (pA) 1,000 1 -50 +125°C +85°C +25°C -40°C -1.0 -0.9 -0.8 -0.7 -0.6 -0.5 -0.4 -0.3 -0.2 -0.1 0.0 Input Voltage (V) FIGURE 2-17: Input Bias Current vs. Input Voltage (below VSS). TA = +125°C VDD = 5.5V IB 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 IOS -0.5 Input Bias, Offset Currents (pA) VDD = 5.5V Common Mode Input Voltage (V) FIGURE 2-15: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +125°C. © 2008 Microchip Technology Inc. DS22093B-page 9 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. Other DC Voltages and Currents 450 400 VDD = 5.5V Supply Current (µA) VDD = 1.8V VDD – VOH 300 250 200 100 50 VOL – VSS 125 FIGURE 2-20: Output Voltage Headroom vs. Ambient Temperature. DS22093B-page 10 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 15% 10% 5% 0% 1.7 100 20% 1.6 0 25 50 75 Ambient Temperature (°C) 93 Samples 3 Lots TA = +25°C 1.5 VDD = 1.8V 25% 1.4 VDD – VOH 30% Supply Current vs. Power 1.3 VDD = 5.5V -25 FIGURE 2-22: Supply Voltage. 1.2 RL = 20 kΩ -50 Power Supply Voltage (V) Percentage of Occurrences FIGURE 2-19: Output Voltage Headroom vs. Output Current. VOL – VSS 0.0 10 1.1 1 Output Current Magnitude (mA) 1.5 0 10 0.1 +125°C +85°C +25°C -40°C 150 1.0 100 350 0.5 Output Voltage Headroom (mV) 6.5 FIGURE 2-21: Output Short Circuit Current vs. Power Supply Voltage. 1000 Output Headroom (mV) 6.0 Power Supply Voltage (V) FIGURE 2-18: Input Common Mode Voltage Headroom (Range) vs. Ambient Temperature. 12 11 10 9 8 7 6 5 4 3 2 1 0 -40 125 5.5 0 25 50 75 100 Ambient Temperature (°C) 5.0 -25 4.5 -50 -30 4.0 Upper ( VDD – VCMR) -0.35 3.5 -0.30 +125°C +85°C +25°C -40°C -20 3.0 -0.25 0 -10 2.5 Lower (VCMR – VSS) -0.20 10 2.0 -0.15 20 1.5 -0.10 -40°C +25°C +85°C +125°C 30 1.0 -0.05 40 0.5 3 Lots 0.00 0.0 0.05 Output Short Circuit Current (mA) Input Common Mode Voltage Headroom (V) 2.2 POR Trip Voltage (V) FIGURE 2-23: Voltage. Power On Reset Trip © 2008 Microchip Technology Inc. MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. POR Trip Voltage (V) 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-24: Power On Reset Voltage vs. Ambient Temperature. © 2008 Microchip Technology Inc. DS22093B-page 11 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. Frequency Response PSRR+ PSRR- 100 1.0 90 0.8 80 0.6 70 0.4 PM 0.2 CMRR and PSRR vs. -90 10 -150 | AOL | 0 -120 -180 -10 -210 -20 -240 10k 100k 1M 1.E+04 1.E+05 1.E+06 Frequency (Hz) VDD = 5.5V CL = 60 pF 50 0 -120 -150 -180 -210 -20 -240 -270 10M 1.E+07 FIGURE 2-27: Open-Loop Gain vs. Frequency with VDD = 5.5V. DS22093B-page 12 80 70 0.4 60 0.2 1.6 -10 10k 100k 1M 1.E+04 1.E+05 1.E+06 Frequency (Hz) 90 0.6 1.8 -90 | AOL | VDD = 1.8V 100 0.8 -30 30 10 1.0 0 -60 ∠AOL VDD = 5.5V 50 PM 40 FIGURE 2-29: Gain Bandwidth Product and Phase Margin vs. Common Mode Input Voltage. 40 20 110 1.2 Common Mode Input Voltage (V) Open-Loop Phase (°) 60 120 1.4 0.0 -270 10M 1.E+07 130 GBWP -0.5 0.0 20 1.6 Phase Margin (°) 30 ∠AOL 1.8 -30 -60 40 125 FIGURE 2-28: Gain Bandwidth Product and Phase Margin vs. Ambient Temperature. 0 40 -25 0 25 50 75 100 Ambient Temperature (°C) 50 130 GBWP 120 1.4 110 1.2 100 1.0 VDD = 1.8V 0.8 VDD = 5.5V 90 80 0.6 70 0.4 60 0.2 0.0 Phase Margin (°) VDD = 1.8V CL = 60 pF -50 1M 1.E+06 60 VDD = 1.8V 3.0 3.5 4.0 4.5 5.0 5.5 6.0 100k 1.E+05 FIGURE 2-26: Open-Loop Gain vs. Frequency with VDD = 1.8V. Open-Loop Gain (dB) 110 1.2 0.5 1.0 1.5 2.0 2.5 1k 10k 1.E+03 1.E+04 Frequency (Hz) Gain Bandwidth Product (MHz) 100 1.E+02 50 -30 1k 1.E+03 GBWP 1.4 120 VDD = 5.5V 0.0 60 -30 1k 1.E+03 130 1.6 Phase Margin (°) CMRR FIGURE 2-25: Frequency. Open-Loop Gain (dB) Gain Bandwidth Product (MHz) 1.8 Gain Bandwidth Product (MHz) 110 100 90 80 70 60 50 40 30 20 10 0 10 1.E+01 Open-Loop Phase (°) CMRR, PSRR (dB) 2.3 50 PM 40 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) FIGURE 2-30: Gain Bandwidth Product and Phase Margin vs. Output Voltage. © 2008 Microchip Technology Inc. MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 10k 1.E+04 100 VDD = 1.8V 1k 1.E+03 100 1.E+02 G = 1 V/V G = 10 V/V G = 100 V/V 10 1.E+01 1 1.E+00 100k 1.0E+05 1M 10M 1.0E+06 1.0E+07 Frequency (Hz) 50 40 VDD = 1.8V 30 20 1M 1.E+06 Frequency (Hz) 10M 1.E+07 FIGURE 2-33: Channel-to-Channel Separation vs. Frequency. Maximum Output Voltage Swing (V P-P ) Open-Loop Output Impedance ( Ω) VDD = 5.5V 60 10 1k 1.E+03 100 1.E+02 1.E+001 100k 1.0E+05 70 0 100k 1.E+05 100M 1.0E+08 VDD = 5.5V 10 1.E+01 80 10 FIGURE 2-31: Closed-Loop Output Impedance vs. Frequency with VDD = 1.8V. 1.E+04 10k RTI 90 Channel-to-Channel Separation (dB) Open-Loop Output Impedance ( Ω) VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. G = 1 V/V G = 10 V/V G = 100 V/V 1M 10M 1.0E+06 1.0E+07 Frequency (Hz) 100M 1.0E+08 FIGURE 2-32: Closed-Loop Output Impedance vs. Frequency with VDD = 5.5V. © 2008 Microchip Technology Inc. VDD = 5.5V 1 0.1 1k 1.E+03 VDD = 1.8V 10k 100k 1.E+04 1.E+05 Frequency (Hz) 1M 1.E+06 FIGURE 2-34: Maximum Output Voltage Swing vs. Frequency. DS22093B-page 13 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. Input Noise and Distortion 100 eni 100 100 Eni(0 Hz to f) IMD Spectrum, RTI (µV PK) 1,000 Input Noise Voltage Density 1 kHz tone VDD = 5.5V 100 80 VDD = 1.8V 60 40 20 1k 10k 1.E+03 1.E+04 Frequency (Hz) VDD = 1.8V NPBW = 10 Hz NPBW = 1 Hz 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.0 0.5 0.0 -0.5 0 0 10 20 30 40 Common Mode Input Voltage (V) IMD Spectrum, RTI (µV PK) 100 IMD tone at DC 10 residual 1 kHz tone 1 100 1.E+02 GDM = 1 V/V VCM tone = 50 mVPK, f = 1 kHz VDD = 5.5V VDD = 1.8V 1k 10k 1.E+03 1.E+04 Frequency (Hz) 100k 1.E+05 FIGURE 2-37: Inter-Modulation Distortion vs. Frequency with VCM Disturbance (see Figure 1-7). DS22093B-page 14 50 60 t (s) 70 80 90 100 FIGURE 2-39: Input Noise vs. Time with 1 Hz and 10 Hz Filters and VDD =1.8V. VDD = 5.5V Input Noise Voltage; e ni(t) (0.5 µV/div) FIGURE 2-36: Input Noise Voltage Density vs. Input Common Mode Voltage. 100k 1.E+05 FIGURE 2-38: Inter-Modulation Distortion vs. Frequency with VDD Disturbance (see Figure 1-7). Input Noise Voltage; e ni(t) (0.5 µV/div) 120 VDD = 5.5V VDD = 1.8V 10 160 140 GDM = 1 V/V VDD tone = 50 mVPK, f = 1 kHz IMD tone at DC 1 100 1.E+02 10 10 10 100 1k 10k 100k 1.E+01 1.E+02 1.E+03 1.E+04 1.E+05 Frequency (Hz) FIGURE 2-35: vs. Frequency. Input Noise Voltage Density; eni (nV/√ Hz) 1000 VDD = 5.5V VDD = 1.8V 1.5 Input Noise Voltage Density; eni (nV/√Hz) 10,000 Input Noise Voltage; Eni (µV P-P ) 2.4 NPBW = 10 Hz NPBW = 1 Hz 0 10 20 30 40 50 60 t (s) 70 80 90 100 FIGURE 2-40: Input Noise vs. Time with 1 Hz and 10 Hz Filters and VDD =5.5V. © 2008 Microchip Technology Inc. MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. Time Response 0 0 15 4.5 4.0 VDD 10 3.5 5 3.0 0 2.5 -5 2.0 VOS -10 1.5 -15 1.0 -20 0.5 -25 Power Supply Voltage (V) Input Offset Voltage (mV) 5.0 POR Trip Point 20 0.0 VOUT 4 5 10 3 2 1 0 10 12 Time (µs) 14 16 18 20 15 20 25 30 Time (µs) 35 40 45 50 Non-inverting Large Signal VDD = 5.5V G = -1 Output Voltage (10 mV/div) Input, Output Voltages (V) 5 8 Non-inverting Small Signal FIGURE 2-45: Step Response. VDD = 5.5V G=1 VIN 6 VDD = 5.5V G=1 0 FIGURE 2-42: Input Offset Voltage vs. Time at Power Up. 6 4 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 0.0 0.2 0.4 0.6 Time 0.8(200 1.0 µs/div) 1.2 1.4 1.6 1.8 2.0 7 2 FIGURE 2-44: Step Response. FIGURE 2-41: Input Offset Voltage vs. Time with Temperature Change. 25 VDD = 5.5V G=1 Output Voltage (10 mV/div) 50 45 40 35 TPCB 30 25 20 15 VOS 10 5 0 -5 -10 -15 20 40 60 80 100 120 140 160 180 200 Time (s) Temperature increased by using heat gun for 4 seconds. Output Voltage (V) 7 6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6 PCB Temperature (°C) Input Offset Voltage (µV) 2.5 -1 0 1 2 3 4 5 6 Time (ms) 7 8 9 10 FIGURE 2-43: The MCP6V06/7/8 family shows no input phase reversal with overdrive. © 2008 Microchip Technology Inc. 0 1 2 FIGURE 2-46: Response. 3 4 5 6 Time (µs) 7 8 9 10 Inverting Small Signal Step DS22093B-page 15 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VDD = 5.5V G = -1 Output Voltage (V) 5.0 5 10 FIGURE 2-47: Response. 15 20 25 30 Time (µs) 35 40 45 VOUT Inverting Large Signal Step VDD = 5.5V 0.8 Rising Edge 0.7 0.6 0.5 VDD = 1.8V 0.4 Falling Edge 0.2 0.1 0.0 -50 -25 FIGURE 2-48: Temperature. DS22093B-page 16 0 25 50 75 Ambient Temperature (°C) 100 Slew Rate vs. Ambient 125 5 4.0 4 3.0 3 2.0 2 1.0 VOUT G VIN VDD = 5.5V G = -100 V/V 0.5V Overdrive -1.0 50 0.9 Slew Rate (V/µs) G VIN 0.0 0 0.3 6 1 0 Input Voltage × G (V/V) 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -1 Time (50 µs/div) FIGURE 2-49: Output Overdrive Recovery vs. Time with G = -100 V/V. Overdrive Recovery Time (µs) Output Voltage (V) VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. 1000 0.5V Output Overdrive VDD = 5.5V 100 tODR, high 10 tODR, low VDD = 1.8V 1 1 10 100 Inverting Gain Magnitude (V/V) 1000 FIGURE 2-50: Output Overdrive Recovery Time vs. Inverting Gain. © 2008 Microchip Technology Inc. MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. Chip Select Response (MCP6V08 only) 2.0 2.5 3.0 3.5 4.0 4.5 Power Supply Voltage (V) 5.0 5.5 Power Supply Current (µA) 350 300 Op Amp turns on here 250 Op Amp turns off here VDD = 1.8V G=1 VIN = 0.9V VL = 0V 200 150 Hysteresis 100 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Chip Select Voltage (V) 50 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 Chip Select Voltage (V) 1.6 1.8 FIGURE 2-52: Power Supply Current vs. Chip Select Voltage with VDD = 1.8V. Power Supply Current (µA) 600 500 400 300 Op Amp turns on here 200 Op Amp turns off here VDD = 5.5V G=1 VIN = 2.75V VL = 0V Hysteresis 100 0 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Chip Select Voltage (V) FIGURE 2-53: Power Supply Current vs. Chip Select Voltage with VDD = 5.5V. © 2008 Microchip Technology Inc. 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 -0.2 -0.4 -0.6 Chip Select Current vs. Chip VOUT On VOUT Off VOUT Off VDD = 1.8V G = +1 V/V VIN= VDD RL = 10 kΩ tied to VDD/2 CS Time (5 µs/div) 12 11 10 9 8 7 6 5 4 3 2 1 0 FIGURE 2-55: Chip Select Voltage, Output Voltage vs. Time with VDD = 1.8V. Output Voltage (V) 0.0 VDD = 5.5V FIGURE 2-54: Select Voltage. FIGURE 2-51: Chip Select Current vs. Power Supply Voltage. 400 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0.0 Chip Select Voltage (V) 1.5 Chip Select Current (µA) CS = VDD 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 -1.0 VOUT On VOUT Off VOUT Off VDD = 5.5V G = +1 V/V VIN= VDD RL = 10 kΩ tied to VDD/2 CS 0 5 39 36 33 30 27 24 21 18 15 12 9 6 3 0 Chip Select Voltage (V) 1.3 1.2 1.1 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0.0 Output Voltage (V) Chip Select Current (µA) 2.6 10 15 20 25 30 35 40 45 50 Time (5 µs/div) FIGURE 2-56: Chip Select Voltage, Output Voltage vs. Time with VDD = 5.5V. DS22093B-page 17 MCP6V06/7/8 Note: Unless otherwise indicated, TA = +25°C, VDD = +1.8V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, CL = 60 pF, and CS = GND. 7 65% VIH/VDD Pull-down Resistor (MΩ) VDD = 5.5V 60% 55% 50% 45% 40% 35% VIL/VDD VDD = 1.8V 4 3 2 1 FIGURE 2-58: 100 125 6.5 0 25 50 75 Ambient Temperature (°C) 6.0 -25 5.5 0.00 5.0 0.05 4.5 0.10 4.0 VDD = 1.8V 0.15 125 +125°C +85°C +25°C -40°C 3.5 0.20 100 CS = VDD Representative Part 3.0 VDD = 5.5V 0.25 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 2.5 0.35 0 25 50 75 Ambient Temperature (°C) FIGURE 2-60: Chip Select’s Pull-down Resistor (RPD) vs. Ambient Temperature. Power Supply Current (µA) 0.40 -25 2.0 FIGURE 2-57: Chip Select Relative Logic Thresholds vs. Ambient Temperature. 0.30 -50 125 1.5 0 25 50 75 100 Ambient Temperature (°C) 1.0 -25 0.0 -50 Chip Select Hysteresis (V) 5 0 30% -50 6 0.5 Relative Chip Select Logic Levels; Low and High ( ) 70% Power Supply Voltage (V) Chip Select Hysteresis. FIGURE 2-61: Quiescent Current in Shutdown vs. Power Supply Voltage. Chip Select Turn On Time (µs) 16 14 12 VDD = 5.5V 10 8 6 VDD = 1.8V 4 2 0 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-59: Chip Select Turn On Time vs. Ambient Temperature. DS22093B-page 18 © 2008 Microchip Technology Inc. MCP6V06/7/8 3.0 PIN DESCRIPTIONS Descriptions of the pins are listed in Table 3-1. TABLE 3-1: PIN FUNCTION TABLE MCP6V06 MCP6V07 MCP6V08 Symbol Description TDFN SOIC DFN SOIC TDFN SOIC 6 6 1 1 6 6 VOUT, VOUTA Output (op amp A) 2 2 2 2 2 2 VIN–, VINA– Inverting Input (op amp A) 3 3 3 3 3 3 VIN+, VINA+ Non-inverting Input (op amp A) 4 4 4 4 4 4 VSS — — 5 5 — — VINB+ Non-inverting Input (op amp B) Negative Power Supply — — 6 6 — — VINB– Inverting Input (op amp B) — — 7 7 — — VOUTB Output (op amp B) 7 7 8 8 7 7 VDD Positive Power Supply — — — — — 8 CS Chip Select (op amp A) 1, 5, 8 1, 5, 8 — — 1, 5, 8 1, 5 NC No Internal Connection 9 — 9 — 9 — EP Exposed Thermal Pad (EP); must be connected to VSS 3.1 Analog Outputs The analog output pins (VOUT) are low-impedance voltage sources. 3.2 Analog Inputs The non-inverting and inverting inputs (VIN+, VIN–, …) are high-impedance CMOS inputs with low bias currents. 3.3 Power Supply Pins The positive power supply (VDD) is 1.8V to 5.5V higher than the negative power supply (VSS). For normal operation, the other pins are between VSS and VDD. 3.4 Chip Select (CS) Digital Input This pin (CS) is a CMOS, Schmitt-triggered input that places the MCP6V08 op amps into a low power mode of operation. 3.5 Exposed Thermal Pad (EP) There is an internal connection between the Exposed Thermal Pad (EP) and the VSS pin; they must be connected to the same potential on the Printed Circuit Board (PCB). This pad can be connected to a PCB ground plane to provide a larger heat sink. This improves the package thermal resistance (θJA). Typically, these parts are used in a single (positive) supply configuration. In this case, VSS is connected to ground and VDD is connected to the supply. VDD will need bypass capacitors. © 2008 Microchip Technology Inc. DS22093B-page 19 MCP6V06/7/8 NOTES: DS22093B-page 20 © 2008 Microchip Technology Inc. MCP6V06/7/8 4.0 APPLICATIONS 4.1 The MCP6V06/7/8 family of auto-zeroed op amps is manufactured using Microchip’s state of the art CMOS process. It is designed for low cost, low power and high precision applications. Its low supply voltage, low quiescent current and wide bandwidth makes the MCP6V06/7/8 ideal for battery-powered applications. Overview of Auto-zeroing Operation Figure 4-1 shows a simplified diagram of the MCP6V06/7/8 auto-zeroed op amps. This will be used to explain how the DC voltage errors are reduced in this architecture. VIN+ VIN– Main Amp. CFW NC Output Buffer VOUT VREF Null Input Switches φ1 Null Output Switches Null Amp. CH Null Correct Switches φ2 φ1 φ2 POR Oscillator Digital Control Clock Randomization CS FIGURE 4-1: 4.1.1 Simplified Auto-zeroed Op Amp Functional Diagram. BUILDING BLOCKS The Null Amp. and Main Amp. are designed for high gain and accuracy using a differential topology. They have an auxiliary input (bottom left) used for correcting the offset voltages. Both inputs are added together internally. The capacitors at the auxiliary inputs (CFW and CH) hold the corrected values during normal operation. The Output Buffer is designed to drive external loads at the VOUT pin. It also produces a single ended output voltage (VREF is an internal reference voltage). The internal POR ensures the part starts up in a known good state. It also provides protection against power supply brown out events. The Chip Select input places the op amp in a low power state when it is high. When it goes low, it powers the op amp at its normal level and starts operation properly. The Digital Control circuitry takes care of all of the housekeeping details of the switching operation. It also takes care of Chip Select and POR events. All of these switches are make-before-break in order to minimize glitch-induced errors. They are driven by two clock phases (φ1 and φ2) that select between normal mode and auto-zeroing mode. The clock is derived from an internal R-C oscillator running at a rate of fOSC1 = 650 kHz. The oscillator’s output is divided down to the desired rate. It is also randomized to minimize (spread) undesired clock tones in the output. © 2008 Microchip Technology Inc. DS22093B-page 21 MCP6V06/7/8 4.1.2 AUTO-ZEROING ACTION Figure 4-2 shows the connections between amplifiers during the Normal Mode of operation (φ1). The hold capacitor (CH) corrects the Null Amplifier’s input offset. Since the Null Amplifier has very high gain, it dominates the signal seen by the Main Amplifier. This greatly reduces the impact of the Main Amplifier’s input offset voltage on overall performance. Essentially, the Null Amplifier and Main Amplifier behave as a regular op amp with very high gain (AOL) and very low offset voltage (VOS). VIN+ VIN– CFW CH Main Amp. NC Output Buffer VOUT VREF Null Amp. Normal Mode of Operation (φ1); Equivalent Amplifier Diagram. FIGURE 4-2: Figure 4-3 shows the connections between amplifiers during the Auto-zeroing Mode of operation (φ2). The signal goes directly through the Main Amplifier, and the flywheel capacitor (CFW) maintains a constant correction on the Main Amplifier’s offset. Since these corrections happen every 50 µs, or so, we also minimize slow errors, including offset drift with temperature (ΔVOS/ΔTA), 1/f noise, and input offset aging. The Null Amplifier uses its own high open loop gain to drive the voltage across CH to the point where its input offset voltage is almost zero. Because the principal input is connected to VIN+, the auto-zeroing action corrects the offset at the current common mode input voltage (VCM) and supply voltage (VDD). This makes the DC CMRR and PSRR very high also. VIN+ VIN– CFW CH FIGURE 4-3: 4.1.3 Null Amp. Main Amp. NC Output Buffer VOUT VREF Auto-zeroing Mode of Operation (φ2); Equivalent Diagram. INTERMODULATION DISTORTION (IMD) The MCP6V06/7/8 op amps will show intermodulation distortion (IMD), products when an AC signal is present. frequencies. IMD distortion tones are generated about all of the square wave clock’s harmonics. See Figure 2-37 and Figure 2-38. The signal and clock can be decomposed into sine wave tones (Fourier series components). These tones interact with the auto-zeroing circuitry’s non-linear response to produce IMD tones at sum and difference DS22093B-page 22 © 2008 Microchip Technology Inc. MCP6V06/7/8 4.2 Other Functional Blocks 4.2.1 RAIL-TO-RAIL INPUTS The input stage of the MCP6V06/7/8 op amps uses two differential CMOS input stages in parallel. One operates at low common mode input voltage (VCM, which is approximately equal to VIN+ and VIN– in normal operation) and the other at high VCM. With this topology, the input operates with VCM up to 0.2V past either supply rail at +25°C (see Figure 2-18). The input offset voltage (VOS) is measured at VCM = VSS – 0.2V and VDD + 0.2V to ensure proper operation. The transition between the input stages occurs when VCM ≈ VDD – 0.9V (see Figure 2-7 and Figure 2-8). For the best distortion and gain linearity, with non-inverting gains, avoid this region of operation. 220.127.116.11 VDD D1 V1 R1 Input Voltage and Current Limits The ESD protection on the inputs can be depicted as shown in Figure 4-4. This structure was chosen to protect the input transistors, and to minimize input bias current (IB). The input ESD diodes clamp the inputs when they try to go more than one diode drop below VSS. They also clamp any voltages that go too far above VDD; their breakdown voltage is high enough to allow normal operation, and low enough to bypass quick ESD events within the specified limits. VIN+ Bond Pad Input Stage Bond V – IN Pad FIGURE 4-4: Structures. Simplified Analog Input ESD In order to prevent damage and/or improper operation of these amplifiers, the circuit must limit the currents (and voltages) at the input pins (see Section 1.1 “Absolute Maximum Ratings †”). Figure 4-5 shows the recommended approach to protecting these inputs. The internal ESD diodes prevent the input pins (VIN+ and VIN–) from going too far below ground, and the resistors R1 and R2 limit the possible current drawn out of the input pins. Diodes D1 and D2 prevent the input © 2008 Microchip Technology Inc. VOUT VSS – (minimum expected V1) 2 mA VSS – (minimum expected V2) R2 > 2 mA R1 > FIGURE 4-5: Inputs. Protecting the Analog It is also possible to connect the diodes to the left of the resistor R1 and R2. In this case, the currents through the diodes D1 and D2 need to be limited by some other mechanism. The resistors then serve as in-rush current limiters; the DC current into the input pins (VIN+ and VIN–) should be very small. A significant amount of current can flow out of the inputs (through the ESD diodes) when the common mode voltage (VCM) is below ground (VSS); see Figure 2-17. Applications that are high impedance may need to limit the usable voltage range. RAIL-TO-RAIL OUTPUT The output voltage range of the MCP6V06/7/8 auto-zeroed op amps is VDD – 15 mV (minimum) and VSS + 15 mV (maximum) when RL = 20 kΩ is connected to VDD/2 and VDD = 5.5V. Refer to Figure 2-19 and Figure 2-20 for more information. These op amps are designed to drive light loads; use another amplifier to buffer the output from heavy loads. 4.2.3 VSS Bond Pad MCP6V0X R2 4.2.2 VDD Bond Pad D2 V2 Phase Reversal The input devices are designed to not exhibit phase inversion when the input pins exceed the supply voltages. Figure 2-43 shows an input voltage exceeding both supplies with no phase inversion. 18.104.22.168 pins (VIN+ and VIN–) from going too far above VDD, and dump any currents onto VDD. When implemented as shown, resistors R1 and R2 also limit the current through D1 and D2. CHIP SELECT (CS) The single MCP6V08 has a Chip Select (CS) pin. When CS is pulled high, the supply current for the corresponding op amp drops to about 1 µA (typical), and is pulled through the CS pin to VSS. When this happens, the amplifier is put into a high impedance state. By pulling CS low, the amplifier is enabled. If the CS pin is left floating, the internal pull-down resistor (about 5 MΩ) will keep the part on. Figure 1-4 shows the output voltage and supply current response to a CS pulse. DS22093B-page 23 MCP6V06/7/8 4.3 Application Tips 4.3.1 4.3.4 INPUT OFFSET VOLTAGE OVER TEMPERATURE Table 1-1 gives both the linear and quadratic temperature coefficients (TC1 and TC2) of input offset voltage. The input offset voltage, at any temperature in the specified range, can be calculated as follows: EQUATION 4-1: V OS ( T A ) = V OS + TC 1 ΔT + TC 2 ΔT 2 Where: ΔT = TA – 25°C VOS(TA) = input offset voltage at TA VOS = input offset voltage at +25°C TC1 = linear temperature coefficient TC2 = quadratic temperature coefficient 4.3.2 DC GAIN PLOTS SOURCE CAPACITANCE The capacitances seen by the two inputs should be small and matched. The internal switches connected to the inputs dump charges on these capacitors; an offset can be created if the capacitances do not match. 4.3.5 CAPACITIVE LOADS Driving large capacitive loads can cause stability problems for voltage feedback op amps. As the load capacitance increases, the feedback loop’s phase margin decreases and the closed-loop bandwidth is reduced. This produces gain peaking in the frequency response, with overshoot and ringing in the step response. These auto-zeroed op amps have a different output impedance than most op amps, due to their unique topology. When driving a capacitive load with these op amps, a series resistor at the output (RISO in Figure 4-6) improves the feedback loop’s phase margin (stability) by making the output load resistive at higher frequencies. The bandwidth will be generally lower than the bandwidth with no capacitive load. Figure 2-9, Figure 2-10 and Figure 2-11 are histograms of the reciprocals (in units of µV/V) of CMRR, PSRR and AOL, respectively. They represent the change in input offset voltage (VOS) with a change in common mode input voltage (VCM), power supply voltage (VDD) and output voltage (VOUT). 4.3.3 SOURCE RESISTANCES The input bias currents have two significant components; switching glitches that dominate at room temperature and below, and input ESD diode leakage currents that dominate at +85°C and above. Make the resistances seen by the inputs small and equal. This minimizes the output offset caused by the input bias currents. The inputs should see a resistance on the order of 10Ω to 1 kΩ at high frequencies (i.e., above 1 MHz). This helps minimize the impact of switching glitches, which are very fast, on overall performance. In some cases, it may be necessary to add resistors in series with the inputs to achieve this improvement in performance. DS22093B-page 24 VOUT CL MCP6V0X FIGURE 4-6: Output Resistor, RISO, Stabilizes Capacitive Loads. Figure 4-7 gives recommended RISO values for different capacitive loads and is independent of the gain. 10000 10k Recommended R ISO (Ω) The 1/AOL histogram is centered near 0 µV/V because the measurements are dominated by the op amp’s input noise. The negative values shown represent noise, not unstable behavior. We validate the op amps’ stability by making multiple measurements of VOS; instability would manifest itself as a greater unexplained variability in VOS or as the railing of the output. RISO GN < 2 1k 1000 100 100 10 10 1p 1.E-12 GN = 5 GN = 10 10p 1.E-11 100p 1n 1.E-10 1.E-09 CL (F) 10n 1.E-08 100n 1.E-07 FIGURE 4-7: Recommended RISO values for Capacitive Loads. © 2008 Microchip Technology Inc. MCP6V06/7/8 After selecting RISO for your circuit, double check the resulting frequency response peaking and step response overshoot. Modify RISO's value until the response is reasonable. Bench evaluation and simulations with the MCP6V06 SPICE macro model (good for all of the MCP6V06/7/8 op amps) are helpful. 4.3.6 STABILIZING OUTPUT LOADS This family of auto-zeroed op amps has an output impedance (Figure 2-31 and Figure 2-32) that has a double zero when the gain is low. This can cause a large phase shift in feedback networks that have low resistance near the part’s bandwidth. This large phase shift can cause stability problems. Figure 4-8 shows one circuit example that has low resistance near the part’s bandwidth. RF and CF set a pole at 0.16 kHz, so the noise gain (GN) is 1 V/V at the circuit’s bandwidth (roughly 1.3 MHz). The load seen by the op amp’s output at 1.3 MHz is RG||RL (99Ω). This is low enough to be a real concern. VIN RN 100Ω MCP6V0X RG 100Ω RF 10.0 kΩ VOUT RL 10.0 kΩ CF 0.1 µF FIGURE 4-8: Output Load Issue. To solve this problem, increase the resistive load to at least 3 kΩ. Methods to accomplish this task include: • Increase RG • Remove CF (relocate the filter) • Add a 3 kΩ resistor at the op amp’s output that is not in the signal path; see Figure 4-9 VIN RN 100 Ω MCP6V0X RG 100 Ω RF 10.0 kΩ RX 3.01 kΩ VOUT RL 10.0 kΩ CF 0.1 µF FIGURE 4-9: Load Issue. 4.3.7 REDUCING UNDESIRED NOISE AND SIGNALS Reduce undesired noise and signals with: • Low bandwidth signal filters: - Minimizes random analog noise - Reduces interfering signals • Good PCB layout techniques: - Minimizes crosstalk - Minimizes parasitic capacitances and inductances that interact with fast switching edges • Good power supply design: - Isolation from other parts - Filtering of interference on supply line(s) 4.3.8 SUPPLY BYPASSING AND FILTERING With this family of operational amplifiers, the power supply pin (VDD for single supply) should have a local bypass capacitor (i.e., 0.01 µF to 0.1 µF) within 2 mm of the pin for good high-frequency performance. These parts also need a bulk capacitor (i.e., 1 µF or larger) within 100 mm to provide large, slow currents. This bulk capacitor can be shared with other low noise, analog parts. Additional filtering of high frequency power supply noise (e.g., switched mode power supplies) can be achieved using resistors. The resistors need to be small enough to prevent a large drop in VDD for the op amp, which would cause a reduced output range and possible load-induced power supply noise. The resistors also need to be large enough to dissipate little power when VDD is turned on and off quickly. The circuit in Figure 4-10 gives good rejection out to 1 MHz for switched mode power supplies. Smaller resistors and capacitors are a better choice for designs where the power supply is reasonably quiet. VS_ANA 143Ω 1/4W 143Ω 1/10W 100 µF 100 µF to other analog parts FIGURE 4-10: 0.1 µF MCP6V0X Additional Supply Filtering. One Solution To Output © 2008 Microchip Technology Inc. DS22093B-page 25 MCP6V06/7/8 4.3.9 PCB DESIGN FOR DC PRECISION In order to achieve DC precision on the order of ±1 µV, many physical errors need to be minimized. The design of the Printed Circuit Board (PCB), the wiring, and the thermal environment has a strong impact on the precision achieved. A poor PCB design can easily be more than 100 times worse than the MCP6V06/7/8 op amps minimum and maximum specifications. 22.214.171.124 Thermo-junctions Any time two dissimilar metals are joined together, a temperature dependent voltage appears across the junction (the Seebeck or thermo-junction effect). This effect is used in thermocouples to measure temperature. The following are examples of thermo-junctions on a PCB: • Components (resistors, op amps, …) soldered to a copper pad • Wires mechanically attached to the PCB • Jumpers • Solder joints • PCB vias 126.96.36.199 Non-inverting and Inverting Amplifier Layout for Thermo-junctions Figure 4-11 shows the recommended non-inverting and inverting gain amplifier circuits on one schematic. Usually, to minimize the input bias current related offset, R1 is chosen to be R2||R3. The guard traces (with ground vias at the ends) help minimize the thermal gradients. The resistor layout cancels the resistor thermal voltages, assuming the temperature gradient is constant near the resistors: EQUATION 4-2: VOUT ≈ VPGP, ≈ -VMGM, • Minimize thermal gradients • Cancel thermo-junction voltages • Minimize difference in thermal potential between metals VP = GND Where: GM = R3/R2, inverting gain magnitude GP = 1 + GM, non-inverting gain magnitude VOS is neglected Typical thermo-junctions have temperature to voltage conversion coefficients of 10 to 100 µV/°C (sometimes higher). There are three basic approaches to minimizing thermo-junction effects: VM = GND R3 R2 VM VP U1 VOUT R1 R2 R3 VM U1 MCP6V06 VP VOUT R1 FIGURE 4-11: PCB Layout and Schematic for Single Non-inverting and Inverting Amplifiers. Note: DS22093B-page 26 Changing the orientation of the resistors will usually cause a significant decrease in the cancellation of the thermal voltages. © 2008 Microchip Technology Inc. MCP6V06/7/8 188.8.131.52 Difference Amplifier Layout for Thermo-junctions Figure 4-12 shows the recommended difference amplifier circuit. Usually, we choose R1 = R2 and R3 = R4. The guard traces (with ground vias at the ends) help minimize the thermal gradients. The resistor layout cancels the resistor thermal voltages, assuming the temperature gradient is constant near the resistors: VOUT ≈ VREF + (VP – VM)GDM Where: The dual op amp amplifiers shown in Figure 4-16 and Figure 4-17 produce a non-inverting difference gain greater than 1, and a common mode gain of 1 .They can use the layout shown in Figure 4-13. The gain setting resistors (R2) between the two sides are not combined so that the thermal voltages can be canceled. EQUATION 4-4: Thermal voltages are approximately equal = Dual Non-inverting Amplifier Layout for Thermo-junctions The guard traces (with ground vias at the ends) help minimize the thermal gradients. The resistor layout cancels the resistor thermal voltages, assuming the temperature gradient is constant near the resistors: EQUATION 4-3: GDM 184.108.40.206 R3/R1 = R4/R2, difference gain VOS is neglected Where: (VOA – VOB) ≈ (VIA – VIB)GDM (VOA + VOB)/2 ≈ (VIA + VIB)/2 Thermal voltages are approximately equal R4 R2 VM VP VOUT U1 = 1 + R3/R2, differential mode gain 1, common mode gain VOB VOA R3 R4 U1 MCP6V06 U1 R3 R2 VOUT R2 R1 R1 VREF R1 R3 FIGURE 4-12: PCB Layout and Schematic for Single Difference Amplifier. Note: GCM VREF VM VP = VOS is neglected R1 R3 R2 GDM VIA VIA VIB R1 VOA ½ MCP6V07 U1 Changing the orientation of the resistors will usually cause a significant decrease in the cancellation of the thermal voltages. R2 R3 R2 R3 U1 ½ MCP6V07 VIB VOB R1 FIGURE 4-13: PCB Layout and Schematic for Dual Non-inverting Amplifier. Note: © 2008 Microchip Technology Inc. Changing the orientation of the resistors will usually cause a significant decrease in the cancellation of the thermal voltages. DS22093B-page 27 MCP6V06/7/8 220.127.116.11 Other PCB Thermal Design Tips In cases where an individual resistor needs to have its thermo-junction voltage cancelled, it can be split into two equal resistors as shown in Figure 4-14. To keep the thermal gradients near the resistors as small as possible, the layouts are symmetrical with a ring of metal around the outside. Make R1A = R1B = R1/2 and R2A = R2B = 2R2. R1A R1B R2A R2B R1A R1B R2A R2B FIGURE 4-14: Resistors. Note: PCB Layout for Individual Changing the orientation of the resistors will usually cause a significant decrease in the cancellation of the thermal voltages. Minimize temperature gradients at critical components (resistors, op amps, heat sources, etc.): • Minimize exposure to gradients - Small components - Tight spacing - Shield from air currents • Align with constant temperature (contour) lines - Place on PCB center line • Minimize magnitude of gradients - Select parts with lower power dissipation - Use same metal junctions on thermo-junctions that need to match - Use metal junctions with low temperature to voltage coefficients - Large distance from heat sources - Ground plane underneath (large area) - FR4 gaps (no copper for thermal insulation) - Series resistors inserted into traces (adds thermal and electrical resistance) - Use heat sinks 18.104.22.168 Crosstalk DC crosstalk causes offsets that appear as a larger input offset voltage. Common causes include: • Common mode noise (remote sensors) • Ground loops (current return paths) • Power supply coupling Interference from the mains (usually 50 Hz or 60 Hz), and other AC sources, can also affect the DC performance. Non-linear distortion can convert these signals to multiple tones, included a DC shift in voltage. When the signal is sampled by an ADC, these AC signals can also be aliased to DC, causing an apparent shift in offset. To reduce interference: - Keep traces and wires as short as possible Use shielding (e.g., encapsulant) Use ground plane (at least a star ground) Place the input signal source near to the DUT Use good PCB layout techniques Use a separate power supply filter (bypass capacitors) for these auto-zeroed op amps 22.214.171.124 Miscellaneous Effects Keep the resistances seen by the input pins as small and as near to equal as possible to minimize bias current related offsets. Make the (trace) capacitances seen by the input pins small and equal. This is helpful in minimizing switching glitch-induced offset voltages. Bending a coax cable with a radius that is too small causes a small voltage drop to appear on the center or (the tribo-electric effect). Make sure the bending radius is large enough to keep the conductors and insulation in full contact. Mechanical stresses can make some capacitor types (such as ceramic) to output small voltages. Use more appropriate capacitor types in the signal path and minimize mechanical stresses and vibration. Humidity can cause electro-chemical potential voltages to appear in a circuit. Proper PCB cleaning helps, as does the use of encapsulants. Make the temperature gradient point in one direction: • Add guard traces - Constant temperature curves follow the traces - Connect to ground plane • Shape any FR4 gaps - Constant temperature curves follow the edges DS22093B-page 28 © 2008 Microchip Technology Inc. MCP6V06/7/8 4.4 Typical Applications 4.4.1 4.4.2 WHEATSTONE BRIDGE Many sensors are configured as Wheatstone bridges. Strain gauges and pressure sensors are two common examples. These signals can be small and the common mode noise large. Amplifier designs with high differential gain are desirable. RTD SENSOR The ratiometric circuit in Figure 4-17 conditions a three wire RTD. It corrects for the sensor’s wiring resistance by subtracting the voltage across the middle RW. The top R1 does not change the output voltage; it balances the op amp inputs. Failure (open) of the RTD is detected by an out-of-range voltage. Figure 4-15 shows how to interface to a Wheatstone bridge with a minimum of components. Because the circuit is not symmetric, the ADC input is single ended, and there is a minimum of filtering, the CMRR is good enough for moderate common mode noise. 0.01C VDD R R R 0.2R R 3 kΩ 100R VDD ½ MCP6V07 2.49 kΩ VDD RW ADC 10 nF RRTD 100Ω 0.2R MCP6V06 FIGURE 4-15: Simple Design. R1 2.49 kΩ 1 µF 10 nF RW Figure 4-16 shows a higher performance circuit for Wheatstone bridges. This circuit is symmetric and has high CMRR. Using a differential input to the ADC helps with the CMRR. 100 nF RT 20 kΩ R1 2.49 kΩ RB 20 kΩ R3 100 kΩ R2 2.55 kΩ VDD R R R3 100 kΩ 2.49 kΩ ½ MCP6V07 R R 3 kΩ 1 µF 200Ω 10 nF VDD ADC 3 kΩ 20 kΩ 1 µF 200 Ω ½ MCP6V07 FIGURE 4-16: RTD Sensor. The voltages at the input of the ADC can be calculated with the following: 20 kΩ 200Ω 3 kΩ 100 nF RW FIGURE 4-17: 1 µF 10 nF ADC R2 2.55 kΩ ½ MCP6V07 200 Ω 3 kΩ VDD G RTD = 1 + 2 ⋅ R 3 ⁄ R 2 G W = G RTD – R 3 ⁄ R 1 V DM = G RTD ( V T – V B ) + G W V W V T + V B + ( G RTD + 1 – G W )V W V CM = -----------------------------------------------------------------------------2 Where: VT = Voltage at the top of RRTD VB = Voltage at the bottom of RRTD VW = Voltage across top and middle RW’s VCM = ADC’s common mode input VDM = ADC’s differential mode input High Performance Design. © 2008 Microchip Technology Inc. DS22093B-page 29 MCP6V06/7/8 4.4.3 THERMOCOUPLE SENSOR Figure 4-18 shows a simplified diagram of an amplifier and temperature sensor used in a thermocouple application. The type K thermocouple senses the temperature at the hot junction (THJ), and produces a voltage at V1 proportional to THJ (in °C). The amplifier’s gain is is set so that V4/THJ is 10 mV/°C. V3 represents the output of a temperature sensor, which produces a voltage proportional to the temperature (in °C) at the cold junction (TCJ), and with a 0.50V offset. V2 is set so that V4 is 0.50V when THJ – TCJ is 0°C. EQUATION 4-5: V1 ≈ THJ(40 µV/°C) V2 = (1.00V) The MCP9700A senses the temperature at its physical location. It needs to be at the same temperature as the cold junction (TCJ), and produces V3 (Figure 4-16). The MCP1541 produces a 4.10V output, assuming VDD is at 5.0V. This voltage, tied to a resistor ladder of 4.100(RTH) and 1.3224(RTH), would produce a Thevenin equivalent of 1.00V and 250(RTH). The 1.3224(RTH) resistor is combined in parallel with the top right RTH resistor (in Figure 4-18), producing the 0.5696(RTH) resistor. V4 should be converted to digital, then corrected for the thermocouple’s non-linearity. The ADC can use the MCP1541 as its voltage reference. Alternately, an absolute reference inside a PICmicro® can be used instead of the MCP1541. V3 = TCJ(10 mV/°C) + (0.50V) 4.4.4 V4 = 250V1 + (V2 – V3) Figure 4-20 shows a MCP6V06 correcting the input offset voltage of another op amp. R2 and C2 integrate the offset error seen at the other op amp’s input; the integration needs to be slow enough to be stable (with the feedback provided by R1 and R3). ≈ (10 mV/°C) (THJ – TCJ) + (0.50V) (hot junction RTH = Thevenin Equivalent Resistance at THJ) (RTH) (RTH) V2 40 µV/°C C Type K Thermocouple (RTH)/250 MCP6V06 V1 V4 (RTH)/250 (cold junction C at TCJ) V3 (RTH) (RTH) FIGURE 4-18: Simplified Circuit. Thermocouple Sensor; VIN R1 R3 R2 VOUT C2 3 kΩ R2 VDD/2 MCP6XXX MCP6V06 FIGURE 4-20: 4.4.5 Figure 4-19 shows a more complete implementation of this circuit. The dashed red arrow indicates a thermally conductive connection between the thermocouple and the MCP9700A; it needs to be very short and have low thermal resistance. OFFSET VOLTAGE CORRECTION Offset Correction. PRECISION COMPARATOR Use high gain before a comparator to improve the latter’s performance. Do not use MCP6V06/7/8 as a comparator by itself; the VOS correction circuitry does not operate properly without a feedback loop. MCP6V06 RTH = Thevenin Equivalent Resistance (e.g., 10 kΩ) VDD 4.100(RTH) 0.5696(RTH) MCP1541 C (R )/250 TH Type K MCP6V06 V1 VDD VIN R1 R2 FIGURE 4-19: DS22093B-page 30 R5 1 kΩ VOUT MCP6541 C (RTH) R4 VDD/2 (RTH)/250 MCP9700A R3 (RTH) 3 kΩ V4 FIGURE 4-21: Precision Comparator. Thermocouple Sensor. © 2008 Microchip Technology Inc. MCP6V06/7/8 5.0 DESIGN AIDS Microchip provides the basic design aids needed for the MCP6V06/7/8 family of op amps. 5.1 SPICE Macro Model The latest SPICE macro model for the MCP6V06/7/8 op amps is available on the Microchip web site at www.microchip.com. This model is intended to be an initial design tool that works well in the op amp’s linear region of operation over the temperature range. See the model file for information on its capabilities. Bench testing is a very important part of any design and cannot be replaced with simulations. Also, simulation results using this macro model need to be validated by comparing them to the data sheet specifications and characteristic curves. 5.2 FilterLab® Software Microchip’s FilterLab® software is an innovative software tool that simplifies analog active filter (using op amps) design. Available at no cost from the Microchip web site at www.microchip.com/filterlab, the Filter-Lab design tool provides full schematic diagrams of the filter circuit with component values. It also outputs the filter circuit in SPICE format, which can be used with the macro model to simulate actual filter performance. 5.3 Mindi™ Circuit Designer & Simulator 5.5 Analog Demonstration and Evaluation Boards Microchip offers a broad spectrum of Analog Demonstration and Evaluation Boards that are designed to help customers achieve faster time to market. For a complete listing of these boards and their corresponding user’s guides and technical information, visit the Microchip web site at www.microchip.com/analog tools. Some boards that are especially useful are: • MCP6V01 Thermocouple Auto-Zeroed Reference Design • MCP6XXX Amplifier Evaluation Board 1 • MCP6XXX Amplifier Evaluation Board 2 • MCP6XXX Amplifier Evaluation Board 3 • MCP6XXX Amplifier Evaluation Board 4 • Active Filter Demo Board Kit • P/N SOIC8EV: 8-Pin SOIC/MSOP/TSSOP/DIP Evaluation Board • P/N SOIC14EV: 14-Pin SOIC/TSSOP/DIP Evaluation Board 5.6 Application Notes The following Microchip Application Notes are available on the Microchip web site at www.microchip. com/appnotes and are recommended as supplemental reference resources. ADN003: “Select the Right Operational Amplifier for your Filtering Circuits”, DS21821 Microchip’s Mindi™ Circuit Designer & Simulator aids in the design of various circuits useful for active filter, amplifier and power management applications. It is a free online circuit designer & simulator available from the Microchip web site at www.microchip.com/mindi. This interactive circuit designer & simulator enables designers to quickly generate circuit diagrams, and simulate circuits. Circuits developed using the Mindi Circuit Designer & Simulator can be downloaded to a personal computer or workstation. AN722: “Operational Amplifier Topologies and DC Specifications”, DS00722 5.4 “Signal Chain Design Guide”, DS21825 Microchip Advanced Part Selector (MAPS) AN723: “Operational Amplifier AC Specifications and Applications”, DS00723 AN884: “Driving Capacitive Loads With Op Amps”, DS00884 AN990: “Analog Sensor Conditioning Circuits – An Overview”, DS00990 These application notes and others are listed in the design guide: MAPS is a software tool that helps efficiently identify Microchip devices that fit a particular design requirement. Available at no cost from the Microchip website at www.microchip.com/maps, the MAPS is an overall selection tool for Microchip’s product portfolio that includes Analog, Memory, MCUs and DSCs. Using this tool, a customer can define a filter to sort features for a parametric search of devices and export side-by-side technical comparison reports. Helpful links are also provided for Data sheets, Purchase and Sampling of Microchip parts. © 2008 Microchip Technology Inc. DS22093B-page 31 MCP6V06/7/8 6.0 PACKAGING INFORMATION 6.1 Package Marking Information Example 8-Lead DFN (4x4) (MCP6V07) XXXXXX XXXXXX YYWW NNN 6V07 e3 E/MD^^ 0823 256 Example: 8-Lead SOIC (150 mil) XXXXXXXX XXXXYYWW MCP6VO6E SN e3 0823 256 NNN Example: 8-Lead TDFN (2x3) (MCP6V06, MCP6V08) XXX YWW NN Device Code MCP6V06 AAC MCP6V08 AAD AAC 838 25 Note: Applies to 8-Lead 2x3 TDFN Legend: XX...X Y YY WW NNN e3 * Note: DS22093B-page 32 Customer-specific information Year code (last digit of calendar year) Year code (last 2 digits of calendar year) Week code (week of January 1 is week ‘01’) Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn) This package is Pb-free. The Pb-free JEDEC designator ( e3 ) can be found on the outer packaging for this package. In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information. © 2008 Microchip Technology Inc. MCP6V06/7/8 /HDG3ODVWLF'XDO)ODW1R/HDG3DFNDJH0'±[[PP%RG\>')1@ 1RWH )RUWKHPRVWFXUUHQWSDFNDJHGUDZLQJVSOHDVHVHHWKH0LFURFKLS3DFNDJLQJ6SHFLILFDWLRQORFDWHGDW KWWSZZZPLFURFKLSFRPSDFNDJLQJ D e b N N L E E2 K EXPOSED PAD 2 2 1 NOTE 1 1 NOTE 1 D2 TOP VIEW BOTTOM VIEW A3 A A1 NOTE 2 8QLWV 'LPHQVLRQ/LPLWV 1XPEHURI3LQV 0,//,0(7(56 0,1 1 120 0$; 3LWFK H 2YHUDOO+HLJKW $ 6WDQGRII $ &RQWDFW7KLFNQHVV $ 2YHUDOO/HQJWK ' ([SRVHG3DG:LGWK ( 2YHUDOO:LGWK ( ([SRVHG3DG/HQJWK %6& 5() %6& %6& ' &RQWDFW:LGWK E &RQWDFW/HQJWK / &RQWDFWWR([SRVHG3DG . ± ± 1RWHV 3LQYLVXDOLQGH[IHDWXUHPD\YDU\EXWPXVWEHORFDWHGZLWKLQWKHKDWFKHGDUHD 3DFNDJHPD\KDYHRQHRUPRUHH[SRVHGWLHEDUVDWHQGV 3DFNDJHLVVDZVLQJXODWHG 'LPHQVLRQLQJDQGWROHUDQFLQJSHU$60(<0 %6& %DVLF'LPHQVLRQ7KHRUHWLFDOO\H[DFWYDOXHVKRZQZLWKRXWWROHUDQFHV 5() 5HIHUHQFH'LPHQVLRQXVXDOO\ZLWKRXWWROHUDQFHIRULQIRUPDWLRQSXUSRVHVRQO\ 0LFURFKLS 7HFKQRORJ\ 'UDZLQJ && © 2008 Microchip Technology Inc. DS22093B-page 33 MCP6V06/7/8 /HDG3ODVWLF6PDOO2XWOLQH61±1DUURZPP%RG\>62,&@ 1RWH )RUWKHPRVWFXUUHQWSDFNDJHGUDZLQJVSOHDVHVHHWKH0LFURFKLS3DFNDJLQJ6SHFLILFDWLRQORFDWHGDW KWWSZZZPLFURFKLSFRPSDFNDJLQJ D e N E E1 NOTE 1 1 2 3 α h b h A2 A c φ L A1 L1 8QLWV 'LPHQVLRQ/LPLWV 1XPEHURI3LQV β 0,//,0(7(56 0,1 1 120 0$; 3LWFK H 2YHUDOO+HLJKW $ ± %6& ± 0ROGHG3DFNDJH7KLFNQHVV $ ± ± 6WDQGRII $ ± 2YHUDOO:LGWK ( 0ROGHG3DFNDJH:LGWK ( %6& 2YHUDOO/HQJWK ' %6& %6& &KDPIHURSWLRQDO K ± )RRW/HQJWK / ± )RRWSULQW / 5() )RRW$QJOH ± /HDG7KLFNQHVV F ± /HDG:LGWK E ± 0ROG'UDIW$QJOH7RS ± 0ROG'UDIW$QJOH%RWWRP ± 1RWHV 3LQYLVXDOLQGH[IHDWXUHPD\YDU\EXWPXVWEHORFDWHGZLWKLQWKHKDWFKHGDUHD 6LJQLILFDQW&KDUDFWHULVWLF 'LPHQVLRQV'DQG(GRQRWLQFOXGHPROGIODVKRUSURWUXVLRQV0ROGIODVKRUSURWUXVLRQVVKDOOQRWH[FHHGPPSHUVLGH 'LPHQVLRQLQJDQGWROHUDQFLQJSHU$60(<0 %6& %DVLF'LPHQVLRQ7KHRUHWLFDOO\H[DFWYDOXHVKRZQZLWKRXWWROHUDQFHV 5() 5HIHUHQFH'LPHQVLRQXVXDOO\ZLWKRXWWROHUDQFHIRULQIRUPDWLRQSXUSRVHVRQO\ 0LFURFKLS 7HFKQRORJ\ 'UDZLQJ &% DS22093B-page 34 © 2008 Microchip Technology Inc. MCP6V06/7/8 /HDG3ODVWLF6PDOO2XWOLQH61±1DUURZPP%RG\>62,&@ 1RWH )RUWKHPRVWFXUUHQWSDFNDJHGUDZLQJVSOHDVHVHHWKH0LFURFKLS3DFNDJLQJ6SHFLILFDWLRQORFDWHGDW KWWSZZZPLFURFKLSFRPSDFNDJLQJ © 2008 Microchip Technology Inc. DS22093B-page 35 MCP6V06/7/8 /HDG3ODVWLF'XDO)ODW1R/HDG3DFNDJH01±[[PP%RG\>7')1@ 1RWH )RUWKHPRVWFXUUHQWSDFNDJHGUDZLQJVSOHDVHVHHWKH0LFURFKLS3DFNDJLQJ6SHFLILFDWLRQORFDWHGDW KWWSZZZPLFURFKLSFRPSDFNDJLQJ DS22093B-page 36 © 2008 Microchip Technology Inc. MCP6V06/7/8 /HDG3ODVWLF'XDO)ODW1R/HDG3DFNDJH01±[[PP%RG\>7')1@ 1RWH )RUWKHPRVWFXUUHQWSDFNDJHGUDZLQJVSOHDVHVHHWKH0LFURFKLS3DFNDJLQJ6SHFLILFDWLRQORFDWHGDW KWWSZZZPLFURFKLSFRPSDFNDJLQJ © 2008 Microchip Technology Inc. DS22093B-page 37 MCP6V06/7/8 NOTES: DS22093B-page 38 © 2008 Microchip Technology Inc. MCP6V06/7/8 APPENDIX A: REVISION HISTORY Revision B (December 2008) The following is the list of modifications: 1. 2. 3. 4. 5. Added the 8-lead, 2x3 TDFN package for the MCP6V01 and MCP6V03 devices. Added 8-lead, 2x3 TDFN package information to Thermal Characteristic table. Added information on the Exposed Thermal Pad (EP) for the 8-lead, 2x3 TDFN and 8-lead, 4x4 DFN packages. Added Section 4.3.6 “Stabilizing Output Loads”. Other minor typographical corrections. Revision A (June 2008) • Original Release of this Document. © 2008 Microchip Technology Inc. DS22093B-page 39 MCP6V06/7/8 APPENDIX B: OFFSET RELATED TEST SCREENS We use production screens to ensure the quality of our outgoing products. These screens are set at wider limits to eliminate any fliers; see Table B-1. Input offset voltage related specifications in the DC spec table (Table 1-1) are based on bench measurements (see Section 2.1 “DC Input Precision”). These measurements are much more accurate because: • • • • More compact circuit Soldered parts on the PCB More time spent averaging (reduces noise) Better temperature control - Reduced temperature gradients - Greater accuracy TABLE B-1: OFFSET RELATED TEST SCREENS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +1.8V to +5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 20 kΩ to VL, and CS = GND (refer to Figure 1-5 and Figure 1-6). Parameters Sym Min Max Units Conditions Input Offset Voltage VOS -10 +10 Input Offset Voltage Drift with Temperature (linear Temp. Co.) TC1 — — PSRR 115 — dB (Note 1) CMRR 106 — dB VDD = 1.8V, VCM = -0.2V to 2.0V (Note 1) CMRR 116 — dB VDD = 5.5V, VCM = -0.2V to 5.7V (Note 1) AOL 114 — dB VDD = 1.8V, VOUT = 0.2V to 1.6V (Note 1) AOL 122 — dB VDD = 5.5V, VOUT = 0.2V to 5.3V (Note 1) Input Offset Power Supply Rejection µV TA = +25°C (Note 1, Note 2) nV/°C TA = -40 to +125°C (Note 3) Common Mode Common Mode Rejection Open-Loop Gain DC Open-Loop Gain (large signal) Note 1: 2: 3: Due to thermal junctions and other errors in the production environment, these specifications are only screened in production. VOS is also sample screened at +125°C. TC1 is not measured in production. DS22093B-page 40 © 2008 Microchip Technology Inc. MCP6V06/7/8 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. PART NO. –X /XXX Device Temperature Range Package Device: MCP6V06 Single Op Amp MCP6V06T Single Op Amp (Tape and Reel for 2x3 TDFN and SOIC) MCP6V07 Dual Op Amp MCP6V07T Dual Op Amp (Tape and Reel for 4×4 DFN and SOIC) MCP6V08 Single Op Amp with Chip Select MCP6V08T Single Op Amp with Chip Select (Tape and Reel for SOIC) Temperature Range: E Package: MD Examples: a) b) a) MCP6V06T-E/SN: Extended temperature, 8LD SOIC package. MCP6V06-E/MNY: Extended temperature, 8LD 2x3 TDFN package. b) Extended temperature, 8LD 4x4 DFN package.. MCP6V07T-E/SN: Tape and Reel, Extended temperature, 8LD SOIC package. a) MCP6V08-E/SN: b) MCP6V07-E/MD: Extended temperature, 8LD SOIC package. MCP6V08-E/MNY:Extended temperature, 8LD 2x3 TDFN package. = -40°C to +125°C = Plastic Dual Flat, No-Lead (4×4x0.9 mm), 8-lead (MCP6V07 only) MNY * = Plastic Dual Flat, No-Lead (2×3x0.75 mm), 8-lead (MCP6V06, MCP6V08) SN = Plastic SOIC (150mil Body), 8-lead * Y = nickel palladium gold manufacturing designator. Only available on the TDFN package. © 2008 Microchip Technology Inc. DS22093B-page 41 MCP6V06/7/8 NOTES: DS22093B-page 42 © 2008 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART, rfPIC, SmartShunt and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. FilterLab, Linear Active Thermistor, MXDEV, MXLAB, SEEVAL, SmartSensor and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, In-Circuit Serial Programming, ICSP, ICEPIC, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, mTouch, PICkit, PICDEM, PICDEM.net, PICtail, PIC32 logo, PowerCal, PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, Select Mode, Total Endurance, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2008, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. © 2008 Microchip Technology Inc. DS22093B-page 43 WORLDWIDE SALES AND SERVICE AMERICAS ASIA/PACIFIC ASIA/PACIFIC EUROPE Corporate Office 2355 West Chandler Blvd. Chandler, AZ 85224-6199 Tel: 480-792-7200 Fax: 480-792-7277 Technical Support: http://support.microchip.com Web Address: www.microchip.com Asia Pacific Office Suites 3707-14, 37th Floor Tower 6, The Gateway Harbour City, Kowloon Hong Kong Tel: 852-2401-1200 Fax: 852-2401-3431 India - Bangalore Tel: 91-80-4182-8400 Fax: 91-80-4182-8422 India - New Delhi Tel: 91-11-4160-8631 Fax: 91-11-4160-8632 Austria - Wels Tel: 43-7242-2244-39 Fax: 43-7242-2244-393 Denmark - Copenhagen Tel: 45-4450-2828 Fax: 45-4485-2829 India - Pune Tel: 91-20-2566-1512 Fax: 91-20-2566-1513 France - Paris Tel: 33-1-69-53-63-20 Fax: 33-1-69-30-90-79 Japan - Yokohama Tel: 81-45-471- 6166 Fax: 81-45-471-6122 Germany - Munich Tel: 49-89-627-144-0 Fax: 49-89-627-144-44 Atlanta Duluth, GA Tel: 678-957-9614 Fax: 678-957-1455 Boston Westborough, MA Tel: 774-760-0087 Fax: 774-760-0088 Chicago Itasca, IL Tel: 630-285-0071 Fax: 630-285-0075 Dallas Addison, TX Tel: 972-818-7423 Fax: 972-818-2924 Detroit Farmington Hills, MI Tel: 248-538-2250 Fax: 248-538-2260 Kokomo Kokomo, IN Tel: 765-864-8360 Fax: 765-864-8387 Los Angeles Mission Viejo, CA Tel: 949-462-9523 Fax: 949-462-9608 Santa Clara Santa Clara, CA Tel: 408-961-6444 Fax: 408-961-6445 Toronto Mississauga, Ontario, Canada Tel: 905-673-0699 Fax: 905-673-6509 Australia - Sydney Tel: 61-2-9868-6733 Fax: 61-2-9868-6755 China - Beijing Tel: 86-10-8528-2100 Fax: 86-10-8528-2104 China - Chengdu Tel: 86-28-8665-5511 Fax: 86-28-8665-7889 Korea - Daegu Tel: 82-53-744-4301 Fax: 82-53-744-4302 China - Hong Kong SAR Tel: 852-2401-1200 Fax: 852-2401-3431 Korea - Seoul Tel: 82-2-554-7200 Fax: 82-2-558-5932 or 82-2-558-5934 China - Nanjing Tel: 86-25-8473-2460 Fax: 86-25-8473-2470 Malaysia - Kuala Lumpur Tel: 60-3-6201-9857 Fax: 60-3-6201-9859 China - Qingdao Tel: 86-532-8502-7355 Fax: 86-532-8502-7205 Malaysia - Penang Tel: 60-4-227-8870 Fax: 60-4-227-4068 China - Shanghai Tel: 86-21-5407-5533 Fax: 86-21-5407-5066 Philippines - Manila Tel: 63-2-634-9065 Fax: 63-2-634-9069 China - Shenyang Tel: 86-24-2334-2829 Fax: 86-24-2334-2393 Singapore Tel: 65-6334-8870 Fax: 65-6334-8850 China - Shenzhen Tel: 86-755-8203-2660 Fax: 86-755-8203-1760 Taiwan - Hsin Chu Tel: 886-3-572-9526 Fax: 886-3-572-6459 China - Wuhan Tel: 86-27-5980-5300 Fax: 86-27-5980-5118 Taiwan - Kaohsiung Tel: 886-7-536-4818 Fax: 886-7-536-4803 China - Xiamen Tel: 86-592-2388138 Fax: 86-592-2388130 Taiwan - Taipei Tel: 886-2-2500-6610 Fax: 886-2-2508-0102 China - Xian Tel: 86-29-8833-7252 Fax: 86-29-8833-7256 Thailand - Bangkok Tel: 66-2-694-1351 Fax: 66-2-694-1350 Italy - Milan Tel: 39-0331-742611 Fax: 39-0331-466781 Netherlands - Drunen Tel: 31-416-690399 Fax: 31-416-690340 Spain - Madrid Tel: 34-91-708-08-90 Fax: 34-91-708-08-91 UK - Wokingham Tel: 44-118-921-5869 Fax: 44-118-921-5820 China - Zhuhai Tel: 86-756-3210040 Fax: 86-756-3210049 01/02/08 DS22093B-page 44 © 2008 Microchip Technology Inc.