TI UCC28019P

UCC28019
www.ti.com
SLUS755 – APRIL 2007
8-Pin Continuous Conduction Mode (CCM) PFC Controller
FEATURES
•
•
•
•
•
•
•
•
•
DESCRIPTION
8-pin Solution Without Sensing Line Voltage
Reduces External Components
Wide-Range Universal AC Input Voltage
Fixed 65-kHz Operating Frequency
Maximum Duty Cycle of 97%
Output Over/Under-Voltage Protection
Input Brown-Out Protection
Cycle-by-Cycle Peak Current Limiting
Open Loop Detection
Low-Power User Controlled Standby Mode
APPLICATIONS
•
•
•
•
•
CCM Boost Power Factor Correction Power
Converters in the 100 W to >2 kW Range
Server and Desktop Power Supplies
Telecom Rectifiers
Industrial Electronics
Home Electronics
CONTENTS
•
•
•
•
•
Electrical Characteristics 3
Device Information 10
Application Information 12
Design Example 23
Additional References 43
The UCC28019 8-pin active Power Factor Correction
(PFC) controller uses the boost topology operating in
Continuous Conduction Mode (CCM). The controller
is suitable for systems in the 100 W to >2 kW range
over a wide-range universal AC line input. Startup
current during under-voltage lockout is less than 200
µA. The user can control low power standby mode
by pulling the VSENSE pin below 0.77 V.
Low-distortion wave-shaping of the input current
using average current mode control is achieved
without input line sensing, reducing the Bill of
Materials component count. Simple external
networks allow for flexible compensation of the
current and voltage control loops. The switching
frequency is internally fixed and trimmed to better
than ±5% accuracy at 25°C. Fast 1.5-A gate peak
current drives the external switch.
Numerous system-level protection features include
peak current limit, soft over-current, open-loop
detection,
input
brown-out,
output
over/under-voltage, a no-power discharge path on
VCOMP, and overload protection on ICOMP.
Soft-Start limits boost current during start-up. A
trimmed internal reference provides accurate
protection thresholds and regulation set-point. An
internal clamp limits the gate drive voltage to 12.5 V.
TYPICAL APPLICATION DIAGRAM
VOUT
EMI Filter
LINE
INPUT
–
Bridge
Rectifier
+
1
GND
2
ICOMP
3
ISENSE
4
VINS
GATE
8
VCC
7
VSENSE
6
VCOMP
5
Auxilary
Supply
Rload
UCC28019
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007, Texas Instruments Incorporated
UCC28019
www.ti.com
SLUS755 – APRIL 2007
ORDERING INFORMATION
(1)
OPERATING TEMPERATURE
RANGE, TA
PART NUMBER
PACKAGE (1)
UCC28019D
SOIC 8-Pin (D) ead (Pb)-Free/Green
UCC28019P
Plastic DIP 8 Pin (P) Lead
(Pb)-Free/Green
–40°C to 125°C
SOIC (D) package is available taped and reeled by adding "R" suffix the the above part number, reeled quantities are 2500 devices per
reel.
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
Input voltage range
Input current range
Junction temperature, TJ
Lead temperature, TSOL
(1)
VCC
–0.3 to 22
GATE
–0.3 to 16
VINS, VSENSE, VCOMP, ICOMP
–0.3 to 7
ISENSE
–24 to 7
VSENSE, ISENSE
–1 to 1
Operating
–55 to 150
Storage
–65 to 150
Soldering, 10s
UNIT
V
mA
°C
300°
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other condition beyond those included under “Recommended Operating
Conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods of time may affect device reliability.
DISSIPATION RATINGS (1)
(1)
PACKAGE
THERMAL IMPEDANCE
JUNCTION TO AMBIENT
(°C/W)
TA = 25°C POWER RATING (W) TA = 85°C POWER RATING (W)
SOIC-8 (D)
160
0.65
0.25
PDIP-8 (P)
110
1
0.36
Tested per JEDEC EIA/JESD 51-1. Thermal resistance is a strong function of board construction and layout. Air flow reduces thermal
resistance. This number is only a general guide. See TI document SPRA953 Thermal Metrics.
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
PARAMETER
MIN
VCC input voltage from a low-impedance source
TJ
MAX
UNIT
VCCOFF(max) + 1 V
21
V
–40
125
°C
Operating junction temperature
ELECTROSTATIC DISCHARGE (ESD) PROTECTION
over operating free-air temperature range (unless otherwise noted)
PARAMETER
RATING
Human Body Model (HBM)
Charged Device Model (CDM)
2
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UNIT
2
kV
500
V
UCC28019
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SLUS755 – APRIL 2007
ELECTRICAL CHARACTERISTICS
Unless otherwise noted, VCC = 15 VDC, 0.1 µF from VCC to GND, -40°C ≤ TJ = TA ≤ 125°C. All voltages are with respect to
GND. Currents are positive into and negative out of the specified terminal.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VCC Bias Supply
IVCC(start)
Pre-start current
VCC = VCCON– 0.1 V
25
100
200
IVCC(stby)
Standby current
VSENSE = 0.5 V
1.0
2.1
2.9
IVCC(on_load)
Operating current
VSENSE = 4.5 V, CGATE = 4.7 nF
4
7
10
10.0
10.5
11.0
µA
mA
Under Voltage Lockout (UVLO)
VCCON
Turn on threshold
VCCOFF
Turn off threshold
UVLO
Hysteresis
9
9.5
10
0.8
1.0
1.2
61.7
65.0
68.3
59
65
71
V
Oscillator
fSW
TA = 25°C
Switching frequency,
– 40°C ≤ TA ≤ 125°C
kHz
PWM
DMIN
Minimum duty cycle
VCOMP = 0 V, VSENSE = 5 V,
ICOMP = 6.4 V
DMAX
Maximum duty cycle
VSENSE = 4.95 V
tOFF(min)
Minimum off time
VSENSE = 3 V, ICOMP = 1 V
0%
94%
97%
99.3%
100
250
600
ns
System Protection
VSOC
ISENSE threshold, soft over current
(SOC) ,
-0.66
-0.73
-0.79
VPCL
ISENSE threshold, peak current Limit
(PCL) ,
-1.00
-1.08
-1.15
VOLP
VSENSE threshold, open loop
protection (OLP),
ICOMP = 1 V, ISENSE = 0 V,
VCOMP = 1 V
0.77
0.82
0.86
Open loop protection (OLP) internal
pull-down current
VSENSE = 0.5 V
100
250
4.63
4.75
4.87
5.12
5.25
5.38
VUVD
VSENSE threshold, output
under-voltage detection (UVD),
VOVP
VSENSE threshold, output
over-voltage protection (OVP),
VINSBROWNOUT_th
Input brown-out detection (IBOP)
high-to-low threshold
0.76
0.82
0.88
VINSENABLE_th
Input brown-out Detection (IBOP)
low-to-high threshold
1.4
1.5
1.6
IVINS_0
VINS bias current
0
±0.1
V
ISENSE = -0.2 V
V
nA
V
VINS = 0 V
ICOMP threshold, external overload
protection
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0.6
µA
V
3
UCC28019
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SLUS755 – APRIL 2007
ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise noted, VCC = 15 VDC, 0.1 µF from VCC to GND, -40°C ≤ TJ = TA ≤ 125°C. All voltages are with respect to
GND. Currents are positive into and negative out of the specified terminal.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Current Loop
gmi
Transconductance gain
TA = 25°C
0.75
Output linear range
0.95
1.15
±50
ICOMP voltage during OLP
VSENSE = 0.5 V
VREF
Reference voltage
-40°C ≤ TA ≤ 125°C
gmv
Transconductance gain
mS
µA
3.7
4.0
4.3
V
4.90
5.00
5.10
V
31.5
42
52.5
µS
21
30
38
Voltage Loop
Maximum sink current under normal
operation
VSENSE = 6 V, VCOMP = 4 V
Source current under soft start
VSENSE = 4 V, VCOMP = 0 V
–21
-30
-38
Maximum source current under EDR
operation
VSENSE = 4 V, VCOMP = 0 V
–100
–170
–250
VSENSE = 4 V, VCOMP = 4 V
–60
–100
–140
4.63
4.75
4.87
V
100
250
nA
0.2
0.4
V
Enhanced dynamic response, VSENSE
low threshold, falling
VSENSE input bias current
1 V ≤ VSENSE ≤ 5 V
VCOMP voltage during OLP
VSENSE = 0.5 V, IVCOMP = 0.5 mA
GATE current, peak, sinking (1)
CGATE = 4.7 nF
2.0
CGATE = 4.7 nF
–1.5
0
µA
GATE Driver
GATE current, peak,
(1)
4
sourcing (1)
A
GATE rise time
CGATE = 4.7 nF, GATE = 2 V to 8 V
40
60
GATE fall time
CGATE = 4.7 nF, GATE = 8 V to 2 V
25
40
GATE low voltage, no load
GATE = 0 A
0
0.05
GATE low voltage, sinking
GATE = 20 mA
0.3
0.8
GATE low voltage, sourcing
GATE = -20 mA
–0.3
–0.8
GATE low voltage, sinking
VCC = 5 V, GATE = 5 mA
0.2
0.75
1.2
GATE low voltage, sinking
VCC = 5 V, GATE = 20 mA
0.2
0.9
1.5
GATE high voltage
VCC = 20 V, CGATE = 4.7 nF
11
12.5
14
GATE high voltage
VCC = 11 V, CGATE = 4.7 nF
9.5
10.5
11.0
GATE high voltage
VCC = VCCOFF + 0.2 V,
CGATE = 4.7 nF
8.0
9.0
10.2
Not tested. Characterized by design.
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ns
V
UCC28019
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SLUS755 – APRIL 2007
TYPICAL CHARACTERISTICS
Unless otherwise noted, VCC = 15 VDC, 0.1 µF from VCC to GND, TJ = TA = 25°C. All voltages are with respect
to GND. Currents are positive into and negative out of the specified terminal.
SUPPLY CURRENT
vs
BIAS SUPPLY VOLTAGE
UVLO THRESHOLDS
vs
TEMPERATURE
4.0
12.0
VSENSE = VINS = 3V
No Gate Load
11.0
VCC Turn ON (VCCON)
3.0
IVCC - Supply Current - mA
VCCON/VCCOFF - UVLO Threshold - V
3.5
10.0
2.5
2.0
IVCC Turn OFF
IVCC Turn ON
1.5
1.0
VCC Turn OFF (VCCOFF)
9.0
0.5
0
8.0
0
-60
-35
-10
15
40
65
90
115
5
140
10
15
20
VCC - Bias Supply Voltage - V
TJ - Temperature - °C
Figure 1.
Figure 2.
SUPPLY CURRENT
vs
TEMPERATURE
SUPPLY CURRENT
vs
TEMPERATURE
10
0.5
VCC = VCCON - 0.1 V
9
7
IVCC(start) - Supply Current - mA
IVCC - Supply Current - mA
8
Operating, GATE Load = 4.7 nF
IVCC(on_load)
6
5
4
Standby
IVCC(stby)
3
2
0.4
0.3
0.2
Pre-Start
(IVCC(start))
0.1
1
0
0
-60
-35
-10
15
40
65
90
115
140
-60
TJ - Temperature - °C
Figure 3.
-35
-10
15
40
65
90
TJ - Temperature - °C
115
140
Figure 4.
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TYPICAL CHARACTERISTICS (continued)
OSCILLATOR FREQUENCY
vs
BIAS SUPPLY VOLTAGE
75
75
73
73
71
71
fSW - Switching Frequency - kHz
fSW - Switching Frequency - kHz
OSCILLATOR FREQUENCY
vs
TEMPERATURE
69
Switching Frequency
67
65
63
61
59
69
67
Switching Frequency
65
63
61
59
57
57
55
55
-60
-35
-10
15
40
65
90
115
140
12
10
TJ - Temperature - °C
Figure 5.
50
1.8
48
1.6
46
1.4
44
gmv - Gain - µA/V
gmi - Gain - mA/V
20
VOLTAGE ERROR AMPLIFIER
TRANSCONDUCTANCE
vs
TEMPERATURE
Gain
1.2
1.0
0.8
Gain
42
40
38
0.6
36
0.4
34
0.2
32
0
30
-60
-35
-10
15
40
65
90
115
140
-60
TJ - Temperature - °C
-35
-10
15
40
65
TJ - Temperature - °C
Figure 7.
6
18
16
Figure 6.
CURRENT AVERAGING
AMPLIFIER TRANSCONDUCTANCE
vs
TEMPERATURE
2.0
14
VCC - Bias Supply Voltage - V
Figure 8.
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90
115
140
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TYPICAL CHARACTERISTICS (continued)
VSENSE THRESHOLD
vs
TEMPERATURE
VSENSE THRESHOLD
vs
TEMPERATURE
5.50
2.0
VOVP / VUVD- VSENSE Threshold - V
VOLP – VSENSE Threshold - V
1.8
1.6
1.4
1.2
1.0
Open Loop Protection (VOLP)
0.8
0.6
0.4
5.25
Over-Voltage Protection (VOVP)
5.00
4.75
Under-Voltage Protection (VUVD)
0.2
4.50
0
-60
-35
-10
15
40
65
90
TJ - Temperature - °C
115
-60
140
-35
-10
Figure 9.
0
1.8
-0.1
1.6
-0.2
VSOC - ISENSE Threshold - V
VINSENABLE_TH / VINSBROUWNOUT_TH – VINS Threshold - V
140
115
140
ISENSE THRESHOLD
vs
TEMPERATURE
2.0
VINS Enable (VINSENABLE_TH)
1.2
1.0
0.8
0.6
115
Figure 10.
VINS THRESHOLD
vs
TEMPERATURE
1.4
15
40
65
90
TJ - Temperature - °C
Input Brown-Out Protection (VINSBROWNOUT_TH)
-0.3
-0.4
-0.5
-0.6
Soft Over-Current Protection (SOC)
-0.7
-0.8
0.4
-0.9
0.2
-1.0
0
-60
-35
-10
15
40
65
90
TJ - Temperature - °C
115
140
-60
Figure 11.
-35
-10
15
40
65
90
TJ - Temperature - °C
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
MINIMUM OFF TIME
vs
TEMPERATURE
GATE DRIVE SWITCHING
vs
TEMPERATURE
600
50
VSENSE = 3 V
ICOMP = 1 V
500
40
450
35
400
30
350
300
CGATE = 4.7 nF
VGATE = 2 V - 8 V
45
t - Time - ns
t - Time - ns
550
tOFF(min)
25
20
250
15
200
10
105
5
100
Fall Time
Rise Time
0
-60
-35
-10
15
40
65
90
115
-60
140
-35
-10
TJ - Temperature - °C
15
Figure 13.
50
115
140
115
140
VCC = 5 V
IVCC = 20 mA
1.8
VGATE – Gate Low Voltage - V
1.6
35
30
Rise Time
25
20
Fall Time
15
10
1.4
1.2
VGATE
1.0
0.8
0.6
0.4
5
0.2
0
10
12
14
16
18
VCC - Bias Supply Voltage - V
20
0
-60
-35
-10
15
40
65
TJ - Temperature - °C
Figure 15.
8
90
GATE LOW VOLTAGE
WITH DEVICE OFF
vs
TEMPERATURE
2.0
CGATE = 4.7 nF
VGATE = 2 V - 8 V
40
t - Time - ns
65
Figure 14.
GATE DRIVE SWITCHING
vs
BIAS SUPPLY VOLTAGE
45
40
TJ - Temperature - °C
Figure 16.
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90
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TYPICAL CHARACTERISTICS (continued)
REFERENCE VOLTAGE
vs
TEMPERATURE
5.50
VREF - Reference Voltage - V
VCC = 15V
5.25
Reference Voltage
5.00
4.75
4.50
-60
-35
-10
15
40
65
90
TJ - Temperature - °C
115
140
Figure 17.
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DEVICE INFORMATION
Connection Diagram
UCC28019 Top View (SOIC-8, PDIP-8)
GATE
8
2 ICOMP
VCC
7
3 ISENSE
VSENSE 6
VINS
VCOMP 5
1 GND
4
Pin Descriptions
Terminal Functions
TERMINAL
NAME
#
GATE
8
GND
1
ICOMP
2
ISENSE
VCC
VCOMP
VINS
VSENSE
10
3
I/O
FUNCTION
O
Gate drive: Integrated push-pull gate driver for one or more external power MOSFETs. 2.0-A sink and 1.5-A
source capability. Output voltage is clamped at 12.5 V.
Ground: Device ground reference.
O
Current loop compensation: Transconductance current amplifier output. A capacitor connected to GND
provides compensation and averaging of the current sense signal in the current control loop. The controller is
disabled if the voltage on ICOMP is less than 0.6 V.
I
Inductor current sense: An input for the voltage across the external current sense resistor, which
represents the instantaneous current through the PFC boost inductor. This voltage is averaged to eliminate
the effects of noise and ripple. Soft Over Current (SOC) limits the average inductor current. Cycle-by-cycle
peak current limit (PCL) immediately shuts off the GATE drive if the peak-limit voltage is exceeded. Use a
220-Ω resistor between this pin and the current sense resistor to limit inrush-surge currents into this pin.
7
Device supply: External bias supply input. Under Voltage Lock Out (UVLO) disables the controller until VCC
exceeds a turn-on threshold of 10.5 V. Operation continues until VCC falls below the turn-off (UVLO)
threshold of 9.5 V. A ceramic by-pass capacitor of 0.1 µF minimum value should be connected from VCC to
GND as close to the device as possible for high frequency filtering of the VCC voltage.
5
O
Voltage loop compensation: Transconductance voltage error amplifier output. A resistor-capacitor network
connected from this pin to GND provides compensation. VCOMP is held at GND until VCC, VINS, and
VSENSE all exceed their threshold voltages. Once these conditions are satisfied, VCOMP is charged until
the VSENSE voltage reaches 95% of its nominal regulation level. When the Enhanced Dynamic Response
(EDR) is engaged, additional current is applied to VCOMP to reduce the charge time. EDR additional current
is inhibited during soft-start. Soft-start is programmed by the capacitance on this pin.
I
Input ac voltage sense: Input Brown Out Protection (IBOP) detects when the system ac-input voltage is
above a user-defined normal operating level, or below a user-defined “brown-out” level. A filtered
resistor-divider network connects from this pin to the rectified-mains node. At startup the controller is disabled
until the VINS voltage exceeds a threshold of 1.5 V, initiating a soft-start. The controller is also disabled if
VINS drops below the brown-out threshold of 0.8 V. Operation will not resume until both VINS and VSENSE
voltages exceed their enable thresholds, initiating another soft-start.
I
Output voltage sense: An external resistor-divider network connected from this pin to the PFC output
voltage provides feedback sensing for output voltage regulation. A small capacitor from this pin to GND filters
high-frequency noise. Standby disables the controller and discharges VCOMP when the voltage at VSENSE
drops below the enable threshold of 0.8V. An internal 100nA current source pulls VSENSE to GND for
Open-Loop Protection (OLP), including pin disconnection. Output over-voltage protection (OVP) disables the
GATE output when VSENSE exceeds 105% of the reference voltage. Enhanced Dynamic Response (EDR)
rapidly returns the output voltage to its normal regulation level when a system line or load step causes
VSENSE to fall below 95% of the reference voltage.
4
6
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EMI Filter
LBST
LINE
INPUT
–
Bridge
Rectifier
DBST
VOUT
+
RVINS1
QBST
CIN
RFB1
RGATE
COUT
RVINS2
RLOAD
RFB2
10k
RSENSE
ICOMP
2
CICOMP
Current
Amplifier
+
FAULT
gmi
VCC
PWM
Comparator
KPC(s)
ICOMP
Gate Driver
S
Q
R
Q
+
GAIN
M1
Fault
IBOP
PWM
RAMP
M2
UVLO
Fault
Logic
GATE
OLP
Min Off Time
65kHz
Oscillator
PCL
8
S
Q
R
Q
OVP
Clock
M2
M1
SOC
RISENSE
Pre-Drive and
Clamp Circuit
VCOMP
UVLO
EDR
VCC
+
ISENSE
40k
40k
3
-1x
CISENSE
Q
S
Q
R
VCCON
10.5V
Peak Current Limit (PCL)
300ns
Leading Edge
Blanking
VPCL
1.08V
PCL
Auxilary
Supply
7
CVCC
GND
UVLO
+
VCCOFF
9.5V
1
+
+
+
OVERVOLTAGE
5.25V
OVP
Soft Over Current (SOC)
SOC
VSOC 0.73V
+
OLP/STANDBY
0.82V
+
UNDERVOLTAGE
4.75V
+
5V
OLP/STANDBY
+
VINS
20k
EDR
Input Brown-Out Protection
(IBOP)
4
+
CVINS
S
VINENABLE_th 1.5V
5V
VINBROWNOUT_th 0.82V
SS
EDR
Q
IBOP
+
R
VSENSE
gmv
Q
FAULT
Voltage Error
Amplifier
6
CVSENSE
100µA
VCOMP
5
RVCOMP
CVCOMP-P
CVCOMP
Figure 18. Block Diagram
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APPLICATION INFORMATION
UCC28019 Operation
The UCC28019 is a switch-mode controller used in boost converters for power factor correction operating at a
fixed frequency in continuous conduction mode. The UCC28019 requires few external components to operate as
an active PFC pre-regulator. Its trimmed oscillator provides a nominal fixed switching frequency of 65 kHz,
ensuring that both the fundamental and second harmonic components of the conducted-EMI noise spectrum are
below the EN55022 conducted-band 150-kHz measurement limit.
Its tightly-trimmed internal 5-V reference voltage provides for accurate output voltage regulation over the typical
world-wide 85 VAC to 265 VAC mains input range from zero to full output load. The usable system load ranges
from 100 W to 2 kW and may be extended in special situations.
Regulation is accomplished in two loops. The inner current loop shapes the average input current to match the
sinusoidal input voltage under continuous inductor current conditions. Under extremely light load conditions,
depending on the boost inductor value, the inductor current may go discontinuous but still meet Class-D
requirements of IEC 1000-3-2 despite the higher harmonics. The outer voltage loop regulates the output voltage
on VCOMP (dependent upon the line and load conditions) which determines the internal gain parameters for
maintaining a low-distortion steady-state input current waveshape.
12
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APPLICATION INFORMATION (continued)
Power Supply
The UCC28019 operates from an external bias supply. It is recommended that the device be powered from a
regulated auxiliary supply. This device is not intended to be used from a bootstrap bias supply. A bootstrap bias
supply is fed from the input high voltage through a resistor with sufficient capacitance on VCC to hold up the
voltage on VCC until current can be supplied from a bias winding on the boost inductor. The minimal hysteresis
on VCC would require an unreasonable value of hold-up capacitance.
During normal operation, when the output is regulated, current drawn by the device includes the nominal run
current plus the current supplied to the gate of the external boost switch. Decoupling of the bias supply must
take switching current into account in order to keep ripple voltage on VCC to a minimum. A ceramic capacitor
with a minimum value of 0.1 µF is recommended from VCC to GND with short, wide traces.
VCC
VCCON 10.5V
VCCOFF 9.5V
IVCC
IVCC(ON)
IVCC(stby) <2.9mA
IVCC(start) <200µA
Controller
State
PWM
State
UVLO
Soft-Start
Run
Fault/Standby
OFF
Ramp
Regulated
OFF
SoftStart
Run
Ramp Regulated
UVLO
OFF
Figure 19. Device Supply States
The device bias operates in several states. During startup, VCC Under-Voltage LockOut (UVLO) sets the
minimum operational dc input voltage of the PFC controller. There are two UVLO thresholds. When the UVLO
turn-on threshold is exceeded, the controller turns ON. If VCC falls below the UVLO lower turn-off threshold, the
controller turns OFF. During UVLO, current drawn by the device is minimal. After the device turns on, Soft Start
(SS) is initiated and the output is ramped up in a controlled manner to reduce the stress on the external
components and prevents output voltage overshoot. During soft start and after the output is in regulation, the
device draws its normal run current. If any of several fault conditions is encountered or if the device is put in
Standby with an external signal, the device draws a reduced standby current.
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APPLICATION INFORMATION (continued)
Soft Start
VCOMP, the output of the voltage loop transconductance amplifier, is pulled low during UVLO, IBOP, and
OLP(Open-Loop Protection)/STANDBY. After the fault condition is released, soft start controls the rate of rise of
VCOMP in order to obtain a linear control of the increasing duty cycle as a function of time. During soft start a
constant 30 µA of current is sourced into the compensation components causing the voltage on this pin to ramp
linearly until the output voltage reaches 85% of its final value. At this point, the sourcing current begins to
decrease until the output voltage reaches 95% of its final rated voltage. The soft-start time is controlled by the
voltage error amplifier compensation components selected, and is user-programmable based on desired loop
crossover frequency. Once VOUT exceeds 95% of rate voltage, EDR is no longer inhibited.
+
VCOMP
5V
gmv
VSENSE
FAULT
VCOMP
ISS = -30uA
for VSENSE < 4.75V
during Soft-Start
Figure 20. Soft Start
System Protection
System level protection features keep the system in safe operating limits:
OVP 105% VREF
100% VREF
EDR 95% VREF
Feedback
Voltage
OLP/SS 16% VREF
Protection
State
OLP
Soft-Start
(No EDR)
Run
OVP
(No Gate Output)
Run
Figure 21. Output Protection States
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(EDR on)
OLP
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APPLICATION INFORMATION (continued)
VCC Under-Voltage Lockout (UVLO)
During startup, UVLO keeps the device in the off state until VCC rises above the 10.5-V enable threshold,
VCCON. With a typical 1 V of hysteresis on UVLO to eliminate noise, the device turns off when VCC drops to the
9.5-V disable threshold, VCCOFF.
VCC
Auxilary Supply
+
S
VCCON 10.5V
CDECOUPLE
UVLO
R
GND
VCCOFF 9.5V
Q
Q
+
Figure 22. UVLO
Input Brown-Out Protection (IBOP)
The VINS, (sensed input line voltage), input provides a means for the designer to set the desired mains RMS
voltage level at which the PFC pre-regulator should start-up, VAC(turnon), as well as the desired mains RMS level
at which it should shut down, VAC(turnoff). This prevents unwanted sustained system operation at or below a
“brown-out” voltage, where excessive line current could overheat components. In addition, because VCC bias is
not derived directly from the line voltage, IBOP protects the circuit from low line conditions that may not trigger
the VCC UVLO turn-off.
RVINS1
VINS
20k
Input Brown-Out Protection (IBOP)
Rectified AC Line
+
CIN
RVINS2
VINENABLE_th
1.5 V
CVINS
5V
VINBROWNOUT_th
0.82 V
S
Q
R
Q
IBOP
+
Figure 23. Input Brown-Out Protection (IBOP)
Input line voltage is sensed directly from the rectified ac mains voltage through a resistor divider filter network
providing a scaled and filtered value at the VINS input. IBOP puts the device in standby mode when VINS falls
(high-to-low) below 0.8 V, VINSBROWNOUT_th. The device comes out of standby when VINS rises (low-to-high)
above 1.5 V, VINSENABLE_th. IVINS_0 V , bias current sourced from VINS, is less than 0.1 µA. With a bias current
this low, there is little concern for any set-point error caused by this current flowing through the sensing network.
The highest reasonable value resistance for this network should be chosen to minimize power dissipation,
especially in applications requiring low standby power. Be aware that higher resistance values are more
susceptible to noise pickup, but low noise PCB layout techniques can help mitigate this. Also, depending on the
resistor type used and its voltage rating, RVINS1 should be implemented with multiple resistors in series to reduce
voltage stresses.
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APPLICATION INFORMATION (continued)
First, select RVINS1 based on the the highest reasonable resistance value available for typical applications.
Then select RVINS2 based on this value:
RVINS 2 = RVINS 1
VINS ENABLE _ th(max)
2VAC( on ) - VINS ENABLE _ th(max) - VF _ BRIDGE
Where VF_Bridge is the forward voltage drop across the ac rectifier bridge.
Power dissipated in the resistor network is:
PVINS =
VIN _ RMS 2
RVINS 1 + RVINS 2
The filter capacitor, CVINS, has two functions. First, to attenuate the voltage ripple to levels between the enable
and brown-out thresholds which will prevent the ripple on VINS from falsely triggering IBOP when the converter
is operating at low line. Second, CVINS delays the brown-out protection operation for a desired number of line
half-cycle periods while still having a good response to an actual brown-out event.
The capacitor is chosen so that it will discharge to the VINSBROWNOUT_th level after N number of half line cycles
of delay to accommodate line dropouts.
-tCVIN _ dschg
CVINS =
é
ê
VINS BROWNOUT _ th(min)
RVINS 2 ln ê
RVINS 2
ê 0 .9 V
IN _ RMS (min) (
êë
RVINS 1 + RVINS 2
ù
ú
ú
)ú
úû
Where:
tCVINS _ dschrg =
N half _ cycles
2 f LINE(min)
and VIN_RMS(min) is the lowest normal operating RMS input voltage.
16
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APPLICATION INFORMATION (continued)
Output Over-Voltage Protection (OVP)
VOUT(OVP) is the output voltage exceeding 5% of the rated value, causing VSENSE to exceed a 5.25-V threshold
(5-V reference voltage + 5%), VOVP. The normal voltage control loop is bypassed and the GATE output is
disabled until VSENSE falls below 5.25 V. For example, VOUT(OVP) is 420 V in a system with a 400-V rated
output.
Open Loop Protection/Standby (OLP/Standby)
If the output voltage feedback components were to fail and disconnect (open loop) the signal from the VSENSE
input, then it is likely that the voltage error amp would increase the GATE output to maximum duty cycle. To
prevent this, an internal pull-down forces VSENSE low. If the output voltage falls below 16% of its rated voltage,
causing VSENSE to fall below 0.8 V, the device is put in Standby, a state where the PWM switching is halted
and the device is still on but draws standby current below 3 mA. This shutdown feature also gives the designer
the option of pulling VSENSE low with an external switch.
Output Under-Voltage Detection (UVD) / Enhanced Dynamic Response (EDR)
During large changes in load, Enhanced Dynamic Response (EDR) acts to speed up the slow response of the
low-bandwidth voltage loop.
Output Voltage
RFB1
Standby
VSENSE
RFB2
Optional
+
OVP
OVERVOLTAGE
5.25V
UNDERVOLTAGE
4.75V
+
UVD
OPEN LOOP
PROTECTION/
STANDBY
0.82V
+
OLP/STANDBY
Figure 24. Over Voltage Protection, Open Loop Protection/Standby
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APPLICATION INFORMATION (continued)
Overcurrent Protection
Inductor current is sensed by RSENSE, a low value resistor in the return path of input rectifier. The other side of
the resistor is tied to the system ground. The voltage is sensed on the rectifier side of the sense resistor and is
always negative. There are two over-current protection features; Peak Current Limit (PCL) protects against
inductor saturation and Soft Over Current (SOC) protects against an overload on the output.
Soft Over Current (SOC)
LINE
INPUT
–
SOC
VSOC 0.73V
+
VOUT
+
RSENSE
ISENSE
CISENSE
RISENSE
300ns
Leading Edge
Blanking
VPCL
1.08V
(Optional)
PCL
+
+
-1x
Peak Current Limit (PCL)
Figure 25. Soft Over Current (SOC) / Peak Current Limit (PCL)
Soft Over-Current (SOC)
SOC limits the input current. SOC is activated when the current sense voltage on ISENSE reaches -0.73 V,
affecting the internal VCOMP level, and the control loop is adjusted to reduce the PWM duty cycle.
Peak Current Limit (PCL)
Peak current limit operates on a cycle-by-cycle basis. When the current sense voltage on ISENSE reaches -1.08
V, PCL is activated terminating the active switch cycle. The voltage at ISENSE is amplified by a fixed gain of
-1.0 and then leading-edge blanked to improve noise immunity against false triggering.
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APPLICATION INFORMATION (continued)
Current Sense Resistor, RSENSE
The current sense resistor, RSENSE, is sized using the minimum threshold value of Soft Over Current (SOC),
VSOC(min) = 0.66 V. To avoid triggering this threshold during normal operation, taking into account the gain of the
internal non-linear power limit, resulting in a decreased duty cycle, the resistor is typically sized for an overload
current of 25% more than the peak inductor peak current.
RSENSE £
VSOC(min)
1.25 I L _ PEAK (max)
Since RSENSE sees the average input current, worst-case power dissipation occurs at input low line when input
line current is at its maximum. Power dissipated by the sense resistor is:
PRSENSE = ( I IN _ RMS (max) )2 R SENSE
Peak Current Limit (PCL) protection turns off the output driver when the voltage across the sense resistor
reaches the PCL threshold, VPCL. The absolute maximum peak current, IPCL, is given as:
I PCL =
VPCL
RSENSE
Gate Driver
The GATE output is designed with a current-optimized structure to directly drive large values of total MOSFET
gate capacitance at high turn-on and turn-off speeds. An internal clamp limits voltage on the MOSFET gate to
12.5 V. An external gate drive resistor, RGATE, limits the rise time and dampens ringing caused by parasitic
inductances and capacitances of the gate drive circuit thus reducing EMI. The final value of the resistor depends
upon the parasitic elements associated with the layout and other considerations. A 10-kΩ resistor close to the
gate of the MOSFET, between the gate and ground, discharges stray gate capacitance and protects against
inadvertent dv/dt-triggered turn-on.
VCC
UVLO
From
PWM
Latch
Fault
Logic
OLP
VCC
Rectified
AC
L BST
DBST
VOUT
QBST
IBOP
GATE
COUT
RGATE
PCL
OVP
S
10k
Q
GND
Clock
R
Q
Pre-Drive and
Clamp Circuit
Figure 26. Gate Driver
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APPLICATION INFORMATION (continued)
Current Loop
The overall system current loop consists of the current averaging amplifier stage, the pulse width modulator
(PWM) stage, the external boost inductor stage, and the external current sensing resistor.
ISENSE and ICOMP Functions
The negative polarity signal from the current sense resistor is buffered and inverted at the ISENSE input. The
internal positive signal is then averaged by the current amplifier (gmi), whose output is the ICOMP pin. The
voltage on ICOMP is proportional to the average inductor current. An external capacitor to GND is applied to the
ICOMP pin for current loop compensation and current ripple filtering. The gain of the averaging amplifier is
determined by the internal VCOMP voltage. This gain is non-linear to accommodate the world-wide ac-line
voltage range. ICOMP is connected to 4 V internally whenever the device is in a Fault or Standby condition.
Pulse Width Modulator
The PWM stage compares the ICOMP signal with a periodic ramp to generate a leading-edge-modulated output
signal which is high whenever the ramp voltage exceeds the ICOMP voltage. The slope of the ramp is defined
by a non-linear function of the internal VCOMP voltage.
The PWM output signal always starts low at the beginning of the cycle, triggered by the internal clock. The
output stays low for a minimum off-time, tOFF(min), after which the ramp rises linearly to intersect the ICOMP
voltage. The ramp-ICOMP intersection determines tOFF, and hence DOFF. Since DOFF = VIN/VOUT by the
boost-topology equation, and since VIN is sinusoidal in wave-shape, and since ICOMP is proportional to the
inductor current, it follows that the control loop forces the inductor current to follow the input voltage wave-shape
to maintain boost regulation. Therefore, the average input current is also sinusoidal in wave-shape.
Control Logic
The output of the PWM comparator stage is conveyed to the GATE drive stage, subject to control by various
protection functions incorporated into the IC. The GATE output duty-cycle may be as high as 99%, but will
always have a minimum off-time tOFF(min). Normal duty-cycle operation can be interrupted directly by OVP and
PCL on a cycle-by-cycle basis. UVLO, IBOP and OLP/Standby also terminate the GATE output pulse, and
further inhibit output until the SS operation can begin.
Voltage Loop
The outer control loop of the PFC controller is the voltage loop. This loop consists of the PFC output sensing
stage, the voltage error amplifier stage, and the non-linear gain generation.
Output Sensing
A resistor-divider network from the PFC output voltage to GND forms the sensing block for the voltage control
loop. The resistor ratio is determined by the desired output voltage and the internal 5-V regulation reference
voltage.
Like the VINS input, the very low bias current at the VSENSE input allows the choice of the highest practicable
resistor values for lowest power dissipation and standby current. A small capacitor from VSENSE to GND serves
to filter the signal in a high-noise environment. This filter time constant should generally be less than 100 µs.
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APPLICATION INFORMATION (continued)
Voltage Error Amplifier
The transconductance error amplifier (gmv) generates an output current proportional to the difference between
the voltage feedback signal at VSENSE and the internal 5-V reference. This output current charges or
discharges the compensation network capacitors on the VCOMP pin to establish the proper VCOMP voltage for
the system operating conditions. Proper selection of the compensation network components leads to a stable
PFC pre-regulator over the entire ac-line range and 0-100% load range. The total capacitance also determines
the rate-of-rise of the VCOMP voltage at soft start, as discussed earlier.
The amplifier output VCOMP is pulled to GND during any Fault or Standby condition to discharge the
compensation capacitors to an initial zero state. Usually, the large capacitor has a series resistor which delays
complete discharge by their respective time constant (which may be several hundred milliseconds). If VCC bias
voltage is quickly removed after UVLO, the normal discharge transistor on VCOMP loses drive and the large
capacitor could be left with substantial voltage on it, negating the benefit of a subsequent Soft-Start. The
UCC28019 incorporates a parallel discharge path which operates without VCC bias, to further discharge the
compensation network after VCC is removed.
When output voltage perturbations greater than ±5% appear at the VSENSE input, the amplifier moves out of
linear operation. On an over-voltage, the OVP function acts directly to shut off the GATE output until VSENSE
returns within ±5% of regulation. On an under-voltage, the UVD function invokes EDR which immediately
increases the internal VCOMP voltage by 2 V and increases the external VCOMP charging current typically to
100 µA to 170 µA. This higher current facilitates faster charging of the compensation capacitors to the new
operating level, improving transient response time.
Non-linear Gain Generation
The voltage at VCOMP is used to set the current amplifier gain and the PWM ramp slope. This voltage is
buffered internally and is then subject to modification by the EDR function and the SOC function, as discussed
earlier.
Together the current gain and the PWM slope adjust to the different system operating conditions (set by the
ac-line voltage and output load level) as VCOMP changes, to provide a low-distortion, high-power-factor input
current wave-shape following that of the input voltage.
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APPLICATION INFORMATION (continued)
Layout Guidelines
As with all PWM controllers, the effectiveness of the filter capacitors on the signal pins depends upon the
integrity of the ground return. The pinout of the UCC28019 is ideally suited for separating the high di/dt induced
noise on the power ground from the low current quiet signal ground required for adequate noise immunity. A star
point ground connection at the GND pin of the device can be achieved with a simple cut out in the ground plane
of the printed circuit board. As shown in Figure 27, the capacitors on ISENSE, VINS, VCOMP, and VSENSE
(C11, C12, C15, C17, and C16, respectively) must all be returned directly to the quiet portion of the ground
plane, indicated by Signal GND, and not the high current return path of the converter, shown as the Power GND.
Because the example circuit in Figure 27 uses surface mount components, the ICOMP capacitor, C10, has its
own dedicated return to the GND pin.
Power
GND
Cut out in
ground plane
GND
GATE
ICOMP
VCC
ISENSE
VSENSE
VINS
VCOMP
Signal
GND
Figure 27. Recommended Layout for the UCC28019
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DESIGN EXAMPLE
350-W, Universal Input, 390-VDC Output, PFC Converter
Design Goals
This example illustrates the design process and component selection for a continuous conduction mode power
factor correction boost converter utilizing the UCC28019. The target design is a universal input, 350W PFC
designed for an ATX supply application. This design process is directly tied to the UCC28019 Design Calculator
spreadsheet that can be found in the Tools section of the UCC28019 product folder on the Texas Instruments
website.
Table 1. Design Goal Parameters
PARAMETER
TEST CONDITION
MIN
TYP
MAX
UNIT
Input characteristics
Input voltage
VIN
85
Input frequency
fLINE
47
Brown out voltage
115
VAC(on)
IOUT = 0.9 A
75
VAC(off)
IOUT = 0.9 A
65
265
VAC
63
Hz
VAC
Output characteristics
Output voltage
VOUT
85 VAC ≤ VIN ≤ 265 VAC
47 Hz ≤ fLINE ≤ 63 Hz
0 A ≤ IOUT ≤ 0.9 A
Line regulation
85 VAC ≤ VIN≤ 65VAC
IOUT = 0.440 A
5%
VIN = 115 VAC, fLINE = 60 Hz
0 A ≤ IOUT≤ 0.9 A
5%
VIN = 230 VAC, fLINE = 50 Hz
0 A ≤ IOUT≤ 0.9 A
5%
VRIPPLE(SW)
VIN = 115 VAC, fLINE = 60 Hz
IOUT = 0.9 A
3.9
VRIPPLE(SW)
VIN = 230 VAC , fLINE = 50 Hz
IOUT = 0.9 A
3.9
VRIPPLE(f_LINE)
VIN = 115 VAC, fLINE = 60 Hz,
IOUT = 0.9 A
19.5
VRIPPLE(f_LINE)
VIN = 230 VAC, fLINE = 50 Hz
IOUT = 0.9 A
19.5
Load regulation
High frequency output voltage
ripple
Line frequency output voltage
ripple
370
390
VDC
Vpp
Output load current
IOUT
85 VAC ≤ VIN≤ 265 VAC
47 Hz ≤ fLINE≤ 63 Hz
Output power
POUT
Output over voltage protection
VOUT(OVP)
410
Output under voltage protection
VOUT(UVP)
370
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410
0.9
A
350
W
V
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DESIGN EXAMPLE (continued)
Table 1. Design Goal Parameters (continued)
PARAMETER
TEST CONDITION
MIN
TYP
MAX
UNIT
Control loop characteristics
Switching frequency
fSW, TJ = 25°C
Control loop bandwidth
f(CO)
VIN = 162 VDC, IOUT = 0.45 A
10
Hz
Phase margin
VIN = 162 VDC, IOUT = 0.45 A
70
degrees
Power factor
PF
VIN = 115 VAC, IOUT = 0.9 A
Total harmonic distortion
61.7
68.3
4.13%
10%
THD
VIN = 230 VAC, fLINE = 50 Hz
IOUT = 0.9 A
6.67%
10%
η
VIN = 115 VAC, fLINE = 60 Hz,
IOUT = 0.9 A
Ambient temperature
TAMB
0.92
50
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kHz
0.99
THD
VIN = 115 VAC, fLINE = 60 Hz
IOUT = 0.9 A
Full load efficiency
24
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+
+
The following procedure refers to the schematic shown in Figure 28.
Figure 28. Design Example Schematic
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Current Calculations
First, determine the maximum average output current, IOUT(max):
I OUT (max) =
I OUT (max) =
POUT (max)
VOUT
350 W
@ 0 .9 A
390 V
The maximum input RMS line current, IIN_RMS(max), is calculated using the parameters from Table 1 and the
efficiency and power factor initial assumptions:
POUT (max)
I
IN _ RMS (max)
=
I
IN _ RMS (max)
=
hVIN (min) PF
350W
= 4.52 A
0.92 ´ 85V ´ 0.99
Based upon the calculated RMS value, the maximum peak input current, IIN_PEAK(max), and the maximum average
input current, IIN_AVG(max), assuming the waveform is sinusoidal, can be determined.
I IN _ PEAK (max) = 2 I IN _ RMS (max)
I IN _ PEAK (max) = 2 ´ 4.52 A = 6.39 A
I IN _ AVG(max) =
2 I IN _ PEAK (max)
p
I IN _ AVG(max) =
2 ´ 6.39 A
= 4.07 A
p
Bridge Rectifier
Assuming a forward voltage drop, VF_BRIDGE, of 0.95 V across the rectifier diodes, BR1, the power loss in the
input bridge, PBRIDGE, can be calculated:
PBRIDGE = 2VF _ BRIDGE I IN _ AVG(max)
PBRIDGE = 2 ´ 0.95V ´ 4.07 A = 7.73W
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Input Capacitor
Note that the UCC28019 is a continuous conduction mode controller and as such the inductor ripple current
should be sized accordingly. Allowing an inductor ripple current, IRIPPLE, of 20% and a high frequency voltage
ripple factor, VRIPPLE_IN, of 6%, the maximum input capacitor value, CIN, is calculated by first determining the
input ripple current, IRIPPLE, and the input voltage ripple, VIN_RIPPLE(max):
I
RIPPLE
= DI RIPPLE I IN _ PEAK (max)
DI RIPPLE = 0.2
I
RIPPLE
= 0.2 ´ 6.39 A = 1.28 A
VIN _ RIPPLE(max) = DVRIPPLE _ INVIN _ RECTIFIED(max)
DVRIPPLE _ IN = 0.06
VIN _ RECTIFIED = 2VIN
VIN _ RECTIFIED(max) = 2 ´ 265V = 375V
VIN _ RIPPLE(max) = 0.06 ´ 375V = 7.21V
The value for the input x-capacitor can now be calculated:
CIN =
CIN =
I RIPPLE
8 f SW VIN _ RIPPLE(max)
1.28 A
= 0.341m F
8 ´ 65kHz ´ 7.21V
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Boost Inductor
The boost inductor, LBST, is selected after determining the maximum inductor peak current, IL_PEAK(max):
I L _ PEAK (max) = I IN _ PEAK (max) +
I L _ PEAK (max) = 6.39 A +
I RIPPLE
2
1.28 A
= 7.03 A
2
The minimum value of the boost inductor is calculated based upon a worst case duty cycle of 0.5:
LBST (min) ³
VOUT D( 1 - D )
f SW ( typ ) I RIPPLE
LBST (min) ³
390V ´ 0.5( 1 - 0.5 )
³ 1.17 mH
65kHz ´1.28 A
The actual value of the boost inductor that will be used is 1.25 mH.
The maximum duty cycle, DUTY(max), can be calculated and will occur at the minimum input voltage:
DUTY(max) =
VOUT - VIN _ RECTIFIED(min)
VOUT
VIN _ RECTIFIED(min) = 2 ´ 85V = 120V
DUTY(max) =
390V - 120V
= 0.692
390V
Boost Diode
The diode losses are estimated based upon the forward voltage drop, VF, at 125°C and the reverse recovery
charge, QRR, of the diode. Using a silicone carbide diode, although more expensive, will essentially eliminate the
reverse recovery losses:
PDIODE = VF _125C I OUT (max) + 0.5 f SW ( typ )VOUT QRR
VF _125C = 1.5V
QRR = 0nC
PDIODE = 1.5V ´ 0.897 A + 0.5 ´ 65kHz ´ 390V ´ 0nC = 1.35W
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Switching Element
The conduction losses of the switch are estimated using the RDS(on) of the FET at 125°C , found in the FET data
sheet, and the calculated drain to source RMS current, IDS_RMS:
2
PCOND = I DS
_ RMS RDSon( 125C )
RDSon( 125C ) = 0.35W
I DS _ RMS =
I DS _ RMS =
POUT (max)
VIN _ RECTIFIED(min)
350W
120V
2-
2-
16VIN _ RECTIFIED(min)
3p VOUT
16 ´120V
= 3.54 A
3p ´ 390V
PCOND = 3.54 A2 ´ 0.35W = 4.38W
The switching losses are estimated using the rise time of the gate, tr, and the output capacitance losses.
For the selected device:
tr = 4.5ns
COSS = 780 pF
2
PSW = f SW ( typ ) ( trVOUT I IN _ PEAK (max) + 0.5COSSVOUT
)
PSW = 65kHz( 4.5ns ´ 390V ´ 6.39 A + 0.5 ´ 780 pF ´ 390V 2 ) = 4.59W
Total FET losses:
PCOND + PSW = 4.38W + 4.59W = 8.97W
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Sense Resistor
To accommodate the gain of the internal non-linear power limit, RSENSE, is sized such that it will trigger the soft
over-current at 25% higher than the maximum peak inductor current using the minimum SOC threshold, VSOC, of
ISENSE.
RSENSE =
RSENSE =
VSOC
I L _ PEAK (max) ´1.25
0.66V
= 0.075W
7.03 A ´1.25
Using a parallel combination of available standard value resistors, the sense resistor is chosen.
RSENSE = 0.067W
The power dissipated across the sense resistor, PRsense, must be calculated:
2
PRsense = I IN
_ RMS (max) RSENSE
PRsense = 4.52 A2 ´ 0.067W = 1.36W
The peak current limit, PCL, protection feature will be triggered when current through the sense resistor results
in the voltage across RSENSE to be equal to the VPCL threshold. For a worst case analysis, the maximum VPCL
threshold is used:
I PCL =
VPCL
RSENSE
I PCL =
1.15V
= 17.25 A
0.067W
To protect the device from inrush current, a standard 220-mΩ resistor, RISENSE, is placed in series with the
ISENSE pin. A 1000-pF capacitor is placed close to the device to improve noise immunity on the ISENSE pin.
30
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Output Capacitor
The output capacitor, COUT, is sized to meet holdup requirements of the converter. Assuming the downstream
converters require the output of the PFC stage to never fall below 300 V, VOUT_HOLDUP(min), during one line cycle,
tHOLDUP = 1/fLINE(min), the minimum calculated value for the capacitor is:
COUT (min) ³
COUT (min) ³
2
OUT
V
2 POUT t HOLDUP
2
- VOUT
_ HOLDUP(min)
2 ´ 350W ´ 21.28ms
³ 240 m F
390V 2 - 300V 2
It is advisable to de-rate this capacitor value by 20%; the actual capacitor used is 270 µF.
Setting the maximum peak-to-peak output ripple voltage to be less than 5% of the output voltage will ensure that
the ripple voltage will not trigger the output over-voltage or output under-voltage protection features of the
controller. The maximum peak-to-peak ripple voltage, occurring at twice the line frequency, and the ripple
current of the output capacitor are calculated:
VOUT _ RIPPLE( pp ) < 0.05VOUT
VOUT _ RIPPLE( pp ) < 0.05 ´ 390V < 19.5VPP
VOUT _ RIPPLE( pp ) =
VOUT _ RIPPLE( pp ) =
I OUT
p ( 2 f LINE(min) )COUT
0 .9 A
= 11.26V
p ( 2 ´ 47 Hz ) ´ 270 m F
The required ripple current rating at twice the line frequency is equal to:
I Cout _ 2 fline =
I Cout _ 2 fline =
I OUT (max)
2
0.9 A
= 0.635 A
2
There will also be a high frequency ripple current through the output capacitor:
I Cout _ HF = I OUT (max)
I Cout _ HF = 0.9 A
16VOUT
3p VIN _ RECTIFIED(min)
- 1 .5
16 ´ 390V
- 1 . 5 = 1 .8 A
3p ´120V
The total ripple current in the output capacitor is the combination of both and the output capacitor must be
selected accordingly:
I
Cout _ RMS ( total )
2
2
= I Cout
_ 2 fline + I Cout _ HF
I
Cout _ RMS ( total )
= 0.635 A2 + 1.8 A2 = 1.9 A
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Output Voltage Set Point
For low power dissipation and minimal contribution to the voltage set point error, it is recommended to use 1 MΩ
for the top voltage feedback divider resistor, RFB1. Multiple resistors in series are used due to the maximum
allowable voltage across each. Using the internal 5-V reference, VREF, select the bottom divider resistor, RFB2, to
meet the output voltage design goals.
RFB 2 =
VREF RFB1
VOUT - VREF
RFB 2 =
5V ´1M W
= 13.04k W
390V - 5V
Using 13 kΩ for RFB2 results in a nominal output voltage set point of 391 V.
The over-voltage protection, OVD, will be triggered when the output voltage exceeds 5% of its nominal set-point:
æ R + RFB 2 ö
VOUT ( OVP ) = VSENSEOVP ç FB1
÷
RFB 2
è
ø
æ 1M W + 13k W ö
VOUT ( OVP ) = 5.25V ´ ç
÷ = 410.7V
13k W
è
ø
The under-voltage detection, UVD, will be triggered when the output voltage falls below 5% of its nominal
set-point:
æ R + RFB 2 ö
VOUT ( UVD ) = VSENSEUVD ç FB1
÷
RFB 2
è
ø
æ 1M W + 13k W ö
VOUT ( UVD ) = 4 .75V ´ ç
÷ = 371 .6V
13k W
è
ø
A small capacitor on VSENSE must be added to filter out noise that would trigger the enhanced dynamic
response in a no-load high-line configuration. Limit the value of the filter capacitor such that the RC time
constant is less than 0.1ms so as not to significantly reduce the control response time to output voltage
deviations.
32
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Loop Compensation
The selection of compensation components, for both the current loop and the voltage loop, is made easier by
using the UCC28019 Design Calculator spreadsheet that can be found in the Tools section of the UCC28019
product folder on the Texas Instruments website. The current loop is compensated first by determining the
product of the internal loop variables, M1M2, using the internal controller constants K1 and KFQ:
M 1M 2 =
K FQ =
K FQ =
2
I OUT (max)VOUT
RSENSE K1
h 2VIN2 _ RMS K FQ
1
f SW ( typ )
1
= 15.385m s
65kHz
K1 = 7
M 1M 2 =
0.9 A ´ 390V 2 ´ 0.067W ´ 7
V
= 0.372
2
2
0.92 ´115V ´15.385m s
ms
The VCOMP operating point is found on Figure 29. The Design Calculator spreadsheet enables the user to
iteratively select the appropriate VCOMP value.
M1M2
vs
VCOMP
2.0
1.8
1.6
1.4
M1M2
1.2
1.0
0.8
0.6
0.4
0.2
0
0
1
2
3
4
5
6
7
VCOMP - V
Figure 29. M1M2 vs. VCOMP
For the given M1M2 of 0.372 V/µs, the VCOMP, approximately equal to 4, as shown in Figure 29.
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The individual loop factors, M1 which is the current loop gain factor, and M2 which is the voltage loop PWM ramp
slope, are calculated using the following conditions:
The M1 current loop gain factor:
if : 0 < VCOMP < 2
then : M 1 = 0.064
if : 2 £ VCOMP < 3
then : M 1 = 0.139 ´ VCOMP - 0.214
if : 3 £ VCOMP < 5.5
then : M 1 = 0.279 ´ VCOMP - 0.632
if : 5.5 £ VCOMP < 7
then : M 1 = 0.903
VCOMP = 4
M 1 = 0.279 ´ 4 - 0.632 = 0.484
The M2 PWM ramp slope:
if : 0 < VCOMP < 1.5
V
then : M 2 = 0
ms
if : 1.5 £ VCOMP < 5.6
then : M 2 = 0.1223 ´ (VCOMP - 1.5 )2
V
ms
if : 5.6 £ VCOMP < 7
then : M 2 = 2.056
V
ms
VCOMP = 4
M 2 = 0.1223 ´ ( 4 - 1.5 )2
34
V
V
= 0.764
ms
ms
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Verify that the product of the individual gain factors is approximately equal to the M1M2 factor determined above,
if not, reselect VCOMP and recalculate M1M2.
M 1 ´ M 2 = 0.484 ´ 0.764
0.37
V
V
= 0.37
ms
ms
V
V
@ M 1M 2 = 0.372
ms
ms
The non-linear gain variable, M3, can now be calculated:
if : 0 < VCOMP < 3
then : M 3 = 0.0510 ´ VCOMP 2 - 0.1543 ´ VCOMP - 0.1167
if : 3 £ VCOMP < 7
then : M 3 = 0.1026 ´ VCOMP 2 - 0.3596 ´ VCOMP + 0.3085
VCOMP = 4
M 3 = 0.1026 ´ 42 - 0.3596 ´ 4 + 0.3085 = 0.512
The frequency of the current averaging pole, fIAVG, is chosen to be at 9.5 kHz. The required capacitor on
ICOMP, CICOMP, for this is determined using the transconductance gain, gmi, of the internal current amplifier:
CICOMP =
gmiM 1
K1 2p f IAVG
CICOMP =
0.95mS ´ 0.484
= 1100 pF
7 ´ 2 ´ p ´ 9.5kHz
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UCC28019
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The transfer function of the current loop can be plotted:
GCL ( f ) =
K1 RSENSEVOUT
´
K FQ M 1M 2 LBST
1
s( f )2 K1CICOMP
s( f ) +
gmiM 1
GCLdB ( f ) = 20 log ( GCL ( f ) )
CURRENT AVERAGING CIRCUIT
100
-80
80
60
-100
Phase
40
-120
0
qGCL(f)
GCLdB(f)
20
Gain
-20
-140
-40
-60
-160
-80
-100
-180
10
100
3
1*10
4
1*10
5
1*10
6
1*10
f - Hz
Figure 30. Bode Plot of the Current Averaging Circuit.
36
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The open loop of the voltage transfer function, GVL(f) contains the product of the voltage feedback gain, GFB,
and the gain from the pulse width modulator to the power stage, GPWM_PS, which includes the pulse width
modulator to power stage pole, fPWM_PS. The plotted result is shown in Figure 31.
GFB =
RFB 2
RFB1 + RFB 2
GFB =
13k W
= 0.013
1M W + 13k W
1
f PWM _ PS =
2p
f PWM _ PS =
3
K1 RSENSEVOUT
COUT
2
K FQ M 1M 2VIN ( typ )
1
= 1.589 Hz
7 ´ 0.067W ´ 390V 3 ´ 270 m F
2p
V
15.385m s ´ 0.484 ´ 0.764 ´115V 2
ms
M 3VOUT
M 1M 2 ´1m s
GPWM _ PS ( f ) =
s( f )
1+
2p f PWM _ PS
GVL ( f ) = GFB GPWM _ PS ( f )
GVLdB ( f ) = 20 log ( GVL ( f ) )
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OPEN LOOP VOLTAGE TRANSFER
FUNCTION
0
20
-20
0
-40
qGVL(f)
GVLdB(f)
Gain
Phase
-20
-60
-40
-80
-60
-100
0.01
0.1
1
10
100
1*103
1*104
f - Hz
Figure 31. Bode Plot of the Open Loop Voltage Transfer Function
38
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The voltage error amplifier is compensated with a zero, fZERO, at the fPWM_PS pole and a pole, fPOLE, placed at 20
Hz to reject high frequency noise and roll off the gain amplitude. The overall voltage loop crossover, fV, is
desired to be at 10 Hz. The compensation components of the voltage error amplifier are selected accordingly.
f ZERO =
1
2p RVCOMP CVCOMP
1
RVCOMP CVCOMP CVCOMP _ P
f POLE =
2p
CVCOMP + CVCOMP _ P
é
ê
ê
1 + s( f )RVCOMP CVCOMP
GEA ( f ) = gmv ê
é
æ RVCOMP CVCOMP CVCOMP _ P
ê
1
C
C
s(
f
)
s(
f
)
+
+
ê
ç
(
)
VCOMP
VCOMP
_
P
ê
ç CVCOMP + CVCOMP _ P
è
ëê
ë
ù
ú
ú
ú
öù ú
÷÷ ú ú
ø ûú û
fV = 10 Hz
From Figure 31, and the Design Calculator spreadsheet, the open loop gain of the voltage transfer function at 10
Hz is approximately 0.709 dB. Estimating that the parallel capacitor, CVCOMP_P, is much smaller than the series
capacitor, CVCOMP, the unity gain will be at fV, and the zero will be at fPWM_PS, the series compensation capacitor
is determined:
gmv
CVCOMP =
10
fV
f PWM _ PS
GVLdB ( f )
20
´ 2p fV
10 Hz
1.589 Hz = 3.88 m F
= 0.709 dB
20
´ 2 ´ p ´ 10 Hz
10
42 m S ´
CVCOMP
A 3.3-µF capacitor is used for CVCOMP.
RVCOMP =
1
2p f ZERO CVCOMP
RVCOMP =
1
= 30.36k W
2 ´ p ´1.589 Hz ´ 3.3m F
A 33-kΩ resistor is used for RVCOMP.
CVCOMP _ P =
CVCOMP
2p f POLE RVCOMP CVCOMP - 1
CVCOMP _ P =
3.3m F
= 0.258m F
2 ´ p ´ 20 Hz ´ 33k W ´ 3.3m F - 1
A 0.22-µF capacitor is used for CVCOMP_P.
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The total closed loop transfer function, GVL_total, contains the combined stages and is plotted in Figure 32.
GVL _ total ( f ) = GFB ( f )GPWM _ PS ( f )GEA ( f )
GVL _ totaldB ( f ) = 20 log GVL _ total ( f )
(
)
100
100
50
80
60
0
Gain
40
-50
Phase
-100
20
-150
0
0.01
0.1
1
10
100
1*103
1*104
f - Hz
Figure 32. Closed Loop Voltage Bode Plot
40
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qGVL_total(f)
GVL_totaldB(f)
CLOSED LOOP VOLTAGE TRANSFER
FUNCTION
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Brown Out Protection
Select the top divider resistor into the VINS pin so as not to contribute excessive power loss. The extremely low
bias current into VINS means the value of RVINS1 could be hundreds of megaohms. For practical purposes, a
value less than 10 MΩ is usually chosen. Assuming approximately 150 times the input bias current through the
resistor dividers will result in an RVINS1 that is less than 10 MΩ , so as to not contribute excessive noise, and still
maintain minimal power loss. The brown out protection will turn off the gate drive when the input falls below the
user programmable minimum voltage, VAC(off), and turn on when the input rises above VAC(on).
IVINS = 150 ´ IVINS _ 0V
IVINS = 150 ´ 0.1m A = 150 m A
VAC( on ) = 75V
V AC ( off ) = 65V
RVINS 1 =
RVINS 1 =
2 ´ VAC( on ) - VF _ BRIDGE - VINS ENABLE _ th(max)
IVINS
2 ´ 75V - 0.95V - 1.6V
= 6.9 M W
150 m A
A 6.5-M resistance is chosen.
RVINS 2 =
RVINS 2 =
VINS ENABLE _ th(max)´R VINS 1
2 ´ VAC( on ) - VINS ENABLE _ th(max) - VF _ BRIDGE
1.6V ´ 6.5M W
= 100k W
2 ´ 75V - 1.6V - 0.95V
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The capacitor on VINS, CVINS, is selected so that it's discharge time is greater than the output capacitor hold up
time. COUT was chosen to meet one-cycle hold-up time so CVINS will be chosen to meet 2.5 half-line cycles.
tCVINS _ dischrg =
tCVINS _ dischrg =
CVINS =
C
42
VINS
=
N HALF _ CYCLES
2 ´ f LINE (min)
2 .5
= 25 .6 ms
2 ´ 47 Hz
-tCVINS _ dischrg
é
ù
ê
ú
VINS BROWNOUT _ th(min)
ê
ú
RVINS 2 ´ ln
ê
æ
öú
RVINS 2
ê 0.9 ´ VIN _ RMS (min) ´ ç
÷ú
êë
è RVINS 1 + RVINS 2 ø úû
-25.6ms
é
ù
ê
ú
0.76V
ú
100k W ´ ln ê
100k W
öú
ê 0.9 ´ 85V ´ æ
ç
÷
è 6.5M W + 100k W ø ûú
ëê
= 0.63m F
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REFERENCES
These references, additional design tools, and links to additional references, including design software and
models may be found on the web at http://www.power.ti.com under Technical Documents.
Evaluation Module, 350-W Universal Input, 390-VDC Output PFC Converter, Texas Instruments Literature No.
SLUA272
Design Spreadsheet, UCC28019 Design Calculator, Texas Instruments
RELATED PRODUCTS
The following parts have characteristics similar to the UCC28019 and may be of interest.
Related Products
DEVICE
DESCRIPTION
UCC3817/18
Full-Feature PFC Controller
UC2853A
8-Pin CCM PFC Controller
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PACKAGE MATERIALS INFORMATION
www.ti.com
17-May-2007
TAPE AND REEL INFORMATION
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
Device
UCC28019DR
17-May-2007
Package Pins
D
8
Site
Reel
Diameter
(mm)
Reel
Width
(mm)
A0 (mm)
B0 (mm)
K0 (mm)
P1
(mm)
FMX
330
0
6.4
5.2
2.1
8
TAPE AND REEL BOX INFORMATION
Device
Package
Pins
Site
Length (mm)
Width (mm)
Height (mm)
UCC28019DR
D
8
FMX
342.9
336.6
20.6
Pack Materials-Page 2
W
Pin1
(mm) Quadrant
12
PKGORN
T1TR-MS
P
MECHANICAL DATA
MPDI001A – JANUARY 1995 – REVISED JUNE 1999
P (R-PDIP-T8)
PLASTIC DUAL-IN-LINE
0.400 (10,60)
0.355 (9,02)
8
5
0.260 (6,60)
0.240 (6,10)
1
4
0.070 (1,78) MAX
0.325 (8,26)
0.300 (7,62)
0.020 (0,51) MIN
0.015 (0,38)
Gage Plane
0.200 (5,08) MAX
Seating Plane
0.010 (0,25) NOM
0.125 (3,18) MIN
0.100 (2,54)
0.021 (0,53)
0.015 (0,38)
0.430 (10,92)
MAX
0.010 (0,25) M
4040082/D 05/98
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Falls within JEDEC MS-001
For the latest package information, go to http://www.ti.com/sc/docs/package/pkg_info.htm
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• DALLAS, TEXAS 75265
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