TI TPS40210DGQ

TPS40210, TPS40211
www.ti.com
SLUS772 – MARCH 2008
4.5-V TO 52-V INPUT CURRENT MODE BOOST CONTROLLER
FEATURES
CONTENTS
1
•
•
•
•
•
•
•
•
•
•
•
Suitable for Boost, Flyback, SEPIC, and LED
Driver Topologies
Wide Input Operating Voltage: 4.5 V to 52 V
Adjustable Oscillator Frequency
Fixed Frequency Current Mode Control
Internal Slope Compensation
Integrated Low-Side Driver
Programmable Closed Loop Soft Start
Overcurrent Protection
700-mV Reference (TPS40210)
263-mV Reference (TPS40211)
Low Current Disable Function
2
Electrical Characteristics
3
Typical Characteristics
5
Terminal Information
10
Application Information
12
Additional References
24
Design Examples
25
DESCRIPTION
The TPS40210 and TPS40211 are wide-input voltage
(4.5 V to 52 V), non-synchronous boost controllers.
They are suitable for topologies which require a
grounded source N-channel FET including boost,
flyback, SEPIC and various LED Driver applications.
The device features include programmable soft start,
overcurrent protection with automatic retry and
programmable oscillator frequency. Current mode
control provides improved transient response and
simplified loop compensation. The main difference
between the two parts is the reference voltage to
which the error amplifier regulates the FB pin.
APPLICATIONS
•
•
•
Device Ratings
LED Lighting
Industrial Control Systems
Battery Powered Systems
VIN
TPS40210
1
RC
2
SS
3
VOUT
VDD 10
BP
9
DIS/EN
GDRV
8
4
COMP
ISNS
7
5
FB
GND
6
RSENSE
UDG-07110
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008, Texas Instruments Incorporated
TPS40210, TPS40211
www.ti.com
SLUS772 – MARCH 2008
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
PACKAGE
PACKAGE LEAD
PACKAGE
AVAILABILITY
10-Pin MSOP PowerPAD
DGQ
Available
10-Pin SON
DRC
Preview
TJ
TAPE AND REEL
QUANTITY
PART NUMBER
2500
TPS40210DGQR
80
TPS40210DGQ
3000
TPS40210DRCR
-40°C to 125°C
10-Pin MSOP PowerPAD
DGQ
Available
-40°C to 125°C
10-Pin SON
DRC
Preview
250
TPS40210DRCT
2500
TPS40211DGQR
80
TPS40211DGQ
3000
TPS40211DRCR
250
TPS40211DRCT
DEVICE RATINGS
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted (1)
TPS40210
TPS40211
Input voltage range
Output voltage range
VDD
–0.3 to 52
RC, SS, FB, DIS/EN
–0.3 to 10
ISNS
–0.3 to 8
COMP, BP, GDRV
–0.3 to 9
TJ
Operating junction temperature range
–40 to 150
Tstg
Storage temperature
–55 to 150
(1)
UNIT
V
°C
Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
NOM
MAX
UNIT
VVDD
Input voltage
4.5
52
V
TJ
Operating Junction temperature
-40
125
°C
PACKAGE DISSIPATION RATINGS
PACKAGE
AIRFLOW (LFM)
RθJA High-K Board (1)
(°C/W)
Power Rating (W)
TA = 25°C
Power Rating (W)
TA = 85°C
10-Pin MSOP PowerPAD
(DGQ)
0 (Natural Convection)
57.7
1.73
0.693
10-Pin SON (DRC)
0 (Natural Convection)
47.9
2.08
0.835
(1)
Ratings based on JEDEC High Thermal Conductivity (High K) Board. For more information on the test method, see TI Technical Brief
SZZA017.
ELECTROSTATIC DISCHARGE (ESD) PROTECTION
MIN
TYP
Human Body Model (HBM)
1500
Charged Device Model (CDM)
1500
2
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MAX
UNIT
V
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SLUS772 – MARCH 2008
ELECTRICAL CHARACTERISTICS
TJ = –40°C to 125°C, VVDD= 12 Vdc, all parameters at zero power dissipation (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
TPS40210 COMP = FB, 4.5 ≤ VVDD ≤ 52 V, TJ = 25°C
693
700
707
TPS40211 COMP=FB, 4.5 ≤ VVDD ≤ 52 V, TJ = 25°C
254
260
266
COMP = FB, 4.5 ≤ VVDD ≤ 52 V, -40°C ≤ TJ ≤
TPS40210
125°C
686
700
714
COMP = FB, 4.5 ≤ VVDD ≤ 52 V, -40°C ≤ TJ ≤
125°C
250
260
270
UNIT
VOLTAGE REFERENCE
Feedback voltage range
VFB
TPS40211
mV
INPUT SUPPLY
VVDD
Input voltage range
IVDD
Operating current
4.5
52
V
4.5 ≤ VVDD ≤ 52 V, no switching, VDIS < 0.8
1.5
2.5
mA
2.5 ≤ VDIS ≤ 7 V
10
20
µA
530
µA
VVDD < VUVLO(on), VDIS < 0.8
UNDERVOLTAGE LOCKOUT
VUVLO(on)
Turn on threshold voltage
4.00
4.25
4.50
V
VUVLO(hyst)
UVLO hysteresis
140
195
240
mV
OSCILLATOR
fOSC
VSLP
Oscillator frequency range (1)
35
Oscillator frequency
RRC = 182 kΩ, CRC = 330 pF
Frequency line regulation
4.5 ≤ VDD ≤ 52 V
Slope compensation ramp
260
1000
300
-20%
520
340
kHz
7%
620
720
275
400
mV
PWM
VVDD = 12V (1)
tON(min)
Minimum pulse width
90
200
tOFF(min)
Minimum off time
170
200
VVLY
Valley voltage
1.2
V
700
mV
VVDD = 30V
ns
SOFT-START
VSS(ofst)
Offset voltage from SS pin to error
amplifier input
RSS(chg)
Soft-start charge resistance
320
430
600
RSS(dchg)
Soft-start discharge resistance
840
1200
1600
1.5
3.0
MHz
60
80
dB
kΩ
ERROR AMPLIFIER
GBWP
Unity gain bandwidth product (1)
(1)
AOL
Open loop gain
IIB(FB)
Input bias current (current out of FB
pin)
ICOMP(src)
Output source current
VFB = 0.6 V, VCOMP = 1 V
100
250
µA
ICOMP(snk)
Output sink current
VFB = 1.2 V, VCOMP = 1 V
1.2
2.5
mA
4.5 ≤ VDD < 52 V, -40°C ≤ TJ ≤ 125°C
120
150
100
300
nA
OVERCURRENT PROTECTION
VISNS(oc)
Overcurrent detection threshold (at
ISNS pin)
DOC
Overcurrent duty cycle (1)
VSS(rst)
Overcurrent reset threshold voltage (at
SS pin)
TBLNK
Leading edge blanking (1)
(1)
180
mV
2%
100
150
350
75
mV
ns
Ensured by design. Not production tested.
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ELECTRICAL CHARACTERISTICS (continued)
TJ = –40°C to 125°C, VVDD= 12 Vdc, all parameters at zero power dissipation (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
4..2
5.6
7.2
V/V
1
3
µA
CURRENT SENSE AMPLIFIER
ACS
Current sense amplifier gain
IB(ISNS)
Input bias current
DRIVER
IGDRV(src)
Gate driver source current
VGDRV = 4 V, TJ = 25°C
375
400
IGDRV(snk)
Gate driver sink current
VGDRV = 4 V, TJ = 25°C
330
400
7
8
9
mA
LINEAR REGULATOR
VBP
Bypass voltage output
0 mA < IBP < 15 mA
V
DISABLE/ENABLE
VDIS(en)
Turn on voltage
0.7
1.3
V
VDIS(hys)
Hysteresis voltage
25
130
220
mV
RDIS
DIS pin pulldown resistance
0.7
1.1
1.5
MΩ
4
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SLUS772 – MARCH 2008
TYPICAL CHARACTERISTICS
FREQUENCY
vs
TIMING RESISTANCE
SWITCHING FREQUENCY
vs
DUTY CYCLE
1200
1200
68 pF
CT(pF)
470
220
100
68
33
fSW - Frequency - kHz
100
800
100pF
600
220 pF
400
1000
fSW - Frequency - kHz
33pF
800
600
400
200
200
470 pF
0
100
0
200
300 400 500 600 700 800
RT - Timing Resistance - kW
0
900 1000
0.2
0.4
0.6
0.8
D - Duty Cycle
Figure 1.
Figure 2.
QUIESCENT CURRENT
vs
JUNCTION TEMPERATURE
SHUTDOWN CURRENT
vs
JUNCTION TEMPERATURE
1.0
1.2
6
1.4
52 V
4.5 V
1.0
12 V
0.8
0.6
0.4
VVDD
12 V
4.5 V
52 V
0.2
5
IVDD – Shutdown Current – mA
IVDD – Quiescent Current – mA
1.2
4
3
2
1
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
Figure 3.
Figure 4.
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TYPICAL CHARACTERISTICS (continued)
REFERENCE VOLTAGE CHANGE
vs
JUNCTION TEMPERATURE
REFERENCE VOLTAGE CHANGE
vs
INPUT VOLTAGE
0.5
0.4
0.4
0.2
0.0
-0.2
-0.4
-0.6
4.5 V
VVDD
12 V
4.5 V
52 V
12 V
VFB – Reference Voltage Change – %
VFB – Reference Voltage Change – %
52 V
0.1
0.0
-0.1
-0.2
-0.3
-0.5
0
10
20
30
40
VVDD – Input Voltage – V
Figure 5.
Figure 6.
UNDERVOLTAGE LOCKOUT THRESHOLD
vs
JUNCTION TEMPERATURE
OVERCURRENT THRESHOLD
vs
JUNCTION TEMPERATURE
UVLO
60
4.5 V
Off
On
UVLO On
4.15
4.10
4.05
VISNS(OC) – Overcurrent Threshold – mV
VVDD
4.20
154
153
4.5 V
7.5 V
12 V & 20 V
30 V
7.5 V
152
30 V
151
150
12 V & 20 V
149
148
UVLO Off
4.00
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
147
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
Figure 7.
6
50
155
4.30
VUVLO – Undervoltage Lockout Threshold – V
0.2
-0.4
-0.8
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
4.25
0.3
Figure 8.
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TYPICAL CHARACTERISTICS (continued)
SWITCHING FREQUENCY CHANGE
vs
JUNCTION TEMPERATURE
155
5
154
4
fOSC – Switching Frequency Change – %
VISNS(OC) – Overcurrent Threshold – mV
OVERCURRENT THRESHOLD
vs
INPUT VOLTAGE
153
152
151
150
149
148
147
146
145
5
10
15
20
25
30 35
VVDD – Input Voltage – V
40
Slope Compensation Ratio (VVDD/VSLP)
4.5 V
1
12 V
0
-1
30 V
-2
VVDD (V)
4.5 V
12 V
30 V
-3
-4
Figure 9.
Figure 10.
OSCILLATOR AMPLITUDE
vs
JUNCTION TEMPERATURE
SOFT-START CHARGE/DISCHARGE RESISTANCE
vs
JUNCTION TEMPERATURE
1400
29
27
2
-5
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
45
4.5 V
RSS – Soft Start Charge/Discharge Resistance - kW
0
3
RSS(DSCH) Discharge
1200
25
1000
23
24 V
12 V
21
19
VVDD (V)
36 V
12 V
24 V
36 V
4.5 V
17
15
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
800
600
400
200
RSS(CHG) Charge
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 11.
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
FB BIAS CURRENT
vs
JUNCTION TEMPERATURE
COMPENSATION SOURCE CURRENT
vs
JUNCTION TEMPERATURE
300
ICOMP(SRC) – Compensation Source Current – mA
180
IIB(FB) – Feedback Bias Current – nA
160
140
120
100
80
60
40
20
150
100
50
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 13.
Figure 14.
COMPENSATION SINK CURRENT
vs
JUNCTION TEMPERATURE
VALLEY VOLTAGE CHANGE
vs
JUNCTION TEMPERATURE
5
4
250
VVLY – Valley Voltage Change – %
ICOMP(SNK) – Compensation Sink Current – mA
200
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
300
8
250
200
150
100
50
3
2
1
0
-1
-2
-3
-4
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
-5
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
Figure 15.
Figure 16.
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TYPICAL CHARACTERISTICS (continued)
REGULATOR VOLTAGE
vs
JUNCTION TEMPERATURE
DIS/EN TURN-ON THRESHOLD
vs
JUNCTION TEMPERATURE
1.10
8.8
VDIS(EN) – DIS/EN Turn-On Threshold – mV
1.09
VBP – Regulator Voltage – V
8.6
1.08
ILOAD = 0 mA
8.4
1.07
1.06
8.2
1.05
8.0
7.8
1.06
1.03
ILOAD = 5 mA
1.02
7.6
1.01
7.4
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
1.00
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
Figure 17.
Figure 18.
CURRENT SENSE AMPLIFIER GAIN
vs
JUNCTION TEMPERATURE
ACS – Current Sense Amplifier Gain – V/V
7
6
5
4
3
2
1
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Juncton Temperature – ° C
Figure 19.
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DEVICE INFORMATION
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
4
O
Error amplifier output. Connect control loop compensation network between COMP pin and FB pin.
DIS/EN
3
I
Disable pin. Pulling this pin high, places the part into a shutdown mode. Shutdown mode is characterized by
a very low quiescent current. While in shutdown mode, the functionality of all blocks is disabled and the BP
regulator is shut down. This pin has an internal 1-MΩ pull-down resistor to GND. Leaving this pin
unconnected enables the device.
FB
5
I
Error amplifier inverting input. Connect a voltage divider from the output to this pin to set output voltage.
Compensation network is connected between this pin and COMP.
GDRV
8
O
Connect the gate of the power N channel MOSFET to this pin.
GND
6
-
Device ground.
ISNS
7
I
Current sense pin. Connect an external current sensing resistor between this pin and GND. The voltage on
this pin is used to provide current feedback in the control loop and detect an overcurrent condition. An
overcurrent condition is declared when ISNS pin voltage exceeds the overcurrent threshold voltage, 150 mV
typical.
RC
1
I
Switching frequency setting pin. Connect capacitor from RC pin to GND. Connect a resistor from RC pin
toVDD of the IC power supply and a capacitor from RC to GND.
SS
2
I
Soft-start time programming pin. Connect capacitor from SS pin to GND to program converter soft-start time.
This pin also functions as a timeout timer when the power supply is in an overcurrent condition.
BP
9
O
Regulator output pin. Connect a 1.0-µF bypass capacitor from this pin to GND.
VDD
10
I
System input voltage. Connect a local bypass capacitor from this pin to GND. Depending on the amount of
required slope compensation, this pin can be connected to the converter output. See Application Information
section for additional details.
NAME
NO.
COMP
10
DGQ PowerPAD PACKAGE
(TOP VIEW)
DRC PACKAGE
(TOP VIEW)
DGQ PowerPAD PACKAGE
(Top View)
DRC SURFACE MOUNT PACKAGE
(Top View)
RC
1
10
VDD
SS
2
9
BP
DIS/EN
3
8
COMP
4
FB
5
1
10
RC
1
SS
2
9
BP
GDRV
DIS/EN
3
8
GDRV
7
ISNS
COMP
4
7
ISNS
6
GND
FB
5
6
GND
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FUNCTIONAL BLOCK DIAGRAM
DIS/EN
3
COMP
4
FB
5
10 VDD
+
+
SS
2
LDO
OC Fault
Soft Start
and
Overcurrent
E/A
SS Ref
9
BP
8
GDRV
6
GND
7
ISNS
700 mV
Driver
Enable E/A
PWM
Logic
Gain = 6
+
RC
Oscillator
and
Slope
Compensation
1
OC Fault
150 mV
UVLO
+
LEB
UDG-07107
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APPLICATION INFORMATION
Minimum On-Time and Off Time Considerations
The TPS40210 has a minimum off time of approximately 200 ns and a minimum on time of 300 ns. These two
constraints place limitations on the operating frequency that can be used for a given input to output conversion
ratio. See Figure 2 for the maximum frequency that can be used for a given duty cycle.
The duty cycle at which the converter operates is dependent on the mode in which the converter is running. If the
converter is running in discontinuous conduction mode, the duty cycle varies with changes to the load much
more than it does when running in continuous conduction mode.
In continuous conduction mode, the duty cycle is related primarily to the input and output voltages.
VOUT + VD
1
=
VIN
1- D
æ æ
VIN
D = ç1 - ç
ç
è è VOUT + VD
(1)
öö
÷ ÷÷
øø
(2)
In discontinuous mode the duty cycle is a function of the load, input and output voltages, inductance and
switching frequency.
D=
2 ´ (VOUT + VD )´ IOUT ´ L ´ fSW
(VIN )2
(3)
All converters using a diode as the freewheeling or catch component have a load current level at which they
transition from discontinuous conduction to continuous conduction. This is the point where the inductor current
just falls to zero. At higher load currents, the inductor current does not fall to zero but remains flowing in a
positive direction and assumes a trapezoidal wave shape as opposed to a triangular wave shape. This load
boundary between discontinuous conduction and continuous conduction can be found for a set of converter
parameters as follows.
2
VOUT + VD - VIN )´ (VIN )
(
IOUT(crit) =
2
2 ´ (VOUT + VD ) ´ fSW ´ L
(4)
For loads higher than the result of Equation 4, the duty cycle is given by Equation 2 and for loads less that the
results of Equation 4, the duty cycle is given Equation 3. For Equations 1 through 4, the variable definitions are
as follows.
• VOUT is the output voltage of the converter in V
• VD is the forward conduction voltage drop across the rectifier or catch diode in V
• VIN is the input voltage to the converter in V
• IOUT is the output current of the converter in A
• L is the inductor value in H
• fSW is the switching frequency in Hz
12
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Setting the Oscillator Frequency
The oscillator frequency is determined by a resistor and capacitor connected to the RC pin of the TPS40210. The
capacitor is charged to a level of approximately VVDD/20 by current flowing through the resistor and is then
discharged by a transistor internal to the TPS40210. The required resistor for a given oscillator frequency is
found from either Figure 1 or Equation 5.
1
RT =
5.8 ´ 10
-8
´ fSW ´ C T + 8 ´ 10
- 10
2
´ fSW + 1.4 ´ 10
-7
´ fSW - 1.5 ´ 10 - 4 + 1.7 ´ 10 - 6 ´ C T - 4 ´ 10 - 9 ´ C T 2
(5)
where
•
•
•
RT is the timing resistance in kΩ
fSW is the switching frequency in kHz
CT is the timing capacitance in pF
For most applications a capacitor in the range of 68 pF to 120 pF gives the best results. Resistor values should
be limited to between 100 kΩ and 1 MΩ as well. If the resistor value falls below 100 kΩ, decrease the capacitor
size and recalculate the resistor value for the desired frequency. As the capacitor size decreases below 47 pF,
the accuracy of Equation 5 degrades and empirical means may be needed to fine tune the timing component
values to achieve the desired switching frequency.
Current Sense and Overcurrent
The tps40210 and TPS40211 are current mode controllers and use a resistor in series with the source terminal
power FET to sense current for both the current mode control and overcurrent protection. The device enters a
current limit state if the voltage on the ISNS pin exceeds the current limit threshold voltage VISNS(oc) from the
electrical specifications table. When this happens the controller discharges the SS capacitor through a relatively
high impedance and then attempt to restart. The amount of output current that causes this to happen is
dependent on several variables in the converter.
TPS40120/11
VIN
10 VDD
TPS40210/11
L
RT
VOUT
VDD 10
1
RC
GDRV
8
ISNS
7
CT
RIFLT
6
GND
CIFLT
GND
RISNS
6
UDG-07119
UDG-07120
Figure 20. Oscillator Components
Figure 21. Current Sense Components
The load current overcurrent threshold is set by proper choice of RISNS. If the converter is operating in
discontinuous mode the current sense resistor is found in Equation 6.
RISNS =
fSW ´ L ´ VISNS(oc)
2 ´ L ´ fSW ´ IOUT(oc) ´ (VOUT + VD - VIN )
(6)
If the converter is operating in continuous conduction mode RISNS can be found in Equation 7.
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RISNS =
VISNS
VISNS
=
æ IOUT ö æ IRIPPLE ö æ IOUT ö æ D ´ VIN ö
+
ç 1- D ÷ + ç
2 ÷ø çç (1 - D ) ÷÷ ç 2 ´ fSW ´ L ÷
è
ø è
ø
è
ø è
(7)
where
•
•
•
•
•
•
•
•
RISNS is the value of the current sense resistor in Ω.
VISNS(oc) is the overcurrent threshold voltage at the ISNS pin (from electrical specifications)
D is the duty cycle (from Equation 2)
fSW is the switching frequency in Hz
VIN is the input voltage to the power stage in V (see text)
L is the value of the inductor in H
IOUT(oc) is the desired overcurrent trip point in A
VD is the drop across the diode in Figure 21
The TPS40210/11 has a fixed undervoltage lockout (UVLO) that allows the controller to start at a typical input
voltage of 4.25 V. If the input voltage is slowly rising, the converter might have less than its designed nominal
input voltage available when it has reached regulation. As a result, this may decreases the apparent current limit
load current value and must be taken into consideration when selecting RISNS. The value of VIN used to calculate
RISNS must be the value at which the converter finishes startup. The total converter output current at startup is
the sum of the external load current and the current required to charge the output capacitor(s). See the Soft Start
section of this datasheet for information on calculating the required output capacitor charging current.
The topology of the standard boost converter has no method to limit current from the input to the output in the
event of a short circuit fault on the output of the converter. If protection from this type of event is desired, it is
necessary to use some secondary protection scheme such a fuse or rely on the current limit of the upstream
power source.
Current Sense and Sub-Harmonic Instability
A characteristic of peak current mode control results in a condition where the current control loop can exhibit
instability. This results in alternating long and short pulses from the pulse width modulator. The voltage loop
maintains regulation and dioes not oscillate, but the output ripple voltage increases. The condition occurs only
when the converter is operating in continuous conduction mode and the duty cycle is 50% or greater. The cause
of this condition is described in Texas Instruments literature number SLUA101, available at www.ti.com. The
remedy for this condition is to apply a compensating ramp from the oscillator to the signal going to the pulse
width modulator. In the TPS40210/11 the oscillator ramp is applied in a fixed amount to the pulse width
modulator. The slope of the ramp is given in Equation 8.
æV
ö
s e = fSW ´ ç VDD ÷
20
è
ø
(8)
To ensure that the converter does not enter into sub-harmonic instability, the slope of the compensating ramp
signal must be at least half of the down slope of the current ramp signal. Since the compensating ramp is fixed in
the TPS40210/11, this places a constraint on the selection of the current sense resistor.
The down slope of the current sense wave form at the pulse width modulator is described in Equation 9.
m2 =
A CS ´ RISNS ´ (VOUT + VD - VIN )
L
(9)
Since the slope compensation ramp must be at least half, and preferably equal to the down slope of the current
sense waveform seen at the pulse width modulator, a maximum value is placed on the current sense resistor
when operating in continuous mode at 50% duty cycle or greater. For design purposes, some margin should be
applied to the actual value of the current sense resistor. As a starting point, the actual resistor chosen should be
80% or less that the value calculated in Equation 10. This equation calculates the resistor value that makes the
slope compensation ramp equal to one half of the current ramp downslope. Values no more than 80% of this
result would be acceptable.
14
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RISNS(max) =
VVDD ´ L ´ fSW
60 ´ (VOUT + VD - VIN )
(10)
where
•
•
•
•
•
•
•
Se is the slope of the voltage compensating ramp applied to the pulse width modulator in V/s
fSW is the switching frequency in Hz
VVDD is the voltage at the VDD pin in V
m2 is the down slope of the current sense waveform seen at the pulse width modulator in V/s
RISNS is the value of the current sense resistor in Ω
VOUT is the converter output voltage VIN is the converter power stage input voltage
VD is the drop across the diode in Figure 21
It is possible to increase the voltage compensation ramp slope by connecting the VDD pin to the output voltage
of the converter instead of the input voltage as shown in Figure 21. This can help in situations where the
converter design calls for a large ripple current value in relation to the desired output current limit setting.
NOTE:
Connecting the VDD pin to the output voltage of the converter affects the startup
voltage of the converter since the controller undervoltage lockout (UVLO) circuit
monitors the VDD pin and senses the input voltage less the diode drop before startup.
The effect is to increase the startup voltage by the value of the diode voltage drop.
If an acceptable RISNS value is not available, the next higher value can be used and the signal from the resistor
divided down to an acceptable level by placing another resistor in parallel with CISNS.
Current Sense Filtering
In most cases, a small filter placed on the ISNS pin improves performance of the converter. These are the
components RIFLT and CIFLT in Figure 21. The time constant of this filter should be approximately 10% of the
nominal pulse width of the converter. The pulse width can be found using Equation 11.
tON =
D
fSW
(11)
The suggested time constant is then
RIFLT ´ CIFLT = 0.1´ tON
(12)
The range of RIFLT should be from about 1 kΩ to 5 kΩ for best results. Higher values can be used but this raises
the impedance of the ISNS pin connection more than necessary and can lead to noise pickup issues in some
layouts. CISNS should be located as close as possible to the ISNS pin as well to provide noise immunity.
Soft Start
The soft-start feature of the TPS40200 is a closed loop soft start, meaning that the output voltage follows a linear
ramp that is proportional to the ramp generated at the SS pin. This ramp is generated by an internal resistor
connected from the BP pin to the SS pin and an external capacitor connected from the SS pin to GND. The SS
pin voltage (VSS) is level shifted down by approximately VSS(ofst) (approximately 700 mV) and sent to one of the
“+” (the “+” input with the lowest voltage dominates) inputs of the error amplifier. When this level shifted voltage
(VSSE) starts to rise at time t1 (see Figure 22), the output voltage the controller expects, rises as well. Since VSSE
starts at near 0 V, the controller attempts to regulate the output voltage from a starting point of zero volts. It
cannot do this due to the converter architecture. The output voltage starts from the input voltage less the drop
across the diode (VIN - VD) and rise from there. The point at which the output voltage starts to rise (t2) is the point
where the VSSE ramp passes the point where it is commanding more output voltage than (VIN - VD). This voltage
level is labeled VSSE(1). The time required for the output voltage to ramp from a theoretical zero to the final
regulated value (from t1 to t3) is determined by the time it takes for the capacitor connected to the SS pin (CSS) to
rise through a 700 mV range, beginning at VSS(ofst) above GND.
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TPS40210/11
VSS
RSS(chg)
700 mV REF
Error Amplifier
SS
VSS(ofst)+700 mV
RSS(dchg)
VSSE(1)
t0
+
+
2
VSSE
VSS(ofst)
t1
VIN - VD
VOUT
t2
t3
DIS
UVLO
OC Fault
FB
5
COMP
4
UDG-07121
Figure 22. SS Pin Voltage adn Output Voltage
Figure 23. SS Pin Functional Circuit
The required capacitance for a given soft start time t3 – t1 in Figure 22 is calculated in Equation 13.
CSS =
tSS
æ
VBP - VSS(ofst)
RSS ´ ln ç
çV - V
SS(ofst) + VFB
è BP
(
ö
÷
÷
ø
)
(13)
where
•
•
•
•
•
•
tSS is the soft-start time
RSS(chg) is the SS charging resistance in Ω, typically 500 kΩ
CSS is the value of the capacitor on the SS pin, in F
VBP is the value of the voltage on the BP pin in V
VSS(ofst) is the approximate level shift from the SS pin to the error amplifier (~700 mV)
VFB is the error amplifier reference voltage, 700m V typical
Note that tSS is the time it takes for the output voltage to rise from 0 V to the final output voltage. Also note the
tolerance on RSS(chg) given in the electrical specifications table. This contributes to some variability in the output
voltage rise time and margin must be applied to account for it in design.
Also take note of VBP. Its value varies depending on input conditions. For example, a converter operating from a
slowly rising input initializes VBP at a fairly low value and increases during the entire startup sequence. If the
controller has a voltage above 8 V at the input and the DIS pin is used to stop and then restart the converter, VBP
is approximately 8 V for the entire startup sequence. The higher the voltage on BP, the shorter the startup time is
and conversely, the lower the voltage on BP, the longer the startup time is.
16
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The soft-start time (tSS) must be chosen long enough so that the converter can start up without going into an
overcurrent state. Since the over current state is triggered by sensing the peak voltage on the ISNS pin, that
voltage must be kept below the overcurrent threshold voltage VISNS(oc). The voltage on the ISNS pin is a function
of the load current of the converter, the rate of rise of the output voltage and the output capacitance, and the
current sensing resistor. The total output current that must be supported by the converter is the sum of the
charging current required by the output capacitor and any external load that must be supplied during startup. This
current must be less than the IOUT(oc) value used in Equation 6 or Equation 7 (depending on the operating mode
of the converter) to determine the current sense resistor value. In these equations, the actual input voltage at the
time that the controller reaches the final output voltage is the important input voltage to use in the calculations. If
the input voltage is slowly rising and is at less than the nominal input voltage when the startup time ends, the
output current limit is less than IOUT(oc) at the nominal input voltage. The output capacitor charging current must
be reduced (decrease COUT or increase the tSS) or IOUT(oc) must be increased and a new value for RISNS
calculated.
æC
ö
IC(chg) = ç OUT ÷
è tSS ø
æ
COUT
t SS > ç
ç I
-I
è OUT(oc ) EXT
(
(14)
ö
÷
÷
ø
)
(15)
where
•
•
•
•
•
IC(chg) is the output capacitor charging current in A
COUT is the total output capacitance in F
tSS is the soft start time from Equation 13
IOUT(oc) is the desired over current trip point in A
IEXT is any external load current in A
The capacitor on the SS pin (CSS) also plays a role in overcurrent functionality. It is used as the timer between
restart attempts. The SS pin is connected to GND through a resistor, RSS(dchg), whenever the controller senses an
overcurrent condition. Switching stops and nothing else happens until the SS pin discharges to the soft-start
reset threshold, VSS(rst). At this point, the SS pin capacitor is allowed to charge again through the charging
resistor RSS(chg), and the controller restarts from that point. The shortest time between restart attempts occurs
when the SS pin discharges from VSS(ofst) (approximately 700 mV) to VSS(rst) (150 mV) and then back to VSS(ofst)
and switching resumes. In actuality, this is a conservative estimate since switching does not resume until the
VSSE ramp rises to a point where it is commanding more output voltage than exists at the output of the controller.
This occurs at some SS pin voltage greater than VSS(ofst) and depends on the voltage that remains on the output
overvoltage the converter while switching has been halted. The fastest restart time can be calculated by using
Equation 16, Equation 17 and Equation 18.
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æ VSS(ofst)
tDCHG = RSS(dchg) ´ CSS ´ ln ç
ç VSS(rst)
è
(
(
ö
÷
÷
ø
(16)
) ö÷
)÷ø
æ V -V
BP
SS(rst)
tCHG = RSS(chg) ´ CSS ´ ln ç
ç V -V
SS(ofst)
è BP
(17)
tRSTRT(min ) = tCHG + tDCHG
(18)
VBP
VSS
tRSTR(min)
VSS(ofst)
VSS(rst)
T - Time
Figure 24. Soft Start During Overcurrent
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BP Regulator
The TPS40210/11 has an on board linear regulator the supplies power for the internal circuitry of the controller,
including the gate driver. This regulator has a nominal output voltage of 8 V and must be bypassed with a 1-µF
capacitor. If the voltage at the VDD pin is less than 8 V, the voltage on the BP pin ia also be less and the gate
drive voltage to the external FET ia reduced from the nominal 8 V. This should be considered when choosing a
FET for the converter.
Connecting external loads to this regulator can be done, but care must be taken to ensure that the thermal rating
of the device is observed since is no thermal shutdown feature in this controller. Exceeding the thermal ratings
cause out of specification behavior and can lead to reduced reliability. The controller dissipates more power
when there is an external load on the BP pin and is tested for dropout voltage for up to 5-mA load. When the
controller is in the disabled state, the BP pin regulator also shuts off so loads connected there power down as
well. When the controller is disabled with the DIS/EN pin, this regulator is turned off.
The total power dissipation in the controller can be calculated as follows. The total power is the sum of PQ, PG
and PE.
PQ = VVDD ´ IVDD(en)
(19)
PG = VVDD ´ Qg ´ fSW
(20)
PE = VVDD ´ IEXT
(21)
where
•
•
•
•
•
•
•
•
PQ is the quiescent power of the device in W
VVDD is the VDD pin voltage in V
IVDD(en) is the quiescent current of the controller when enabled but not switching in A
PG is the power dissipated by driving the gate of the FET in W
Qg is the total gate charge of the FET at the voltage on the BP pin in C
fSW is the switching frequency in Hz
PE is the dissipation caused be external loading of the BP pin in W
IEXT is the external load current in A
Shutdown (DIS/EN Pin)
The DIS/EN pin is an active high shutdown command for the controller. Pulling this pin above 1.2 V causes the
controller to completely shut down and enter a low current consumption state. In this state, the regulator
connected to the BP pin is turned off. There is an internal 1.1-MΩ pull-down resistor connected to this pin that
keeps the pin at GND level when left floating. If this function is not used in an application, it is best to connect
this pin to GND
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Control Loop Considerations
There are two methods to design a suitable control loop for the TPS4021x. The first and preferred if equipment is
available is to use a frequency response analyzer to measure the open loop modulator and power stage gain
and to then design compensation to fit that. The usage of these tools for this purpose is well documented with
the literature that accompanies the tool and is not be discussed here.
The second option is to make an initial guess at compensation, and then evaluate the transient response of the
system to see if the compensation is acceptable to the application or not. For most systems, an adequate
response can be obtained by simply placing a series resistor and capacitor (RFB and CFB) from the COMP pin to
the FB pin as shown in Figure 25.
VIN
TPS40210
1
RC
2
SS
3
DIS/EN
L
VOUT
VDD 10
BP
9
GDRV
8
CHF
CFB
RFB
COUT
RIFLT
4
COMP
ISNS
7
CIFLT
5
FB
GND
ROUT
RSENSE
6
R1
R2
UDG-07177
Figure 25. Basic Compensation Network
The natural phase characteristics of most capacitors used for boost outputs combined with the current mode
control provide adequate phase margin when using this type of compensation. To determine an initial starting
point for the compensation, the desired crossover frequency must be considered when estimating the control to
output gain. The model used is a current source into the output capacitor and load.
When using these equations, the loop bandwidth should be no more than 20% of the switching frequency, fSW. A
more reasonable loop bandwidth would be 10% of the switching frequency. Be sure to evaluate the transient
response of the converter over the expected load range to ensure acceptable operation.
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K CO = gM ´ ZOUT (fCO ) = 19.1A
0.13 ´ L ´
gM =
fSW
ROUT
2
V
´ 0.146 W = 2.80
(22)
0.13 ´ 10 mH ´
=
600kHz
240 W
2
(RISNS ) ´ (120 ´ RISNS + L ´ fSW ) (12mW ) ´ (120 ´ 12mW + 10 mH ´ 600kHz )
(1+ (2p ´ f ´ R
L
ZOUT = ROUT ´
(
2
ESR
2
´ COUT )
2
)
= 19.1 A
V
(23)
)
2
1 + (ROUT ) + 2 ´ ROUT ´ RESR + (RESR ) ´ (2p ´ fL ´ COUT )
(24)
where
•
•
•
•
•
•
•
•
•
•
KCO is the control to output gain of the converter, in V/V
gM is the transconductance of the power stage and modulator, in S
ROUT is the output load equivalent resistance, in Ω
ZOUT is the output impedance, including the output capacitor, in Ω
RISNS is the value of the current sense resistor, in Ω
L is the value of the inductor, in H
COUT is the value of the output capacitance, in µF
RESR is the equivalent series resistance of COUT, in Ω
fSW is the switching frequency, in Hz
fL is the desired crossover frequency for the control loop, in Hz
These equations assume that the operation is discontinuous and that the load is purely resistive. The gain in
continuous conduction can be found by evaluating Equation 23 at the resistance that gives the critical conduction
current for the converter. Loads that are more like current sources give slightly higher gains than predicted here.
To find the gain of the compensation network required for a control loop of bandwidth fL, take the reciprocal of
Equation 22.
K COMP =
1
1
=
= 0.356
K CO
2.80
(25)
The GBWP of the error amplifier is only guaranteed to be at least 1.5 MHz. If KCOMP multiplied by the fL is greater
than 750 kHz, reduce the desired loop crossover frequency until this condition is satisfied. This ensures that the
high-frequency pole from the error amplifier response with the compensation network in place does not cause
excessive phase lag at the fL and decrease phase margin in the loop.
The R-C network connected from COMP to FB places a zero in the compensation response. That zero should be
approximately 1/10th of the desired crossover frequency, fL. With that being the case, RFB and CFB can be found
from Equation 26 and Equation 27
RFB =
CFB =
R1
= R1´ K COMP
K CO
(26)
10
2p ´ fL ´ RFB
(27)
where
•
•
R1 is in fL is the loop crossover frequency desired, in Hz
RFB is the feedback resistor in CFB is the feedback capacitance in µF.
Thought not strictly necessary, it is recommended that a capacitor be added between COMP and FB to provide
high-frequency noise attenuation in the control loop circuit. This capacitor introduces another pole in the
compensation response. The allowable location of that pole frequency determines the capacitor value. As a
starting point, the pole frequency should be 10 × fL. The value of CHF can be found from Equation 28.
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CHF =
1
20p ´ fL ´ RFB
(28)
The error amplifier GBWP will usually be higher, but is ensured by design to be at least 1.5 MHz. If the gain
required in Equation 25 multiplied by 10 times the desired control loop crossover frequency, the high-frequency
pole introduced by CHF is overridden by the error amplifier capability and the effective pole is lower in frequency.
If this is the case, CHF can be made larger to provide a consistent high-frequency roll off in the control loop
design. Equation 29 calculates the required CHF in this case.
CHF =
1
6
2p ´ 1.5 ´ (10 ) ´ RFB
(29)
where
•
•
CHF is the high-frequency roll-off capacitor value in µF
RFB is the mid band gain setting resistor value in Ω
GATE DRIVE CIRCUIT
Some applications benefit from the addition of a resistor connected between the GDRV pin and the gate of the
switching MOSFET. In applications that have particularly stringent load regulation (under 0.75%) requirements
and operate from input voltages above 5 V, or are sensitive to pulse jitter in the discontinuous conduction region,
this resistor is recommended. The recommended starting point for the value of this resistor can be calculated
from Equation 30.
RG =
105
QG
(30)
where
•
•
QG is the MOSFET total gate charge at 8-V VGS in nC
RG is the suggested starting point gate resistance in Ω
VIN
TPS40210/11
L
VDD 10
VOUT
RG
GDRV
8
ISNS
7
GND
6
UDG-07196
Figure 26. Gate Drive Resistor
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TPS40211
The only difference between the TPS40210 and the TPS40211 is the reference voltage that the error amplifier
uses to regulate the output voltage. The TPS40211 uses a 250-mV reference and is intended for applications
where the output is actually a current instead of a regulated voltage. A typical example of an application of this
type is an LED driver. An example schematic is shown in Figure 27.
VIN
IOUT
L
TPS40210/11
1
RC
2
SS
3 DIS/EN
VDD 10
BP
9
GDRV
8
4
COMP
ISNS 7
5
FB
GND
RIFB
6
UDG-07197
Figure 27. Typical LED Drive Schematic
The current in the LED string is set by the choice of the resistor RISNS as shown in Equation 31.
RIFB =
VFB
IOUT
(31)
where
•
•
•
RIFB is the value of the current sense resistor for the LED string in Ω
VFB is the reference voltage for the TPS40211 in V (0.263 V typ)
IOUT is the desired DC current in the LED string in A
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ADDITIONAL REFERENCES
Related Devices
The following devices have characteristics similar to the TPS40210 and may be of interest.
Related Parts
DEVICE
DESCRIPTION
TPS6100X
Single- and Dual-Cell Boost Converter with Strart-up into Full Load
TPS6101X
High Efficiency 1-Cell and 2-Cell Bost Converters
TPS6300X
High Effiency Single Inductor Buck-Boost Converter with 1.8A Switches
References
These references may be found on the web at www.power.ti.com under Technical Documents. Many design
tools and links to additional references, may also be found at www.power.ti.com
1. Design and Application Guide for High Speed MOSFET Gate Drive Circuits, SEM 1400, 2001 Seminar
Series
2. Designing Stable Control Loops, SEM 1400, 2001 Seminar Series
3. Additional PowerPADTM information may be found in Applications Briefs SLMA002 and SLMA004
4. QFN/SON PCB Attachment, Texas Instruments Literature Number SLUA271, June 2002
24
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DESIGN EXAMPLE 1
12-V to 24-V Non-Synchronous Boost Regulator
The following example illustrates the design process and component selection for a 12-V to 24-V
non-synchronous boost regulator using the TPS40210 controller.
+
+
Figure 28. TPS40210 Design Example – 8-V to 24-V at 2-A
TPS40210 Design Example Specifications
PARAMETER
CONDITIONS
MIN NOM MAX
UNIT
INPUT CHARACTERISTICS
VIN
Input voltage
IIN
Input current
8
12
No load input current
VIN(UVLO)
14
4.4
0.05
Input undervoltage lockout
4.5
V
A
V
OUTPUT CHARACTERISTICS
VOUT
Output voltage
23.5
24.0
Line regulation
24.5
V
1%
Load regulation
1%
VOUT(ripple)
Output voltage ripple
500
IOUT
Output current
IOCP
Output overcurrent inception point
8 V ≤ VIN ≤ 14 V
0.2
1
2.0
3.5
mVPP
A
Transient response
ΔI
Load step
1
A
Load slew rate
1
A/µs
500
mV
5
ms
Overshoot threshold voltage
Settling time
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TPS40210 Design Example Specifications (continued)
PARAMETER
CONDITIONS
MIN NOM MAX
UNIT
600
kHz
SYSTEM CHARACTERISTICS
fSW
Switching frequency
ηPK
Peak efficiency
VIN = 12 V, 0.2 A ≤ IOUT ≤ 2 A
95%
η
Full load efficiency
VIN = 12 V, IOUT = 2 A
94%
TOP
Operating temperature range
10 V ≤ VIN ≤ 14 V, 0.2 A ≤ IOUT ≤ 2 A
°C
25
MECHANICAL DIMENSIONS
W
Width
1.5
L
Length
1.5
h
Height
0.5
in
Step-By-Step Design Procedure
Duty Cycle Estimation
The duty cycle of the main switching MOSFET is estimated using Equation 32 and Equation 33.
VOUT - VIN(max) + VFD
DMIN »
VOUT + VFD
DMAX »
24 V - 14 V + 0.5 V
= 42.8%
24 V + 0.5 V
=
(32)
VOUT - VIN(m in) + VFD
24 V - 8 V + 0.5 V
=
= 67.3%
VOUT + VFD
24 V + 0.5 V
(33)
Using and estimated forward drop of 0.5 V for a schottkey rectifier diode, the approximate duty cycle is 42.8%
(minimum) to 67.3% (maximum).
Inductor Selection
The peak-to-peak ripple is limited to 30% of the maximum output current.
ILrip(m ax) = 0.3 ´
IOUT(m ax)
1 - DMIN
= 0.3 ´
2
= 1.05 A
1 - 0.428
(34)
The minimum inductor size can be estimated using Equation 35.
LMIN »
VIN(max)
ILrip(max)
´ DMIN ´
1
fSW
=
14 V
1
´ 0.673 ´
= 9.5 mH
1.05 A
600kHz
(35)
The next higher standard inductor value of 10 µH is selected. The ripple current is estimated by Equation 36.
IRIPPLE »
VIN
1
12 V
1
´D´
=
´ 0.50 ´
= 1.02 A
fSW
L
10 m H
600 kHz
(36)
V
1
8V
1
IRIPPLE(Vinmin) » IN ´ D ´
=
´ 0.673 ´
= 0.89 A
fSW 10 mH
L
600kHz
(37)
The worst case peak-to-peak ripple current occurs at 50% duty cycle and is estimated as 1.02 A. Worst case
RMS current through the inductor is approximated by Equation 38.
ILrms =
(
IL(avg)
2
2
)
+
(
1 I
12 RIPPLE
2
)
æ IOUT(max) ö
» ç
+
ç
÷÷
è 1 - DMAX ø
2
(112IRIPPLE(VINmin) )
2
2
æ
ö
= ç
÷ +
è 1 - 0.673 ø
2
((112)´ 0.817A )
= 6.13 Arms
(38)
The worst case RMS inductor current is 6.13 Arms. The peak inductor current is estimated by Equation 39.
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ILpeak »
IOUT(max)
1 - DMAX
+ (12 )IRIPPLE(Vinmin) =
2
+ (12 )0.718 = 6.57 A
1 - 0.673
(39)
A 10-µH inductor with a minimum RMS current rating of 6.13 A and minimum saturation current rating of 6.57 A
must be selected. A TDK RLF12560T-100M-7R5 7.5-A 10-µH inductor is selected.
This inductor power dissipation is estimated by Equation 40.
2
PL » (ILrms ) ´ DCR
(40)
The TDK RLF12560T-100M-7R5 12.4-mΩ DCR dissipates 466 mW of power.
Rectifier Diode Selection
A low-forward voltage drop schottky diode is used as a rectifier diode to reduce its power dissipation and improve
efficiency. Using 80% derating, on VOUT for ringing on the switch node, the rectifier diode minimum reverse
break-down voltage is given by Equation 41.
V
V(BR)R(min) ³ OUT = 1.25 ´ VOUT = 1.25 ´ 24 V = 30 V
0.8
(41)
The diode must have reverse breakdown voltage greater than 30 V. The rectifier diode peak and average
currents are estimated by Equation 42 and Equation 43.
ID (avg ) » IOUT (m ax ) = 2 A
(42)
ID(peak ) = IL(peak ) = 6.57 A
(43)
For this design, 2-A average and 6.57-A peak is
The power dissipation in the diode is estimated by Equation 44.
PD(max) » VF ´ IOUT(max) = 0.5 V ´ 2 A = 1W
(44)
For this design, the maximum power dissipation is estimated as 1 W. Reviewing 30-V and 40-V schottky diodes,
the MBRS340T3, 40-V, 3-A diode in an SMC package is selected. This diode has a forward voltage drop of
0.48-V at 6-A, so the conduction power dissipation is approximately 960 mW, less than half its rated power
dissipation.
Output Capacitor Selection
Output capacitors must be selected to meet the required output ripple and transient specifications.
COUT = 8
ESR =
IOUT ´ D
æ 2 A ´ 0.673 ö
1
1
´
= 8ç
= 35 mF
÷´
VOUT(ripple) fSW
è 500mV ø 600kHz
(45)
VOUT(ripple )
7
7
500mV
´
= ´
= 95mW
8 IL(peak ) - IOUT 8 6.57 A - 2 A
(46)
A Panasonic EEEFC1V330P 35V 33-µF, 120-mΩ bulk capacitor and 6.8-µF ceramic capacitor is selected to
provide the required capacitance and ESR at the switching frequency. The combined capacitance of 39.8 µF and
60 mΩ are used in compensation calculations.
Input Capacitor Selection
Since a boost converter has continuous input current, the input capacitor senses only the inductor ripple current.
The input capacitor value can be calculated by Equation 47 and Equation 48 .
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CIN >
IL(ripple )
4 ´ VIN(ripple ) ´ fSW
ESR <
VIN(ripple )
2 ´ IL(ripple )
=
=
1.02 A
= 7.0 mF
4 ´ 60mV ´ 600kHz
(47)
60mV
= 30mW
2 ´ 1.02 A
(48)
For this design, to meet a maximum input ripple of 60 mV, a minimum 7.0-µF input capacitor with ESR less than
30 mΩ is needed. A 10-µF X7R ceramic capacitor is selected.
Current Sense and Current Limit
The maximum allowable current sense resistor value is limited by both the current limit and sub-harmonic
stability. These two limitations are given by Equation 49 and Equation 50.
RISNS <
VOCP(min)
(
1.1´ IL(peak ) + IDrive
RISNS <
=
)
110mV
= 14.2mW
1.1´ 6.57 A + 0.50 A
(49)
VDDMAX ´ L ´ fSW
14 V ´ 10 mH ´ 600kHz
=
= 133mW
60 ´ (VOUT + Vfd - VIN ) 60 ´ (24 V + 0.48 V - 14 V)
(50)
The current limit requires a resistor less than 14.2 mΩ and stability requires a sense resistor less than 133 mΩ. A
10-mΩ resistor is selected. Approximately 2-mΩ of routing resistance added in compensation calculations.
Current Sense Filter
To remove switching noise from the current sense, an R-C filter is placed between the current sense resistor and
the ISNS pin. A resistor with a value between 1 kΩ and 5 kΩ is selected and a capacitor value is calculated by
Equation 51.
CIFLT =
0.1´ DMIN
0.1´ 0.428
=
= 71pF
fSW ´ RIFLT 600kHz ´ 1kW
(51)
For a 1-kΩ filter resistor, 71 pF is calculated and a 100-pF capacitor is selected.
Switching MOSFET Selection
The TPS40210 drives a ground referenced N-channel FET. The RDS(on) and gate charge are estimated based on
the desired efficiency target.
æ1 ö
æ1 ö
æ 1
ö
PDISS(total) » POUT ´ ç - 1÷ = VOUT ´ IOUT ´ ç - 1÷ = 24 V ´ 2 A ´ ç
- 1÷ = 2.526 W
0.95
h
h
è
ø
è
ø
è
ø
(52)
For a target of 95% efficiency with a 24 V Input voltage at 2 A, maximum power dissipation is limited to 2.526 W.
The main power dissipating devices are the MOSFET, inductor, diode, current sense resistor and the integrated
circuit, the TPS40210.
PFET < PDISS(total) - PL - PD - PRisns - VIN(max) ´ IVDD
(53)
This leaves 740 mW of power dissipation for the MOSFET. This can likely cause an SO-8 MOSFET to get too
hot, so power dissipation is limited to 500 mW. Allowing half for conduction and half for switching losses, we can
determine a target RDS(on) and QGS for the MOSFET by Equation 54 and Equation 55.
QGS <
3 ´ PFET ´ IDRIVE
3 ´ 0.50 W ´ 0.50 A
=
= 13.0nC
2 ´ VOUT ´ IOUT ´ fSW 2 ´ 24 V ´ 2 A ´ 600kHz
(54)
A target MOSFET gate-to-source charge of less than 13.0 nC is calculated to limit the switching losses to less
than 250 mW.
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RDS(on ) <
PFET
2
2 ´ (IRMS ) ´ D
=
0.50 W
2 ´ 6.132 ´ 0.674
= 9.8mW
(55)
A target MOSFET RDS(on) of 9.8 mΩ is calculated to limit the conduction losses to less than 250 mW. Reviewing
30-V and 40-V MOSFETs, an Si4386DY 9-mΩ MOSFET is selected. A gate resistor was added per equation
(30). The maximum gate charge at Vgs=8 V for the Si4386DY is 33.2 nC, this impiles RG = 3.3 Ω.
Feedback Divider Resistors
The primary feedback divider resistor (RFB) from VOUT to FB should be selected between 10-kΩ and 100-kΩ to
maintain a balance between power dissipation and noise sensitivity. For a 24-V output a high feedback
resistance is desirable to limit power dissipation so RFB = 51.1 kΩ is selected.
RBIAS =
VFB ´ RFB
0.700 V ´ 51.1kW
=
= 1.53kW
VOUT - VFB
24 V - 0.700 V
(56)
RBIAS = 1.50 kΩ is selected.
Error Amplifier Compensation
While current mode control typically only requires Type II compensation, it is desirable to layout for Type III
compensation to increase flexibility during design and development.
Current mode control boost converters have higher gain with higher output impedance, so it is necessary to
calculate the control loop gain at the maximum output impedance, estimated by Equation 57.
ROUT(max ) =
VOUT
IOUT(min )
=
24 V
= 240 W
0.1A
(57)
The transconductance of the TPS40210 current mode control can be estimated by Equation 58.
0.13 ´ L ´
gM =
fSW
ROUT
0.13 ´ 10 mH ´
=
2
600kHz
240 W
2
(RISNS ) ´ (120 ´ RISNS + L ´ fSW ) (12mW ) ´ (120 ´ 12mW + 10 mH ´ 600kHz )
= 19.1 A
V
(58)
The maximum output impedance ZOUT, can be estimated by Equation 59.
(1+ (2p ´ f ´ R
ESR
ZOUT (f ) = ROUT ´
(
2
)
) )´ (2p ´ f ´ C
2
´ COUT )
1 + (ROUT ) + 2 ´ ROUT ´ RESR + (RESR
2
OUT
)2
(59)
(1+ (2p ´ 20kHz ´ 60mW ´ 39.8 mF) )
1 + ((240 W ) + 2 ´ 240 W ´ 60mW + (60mW ) )´ (2p ´ 20kHz ´ 39.8 mF )
2
ZOUT (fCO ) = 240 W ´
2
2
2
= 0.146 W
(60)
The modulator gain at the desired cross-over can be estimated by Equation 61.
K CO = gM ´ ZOUT (fCO ) = 19.1A
V
´ 0.146 W = 2.80
(61)
The feedback compensation network needs to be designed to provide an inverse gain at the cross-over
frequency for unit loop gain. This sets the compensation mid-band gain at a value calculated in Equation 62.
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K COMP =
1
K CO
=
1
= 0.356
2.80
(62)
To set the mid-band gain of the error amplifier to KCOMP use Equation 63.
R4 = R7 ´ K COMP =
R7
51.1kW
=
= 18.2kW
K CO
2.80
(63)
R4 = 18.7 kΩ selected.
Place the zero at 10th the desired cross-over frequency.
C2 =
10
10
=
= 2837pF
2p ´ fL ´ R4 2p ´ 30kHz ´ 18.7kW
(64)
C2 = 2200 pF selected.
Place a high-frequency pole at about 5 times the desired cross-over frequency and less than one-half the unity
gain bandwidth of the error amplifier:
C4 »
C4 >
1
1
=
= 56.74pF
10p ´ fL ´ R4 10p ´ 30kHz ´ 18.7kW
(65)
1
1
=
= 11.35pF
p ´ GBW ´ R4 p ´ 1.5MHz ´ 18.7kW
(66)
C4 = 47 pF selected.
R-C Oscillator
The R-C oscillator calculation is given as shown in Equation 5, in the datasheet substituting 100 for CT and 600
for fSW. For a 600-kHz switching frequency, a 100-pF capacitor is selected and a 262-kΩ resistor is calculated
(261 kΩ selected)
Soft-Start Capacitor
Since VDD > 8V, the soft-start capacitor is selected by using Equation 67 to calculate the value.
CSS = 20 ´ TSS ´ 10-6
(67)
For TSS = 12 ms, CSS = 240 nF, a 220-nF capacitor selected.
Regulator Bypass
A regulator bypass capacitor of 1.0-µF is selected per the datasheet recommendation.
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TEST DATA
GAIN AND PHASE
vs
FREQUENCY
FET VDS and VGS VOLTAGES
vs
TIME
80
180
60
VIN = 8 V
VOUT = 24 V
IOUT = 2 A
135
40
90
20
45
0
0
Gain
-20
-45
-40
-90
-60
-135
-80
100
FET Vds
(20 V/ div)
-180
1M
1000
10 k
100 k
fSW – Frequency – Hz
T – Time – 400 ns
Figure 29.
Figure 30.
EFFICIENCY
vs
LOAD CURRENT
POWER LOSS
vs
LOAD CURRENT
100
6
VIN (V)
14
12
8
96
VIN (V)
14
12
8
VIN = 14 V
5
PLOSS – Power Loss – W
98
94
h – Efficiency – %
GDRV
(5 V/ div)
Phase – °
Gain – dB
Phase
92
VIN = 12 V
90
88
VIN = 8 V
86
VIN = 14 V
4
VIN = 12 V
3
2
VIN = 8 V
84
1
82
80
0
0
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
2.5
0
0.5
Figure 31.
1.0
1.5
2.0
ILOAD – Load Current – A
2.5
Figure 32.
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OUTPUT VOLTAGE
vs
LOAD CURRENT
24.820
VOUT – Output Voltage – V
24.772
24.724
VIN (V)
14
12
8
VIN = 8 V
24.676
24.628
24.580
VIN = 14 V
24.532
VIN = 12 V
24.484
24.436
24.388
24.340
0
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
2.5
Figure 33.
List of Materials
List of Materials
REFERENCE
DESIGNATOR
DESCRIPTION
SIZE
PART
NUMBER
MANUFACTURER
C1
100 µF, aluminum capacitor, SM, ± 20%, 35 V
0.406 x 0.457
EEEFC1V101P
Panasonic
C2
2200 pF, ceramic capacitor, 25 V, X7R, 20%
0603
Std
Std
C3
100 pF, ceramic capacitor, 16 V, C0G, 10%
0603
Std
Std
C4
47 pF, ceramic capacitor, 16V, X7R, 20%
0603
Std
Std
C5
0.22 µF, ceramic capacitor, 16 V, X7R, 20%
0603
Std
Std
C7
1.0 µF, ceramic capacitor, 16 V, X5R, 20%
0603
Std
Std
C8
10 µF, ceramic capacitor, 25 V, X7R, 20%
0805
C3225X7R1E106M
TDK
C9
0.1 µF, ceramic capacitor, 50 V, X7R, 20%
0603
Std
Std
C10
100 pF, ceramic capacitor, 16 V, X7R, 20%
0603
Std
Std
D1
Schottky diode, 3 A, 40 V
SMC
MBRS340T3
On Semi
L1
10 µH, inductor, SMT, 7.5 A, 12.4 mΩ
0.325 x 0.318 inch
RLF12560T-100M-7R5
TDK
Q1
MOSFET, N-channel, 40 V, 14 A, 9mΩ
SO-8
Si4840DY
Vishay
R3
10 kΩ, chip resistor, 1/16 W, 5%
0603
Std
Std
R4
18.7 kΩ, chip resistor, 1/16 W, 1%
0603
Std
Std
R5
1.5 kΩ, chip resistor, 1/16 W, 1%
0603
Std
Std
R6
261 kΩ, chip resistor, 1/16 W, 1%
0603
Std
Std
R7
51.1 kΩ, chip resistor, 1/16 W, 1%
0603
Std
Std
R9
3.3 Ω, chip resistor, 1/16 W, 5%
0603
Std
Std
R10
1.0 kΩ, chip resistor, 1/16 W, 5%
0603
Std
Std
R11
10 mΩ, chip resistor, 1/2 W, 2%
1812
Std
Std
U1
IC, 4.5 V-52 V I/P, current mode boost controller
DGQ10
TPS40210DGQ
TI
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DESIGN EXAMPLE 2
12-V Input, 700-mA LED Driver, Up to 35-V LED String
Application Schematic
L1
VIN
D1
B2100
R2
GDRV
C21
C1
C2
C3
R1
C4
R11
VIN
ISNS
R3
D2
C8
1
U1
TPS40210
VIN 10
RT
C10
C9
Loop
Response
Injection
R23
DIS/EN
C11
BP
9
DIS/EN
GDRV
8
4
COMP
ISNS
7
5
FB
GND
6
2
SS
3
GDRV
LEDC
C6
R13
R4
ISNS
C6
C13
LEDC
R24
R6
D3
R15
C14
COMP
UDG-08016
Figure 34. 12-V Input, 700-mA LED Driver, Up to 35-V LED String
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List of Materials
List of Materials
REFERENCE
DESIGNATOR
TYPE
DESCRIPTION
SIZE
C1,C2
10 µF, 25 V
1206
C3, C4
2.2 µF, 100 V
1210
C5
1 nF, NPO
0603
C6
100 pF, NPO
0603
C8
100 pF
0603
Capacitor
0.1 µF
0603
C10
0.1 µF, 25 V
0805
C11
1 µF, 25 V
1206
C13
100 pF
0603
C14
10 nF, X7R
0603
D1
SHTKY, 100 V, 2 A
SMB
C9
D2
Diode
D3
L1
Inductor
Q1
MOSFET
Q3
43 V
SOD-123
MMBD7000
SOT-23
10 µH, 6 A
12 × 12 × 10 mm
60 V, 31 mΩ
2N7002, 60 V, 0.1 A
SO-8
SOT-23
R1
15 mΩ
2512
R2
3.01 Ω
0805
R3
402 kΩ
0603
R4
14.3 kΩ
0603
0.36 Ω
2512
R11
1 kΩ
0603
R13
30.1 kΩ
0603
R15
49.9 kΩ
0603
R23, R24
10 kΩ
0603
R6
U1
34
Resistor
Integrated circuit
TPS40211
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PACKAGE OPTION ADDENDUM
www.ti.com
18-Mar-2008
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
TPS40210DGQ
ACTIVE
MSOPPower
PAD
DGQ
10
80
TBD
Call TI
Call TI
TPS40210DGQR
ACTIVE
MSOPPower
PAD
DGQ
10
2500
TBD
Call TI
Call TI
TPS40211DGQ
ACTIVE
MSOPPower
PAD
DGQ
10
80
TBD
Call TI
Call TI
TPS40211DGQR
ACTIVE
MSOPPower
PAD
DGQ
10
2500
TBD
Call TI
Call TI
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
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information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
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Following are URLs where you can obtain information on other Texas Instruments products and application solutions:
Products
Amplifiers
Data Converters
DSP
Clocks and Timers
Interface
Logic
Power Mgmt
Microcontrollers
RFID
RF/IF and ZigBee® Solutions
amplifier.ti.com
dataconverter.ti.com
dsp.ti.com
www.ti.com/clocks
interface.ti.com
logic.ti.com
power.ti.com
microcontroller.ti.com
www.ti-rfid.com
www.ti.com/lprf
Applications
Audio
Automotive
Broadband
Digital Control
Medical
Military
Optical Networking
Security
Telephony
Video & Imaging
Wireless
www.ti.com/audio
www.ti.com/automotive
www.ti.com/broadband
www.ti.com/digitalcontrol
www.ti.com/medical
www.ti.com/military
www.ti.com/opticalnetwork
www.ti.com/security
www.ti.com/telephony
www.ti.com/video
www.ti.com/wireless
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
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