SC4530 POWER MANAGEMENT 30V, 300mA Output Micropower Step-Down Switching Regulator Features Description Input Voltage Range: 3V to 30V Low Quiescent Current: Drawing 19mA from VIN when Stepping Down from 12V to 3.3V at No Load High Efficiency from 12V Input to 5V Output > 80% at 650mA > 85% at 10mA - 300mA Up to 300mA Continuous DC Output Current Integrated Power Switch and Schottky Diodes Low Output Ripple <1mA Shutdown Current Hysteretic Current-Mode Control Cycle-by-Cycle Current Limiting Alternating Between Micropower Idling and Switching States at Light Loads to Conserve Power Output Short-Circuit Protection Solution Footprint as Small as 50mm2 Low-Profile 3mm x 2mm MLPD 8-Lead Package Applications Portable Equipment Notebook Computers Distributed Supplies Backup Power Supplies The SC4530 is a micropower hysteretic current-mode step-down switching regulator capable of providing up to 300mA of output current from 3V to 30V input voltage range. It is designed to provide very high standby efficiency while simplifying design. At light loads, the SC4530 switches only as needed to maintain regulation, while idling most of the time, to improve efficiency. Typical quiescent currents from VIN and BIAS are 7mA and 26mA respectively. The control scheme produces less than 10mV of FB voltage ripple at light loads. The SC4530 automatically switches to continuous-conduction mode at heavy loads. The SC4530 has integrated power devices and on-chip control circuitry, simplifying design and enabling a solution footprint as small as 50mm2. Only an inductor and a few passive components are needed to complete a DC-DC regulator. The inductor current hysteretic control of SC4530 makes it inherently short-circuit robust. The wide input voltage range enables the device to operate from a variety of input sources, including single- or multi-cell batteries, system rails and wall transformers. Typical Application Circuit Efficiency vs Load Current 90 VIN = 12V BIAS EN BST VIN SW IN SC4530 C2 4.7µF 80 C3 0.22µF L1 OUT 33µH C4 10pF 5V/0.3A R1 619k C1 22µF FB R2 200k GND Efficiency (%) OFF ON VIN = 24V 70 60 50 VOUT = 5V L1: Coilcraft LPS4018 C1: Murata GRM31CR60J226K C2: Murata GRM32ER71H475K 40 0.1 1.0 10.0 100.0 Load Current (mA) 1000.0 Figure 1. 5V Output Step-Down Converter September 24, 2012 C4=10pF C5=33pF Verified 1/12/2012 SC4530 Pin Configuration Ordering Information FB 1 BIAS 2 BST 3 SW 4 8 EN 9 7 NC 6 IN 5 GND θJA = 80°C/W MLPD: 3mm x 2mm 8 Lead Marking Information 4530 xxxx XXXX - Lot Number Device Package SC4530WLTRT(1) (2) MLPD-W-8 3x2 SC4530EVB Evaluation Board Notes: (1) Available in tape and reel only. A reel contains 3,000 devices. (2) Available in lead-free package only. Device is WEEE and RoHS compliant. SC4530 Absolute Maximum Ratings (1) Recommended Operating Conditions IN………………………………………………… -0.3V to 32V Junction Temperature Range………………… -40°C to +125°C SW………………………………………………… -0.6V to VIN VIN …………………………………………………… 3V to 30V BST ……………………………………………………… 42V Output DC Current……………………………… up to 300mA BST Above SW…………………………………………… 30V FB ……………………………………………… -0.3V to 1.9V Thermal Information BIAS ……………………………………………… -0.3V to VIN Thermal Resistance, Junction to Ambient (3) ………… 80°C/W EN ………………………………………………… -0.3V to VIN Maximum Junction Temperature………………………+150 °C Storage Temperature Range …………………-65°C to +150°C ESD Protection Level (2) …………………………………… 2kV Peak IR Reflow Temperature (10s to 30s)……………… +260°C Exceeding the above specifications may result in permanent damage to the device or the device may malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not recommended. NOTES: (1) Unless noted otherwise, all voltage values in this section are with respect to ground. (2) Tested according to JEDEC standard JESD22-A114-B. (3) Calculated from package in still air, mounted to 3” x 4.5”, 4-layer FR4 PCB with thermal vias under the exposed pad, per JESD51 standards. Electrical Characteristics Unless otherwise noted, TA = 25°C for typical values, -40°C < TA = TJ < 125°C. VIN = VEN = 10V, VBST = 15V, VBIAS = 3V. Parameter Name Conditions VIN Operating Range Min Typ 3 VIN Quiescent Supply Current BIAS Quiescent Supply Current 30 V 0.1 0.5 µA Not Switching 7 11 µA Not Switching , VBIAS = 0 34 50 µA VEN = 0.2V 0.1 0.5 µA Not Switching 26 40 µA Not Switching , VBIAS = 0 0.1 1 µA 2 V EN Pin Input Low Voltage 0.2 V 1 2.5 µA 1.232 1.245 V VIN = 3V to 30V 0.01 0.02 %/V VFB = 1.25V 20 60 nA EN Pin Current VEN = 2.5V Feedback Voltage VFB Falling Feedback Voltage Line Regulation FB Pin Bias Current Maximum Switch Duty Cycle Units VEN = 0.2V EN Pin Input High Voltage Minimum Switch Off-time Max 1.212 TOFF(MIN) DMAX 90 530 ns 96 % SC4530 Electrical Characteristics (continued) Unless otherwise noted, TA = 25°C for typical values, -40°C < TA = TJ < 125°C. VIN = VEN = 10V, VBST = 15V, VBIAS = 3V. Parameter Switch Current Limit Name Conditions Min Typ Max Units ILIM VFB = 0 0.39 0.50 0.66 A Inductor Current Hysteresis (1) Switch Saturation Voltage Switch Leakage Current VFB = 0 65 ISW = -0.3A 200 VSW = 0 mA 300 mV 2 µA Switch Minimum Bootstrap Voltage ISW = -0.3A 1.7 2.2 V BST Pin Current ISW = -0.3A 7.1 12 mA ISW = -0.3A 700 Freewheeling Diode Forward Voltage Freewheeling Diode Reverse Leakage VD VSW = 10V Bootstrap Diode Forward Voltage IBST = 40mA Bootstrap Diode Reverse Leakage VSW = 10V, VBIAS = 0 mV 15 700 µA mV 1 µA Notes: (1) The inductor current hysteresis is the difference between the switch current limit and the freewheeling diode valley current. Pin Descriptions Pin # Pin Name Pin Function 1 FB Inverting input of the error amplifier. The FB pin is tied to a resistive divider between the output and ground. The voltage divider sets the output voltage. 2 BIAS Anode of the internal bootstrap diode. BIAS also powers the internal control circuit if VBIAS > 2.3V. Tie to the output of the DC-DC converter if VOUT > 2.5V. Tie BIAS to IN if VOUT is set below 2.5V. 3 BST Power transistor driver supply. Connect an external bootstrap capacitor from the SW pin to this pin to generate a drive voltage higher than VIN to fully saturate the internal power transistor. 4 SW The power transistor emitter and the cathode of the freewheeling diode. The SW pin is connected to an inductor and a bootstrap capacitor. 5 GND 6 IN Power supply to the SC4530. It must be closely bypassed to the ground pin. 7 NC No Connection. 8 EN The enable pin for the SC4530. Driving this pin below 0.2V completely shuts off the SC4530. Applying more than 2V to this pin enables the SC4530. If not driven from a control circuit, tie this pin to IN. 9 Exposed Pad The exposed pad at the bottom of the package serves as a thermal contact to the circuit board. It is to be soldered to the ground plane of the PC board. Connect this pin to the PC board power ground plane. SC4530 Block Diagram IN 6 R OC IPK + } RS1 PARASITICS VOS BIAS 2 EN 8 D2 530ns Min. tOFF BANDGAP REFERENCE BST 3 530ns 1.232V R Q S Q Q1 SW 4 ICNTL EA DZ D1 + - - 1 + FB VHYS CMP UC RUN / IDLE + IVLY - R RS2 GND 5 Figure 2. SC4530 Block Diagram SC4530 Typical Characteristics Efficiency vs Load Current VOUT = 3.3V 90 Efficiency vs Load Current VOUT = 2.5V 90 Feedback Voltage vs Temperature 1.24 VIN = 10V 70 70 VIN = 12V 60 VIN = 24V VIN = 5V 50 40 60 50 0.1 1.0 10.0 100.0 Load Current (mA) 0.1 1000.0 VBST = 15V 550 500 25 50 75 100 12 75 0.60 VBST = 15V Peak Valley 0.45 0.40 0.35 -25 0 25 50 75 100 125 -50 -25 o 0 25 50 75 100 Temperature ( C) Switch Saturation Voltage vs Switch Current BST Pin Current vs Switch Current Minimum Bootstrap Voltage vs Temperature 15 125 C -55oC 100 2.2 ISW = -0.39A 2.0 125oC 10 VBST - VSW (V) Bootstrap Current (mA) 200 25oC 5 0.1 0.2 0.3 0.4 Switch Current (A) 0.5 0.6 1.8 1.6 1.4 1.2 0 0 125 o -55oC VIN = 10V VBST = 15V o 25oC (1) 0.30 -50 125 300 125 VIN = 10V Temperature ( C) VBST = 15V 0.0 0.1 0.2 0.3 0.4 Switch Current (A) 0.5 0.6 -50 -25 0 25 50 75 Temperature (o C) Notes: (1) Circuit propagation delays and the error amplifier output voltage ripples may cause the actual inductor valley current to differ from its DC value. 100 Temperature ( C) VIN = 10V 0.0 50 VFB = 0 VIN = 10V o 400 25 Thresholds vs Temperature 0.50 13 10 400 0 0 Temperature(o C) VBST = 15V -25 -25 0.55 11 450 -50 -50 Current (A) 600 1.21 1.20 1000.0 14 On Time ( P s) Off Time (ns) 1.0 10.0 100.0 Load Current (mA) Maximum On Time vs Temperature 15 VIN = 10V 650 1.22 Peak and Valley Current DC Minimum Off Time vs Temperature 700 1.23 Coilcraft LPS4018-333ML 30 30 Saturation Voltage (mV) VIN = 24V VIN = 12V 40 Coilcraft LPS4018-333ML Feedback Voltage (V) 80 Efficiency (%) Efficiency (%) VIN = 5V 80 100 125 SC4530 Typical Characteristics (Continued) 125oC 100 25oC 10 10 0.2 0.4 0.6 0.8 5 0 1.0 5 10 20 25 30 o 25 C 1.0 0.1 35 0.2 0.4 Reverse Voltage (V) Voltage (V) Bootstrap Diode Reverse Leakage Current VIN = 10V 125oC Current ( P A) 25oC IIN 30 IBIAS -40oC 20 -40oC 1 10 0 0 IBIAS 25 40 2 1.0 30 125oC 3 0.8 Quiescent Currents vs VIN 50 4 0.6 Voltage (V) Quiescent Currents vs BIAS Voltage 5 Reverse Current ( P A) 15 o -40 C 10.0 0 1 20 15 IIN 10 5 VBIAS = 3V 0 5 10 15 20 25 30 35 40 0 0.0 1.0 2.0 3.0 VBST - VBIAS (V) BIAS Voltage (V) VIN Quiescent Current vs Temperature BIAS Quiescent Current vs Temperature 0 4.0 20 25 30 EN Pin Current vs VEN VBIAS = 3V 10 1.5 Current ( P A) 20 -40oC VBIAS = 3V 30 Current ( P A) 30 20 10 0 25 50 75 o Temperature ( C) 100 125 25oC 1.0 125oC 0.5 0 0 15 VIN = 10V VBIAS = 1V -25 10 2.0 VIN = 10V -50 5 VIN (V) 40 40 Current ( P A) o 125 C Current (mA) -40oC Current ( P A) Current (mA) 125oC 100.0 15 Reverse Current ( P A) 1000 Bootstrap Diode Forward Characteristics Freewheeling Diode Reverse Leakage Current Freewheeling Diode Forward Characteristics 0.0 -50 -25 0 25 50 75 o Temperature ( C) 100 125 0 5 10 15 20 25 30 VEN (V) SC4530 General Description and Operation The SC4530 is a micropower, hysteretic current-mode step-down switching regulator. As shown in the block diagram in Figure 2, the converter is controlled by an error amplifier EA and two current-sensing comparators IPK and IVLY. IPK and IVLY monitor the switch (Q1) collector current and the freewheeling diode (D1) current respectively. The EA amplifies the differential voltage between the FB and the bandgap reference, and produces a current, ICNTL, proportional to its output voltage. ICNTL, in turn, adjusts the switching thresholds of both the peak and valley current comparators. The EA output voltage is high at heavy loads, as is the peak inductor current. The Zener diode DZ clamps the amplifier output and sets the switch peak current limit. When the switch Q1 is turned on, the current through Q1 ramps up until it reaches the peak threshold set by ICNTL. The output of the IPK comparator, OC, goes high. This resets the latch and turns off the switch. With Q1 off, the inductor current ramps down through the freewheeling diode D1. When D1 current ramps below the valley threshold established by ICNTL, the output of the IVLY comparator, UC, goes high. If Q1 has been turned off for more than 530ns, then the latch will be set and Q1 will again turn on, starting a new cycle. The inductor ripple current in continuous-conduction mode is independent of ICNTL and is primarily determined by VOS and VHYS. Continuous mode switching frequency, therefore, depends on VIN, VOUT, the inductance L and the propagation delay times of the current comparators. If the regulator output is shorted to ground, then the amplifier output will rise to DZ clamp voltage. Q1 turns off as the inductor current reaches the peak current limit. With the output shorted to ground, the inductor current ramps down at a slower rate through D1. Q1 turns on again when the inductor current crosses the valley threshold. Therefore, short-circuiting the output merely lowers the converter switching frequency. The inductor current remains bounded by the peak switch current limit. The RUN/IDLE comparator, CMP, monitors the output of the error amplifier. If the EA output falls below the RUN/ IDLE threshold, then Q1 and all control circuits except the reference and EA will be shut off. The output capacitor will then supply the load, causing the output voltage to fall. When the EA output rises above the RUN/IDLE threshold, the control circuit wakes up and the part starts to switch, delivering power to the output. The offset voltage VOS at the input of the IPK comparator ensures that any current pulse delivered to the output has some minimum amplitude. At very light loads, even a single minimum charge packet delivered to the output will cause the FB voltage to rise above the reference voltage. This causes the EA output voltage to fall and the part to idle. The part resumes switching when the output current discharges the FB voltage below the reference. At light loads, the part switches only as needed to keep the output in regulation. By reducing the supply current drawn when idling, high efficiency is maintained at light loads. At heavier loads, it may take a number of consecutive minimum pulses to bring the FB above the reference voltage. The part enters continuous conduction mode when the amplifier output never falls below the RUN/IDLE threshold. Driving the base of the power transistor above the input power supply rail minimizes the power transistor turnon voltage and maximizes efficiency. A bootstrap circuit [formed by an internal bootstrap diode D2 (Figure 2) and an external capacitor connected between BST and SW] generates a voltage higher than VIN at the BST pin. The bootstrapped voltage becomes the supply voltage of the power transistor driver. The internal control circuit takes its power from either the input or from the BIAS pin if VBIAS > 2.3V. For applications with output voltage higher than 2.5V, the BIAS pin should be tied to the regulator output to maximize efficiency. SC4530 Applications Information Setting the Output Voltage mode (CCM) primarily depends on the input and output voltages: D VOUT R1 20nA VOUT VD VIN VD VCESAT (3) where VCESAT = 0.25V is the switch saturation voltage and VD = 0.6V is the forward voltage drop of the freewheeling diode. SC4530 FB R2 Figure 3. R1 and R2 Set the Output Voltage The SC4530 output voltage is programmed using a resistive divider (Figure 3) with its center tap tied to the FB pin. For a given R2, R1 can be determined: § V · R1 R 2 ¨ OUT 1¸ 1 . 232 © ¹ (1) The percentage error due to the input bias current of the error amplifier is: 'VOUT VOUT 20nA 100 (R1°«R 2 ) 1.232V (2) Whenever the power switch is turned off, it is kept off for at least 530ns. Moreover, the control circuit prevents the power transistor from turning on for more than 13.5ms. The inductor current pulls the SW node low as the power switch turns off, allowing the inductor current to charge the bootstrap capacitor. The maximum on-time ensures that the bootstrap capacitor gets replenished after a long switch-on interval. The minimum off-time, together with the maximum on-time, put an upper limit on the achievable duty cycle (≈ 0.96). From Equation (3), the minimum VIN to avoid dropout is: VIN(MIN) VOUT VD VCESAT VD 0.96 (4) Example: Determine the output voltage error caused by the amplifier input bias current in a 5V output converter. If VIN falls below this minimum, then the regulator will not be able to attain its set output voltage regardless load. Using Equation (4), the input supply voltage must be at least 5.5V in order to generate a 5V output. Assuming R2 = 200kW and using Equations (1) and (2), Inductor Selection § 5 · R1 200k: ¨ 1¸ | 619k: © 1.232 ¹ 'VOUT VOUT 20nA 100 (200k°«619k ) 0.25% 1.232V Using large R1 and R2 helps in maintaining light-load efficiency, since the current drawn by the feedback resistive divider is not delivered to the converter output. The simple calculation above shows that relatively large R1 and R2 can be used without introducing more error than that resulting from the tolerance of the standard 1% resistors. Maximum Duty Cycle Limitation The SC4530 is a non-synchronous, step-down switching regulator. Its duty cycle in continuous-conduction The SC4530 uses a hysteretic current-mode control topology. The peak-to-peak inductor ripple current, ∆IL, is theoretically constant. However, propagation delays of the current comparators (IPK and IVLY in Figure 2), as well as the error amplifier (EA) output ripples, will cause the actual inductor ripple current to vary depending on the input voltage and the duty cycle. The inductor should be chosen so that the valley current comparator, not the minimum off-time, determines the switch turn-on instant. To simplify inductance calculation, we will assume that ∆IL is constant and equal to 150mA. Furthermore, we will use 1.5 times the typical tOFF(MIN), to allow for tolerance and temperature variation. LMIN VOUT VD 1.5 t OFF(MIN) 'IL (5) SC4530 Applications Information (Continued) For a given VIN and inductance L, the continuousconduction switching frequency is: ( VOUT VD )(1 D) L 'IL f ( VOUT VD )( VIN VOUT VCESAT ) (6) L ( VIN VD VCESAT )'IL The minimum inductance is first found using Equation (5). Next the switching frequencies are estimated at VIN extremes using Equation (6). The inductance is then adjusted for achieving desired switching frequency. The resulting switch on-time at the maximum VIN must exceed the minimum controllable switch on-time, which can be as high as 180ns. This prevents the inductor current from running away when the output is shorted to ground. Example: Select the inductor for a 3.3V output regulator, with input voltage ranging from 10V to 26V. The desired switching frequency is about 600kHz. The minimum inductance is found using Equation (5): LMIN (3.3 0.6) u 0.53 u 1.5 | 22PH 0.15 Duty cycles and switching frequencies at the input voltage extremes can be found using Equations (3) and (6) respectively. The results for 22mH and 33mH are tabulated (Table 1). Table 1. Estimated Switching Frequencies for 3.3V Output Input Voltage Duty Cycle VIN (V) D (%) 10 26 37.7 14.8 Switching Frequency f (kHz) L = 22µH L = 33µH 740 490 1000 670 The 33mH inductance will be chosen, as it gives the desired switching frequency range. The resulting switch on-time is checked against the minimum controllable switch on-time. The switch ontime can be calculated using Equation (7) below: t ON L 'IL VIN VCESAT VOUT (7) With L = 33µH, the switch on-time at 10V and 26V VIN are 770ns and 220ns respectively, above the 180ns minimum controllable on-time. 10 Table 2 lists some recommended inductor values for various output voltages. Table 2. Recommended Inductor Values Inductor Value (µH) Output Voltage VOUT (V) VIN = 16 V VIN = 30 V 1.8 22 47 2.5 22 33 3.3 22 33 5.0 33 33 12 68 68 18 - 100 The saturation current of the inductor should be at least 20%~30% higher than the peak inductor current. Lowcost inductors with powder iron cores are not suitable for high-frequency switching due to their high core losses. Inductors with ferrite cores are recommended. Input Capacitor Selection A step-down regulator draws pulse current from the input power supply. A capacitor placed between the supply and the converter filters the AC current and keeps the current drawn from the supply to a DC constant. The input capacitance should be high enough to filter the pulse input current. Its equivalent series resistance (ESR) should be low so that power dissipated in the capacitor does not result in significant temperature rise and degrade reliability. Multi-layer ceramic capacitors, which have very low ESR (a few mW) and can easily handle high RMS ripple current, are the ideal choice. A single 4.7µF (X5R or X7R) ceramic capacitor should be enough for most applications. Using a larger capacitor (for example, 10µF) will reduce SW node jitters if the minimum input voltage is less than 0.7V above the output voltage. For applications with high input voltage, a small (1µF ~ 2.2µF) ceramic capacitor can be placed in parallel with a low ESR electrolytic capacitor to satisfy both the ESR and bulk capacitance requirements. SC4530 Applications Information (Continued) Output Capacitor Selection The output ripple voltage ∆VOUT of a step-down regulator in continuous conduction can be expressed as: 'VOUT § t t · 'IL ¨¨ ESR ON OFF ¸¸ 8C OUT ¹ © (8) where COUT is the output capacitance. The first term in Equation (8) results from the equivalent series resistance (ESR) of the output capacitor while the rest is due to the charging and discharging of COUT by the inductor ripple current. Substituting ∆IL = 150mA, tON + tOFF = 2µs, COUT = 22µF and ESR = 3mΩ in Equation (8), we get: 'VOUT 0.15A (3 m: 11.4 m:) 0.45 1.7 2.2 mV Depending on the switching period and the type of the capacitor used, the output voltage ripple resulting from charging/discharging of COUT may be higher than the ripple due to the ESR. The example above also shows that the output voltage ripple in continuous mode is very low. The SC4530 relies on fast amplifier response to reduce the output voltage overshoot during power-up. Neither the error amplifier output nor the reference is ramped during start-up. The Zener diode DZ (refer to page 5) clamps the amplifier output, while the regulator output voltage ramps up. As a result, the switch Q1 is turned off every cycle at the switch current limit, ILIM (typically 0.5A). The regulator thus delivers about 0.5A to its output until VOUT rises to its set value. If the load is light, then the amplifier output voltage will fall below the RUN/IDLE threshold following regulation. This causes the regulator to idle. However the energy previously stored in the inductor still flows to the output, causing the output voltage to rise above its regulation level. The minimum output capacitance required to keep the overshoot to less than 1% of the nominal output voltage is: C OUT ! 2 50 L ILIM VOUT VOUT VD (9) The minimum output capacitance for various output voltages can be estimated from Equation (9) using the inductances given in Table 2. The results are shown in Table 3. Smaller output capacitors may also be used if higher output voltage overshoot is acceptable. Table 3. Calculated Minimum Output Capacitance for 1% VOUT Overshoot during Start-up Minimum COUT (µF) VOUT (V) VIN = 16V VIN = 30V 1.8 2.5 3.3 5.0 12 18 64 36 22 15 5.6 - 96 53 33 15 5.6 3.7 Ceramic capacitors are the best choice for most applications. Sanyo TPE series polymer capacitors in Bcase, which offer large capacitors (>100µF) with slightly higher ESR, are also good alternatives. Ripple current in the output capacitor is not a concern because the inductor current of a step-down converter directly feeds COUT, resulting in very low ripple current. Avoid using Z5U or Y5V ceramic capacitors because these types of capacitors have high temperature and high voltage coefficients. Bootstrapping the Power Transistor To reduce the switch on-state voltage and maximize efficiency, the base of the power transistor should be driven from a power supply higher in voltage than VIN. The required driver supply voltage (at least 2V higher than the SW) is generated with a bootstrap capacitor C3 connected between the BST and the SW nodes (Figure 1) and the bootstrap diode D2 (Figure 2). The D2 anode is connected to the BIAS pin. During startup, the power transistor in the SC4530 is first switched on so the current flows through to the inductor. When the transistor is switched off, the inductor current pulls the SW voltage low, allowing C3 to be charged through the internal bootstrap diode D2. When the power switch is turned on again, the SW voltage goes high. This brings the BST voltage to VSW + VC3, thus back-biasing D2. 11 SC4530 Applications Information (Continued) least voltage stress at the BST pin. The maximum BST pin voltage is about VIN + VOUT. If the output voltage is between 2.5V and 3V, then a 0.33-0.47mF bootstrap capacitor may be needed to reduce droop. In most other cases, a 0.22mF ceramic capacitor is adequate. The C3 voltage increases with each subsequent switching cycle, as does the bootstrapped voltage at the BST pin. After a number of switching cycles, C3 will be fully charged to a voltage approximately equal to that applied to the anode of D2. The minimum BST to SW voltage required to fully saturate the power transistor is shown in the Typical Characteristics (pages 6-7). This difference voltage must be at least 1.72V at room temperature. This is also specified in the Electrical Characteristics (pages 3-4) as the Minimum Bootstrap Voltage. The minimum required VC3 increases as temperature decreases. The bootstrap circuit reaches equilibrium when the base charge drawn from C3 during transistor on-time is equal to the charge replenished during the off interval. Figure 4(b) shows the SC4530 can also be bootstrapped from the input. This way it is not as efficient as the configuration shown in Figure 4(a). However this may be only option if the output voltage is less than 2.5V and there is no other supply with voltage higher than 2.5V. Voltage stress at the BST pin can be somewhat higher than 2VIN. The BST pin voltage should not exceed its absolute maximum rating of 42V. Figure 4(c) shows how to bootstrap the SC4530 from an independent power supply VS with its voltage > 2.5V. Figure 4 summarizes various ways of bootstrapping the SC4530. In Figure 4(a) the BIAS pin is connected to the converter output. The bootstrap charge is obtained from the output of the step-down converter. The inputreferred charge is reduced by the step-down ratio. This is the most efficient configuration and it also results in the To demonstrate the effect of an under-sized bootstrap capacitor, C3 (Figure 1, page 1) is deliberately reduced to 10nF. The BIAS pin is tied to an external power supply similar to Figure 4(c). By adjusting the external supply voltage Max VBST ˜ 2VIN Max VBST ˜ VIN + VOUT C3 BST VIN BST VOUT > 2.5V BIAS SW IN VIN BIAS IN SC4530 C3 SW VOUT < 2.5V SC4530 GND GND (a) (b) Max VBST ˜ VIN + VS VS > 2.5V 0.1µF VIN BIAS BST SW IN C3 VOUT SC4530 GND VIN (c) Figure 4. Methods of Bootstrapping the SC4530 12 SC4530 Applications Information (Continued) VS, the bootstrap voltage can be varied. Figure 5(a) shows the switching waveforms of a correctly bootstrapped 10V to 5V regulator with VS = 2.5V. All three traces share the same ground level. When the power transistor is turned on, VSW should come within a few hundred millivolts of VIN VBST VIN Since the inductor current charges C3, the bootstrap circuit requires some minimum load current to function. Figures 6(a) and 6(b) show the minimum input voltage required to saturate the power transistor and to produce a regulated output as a function of the load current. Once started, the bootstrap circuit is able to sustain itself down to zero load. VSW All Traces 2V/div Minimum Input Voltage 7.0 VOUT = 5V To Start 400ns/div (a) VBST VIN All Traces 2V/div Input Voltage (V) 6.5 6.0 Dropout 5.5 5.0 VSW 1 10 100 1000 Load Current (mA) (a) Minimum Input Voltage 5.5 400ns/div (a) Sufficient Bootstrap Voltage Drives the Power Transistor into Saturation, Minimizing Power Loss. (b) Excessive Droop in Bootstrap Capacitor Voltage fails to keep the Power Transistor Saturated near the End of its Conduction Cycles, Causing Jitters and Low Efficiency. and VBST should have at least 2V of headroom above VSW. As VS is reduced to 1.9V, excessive VBST droop decreases transistor driver headroom, as shown in Figure 5(b). The power transistor can no longer be fully saturated (as evidenced by the round VSW turn-off corners), resulting in high power dissipation. When bootstrapping from a lowvoltage output or supply, checking the bootstrap voltage is a good precaution. 5.0 Input Voltage (V) Figure 5. Switching Waveforms of a 10V to 5V Regulator VOUT = 3.3V To Start (b) 4.5 4.0 Dropout 3.5 3.0 1 10 100 1000 Load Current (mA) (b) Figure 6. The Minimum Input Voltage Required to Start and to Operate Before Dropout (a) VOUT =5V (b) VOUT = 3.3V 13 SC4530 Applications Information (Continued) Feed-Forward Compensation Mode Transition and the FB Pin A feed-forward capacitor C4 (connected across the upper feedback resistor R1) is needed for stability. An initial estimate of C4 can be found using Equation (10) below: If the upper feedback resistor R1 (Figure 3, page 9) is large and is about the same magnitude as R2, then fast switching transients may couple into the FB pin, disturbing or delaying the transition from light-load operating mode to continuous-conduction mode (CCM). As described previously, the output ripple voltage is very low in continuous-conduction mode. Delayed CCM transition extends the load range in which the converter produces larger output voltage ripples. This disturbance becomes more pronounced when VIN is increased above 21V and C4 6.8 u 10 6 R1 (10) The value of C4 can be optimized empirically by observing the inductor current and the output voltage during load transient. Starting with the initial estimate, C4 is tuned until there is no excessive ringing or overshoot in the inductor current or the output voltage during load transient. VOUT 50mV/div AC Coupled VOUT 50mV/div AC Coupled IL1 200mA/div IL1 200mA/div VSW 20V/div VSW 20V/div 4ms/div 4ms/div (a) C4 = 10pF, C5 not Placed SW_OUT_IL_28V to 5V@174mA_C4=10pF_onset VOUT = 5V, IOUT = 174mA CCM=176mA CCM Onset IOUT = 176mA VOUT 50mV/div AC Coupled VOUT 50mV/div AC Coupled IL1 200mA/div IL1 200mA/div VSW 20V/div VSW 20V/div 10ms/div (b) C4 = 10pF, C5 = 33pF V = 5V, I = 110mA CCM Onset IOUT = 126mA SW_OUT_IL_28V CCM=126mA OUT to 5V@110mA_C4=10pF_C5=33pF_onset OUT Figure 7. Switching Waveforms of a 28V to 5V Converter Just Before It Enters ContinuousConduction Mode 14 (a) C4 = 22pF, C5 not Placed SW_OUT_IL_28V to 3.3V@145mA_C4=22pF_onset VOUT = 3.3V, IOUT = 145mA CCM=150mA CCM Onset IOUT = 150mA 10ms/div (b) C4 = 22pF, C5 = 47pF VOUT = 3.3V, IOUT = 118mA CCM Onset IOUT = 121mA SW_OUT_IL_28V to 3.3V@118mA_C4=22pF_C5=47pF_onset CCM=121mA Figure 8. Switching Waveforms of a 28V to 3.3V Converter Just Before It Enters ContinuousConduction Mode SC4530 Applications Information (Continued) when large feedback resistors are used. The regulator becomes insensitive to switching disturbances after it enters continuous-conduction mode. The operating mode transition can be significantly smoothed by filtering the FB node. A capacitor between FB pin and ground (Capacitor C5, as shown in Figures 13(a), page 18) serves this purpose. It should be chosen so that it improves mode transition without significantly slowing down load transients. Switching waveforms of a 5V output regulator (Figure 13(a), page 18) immediately before it enters continuous-conduction mode are shown in Figure 7. The inductor current waveform appears to be more jagged without filtering. Moreover, transition to CCM occurs at an output current of 176mA, instead of 126mA with FB filtering. Figure 8 compares the corresponding switching waveforms of an output 3.3V (Figure 14(a), page 19) regulator. If the converter output voltage is 1.8V or less, or if R2 is reduced to below 2kW, then C5 will not be necessary. C5 is also optional in Figures 13(a) and 14(a) if the maximum VIN never exceeds 21V. Bench testing shows that removing C5 from these converters still results in acceptable transitional behavior, provided that VIN < 21V. C5 can be estimated using the following empirical equation: C5 8 u 10 6 C4 R1°«R 2 (11) FB filtering has no significant impact on the output ripple voltage. However, it improves the converter efficiency by 0.25% to 0.5% around the mode transition point (Figure 9). Regulator efficiencies are slightly lower (< 0.25%) at light loads when filtering the FB voltage. Positive values in Figure 9 imply that FB filtering improves efficiency compared to no filtering. Effect of FB Filtering on Efficiencies vs Load Current 1.0 Efficiency Difference (%) VOUT = 5V 0.5 0.0 -0.5 VOUT = 3.3V VIN = 28V -1.0 1 10 100 Load Current (mA) 1000 Figure 9. Effect of FB Filtering on Converter Efficiency (VIN = 28V) Plotted Efficiency = the Efficiency of a FB-Filtered Converter the Efficiency of the Same Converter without FB-Filtering 15 SC4530 Applications Information (Continued) Reverse Input Protection Consider a circuit board where the input power source supplies several DC-DC converters, including an SC4530 regulator with a large output capacitor. During poweroff, the SC4530 regulator output may be held high by its output capacitor, while VIN is discharged rapidly by other DC-DC converters. If VIN falls to two diode voltages below VOUT, then the parasitic junction diodes inside the SC4530 (see Figure 2, page 5) will draw current from the output through the SW pin to the input. If the load is light and the output capacitor is large, then high reverse current will flow, or even damage the internal circuits. BIAS OFF ON VIN EN BST IN SW D4 OUT SC4530 FB Figure 10 shows two protection schemes. In Figure 10(a), a Schottky diode D4 placed at the input blocks the reverse current. This method has the disadvantage that it lowers the converter efficiency. A PN junction diode placed from the converter output to the input [(as shown in Figure 10(b)] shunts the reverse current away from the part, thus protecting the part. This scheme is not suitable in a power supply system where a backup battery is diode OR-ed with the SC4530 regulator output and with the SC4530 input grounded. Board Layout Considerations In a step-down switching regulator, the input bypass capacitor, the main power switch and the freewheeling power diode carry pulse current with high di/dt (Figure 11). To minimize jittering, the size of the loop formed by these components must be minimized. Since the main power switch and the freewheeling diode are already integrated inside the part, connecting the input bypass capacitor close to the ground pin minimizes size of the switched current loop. GND (a) D4 VOUT 1N4148 OFF ON EN BST OUT SW VIN IN ZL SC4530 BIAS FB GND (b) Figure 10. Reversed Input Protection Schemes (a) D4 Blocks the Reverse Current (b) D4 Shunts the Reverse Current from the Part During Power-off. 16 Figure 11. Heavy Lines Show the Fast Switching Current Paths in a Step-down Converter. The Input Capacitor Should be Placed Close to the Part for Improved Switching Performance. SC4530 Applications Information (Continued) Shortening the traces at the SW and BST nodes reduces the parasitic trace inductance at these nodes. This not only reduces EMI, but also decreases switching voltage spikes at these nodes. Shielding the FB trace from the SW and the BST nodes with ground traces is a good precaution in mitigating switching transient disturbance. Figure 12 shows an example of external component placement around the SC4530. The exposed pad should be soldered to a large power ground plane as the ground copper acts as a heat sink for the device. C1 VOUT R1 R2 C4 C5 GND R5 R4 U1 L1 C3 C2 VIN SW Figure 12. Suggested PCB Layout for the SC4530 17 SC4530 Typical Application Circuits BIAS OFF ON EN BST IN SW VIN 6V - 30V C3 0.22mF SC4530 C2 4.7mF L1 OUT 33mH C4 10pF 5V/0.3A R1 619k C1 22mF FB GND C5 33pF L1: Coilcraft LPS4018 C1: Murata GRM31CR60J226K C2: Murata GRM32ER71H475K R2 200k (a) VOUT 20mV/div AC Coupled C4=10pF C5=33pF Verified 1/12/2012 C4=22pF C5=68pF Verified 1/24/2011 IL1 200mA/div VSW 10V/div VOUT 50mV/div AC Coupled IL1 200mA/div VSW 10V/div 1ms/div 4ms/div (c) (b) SW_OUT_IL_12V to 5V@60mA_C4=10pF_C5=33pF SW_OUT_IL_12V to 5V@10uA_C4=10pF_C5=33pF VOUT 200mV/div AC Coupled VOUT 10mV/div AC Coupled IL1 200mA/div IL1 200mA/div VSW 10V/div 1ms/div (d) SW_OUT_IL_12V to 5V@300mA_C4=10pF_C5=33pF 40ms/div (e) OUT_IL_12V to 5V_ld_tran_0-300mA_C4=10pF_C5=33pF Figure 13. (a) 5V Output Step-Down Converter (b) Switching Waveforms of the Figure 13(a) Circuit. VIN = 12V, IOUT = 10mA (c) VIN = 12V, IOUT = 60mA (d) VIN = 12V, IOUT = 300mA (e) Load Transient. VIN = 12V, IOUT is Switched Between 0 and 300mA 18 SC4530 Typical Application Circuits (Continued) BIAS OFF ON EN BST IN SW VIN 4V - 30V C3 0.22µF SC4530 C2 4.7µF L1 OUT 33µH C4 22pF 3.3V/0.3A R1 332k C1 22µF FB GND C5 47pF L1: Coilcraft LPS4018 C1: Murata GRM31CR60J226K C2: Murata GRM32ER71H475K R2 200k (a) VOUT 50mV/div AC Coupled C4=22pF C5=47pF to reduce start-up overshoot 3/14/2012 VOUT 20mV/div AC Coupled C4=10pF C5=47pF Verified 1/13/2012 IL1 200mA/div C4=22pF C5=47pF Verified 1/26/2011 IL1 200mA/div VSW 20V/div VSW 20V/div 400ms/div 4ms/div (b) (c) SW_OUT_IL_24V to 3.3V@100uA_C4=22pF_C5=47pF VOUT 10mV/div AC Coupled SW_OUT_IL_24V to 3.3V@60mA_C4=22pF_C5=47pF VOUT 200mV/div AC Coupled IL1 200mA/div IL1 200mA/div VSW 20V/div 1ms/div 40ms/div (d) (e) SW_OUT_IL_24V to 3.3V@300mA_C4=22pF_C5=47pF OUT_IL_24V to 3.3V_ld_tran_0-300mA_C4=22pF_C5=47pF Figure 14. (a) 3.3V Output Step-Down Converter (b) Switching Waveforms of the Figure 14(a) Circuit. VIN = 24V, IOUT = 100mA (c) VIN = 24V, IOUT = 60mA (d) VIN = 24V, IOUT = 300mA (e) Load Transient. VIN = 24V, IOUT is Switched Between 0 and 300mA 19 SC4530 Typical Application Circuits (Continued) BIAS OFF ON EN BST VIN C3 0.22µF SW IN 3.3V - 20V BIAS OFF ON 1.23V/0.3A 3.1V - 20V BST IN SW C2 4.7µF L1 OUT 33µH C4 33pF L1: Coilcraft LPS4018 Figure 15. 1.23V Output Step-Down Converter R1 93.1k C1 100µF R2 200k GND C1: Sanyo POSCAP 4TPE150MAUB C2: Murata GRM32ER71H475K 1.8V/0.3A FB GND L1: Coilcraft LPS4018 C3 0.22µF SC4530 C1 150µF FB EN VIN OUT 33µH SC4530 C2 4.7µF L1 C1: Murata GRM31CR60G107M C2: Murata GRM32ER71H475K Figure 16. 1.8V Output Step-Down Converter Verified 2/17/2011 BIAS BIAS OFF ON EN BST VIN SW IN 3.2V - 30V SC4530 C2 10µF OFF ON C3 0.47µF L1 OUT 33µH 2.5V/0.3A C4 33pF C3 0.22mF VIN C2 4.7mF C5 47pF L1: Coilcraft LPS4018 C1: Murata GRM21BR60G476M C2: Murata GRM32ER61H106K C4=33pF C5=47pF to reduce start-up overshoot 3/14/2012 OFF ON EN C4=33pF C5=100pF Verified 1/27/2011 SW SC4530 C2 4.7µF C1 10mF C5 68pF L1: Coilcraft LPS4018 C1: Murata GRM31CR61C106K C2: Murata GRM32ER71H475K C3 0.22µF L1 OUT 68µH C4 10pF 12V/0.3A R1 866k C1 4.7µF C5 100pF GND R2 100k C1: Murata GRM31CR61C475K C2: Murata GRM32ER71H475K Figure 19. 12V Output Step-Down Converter Change C4 from 22pF to 10pF and C5 from 470pF to 100pF, 3/15/2012 L1 changed from 47uH to 68uH 1/16/2012 * LPS4018 68uH saturation current is marginally low 20 R1 931k FB FB L1: Coilcraft LPS6225 9V/0.3A R2 147k Figure 18. 9V Output Step-Down Converter BST IN 14V - 30V C4 6.8pF BIAS VIN OUT GND R2 200k Figure 17. 2.5V Output Step-Down Converter L1 47mH SC4530 C1 47µF GND SW IN 10V - 30V R1 205k FB ENchanged from BST L1 22uH to 33uH 1/17/2012 SC4530 Outline Drawing – MLPD-W-8 3x2 21 SC4530 Land Pattern – MLPD-W-8 3x2 22 SC4530 © Semtech 2012 All rights reserved. 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