RFMD RF2516

RF2516
Preliminary
11
VHF/UHF TRANSMITTER
Typical Applications
• 315/433MHz Band Systems
• Remote Keyless Entry
• Local Oscillator Source
• Wireless Security Systems
• Part 15.231 Applications
• AM/ASK/OOK Transmitter
Product Description
The RF2516 is a monolithic integrated circuit intended for
use as a low-cost AM/ASK transmitter. The device is provided in a 16-pin QSOP-16 package and is designed to
provide a phased locked frequency source for use in local
oscillator or transmitter applications. The chip can be
used in applications in the North American and European
VHF/UHF bands. The integrated VCO, phase detector,
prescaler, and reference oscillator transistor require only
the addition of an external crystal to provide a complete
phase-locked loop. In addition to the standard powerdown mode, the chip also includes an automatic lockdetect feature that disables the transmitter output when
the PLL is out-of-lock.
0.196
0.189
0.025
8° MAX
0°MIN
SiGe HBT
Si CMOS
0.050
0.016
0.0098
0.0075
NOTES:
1. Shaded lead is Pin 1.
2. All dimensions are excluding mold flash.
3. Lead coplanarity - 0.005 with respect to datum "A".
Package Style: QSOP-16
11
Features
RESNTR-
LOOP FLT
OSC E
OSC B
• Fully Integrated PLL Circuit
• Integrated VCO and Reference Oscillator
13
12
14
2
1
• 2.0V to 3.6V Supply Voltage
• Low Current and Power Down Capability
Lock
Detect
Prescaler
32/64
8
15
16
LD FLT
DIV CTRL
5
MOD IN
TX OUT
Phase
Detector &
Charge Pump
• 100MHz to 500MHz Frequency Range
DC
Bias
Functional Block Diagram
Rev A10 010613
3
PD
• Out-of-Lock Inhibit Circuit
Ordering Information
RF2516
RF2516 PCBA
VHF/UHF Transmitter
Fully Assembled Evaluation Board
RF Micro Devices, Inc.
7625 Thorndike Road
Greensboro, NC 27409, USA
Tel (336) 664 1233
Fax (336) 664 0454
http://www.rfmd.com
11-35
TRANSCEIVERS
GaAs MESFET
0.0688
0.0532
0.2440
0.2284
RESNTR+
üSi Bi-CMOS
GaAs HBT
0.0098
0.0040
0.012
0.008
Optimum Technology Matching® Applied
Si BJT
-A-
0.157
0.150
RF2516
Preliminary
Absolute Maximum Ratings
Parameter
Supply Voltage
Power Down Voltage (VPD)
MOD IN
Operating Ambient Temperature
Storage Temperature
Parameter
Rating
Unit
-0.5 to +3.6
-0.5 to VCC
-0.5 to 1.1
-40 to +85
-40 to +150
VDC
V
V
°C
°C
Specification
Min.
Typ.
Max.
Caution! ESD sensitive device.
RF Micro Devices believes the furnished information is correct and accurate
at the time of this printing. However, RF Micro Devices reserves the right to
make changes to its products without notice. RF Micro Devices does not
assume responsibility for the use of the described product(s).
Unit
T=25°C, VCC =2.8V, Freq=433MHz,
RMODIN =3kΩ
Overall
Frequency Range
Modulation
Modulation Frequency
Incidental FM
Output Power
ON/OFF Ratio
Condition
100 to 500
AM/ASK
MHz
1
15
+8.5
+10
75
MHz
kHz p-p
dBm
dB
50Ω load
PLL and Prescaler
Prescaler Divide Ratio
VCO Gain, KVCO
PLL Phase Noise
Harmonics
Reference Frequency
Crystal Frequency Spurs
Max Crystal RS
Max Crystal Motional Inductance
Charge Pump Current
32/64
20
-97
-102
-60
17
TBD
-50
35
60
100
50
MHz/V
dBc/Hz
dBc/Hz
dBc
MHz
dBc
Ω
mH
µA
Frequency and board layout dependent.
10kHz Offset, 50kHz loop bandwidth
100kHz Offset, 50kHz loop bandwidth
With output tuning.
50kHz PLL loop bandwidth
For a typ. 1ms turn-on time.
For a typ. 1ms turn-on time.
KPD =100µA/2π=0.0159mA/rad
Power Down Control
TRANSCEIVERS
11
Power Down “ON”
Power Down “OFF”
Control Input Impedance
Turn On Time
Turn Off Time
VCC -0.3V
V
V
Ω
ms
ms
Voltage supplied to the input; device is “ON”
Voltage supplied to the input; device is “OFF”
3.6
V
V
mA
Specifications
Operating limits
50% Duty Cycle 10kHz Data applied to the
MOD IN input. RMODIN (R10)=3kΩ. Output
power/DC current consumption externally
adjustable by modulation input resistor (see
applicable Application Schematic).
1
uA
PD=0V, MOD IN=0V, DIV CTRL=0V
+0.3
100k
1
1
2
2
Crystal start-up, 13.57734MHz crystal.
Power Supply
Voltage
Current Consumption (Avg.)
Power Down Current
11-36
2.8
2.0
6
0
10.5
Rev A10 010613
RF2516
Preliminary
Pin
1
Function
OSC B
Description
This pin is connected directly to the reference oscillator transistor base.
The intended reference oscillator configuration is a modified Colpitts. A
68pF capacitor should be connected between pin 1 and pin 2. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.
2
OSC E
This pin is connected directly to the emitter of the reference oscillator
transistor. A 33pF capacitor should be connected from this pin to
ground. Diodes shown in the interface schematic provide 3kV electrostatic discharge (ESD) protection using the human body model.
3
PD
Power Down control for all circuitry. When this pin is a logic “low” all circuits are turned off. When this pin is a logic “high”, all circuits are operating normally. A “high” is VCC. Diodes shown in the interface
schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.
4
5
GND
TXOUT
Interface Schematic
VCC
OSC B
OSC E
See pin 1.
VCC
PD
Ground connection for the TX OUT amp. Keep traces physically short
and connect immediately to ground plane for best performance. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.
Transmitter output. This output is an open collector and requires a pullup inductor for bias/matching and a tapped capacitor for matching.
VCC
GND
TX OUT
RF IN
MOD IN
GND1
7
VCC1
8
MOD IN
9
VCC2
10
GND2
Rev A10 010613
Ground connection for the TX output buffer amplifier. Diodes shown in
the interface schematic provide 3kV electrostatic discharge (ESD) protection using the human body model.
This pin is used to supply bias to the TX buffer amplifier. Diodes shown
in the interface schematic provide 3kV electrostatic discharge (ESD)
protection using the human body model.
AM analog or digital modulation can be imparted to the carrier by an
input to this pin. An external resistor is used to bias the output amplifiers through this pin. The voltage at this pin must not exceed 1.1V.
Higher voltages may damage the device. Diodes shown in the interface
schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.
This pin is used to supply DC bias to the VCO, crystal oscillator, prescaler, phase detector, and charge pump. An IF bypass capacitor
should be connected directly to this pin and returned to ground. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.
Digital PLL ground connection. Diodes shown in the interface schematic provide 3kV electrostatic discharge (ESD) protection using the
human body model.
See pin 4.
11
VCC
TRANSCEIVERS
6
VCC1
VCC
TX OUT
1 kΩ
MOD IN
See pin 7.
See pin 4.
11-37
RF2516
Pin
11
12
Function
VREF P
RESNTR-
Preliminary
Description
Interface Schematic
Bias voltage reference pin for bypassing. The bypass capacitor should
be of appropriate size to provide filtering of the reference crystal frequency and be connected directly to this pin. Diodes shown in the interface schematic provide 3kV electrostatic discharge (ESD) protection
using the human body model.
The RESNTR pins are used to supply DC voltage to the VCO, as well
as to tune the center frequency of the VCO. Equal value inductors
should be connected to this pin and pin 13.
VCC
VREF P
RESNTR+
RESNTR-
LOOP FLT
4 kΩ
13
14
15
16
TRANSCEIVERS
11
11-38
RESNTR+
LOOP FLT
LD FLT
DIV CTRL
See pin 12.
Output of the charge pump. An RC network from this pin to ground is
used to establish the PLL bandwidth. Diodes shown in the interface
schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.
This pin is used to set the threshold of the lock-detect circuit. A shunt
capacitor should be used to set an RC time constant with the on-chip
series 1k resistor. This signal is used to clamp (enable or disable) the
MOD IN circuitry. The time constant should be approximately 10 times
the reference period. Diodes shown in the interface schematic provide
3kV electrostatic discharge (ESD) protection using the human body
model.
Logic “High” input selects divide-by-64 prescaler. Logic “Low” input
selects divide-by-32 prescaler. Diodes shown in the interface schematic provide 3kV electrostatic discharge (ESD) protection using the
human body model.
VCC
LOOP FLT
VCC
1 kΩ
LD FLT
VCC
DIV CTRL
Rev A10 010613
RF2516
Preliminary
RF2516 Theory of Operation
The RF2516 Transmitter
The RF2516 is a low-cost AM/ASK VHF/UHF transmitter designed for applications operating within the frequency range of 100MHz to 500MHz. In particular, it is
intended for 315MHz to 433MHz band systems,
remote keyless entry systems, and FCC Part 15.231
periodic transmitters. It can also be used as a local
oscillator signal source. The integrated VCO, phase
detector, prescaler, and reference oscillator require
only the addition of an external crystal to provide a
complete phase-locked loop. In addition to the standard power-down mode, the chip also includes an
automatic lock-detect feature that disables the transmitter output when the PLL is out-of-lock.
The device is manufactured on a 25GHz Silicon Bipolar-CMOS process and packaged in an industry standard SSOP-16 plastic package. This, combined with
the low external parts count, enables the designer to
achieve small-footprint, high-performance, low-cost
designs.
The RF2516 is designed to operate from a supply voltage ranging from 2.0V to 3.6V, accommodating
designs using three NiCd battery cells, two AAA flashlight cells, or a lithium button battery. The device is
Rev A10 010613
capable of providing up to +10dBm output power into a
50Ω load, and is intended to comply with FCC requirements for unlicensed remote control transmitters. ESD
protection is provided on all pins except VCO and TX
OUT.
While this device is intended for OOK operation, it is
possible to use narrowband FM. This is accomplished
by modulating the reference oscillator rather than
applying the data to the MOD IN input pin. The MOD
IN pin should be tied high to cause the device to transmit. The deviation will be set by pulling limits of the
crystal. Deviation sufficient for the transmission of
voice and other low data rate signals can therefore be
accomplished. Refer to the Application Schematic in
the data sheet for details.
The RF2516 Functional Blocks
A PLL consists of a reference oscillator, a phase detector, a loop filter, a voltage controlled oscillator (VCO),
and a programmable divider in the feedback path. The
RF2516 includes all of these internally, except for the
loop filter and the reference oscillator’s crystal and two
feedback capacitors.
The reference oscillator is a Colpitts type oscillator.
Pins 1 (OSC B) and 2 (OSC E) provide connections to
a transistor that is used as the reference oscillator. The
Colpitts configuration is a low parts count topology with
reliable performance and reasonable phase noise.
Alternatively, an external signal could be injected into
the base of the transistor. The drive level should, in
either case, be around 500mVPP. This level prevents
overdriving the device and keeps the phase noise and
reference spurs to a minimum.
The prescaler divides the VCO frequency by either 64
or 32, using a series of flip-flops, depending upon the
logic level present at the DIV CTRL pin. A high logic
level will select the 64 divisor. A low logic level will
select the 32 divisor. This divided signal is then fed into
the phase detector where it is compared with the reference frequency.
The RF2516 contains an onboard phase detector and
charge pump. The phase detector compares the
phase of the reference oscillator to the phase of the
VCO. The phase detector is implemented using flipflops in a topology referred to as either “digital phase/
frequency detector” or “digital tri-state comparator”.
The circuit consists of two D flip-flops whose outputs
are combined with a NAND gate which is then tied to
11-39
11
TRANSCEIVERS
Introduction
Short range radio devices are becoming commonplace
in today’s environment. The most common examples
are the remote keyless entry systems popular on many
new cars and trucks, and the ubiquitous garage door
opener. Other applications are emerging with the
growth in home security, automation and the advent of
various remote control applications. Typically these
devices have been simplex, or one-way, links. They are
also typically built using surface acoustic wave (SAW)
devices as the frequency control elements. This
approach has been attractive because the SAW
devices have been readily available and a transmitter,
for example, could be built with only a few additional
components. Recently however, RF Micro Devices,
Inc. (RFMD), has introduced several new components
that enable a new class of short-range radio devices
based on the use of crystals and phase-locked loops
for frequency control. These devices are superior in
performance and comparable in cost to the traditional
SAW-based designs. The RF2516 is an example of
such a device. The RF2516 is targeted for applications
such as 315MHz and 433MHz band remote keyless
entry systems and wireless security systems, as well
as other remote control applications.
RF2516
the reset on each flip-flop. The outputs of the flip-flops
are also connected to the charge pump. Each flip-flop
output signal is a series of pulses whose frequency is
related to the flip-flop input frequency.
When both inputs of the flip-flops are identical, the signals are both frequency- and phase-locked. If they are
different, they will provide signals to the charge pump
which will either charge or discharge the loop filter, or
enter into a high impedance state. The name “tri-state
comparator” comes from this.
The main benefit of this type of detector is the ability to
correct for errors in both phase and frequency. When
locked, the detector uses phase error for correction.
When unlocked, it uses frequency error for correction.
This type of detector will lock under all conditions.
The charge pump consists of two transistors, one for
charging the loop filter and the other for discharging
the loop filter. Its inputs are the outputs of the phase
detector flip-flops. Since there are two flip-flops, there
are four possible states. If both amplifier inputs are low,
then the amplifier pair goes into a high impedance
state, maintaining the charge on the loop filter. The
state where both inputs are high will not occur. The
other states are either charging or discharging the loop
filter. The loop filter integrates the pulses coming from
the charge pump to create a control voltage for the
voltage controlled oscillator.
TRANSCEIVERS
11
The VCO is a tuned differential amplifier with the bases
and collectors cross-coupled to provide positive feedback and a 360° phase shift. The tuned circuit is
located in the collectors, and is comprised of internal
varactors and external inductors. The designer selects
the inductors for the desired frequency of operation.
These inductors also provide DC bias for the VCO.
The output of the VCO is buffered and applied to the
prescaler circuit, where it is divided by either 32 or 64,
as selected by the designer, and compared to the reference oscillator frequency.
The transmit amplifier is a two-stage amplifier consisting of a driver and an open collector final stage. It is
capable of providing 10dBm of output power into a
50Ω load while operating from a 3.6V power supply.
11-40
Preliminary
The lock-detect circuitry connects to the output of the
phase detector circuitry and is used to disable the
transmitter when the VCO is not phase-locked to the
reference oscillator. This is necessary to avoid
unwanted out-of-band transmission and to provide
compliance with regulatory limits during an unlocked
condition.
There are many possible reasons for the PLL not to be
locked. For instance, there is a short period during the
start of any VCO in which the VCO begins oscillating
and the reference oscillator builds up to full amplitude.
During this period, the frequency will likely be outside
the authorized band. Typically, the VCO starts much
faster than the reference oscillator. Once both VCO
and reference oscillators are running, the phase detector can start slewing the VCO to the correct frequency,
slowly sliding across 200MHz of occupied spectrum. In
competitive devices, the VCO radiates at full power
under all of these conditions.
The lock protection circuit in the RF2516 is intended to
stabilize quickly after power is applied to the chip, and
to disable the base drive to the transmit amplifier. This
attenuates the output to levels that will be generally
acceptable to regulatory boards as spurious emissions. Once the phase detector has locked the oscillators, then the lock circuit enables the MOD IN pin for
transmission of the desired data. There is no need for
an external microprocessor to monitor the lock status,
although that can be done with a low current A/D converter in a system micro, if needed. The lock-detect circuitry contains an internal resistor which, combined
with a designer-chosen capacitor for a particular RC
time constant, filters the lock-detect signal. This signal
is then passed through an internal Schmitt trigger and
used to enable or disable the transmit amplifier.
If the oscillator unlocks, even momentarily, the protection circuit quickly disables the output until the lock is
stable. These unlocks can be caused by low battery
voltage, poor power supply regulation, severe shock of
the crystal or VCO, antenna loading, component failure, or a myriad of unexpected single-point failures.
The RF2516 contains onboard band gap reference
voltage circuitry which provides a stable DC bias over
varying temperature and supply voltages. Additionally,
the device features a power-down mode, eliminating
battery disconnect switches.
Rev A10 010613
RF2516
Preliminary
Designing With the RF2516
The reference oscillator is built around the onboard
transistor at pins 1 and 2. The intended topology is that
of a Colpitts oscillator. The Colpitts oscillator is quite
common and requires few external components, making it ideal for low-cost solutions. The topology of this
type of oscillator is as seen in the following figure.
VCC
Additionally, by placing a variable capacitor in series
with the crystal, one is able to adjust the frequency.
This will also alter the drive level, so it should be
checked again.
An important part of the overall design is the voltage
controlled oscillator. The VCO is configured as a differential amplifier. The VCO is tuned via internal varactors. The varactors are tuned by the loop filter output
voltage through a 4kΩ resistor.
RESNTR+
X1
RESNTR-
C2
L
C1
L
LOOP FLT
4 kΩ
This type of oscillator is a parallel resonant circuit for a
fundamental mode crystal. The transistor amplifier is
an emitter follower and the voltage gain is developed
by the tapped capacitor impedance transformer. The
series combination of C1 and C2 act in parallel with the
input capacitance of the transistor to capacitively load
the crystal.
The nominal capacitor values can be calculated with
the following equations6:
The load capacitance is usually 32pF. The variable freq
is the oscillator frequency in MHz. The frequency can
be adjusted by either changing C2 or by placing a variable capacitor in series with the crystal. As an example, assume a desired frequency of 14MHz and a load
capacitance of 32pF. C1 =137.1pF and C2 =41.7pF.
These capacitor values provide a starting point. The
drive level of the oscillator should be checked by looking at the signal at pin 2 (OSC E). It has been found
that the level at this pin should generally be around
500mVPP or less. This will reduce the reference spur
levels and reduce noise from distortion. If this level is
higher than 500mVPP then decrease the value of C1.
The values of these capacitors are usually tweaked
during design to meet performance goals, such as
minimizing the start-up time.
Rev A10 010613
As mentioned earlier, the inductors and the varactors
are tuning a differential amplifier. To tune the VCO the
designer only needs to calculate the value of the inductors connected to pins 12 and 13 (RESNTR- and
RESNTR+). The inductor value is determined by the
equation:
2 1 1
1
L = æ ----------------ö ⋅ ---- ⋅ --è 2 ⋅ π ⋅ fø C 2
In this equation, f is the desired operating frequency
and L is the value of the inductor required. The value C
is the amount of capacitance presented by the varactors and parasitics. For calculation purposes 1.5pF
should be used. The factor of one-half is due to the
inductors being in each leg. As an example, assume
an operating frequency of 433MHz. The calculated
value of each inductor is 45nH. A 47nH inductor would
be appropriate as the closest available value.
11-41
11
TRANSCEIVERS
60 ⋅ C load
1
C 1 = ------------------------ and C 2 = -------------------------1
freq MHz
1
------------- – -----C load C 1
RF2516
Preliminary
The setup of the VCO can be summarized as follows.
First, open the loop. Next, get the VCO to run on the
desired frequency by selecting the proper inductor and
capacitor values. The capacitor value will need to
include the varactor and circuit parasitics.
where the time constants are defined as:
After the VCO is running at the desired frequency, set
the VCO sensitivity. The sensitivity is determined by
connecting the control voltage input point to ground
and noting the frequency.
The frequency at which unity gain occurs is given by:
Connect the same point to the supply, and again note
the frequency. The difference between these two frequencies divided by the supply voltage is the VCO sensitivity expressed in Hz/V. Increasing the inductor value
while decreasing the capacitor value will increase the
sensitivity. Decreasing the inductor value while
increasing the capacitor value will lower the sensitivity.
When increasing or decreasing component values,
make sure that the center frequency remains constant.
Finally, close the loop.
TRANSCEIVERS
11
External to the part, the designer needs to implement a
loop filter to complete the PLL. The loop filter converts
the output of the charge pump into a voltage that is
used to control the VCO. Internally, the VCO is connected to the charge pump output through a 4kΩ resistor. The loop filter is then connected in parallel to this
point at pin 14 (LOOP FLT). This limits the loop filter
topology to a second order filter usually consisting of a
shunt capacitor and a shunt series RC. A passive filter
is most common, as it is a low-cost and low-noise
design. An additional pole could be used for reducing
the reference spurs, however there is not a way to add
the series resistor. However, this should not be a reason for concern.
The schematic of the loop filter is:
Charge Pump
VCC
Loop Filter
VCO
R2
C1
C2
C1 ⋅ C2
τ 2 = R 2 ⋅ C 2 and τ 1 = R 2 ⋅ æ ------------------- ö
è C1 + C 2 ø
1
ω LBW = ------------------τ1 ⋅ τ2
This is also the loop bandwidth.
If the phase margin (PM) and the loop bandwidth
(ωLBW) are known, it is possible to calculate the time
constants. These are found using the equations4:
sec ( PM ) – tan ( PM )
1
τ 1 = -------------------------------------------------- and τ 2 = -----------------------2
ω LBW
ω LBW ⋅ τ 1
With these known, it is then possible to determine the
values of the filter components.4
2
τ 1 K PD ⋅ K VCO 1 + ( ω LBW ⋅ τ 2 )
- ⋅ ---------------------------------------C 1 = ----- ⋅ ---------------------------2
τ2 ω2
⋅
N
1 + ( ω LBW ⋅ τ 1 )
LBW
τ2
C 2 = C 1 ⋅ æ ----- – 1ö
è τ1 ø
τ2
R 2 = -----C2
As an example, consider a loop bandwidth of 50kHz, a
phase margin of 45°, a divide ratio of 64, a KVCO of
20MHz/V, and a KPD of 0.01592mA/2πrad. Time constant τ1 is 1.31848µs, time constant τ2 is 7.68468µs,
C1 is 131.15pF, C2 is 633.26pF, and R2 is 12.14kΩ.
In order to perform these calculations, one will need to
know the value of two constants, KVCO and KPD. KPD is
calculated by dividing the charge pump current by 2π.
For the RF2516, the charge pump current is 100µA.
KVCO is best found empirically as it will change with
frequency and board parasitics. By briefly connecting
pin 14 (LOOP FLT) to VCC and then to ground, the frequency tuning range of the VCO can be seen. Dividing
the difference between these two frequencies by the
difference in the voltage gives KVCO in MHz/V.
The transfer function is:
s ⋅ τ2 + 1
F ( s ) = R 2 ⋅ ------------------------------------------s ⋅ τ2 ⋅ ( s ⋅ τ1 + 1 )
11-42
Rev A10 010613
RF2516
Preliminary
Pin 15 (LD FLT) is used to set the threshold of the
lock-detect circuit. A shunt capacitor is used to set an
RC time constant with an on-chip series 1kΩ resistor.
The time constant should be approximately 10 times
the reference period.
General RF bypassing techniques must be observed
to get the best performance. Choose capacitors such
that they are series resonant near the frequency of
operation.
Board layout is always an area in which great care
must be taken. The board material and thickness are
used in calculating the RF line widths. The use of vias
for connection to the ground plane allows one to connect to ground as close as possible to ground pins.
When laying out the traces around the VCO, it is desirable to keep the parasitics equal between the two legs.
This will allow equal valued inductors to be used.
Pre-compliance testing should be performed during
the design process. This can be done with a GTEM cell
or at a compliance testing laboratory. It is recommended that pre-compliance testing be performed so
that there are no surprises during final compliance
testing. This will help keep the product development
and release on schedule.
Working with a laboratory offers the benefit of years of
compliance testing experience and familiarity with the
regulatory issues. Also, the laboratory can often provide feedback that will help the designer make the
product compliant.
On the other hand, having a GTEM cell or an open air
test site locally offers the designer the ability to rapidly
determine whether or not design changes impact the
product's compliance. Set-up of an open air test site
and the associated calibration is not trivial. An alternative is to use a GTEM test cell.
After the design has been completed and passes compliance testing, application will need to be made with
the respective regulatory bodies for the geographic
region in which the product will be operated to obtain
final certifications.
Rev A10 010613
RF2516 Typical Applications
FCC Part 15.231 Periodic Transmitter - 315MHz Automotive Keyless Entry Transmitter
The following information is taken or paraphrased from
the Code of Federal Regulations Title 47, Part 15, Section 231 (47 CFR 15.231). Part 15 discusses radio frequency devices and section 231 discusses periodic
transmissions. Please refer to the regulation itself as
the final authority. Additional information may be found
on the Internet at www.fcc.gov.
To highlight the main guidelines outlined by this section, there are five main limitations: operating frequency, transmission content, transmission duration,
emission bandwidth, and spurious emissions.
Part 15.231 allows operation in two bands: 40.66MHz
to 40.70MHz and above 70MHz. Transmission is limited to control signals such as alarm systems, door
openers, remote switches, etc. Radio control of toys is
not permitted, nor is continuous transmission such as
voice or video. Data transmission other than a recognition code is not permitted. Transmission time is limited
to 5 seconds (paragraph a) or for 1 second with greater
than ten seconds off (paragraph e).
Emission bandwidth between 70MHz and 900MHz
can not be more than 0.25% of the center frequency.
Above 900MHz, the emission bandwidth cannot be
greater than 0.50% of the center frequency. The emission bandwidth is determined from the points that are
20dB down from the modulated carrier. This corresponds to an occupied bandwidth of 4.5MHz at a center frequency of 902MHz, 1.1MHz at 433MHz, and
788kHz at 315MHz.
Spurious emissions limits are listed in tabular form for
various frequency ranges in the Section 231. Above
470MHz with a manually activated transmitter, the fundamental field strength at a distance of 3 meters shall
not exceed 12,500microvolts/meter. The spurious
emissions shall not exceed 1,250microvolt/meter at a
distance of 3meters above 470MHz. Refer to Appendix A for a method of converting field strength to power.
In the frequency range of 260MHz to 470MHz, one
needs to linearly interpolate the maximum emissions
level for both the fundamental and spurious emissions.
The equation for this line is given by:
2
1
E µV = 41 --- ⋅ FreqMHz – 7083 --3
3
------m
11-43
11
TRANSCEIVERS
The control lines provide an interface for connecting
the device to a microcontroller or other signal generating mechanism. The designer can treat pin 8 (MOD
IN), pin 16 (DIV CTRL), and pin 3 (PD) as control pins
whose voltage level can be set. The lock-detect voltage
at pin 15 (LD FLT) is an output that can be monitored
by the microcontroller.
RF2516
Preliminary
This equation is derived from the endpoints of the frequency range and their respective field strengths. Note
that the field strength is in microvolts per meter and the
frequency is in megahertz. To determine the spurious
level, divide the level calculated above for the spurious
frequency by ten.
As an example, assume the fundamental is 315MHz
and the reference frequency is 9.8MHz. The field
strengths of the fundamental, the reference spurs, and
the harmonics of the fundamental up through the tenth
harmonic are calculated in the following table The
occupied bandwidth limit is 787.5kHz. As shown in
Table A, the fifth, seventh, and ninth harmonics fall into
restricted bands as called out in section 15.205. The
limits for these restricted bands are called out in section 15.209. The power level in the last column is the
level if the output is connected directly to a spectrum
analyzer. Refer to Appendix A as to how this column
was calculated.
Since the RF2516 has a phase-locked VCO, it can be
used as a signal source. The device is an ASK/OOK
transmitter, with the data provided at the MOD IN pin.
When the MOD IN is a high logic level, the carrier is
transmitted. When MOD IN is a low logic level, then the
carrier is not transmitted. Therefore, to use the RF2516
as signal source, simply tie the MOD IN pin to the supply voltage, through a suitable series resistor (minimum 3kΩ).
Conclusions
The RF2516 is an AM/OOK VHF/UHF transmitter that
features a phase-locked output. This device is suitable
for use in a CFR Part 15.231 compliant product as well
as a local oscillator signal source. Two examples showing these applications were discussed.
The RF2516 is packaged in a low-cost plastic package
and requires few external parts, thus making it suitable
for low-cost designs.
Local Oscillator Source
Ref Spur
1
Ref Spur
2
3
4
5
6
7
8
9
10
TRANSCEIVERS
11
Frequency
(MHz)
305.2
315.0
324.8
630.0
945.0
1260.0
1575.0
1890.0
2205.0
2520.0
2835.0
3150.0
15.205 Limits
(µV/m@ 3m)
500
500
500
-
15.231 Limits
(µV/m@3m)
604.17
6041.67
604.17
604.17
604.17
604.17
604.17
604.17
604.17
Final FCC Mask
(µV/m@ 3m)
604.17
6041.67
604.17
604.17
604.17
604.17
500.00
604.17
500.00
604.17
500.00
604.17
Final FCC Mask
(µV/m@3m)
55.62
75.62
55.62
55.62
55.62
55.62
53.98
55.62
53.98
55.62
53.98
55.62
Power Level
(dBm, 50 Ω)
-39.61
-19.61
-39.61
-39.61
-39.61
-39.61
-41.25
-39.61
-41.25
-39.61
-41.25
-39.61
Table A
11-44
Rev A10 010613
RF2516
Preliminary
Pin Out
OSC B
1
16 DIV CTRL
OSC E
2
15 LD FLT
PD
3
14 LOOP FLT
GND
4
13 RESNTR+
TX OUT
5
12 RESNTR-
GND1
6
11 VREFP
VCC1
7
10 GND2
MOD IN
8
9
VCC2
TRANSCEIVERS
11
Rev A10 010613
11-45
RF2516
Preliminary
Application Schematic
315MHz
9.83 M H z
1 nF
68 pF
1
O SC B
D IV C TR L
16
2
O SC E
LD F LT
15
3
PD
LO O P F LT
14
4
GND
RESNTR+
13
5
TX O U T
RESNTR-
12
6
GND1
VREFP
11
7
VCC1
G ND2
10
8
M O D IN
VCC2
9
2.2 nF
4 .3 kΩ
33p F
VC
10 Ω
C
S1
C A S -12 0 B
J1
TX OUT
50 Ω µs trip
4 pF
10 Ω
50 Ω µstrip
56 nH
TX V C C
82 n H
82 n H
VC
2 kΩ
C
1 0 nF
VC
2 20 pF
C
*N ot populated on standa rd E valuation B o ard .
16 k Ω
100 pF
M O D IN
TRANSCEIVERS
11
11-46
Rev A10 010613
RF2516
Preliminary
Application Schematic
315MHz
1 50 kΩ
R F2516 A ud io T ransm itter
9 .8 3 M H z
A U D IO
D1
S M V 1 2 49 -0 11
6 8 pF
1 nF
1
OSC B
D IV C T R L
16
2
OSC E
LD F LT
15
3
PD
L O O P F LT
14
4
GND
R ES N T R +
13
5
TX O UT
R ES N T R -
12
6
GND1
V R E FP
11
7
VCC1
GND2
10
8
M O D IN
VCC2
9
4.3 kΩ
2.2 nF
3 3p F
V CC
10 Ω
S1
C A S -120 B
J1
TX OUT
50 Ω µstrip
4 pF
10 Ω
5 0 Ω µstrip
56 n H
TX VCC
2 2 0 pF
16 kΩ
82 nH
2 kΩ
82 nH
V CC
10 n F
V CC
1 0 0 pF
V CC
TRANSCEIVERS
11
Rev A10 010613
11-47
RF2516
Preliminary
Application Schematic
433MHz
DIV CTRL
13.57734 MHz
*
68 pF
1
OSC B
2
OSC E
3
PD
4
GND
RESNTR+
13
5
TX OUT
6
DIV CTRL
16
LD FLT
15
LOOP FLT
14
1 nF
33pF
220 pF
2.2 nF
4.3 kΩ
PWR DWN
50 Ω µstrip
2 pF
68 nH
4 pF
50 Ω µstrip
50 Ω µstrip
TX OUT
15 pF
10 nH
10 nH
15 pF
RESNTR-
12
GND1
VREFP
11
7
VCC1
GND2
10
8
MOD IN
VCC2
9
22 nH
VCC
10 nF
10 Ω
39 nH
2 kΩ
39 nH
220 pF
3 kΩ
10 nF
10 nF
220 pF
220 pF
VCC
10 nF
10 nF
MOD IN
VCC
TRANSCEIVERS
11
11-48
10 Ω
10 nF
*Not populated on standard Evaluation Board.
220 pF
VCC (V) Mod. in Res. Value
(R5)
1k
2.0
3k
5k
7k
9k
11k
13k
15k
17k
19k
21k
ICC
(mA)
11.08
10.83
4.61
4.00
3.63
3.42
3.26
3.15
3.07
3.01
2.95
POUT
(dBm)
-6.23
-4.40
-5.61
-6.66
-8.08
-8.93
-10.04
-10.71
-11.58
-12.32
-13.10
VCC (V) Mod. in Res. Value
(R5)
1k
2.4
3k
5k
7k
9k
11k
13k
15k
17k
19k
21k
ICC
(mA)
14.05
9.00
7.48
6.73
6.16
5.79
5.53
5.29
5.13
4.98
4.86
POUT
(dBm)
7.94
7.63
5.95
4.64
3.35
2.40
1.47
0.75
0.05
-0.60
-1.26
VCC (V) Mod. in Res. Value
(R5)
1k
3.2
3k
5k
7k
9k
11k
13k
15k
17k
19k
21k
ICC
(mA)
20.90
12.12
9.66
8.95
8.23
7.75
7.42
7.10
6.89
6.68
6.52
POUT
(dBm)
6.77
9.70
8.30
7.11
5.91
5.02
4.16
3.51
2.89
2.26
1.66
VCC (V) Mod. in Res. Value
(R5)
1k
3.6
3k
5k
7k
9k
11k
13k
15k
17k
19k
21k
ICC
(mA)
24.68
13.88
10.94
10.14
9.34
8.81
8.44
8.09
7.86
7.63
7.44
POUT
(dBm)
5.78
10.42
9.18
8.08
6.88
6.02
5.19
4.52
3.93
3.35
2.72
VCC (V) Mod. in Res. Value
(R5)
1k
2.8
3k
5k
7k
9k
11k
13k
15k
17k
19k
21k
ICC
(mA)
17.38
10.51
8.68
7.82
7.18
6.75
6.45
6.18
5.99
5.80
5.66
POUT
(dBm)
7.45
8.78
7.23
6.00
4.73
3.81
2.98
2.30
1.63
1.00
0.35
Rev A10 010613
RF2516
Preliminary
Evaluation Board Schematic
315MHz
(Download Bill of Materials from www.rfmd.com.)
P1
P1-1
P1-3
1
VCC1
2
GND
3
MOD IN
VCC
+
B1
LITH BATT
CON3
-
Y1
9.83 MHz
C7
33pF
R2
10 Ω
C3
1 nF
C8
68 pF
CAS-120B
J1
TX OUT
50 Ω µstrip
R5
10 Ω
OSC B
DIV CTRL
16
2
OSC E
LD FLT
15
3
PD
LOOP FLT
14
4
GND
RESNTR+
13
5
TX OUT
RESNTR-
12
6
GND1
VREFP
11
7
VCC1
GND2
10
8
MOD IN
VCC2
9
R3
4.3 kΩ
C2
2.2 nF
S1
VCC
C6
4 pF
1
L1
82 nH
R1
2 kΩ
VCC
50 Ω µstrip
L3
56 nH
TX VCC
C5
220 pF
L2
82 nH
C1
1 µF
VCC
2516400, rev A
R4
16 kΩ
C4
100 pF
*Not populated on standard Evaluation Board.
11
TRANSCEIVERS
MOD IN
Rev A10 010613
11-49
RF2516
Preliminary
Evaluation Board Schematic
433MHz
P2
P1
1
NC
1
PWR DWN
2
GND
2
GND
3
VCC
3
DIV CTRL
CON3
C17*
C1
33pF
PWR DWN
J1
TX OUT
DIV CTRL
X1
13.57734 MHz
CON3
C2
68 pF
1
OSC B
2
OSC E
3
50 Ω µstrip
C14
15 pF
C15
2 pF
L3
10 nH
L4
68 nH
L5
10 nH
50 Ω µstrip
C3
4 pF
PD
DIV CTRL
16
LD FLT
15
LOOP FLT
14
4
GND
RESNTR+
13
5
TX OUT
RESNTR-
12
6
GND1
VREFP
11
7
VCC1
GND2
10
8
MOD IN
VCC2
9
50 Ω µstrip
C16
15 pF
L6
22 nH
C6
1 nF
C7
220 pF
R2
4.3k Ω
C8
2.2 nF
VCC
L1
39 nH
L2
39 nH
R3
10 Ω
R4
2k Ω
C9
220 pF
C10
10 nF
C11
10 nF
C12
220 pF
C13
10 nF
VCC
C19
10 nF
C18
220 pF
R5
3k Ω
C20
10 nF
J2
MOD IN
R1
10 Ω
C4
10 nF
C5
220 pF
*Not populated on standard Evaluation Board.
VCC
TRANSCEIVERS
11
11-50
Rev A10 010613
RF2516
Preliminary
Evaluation Board Layout (315MHz)
Board Size 1.285” x 1.018”
Board Thickness 0.062”, Board Material FR-4
TRANSCEIVERS
11
Rev A10 010613
11-51
RF2516
Preliminary
Evaluation Board Layout (433MHz)
Board Size 1.392” x 1.392”
Board Thickness 0.031”, Board Material FR-4
TRANSCEIVERS
11
11-52
Rev A10 010613
RF2516
Preliminary
433MHz Phase Noise
0.8
Swp Max
1GHz
2.
0
0.6
VCC = 2 V
1.0
RF2516 Output Z
VCC = 3 V
VCC = 3.3 V
0.
4
0
3.
4.0
5.0
0.2
10.0
4.0
5.0
3.0
2.0
1.0
0.8
0.6
0.4
0.2
0
10.0
1.0 GHz
0.1 GHz
11
-10.0
-0.2
TRANSCEIVERS
-4.
0
-5.0
-3
.0
Rev A10 010613
-2
-1.0
-0.8
-0.
6
.0
.4
-0
Swp Min
0.1GHz
11-53
RF2516
Preliminary
TRANSCEIVERS
11
11-54
Rev A10 010613