TPS55340 www.ti.com SLVSBD4 – MAY 2012 Integrated 5-A 40-V Wide Input Range Boost/SEPIC/Flyback DC-DC Regulator Check for Samples: TPS55340 FEATURES DESCRIPTION • • • • • • • • • • The TPS55340 is a monolithic non-synchronous switching regulator with integrated 5-A, 40-V power switch. It can be configured in several standard switching-regulator topologies, including boost, SEPIC and isolated flyback. The device has a wide input voltage range to support applications with input voltage from multi-cell batteries or regulated 3.3-V, 5V, 12-V, and 24-V power rails. 1 2 • • • Internal 5-A, 40-V Low-Side MOSFET Switch 2.9-V to 32-V Input Voltage Range ±0.7% Reference Voltage 0.5mA Operating Quiescent Current 2.7µA Shutdown Supply Current Fixed Frequency Current Mode PWM Control Frequency Adjustable from 100kHz to 1.2MHz Synchronization Capability to External Clock Adjustable Soft-Start Time Pulse-Skipping for Higher Efficiency at Light Loads Cycle-by-Cycle Current Limit, Thermal Shutdown, and UVLO Protection QFN-16 (3mmx3mm) Package with PowerPad™ Wide –40°C to 150°C Operating TJ Range • • The device features a programmable soft-start function to limit inrush current during start-up and has other built-in protection features including cycle-bycycle over current limit and thermal shutdown. The TPS55340 is available in a small 3mm x 3mm 16-pin QFN package with PowerPad™ for enhanced thermal performance. APPLICATIONS • • • The TPS55340 regulates the output voltage with current mode PWM (pulse width modulation) control, and has an internal oscillator. The switching frequency of PWM is set by either an external resistor or by synchronizing to an external clock signal. The user can program the switching frequency from 100 kHz to 1.2 MHz. 3.3-V, 5-V, 12-V, 24-V Power Conversion Boost, SEPIC, and Flyback Topologies Thunderbolt Port, Power Docking for Tablets and Portable PCs Industrial Power Systems ADSL Modems TYPICAL APPLICATION (BOOST) L VIN D 100 VOUT 95 TPS55340 VIN SW EN SW FREQ SW RSH SS FB COMP PGND SYNC PGND AGND PGND CSS RFREQ RC CC 90 CO Efficiency (%) CI 85 VOUT = 24 V fSW = 600 kHz 80 75 70 65 RSL VIN = 15 V VIN = 12 V VIN = 5 V 60 55 50 0 0.4 0.8 1.2 1.6 Output Current (A) 2 2.4 G031 √ 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPad is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2012, Texas Instruments Incorporated TPS55340 SLVSBD4 – MAY 2012 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ORDERING INFORMATION (1) (1) TJ PART NUMBER PACKAGE -40°C to 150°C TPS55340RTE QFN-16 For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com. PIN ASSIGNMENTS SW 1 VIN 2 EN SS SW SW NC PGND QFN-16 PACKAGE (Top View) 16 15 14 13 12 PGND 11 PGND 3 10 NC 4 9 7 FREQ 8 FB 6 COMP SYNC 5 AGND PowerPAD (17) PIN FUNCTIONS PIN NAME DESCRIPTION NO. VIN 2 SW 1, 15, 16 FB 8 Error amplifier input and feedback pin for positive voltage regulation. Connect to the center tap of a resistor divider to program the output voltage. EN 3 Enable pin. When the voltage of this pin falls below the enable threshold for more than 1ms, the IC turns off. COMP 7 Output of the transconductance error amplifier. An external RC network connected to this pin compensates the regulator feedback loop. SS 4 Soft-start programming pin. A capacitor between the SS pin and AGND pin programs soft-start timing. FREQ 9 Switching frequency program pin. An external resistor connected between the FREQ pin and AGND sets the switching frequency. AGND 6 Signal ground of the IC. PGND 11, 12, 13 SYNC 5 NC 2 10, 14 The input supply pin to the IC. Connect VIN to a supply voltage between 2.9V and 32V. It is acceptable for the voltage on the pin to be different from the boost power stage input. SW is the drain of the internal power MOSFET. Connect SW to the switched side of the boost or SEPIC inductor or the flyback transformer. Power ground of the IC. It is connected to the source of the internal power MOSFET switch. Switching frequency synchronization pin. An external clock signal can be used to set the switching frequency between 200kHz and 1.0MHz. If not used, this pin should be tied to AGND. Reserved pin that must be connected to ground. Copyright © 2012, Texas Instruments Incorporated TPS55340 www.ti.com SLVSBD4 – MAY 2012 PIN FUNCTIONS (continued) PIN NAME PowerPAD DESCRIPTION NO. 17 The PowerPAD should be soldered to the AGND. If possible, use thermal vias to connect to internal ground plane for improved power dissipation. FUNCTIONAL BLOCK DIAGRAM VIN SW FB Error Amp EN 1.229V Reference COMP PWM Control Ramp Generator Gate Driver Lossless Current Sense S Oscillator SS FREQ Copyright © 2012, Texas Instruments Incorporated SYNC AGND PGND 3 TPS55340 SLVSBD4 – MAY 2012 www.ti.com ABSOLUTE MAXIMUM RATINGS (1) over operating temperature range (unless otherwise noted) VALUE UNIT MIN MAX Supply voltages on pin VIN (2) –0.3 34 V Voltage on pin EN (2) –0.3 34 V Voltage on pins FB, FREQ, and COMP (2) –0.3 3 V Voltage on pin SS (2) –0.3 5 V Voltage on pin SYNC (2) –0.3 7 V Voltage on pin SW (2) –0.3 40 V Operating junction temperature range –40 150 °C Storage temperature range –65 150 °C 2 kV 500 V (HBM) QSS 009-105 (JESD22-A114A) Electrostatic discharge (1) (2) (CDM) QSS 009-147 (JESD22-C101B 01) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal THERMAL INFORMATION TPS55340 THERMAL METRIC (1) QFN (16-PINS) θJA Junction-to-ambient thermal resistance 43.3 θJCtop Junction-to-case (top) thermal resistance 38.7 θJB Junction-to-board thermal resistance 14.5 ψJT Junction-to-top characterization parameter 0.4 ψJB Junction-to-board characterization parameter 14.5 θJCbot Junction-to-case (bottom) thermal resistance 3.5 UNITS °C/W space (1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953. RECOMMENDED OPERATING CONDITIONS MIN NOM MAX UNIT VIN Input voltage range 2.9 32 V VOUT Output voltage range VIN 38 V VEN EN voltage range 0 32 V VSYN External switching frequency logic input range 0 5 V TA Operating free-air temperature –40 125 °C TJ Operating junction temperature –40 150 °C MAX UNIT ELECTRICAL CHARACTERISTICS Vin=5V, TJ = –40°C to +150°C, unless otherwise noted. Typical values are at TA = 25°C. PARAMETER TEST CONDITIONS MIN TYP SUPPLY CURRENT VIN Input voltage range IQ Operating quiescent current into Vin Device non-switching, VFB = 2 V 0.5 ISD Shutdown current EN = GND 2.7 VUVLO Under-voltage lockout threshold VIN falling Vhys Under-voltage lockout hysteresis 4 2.9 120 Submit Documentation Feedback 32 V mA 10 µA 2.5 2.7 V 140 160 mV Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 ELECTRICAL CHARACTERISTICS (continued) Vin=5V, TJ = –40°C to +150°C, unless otherwise noted. Typical values are at TA = 25°C. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V ENABLE AND REFERENCE CONTROL VEN EN threshold voltage 0.9 1.08 1.30 VENh EN threshold hysteresis EN rising input 0.1 0.16 0.22 V REN EN pull down resistor 400 950 1600 kΩ Toff Shutdown delay, SS discharge VSYNh SYN logic high voltage VSYNl SYN logic low voltage EN high to low 1.0 ms 1.2 0.4 V VOLTAGE AND CURRENT CONTROL 1.204 1.229 1.254 1.220 1.229 1.238 TA = 25°C 1.6 20 VFB = VREF+200 mV, VCOMP = 1 V 42 µA 42 µA 3.1 0.75 V VREF Voltage feedback regulation voltage IFB Voltage feedback input bias current Isink Comp pin sink current Isource Comp pin source current VFB = VREF–200 mV, VCOMP = 1 V VCCLP Comp pin Clamp Voltage High Clamp, VFB = 1 V Low Clamp, VFB = 1.5 V VCTH Comp pin threshold Duty cycle = 0% 1.04 V Gea Error amplifier transconductance Rea Error amplifier output resistance fea Error amplifier crossover frequency TA = 25°C 240 360 V nA 440 µmho 10 MΩ 500 kHz FREQUENCY fSW Frequency Dmax Maximum duty cycle VFREQ FREQ pin voltage Tmin_on Minimum on pulse width RFREQ = 480 kΩ 75 94 RFREQ = 80 kΩ 460 577 740 RFREQ = 40 kΩ 920 1140 1480 89% 96% VFB = 1.0 V, RFREQ = 80 kΩ 130 kHz 1.25 V RFREQ = 80 kΩ 77 ns 60 70 POWER SWITCH RDS(ON) N-channel MOSFET on-resistance VIN = 5 V VIN = 3 V ILN_NFET N-channel leakage current VDS = 25 V, TA = 25°C 110 120 mΩ 2.1 µA 7.75 A OCP and SS ILIM N-Channel MOSFET current limit D = Dmax ISS Soft-start bias current Vss = 0 V 5.25 6.6 6 µA 165 °C 15 °C THERMAL SHUTDOWN Tshutdown Thermal shutdown threshold Thysteresis Thermal shutdown threshold hysteresis Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 5 TPS55340 SLVSBD4 – MAY 2012 www.ti.com TYPICAL CHARACTERISTICS Vin = 5 V, TA = 25°C (unless otherwise noted) 8 Current Limit Threshold (A) Transconductance (µA/V) 400 380 360 340 320 7 6 5 4 3 2 300 −50 −25 0 25 50 75 Temperature (°C) 100 125 1 −50 150 Figure 1. Error Amplifier Transconductance vs Temperature 125 150 G002 VIN = 3 V Resistance (mΩ) Voltage Reference (V) 100 100 1.228 1.226 1.224 80 60 VIN = 12 V 40 VIN = 5 V 20 1.22 −50 −25 0 25 50 75 Temperature (°C) 100 125 0 −50 150 −25 0 G003 Figure 3. Feedback Voltage Reference vs Temperature 1600 1400 1400 1200 Frequency (kHz) 1000 800 600 400 25 50 75 Temperature (°C) 30 100 Resistance (kΩ) 500 125 150 G004 1000 800 600 RFREQ = 40 kΩ RFREQ = 80 kΩ RFREQ = 480 kΩ 400 200 200 100 Figure 4. RDS(ON) vs Temperature 1200 Frequency (kHz) 25 50 75 Temperature (°C) 120 1.222 0 −50 G005 Figure 5. Frequency vs FREQ Resistance 6 0 Figure 2. Switch Current Limit vs Temperature 1.23 0 −25 G001 Submit Documentation Feedback −25 0 25 50 75 Temperature (°C) 100 125 150 G006 Figure 6. Frequency vs Temperature Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 TYPICAL CHARACTERISTICS (continued) Vin = 5 V, TA = 25°C (unless otherwise noted) 700 3.5 RFREQ = 80 kΩ 3 500 COMP Voltage (V) Frequency (kHz) 600 400 Non-Foldback Foldback 300 200 2.5 1.5 1 100 0 −50 −25 0 25 50 75 Temperature (°C) 100 125 0.5 −50 150 0 25 50 75 Temperature (°C) 100 125 150 G008 Figure 8. COMP Clamp Voltage vs Temperature 2.7 1.3 EN Voltage Rising EN Voltage Falling 1.2 Enable Voltage (V) 2.66 2.62 UVLO Start UVLO Stop 2.58 2.54 1.1 1 0.9 0.8 2.5 −50 −25 0 25 50 75 Temperature (°C) 100 125 0.7 −50 150 0 25 50 75 Temperature (°C) 100 125 150 G010 Figure 10. Enable Voltage vs Temperature 100 100 RFREQ = 80 kΩ RFREQ = 80 kΩ 95 Minimum On Time (ns) 99 98 97 96 95 94 −50 −25 G009 Figure 9. Input Voltage UVLO vs Temperature Maximum Duty Cycle (%) −25 G007 Figure 7. Non-Foldback Frequency vs Foldback Frequency Input Voltage (V) COMP Pin Clamp High COMP Pin Clamp Low 2 90 85 80 75 −25 0 25 50 75 Temperature (°C) 100 125 150 70 −50 G011 Figure 11. Maximum Duty Cycle vs Temperature −25 0 25 50 75 Temperature (°C) 100 125 150 G012 Figure 12. Minimum On Time vs Temperature Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 7 TPS55340 SLVSBD4 – MAY 2012 www.ti.com TYPICAL CHARACTERISTICS (continued) 8 2.1 7 1.8 Supply Current (mA) Shutdown Current (µA) Vin = 5 V, TA = 25°C (unless otherwise noted) 6 5 4 3 Switching Non-Switching 1.2 0.9 0.6 2 1 −50 −25 0 25 50 75 Temperature (°C) 100 125 150 0.3 −50 −25 G013 Figure 13. Shutdown Current vs Temperature 8 1.5 0 25 50 75 Temperature (°C) 100 125 150 G014 Figure 14. Supply Current vs Temperature Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 DETAILED DESCRIPTION OPERATION The TPS55340 integrates a 5 A, 40 V low side n-channel MOSFET for boost converter output up to 38V. The TPS55340 regulates the output with current mode PWM (pulse width modulation) control. The PWM control circuitry turns on the switch at the beginning of each oscillator clock cycle. The input voltage is applied across the inductor and stores the energy as inductor current ramps up. During this portion of the switching cycle, the load current is provided by the output capacitor. When the inductor current reaches a threshold level set by the error amplifier output, the power switch turns off and the external Schottky diode is forward biased to allow the inductor current to flow to the output. The inductor transfers stored energy to replenish the output capacitor and supply the load current. This operation repeats every switching cycle. The duty cycle of the converter is determined by the PWM control comparator which compares the error amplifier output and the current signal. The oscillator frequency is programmed by the external resistor or synchronized to an external clock signal. A ramp signal from the oscillator is added to the inductor current ramp to provide slope compensation. Slope compensation is necessary to avoid sub-harmonic oscillation that is intrinsic to peak current mode control at duty cycles higher than 50%. If the inductor value is too small, the internal slope compensation may not be adequate to maintain stability. The PWM control feedback loop regulates the FB pin to a reference voltage through a transconductance error amplifier. The output of the error amplifier is connected to the COMP pin. An external RC compensation network connected to the COMP pin is chosen for feedback loop stability and optimum transient response. SWITCHING FREQUENCY The switching frequency is set by a resistor (RFREQ) connected to the FREQ pin of the TPS55340. The relationship between the timing resistance RFREQ and frequency is shown in the Figure 5. Do not leave this pin open. A resistor must always be connected from the FREQ pin to ground for proper operation. The resistor value required for a desired frequency can be calculated using Equation 1. RFREQ(kΩ) = 57500 × ƒsw(kHz)–1.03 (1) For the given resistor value, the corresponding frequency can be calculated by Equation 2. ƒsw(kHz) = 41600 × RFREQ(kΩ)–0.97 (2) The TPS55340 switching frequency can be synchronized to an external clock signal that is applied to the SYNC pin. The required logic levels of the external clock are shown in the specification table. The recommended duty cycle of the clock is in the range of 10% to 90%. A resistor must be connected from the FREQ pin to ground when the converter is synchronized to the external clock and the external clock frequency must be within ±20% of the corresponding frequency set by the resistor. For example, if the frequency programmed by the FREQ pin resistor is 600kHz, the external clock signal should be in the range of 480kHz to 720kHz. VOLTAGE REFERENCE AND SETTING OUTPUT VOLTAGE An internal voltage reference provides a precise 1.229 V voltage reference at the error amplifier non-inverting input. To set the output voltage, select the FB pin resistor RSH and RSL according to Equation 3. æR ö VOUT = 1.229V ´ ç SH + 1÷ è RSL ø (3) SOFT-START The TPS55340 has a built-in soft-start circuit which significantly reduces the start-up current spike and output voltage overshoot. When the IC is enabled, an internal bias current source (6 µA typical) charges a capacitor (CSS) on the SS pin. The voltage at the capacitor clamps the output of the internal error amplifier that determines the peak current and duty cycle of PWM controller. Limiting the peak switch current during start-up with a slow ramp on the SS pin will reduce in-rush current and output voltage overshoot. Once the capacitor reaches 1.8V, the soft-start cycle is completed and the soft-start voltage no longer clamps the error amplifier output. When the EN is pulled low for at least 1ms, the IC enters the shutdown mode and the SS capacitor is discharged through a 5kΩ resistor to prepare for the next soft-start sequence. Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 9 TPS55340 SLVSBD4 – MAY 2012 www.ti.com SLOPE COMPENSATION The TPS55340 has internal slope compensation to prevent sub-harmonic oscillations. The sensed current slope of boost converter can be expressed as Equation 4: V Sn = IN ´ RSENSE (4) L The slope compensation dv/dt can be calculated using Equation 5. 0.32 V RFREQ 0.5 mA Se = + 16 ´ (1 - D) ´ 6 pF 6 pF (5) In a converter with current mode control, in addition to the output voltage feedback loop, the inner current loop including the inductor current sampling effect as well as the slope compensation on the small signal response should be taken into account, which can be modeled as seen in Equation 6: 1 He(s) = éæ ù S ö s ´ êç 1 + e ÷ ´ (1 - D) - 0.5 ú Sn ÷ø êëçè úû s2 1+ + 2 fsw p ´ fsw (6) ( ) Where RSENSE (15mΩ) is the equivalent current sense resistor, RFREQ is timing resistor used to set frequency, and D is the duty cycle. Note that if Sn << Se, the converter operates in voltage mode control rather than current mode control, and Equation 6 is no longer valid. OVER-CURRENT PROTECTION AND FREQUENCY FOLDBACK The TPS55340 provides cycle-by-cycle over-current protection that turns off the power switch once the inductor current reaches the over-current limit threshold. The PWM circuitry resets itself at the beginning of the next switch cycle. During an over-current event, the output voltage begins to droop as a function of the load on the output. When the FB voltage through the feedback resistors, drops lower than 0.9 V, the switching frequency is automatically reduced to 1/4 of the normal value. Figure 7 shows the non-foldback frequency with an 80kΩ timing resistor and the corresponding foldback frequency. The switching frequency does not return to normal until the over-current condition is removed and the FB voltage increases above 0.9 V. The frequency foldback feature is disabled during soft-start. ENABLE AND THERMAL SHUTDOWN The TPS55340 enters shutdown when the EN voltage is less than 0.68 V (min) for more than 1ms. In shutdown, the input supply current for the device is less than 10µA (max). The EN pin has an internal 950kΩ pull down resistor to disable the device if the pin is floating. An internal thermal shutdown turns off the device when the junction temperature exceeds 165°C (typical). The device will restart when the junction temperature drops by 15°C. UNDER-VOLTAGE LOCKOUT (UVLO) An under-voltage lockout circuit prevents mis-operation of the device at input voltages below 2.5 V (typical). When the input voltage is below the UVLO threshold, the device remains off and the internal power MOSFET is turned off. The UVLO threshold is set below minimum operating voltage of 2.9 V to ensure that a transient VIN dip will not cause the device to reset. For the input voltages between UVLO threshold and 2.9 V, the device attempts to operate, but the electrical specifications are not guaranteed. 10 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 MINIMUM ON TIME AND PULSE SKIPPING The TPS55340 PWM control system has a minimum PWM pulse width of 77ns (typical). This minimum on-time determines the minimum duty cycle of the PWM, for any set switching frequency. When the voltage regulation loop of the TPS55340 requires a minimum on-time pulse width less than 77ns, the IC enters pulse-skipping mode. In this mode, the device will hold the power switch off for several switching cycles to prevent the output voltage from rising above the desired regulated voltage. This operation typically occurs in light load conditions when the PWM operates in discontinuous conduction mode. Pulse skipping increases the output ripple as shown in Figure 21. LAYOUT COSIDERATIONS As for all switching power supplies, especially those with high frequency and high switch current, printed circuit board (PCB) layout is an important design step. If the layout is not carefully designed, the regulator could suffer from instability as well as noise problems. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent radiation of high frequency resonance problems, proper layout of the high frequency switching path is essential. Minimize the length and area of all traces connected to the SW pin and always use a ground plane under the switching regulator to minimize inter-plane coupling. The high current path including the internal MOSFET switch, Schottky diode, and output capacitor, contains nanosecond rise and fall times and should be kept as short as possible. The input capacitor needs not only to be close to the VIN pin, but also to the AGND pin in order to reduce the IC supply ripple. THERMAL CONSIDERATIONS The maximum IC junction temperature should be restricted to 150°C under normal operating conditions. This restriction limits the power dissipation of the TPS55340. The TPS55340 features a thermally enhanced QFN package. This package includes a PowerPad™ that improves the thermal capabilities of the package. The thermal resistance of the QFN package in any application greatly depends on the PCB layout and the PowerPad™ connection. The PowerPad™ must be soldered to the analog ground on the PCB. Use thermal vias underneath the PowerPad™ to achieve good thermal performance. Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 11 TPS55340 SLVSBD4 – MAY 2012 www.ti.com DESIGN GUIDE-STEP-BY-STEP DESIGN PROCEDURE OF BOOST CONVERTER The following section provides a step-by-step design approach for configuring the TPS55340 as a voltage regulating boost converter, as shown in Figure 15. When configured as SEPIC or flyback converter, a different design approach is required. A design example of SEPIC converter is provided in the next section. 100 pF Figure 15. Boost Converter Application Schematic A few parameters must be known in order to start the design process. These parameters are typically determined at the system level. For this example, we will start with the following known parameters: Table 1. Key Parameters of Boost Converter Example PARAMETER VALUE Output Voltage 24 V Input Voltage 5 V to 12 V Maximum Output Current 800 mA Transient Response 50% load step (ΔVOUT = 3%) 960 mV Output Voltage Ripple (0.5% of VOUT) 120 mV SELECTING THE SWITCHING FREQUENCY (R4) The first step is to decide on a switching frequency for the regulator. There are tradeoffs to consider for a higher or lower switching frequency. A higher switching frequency allows for lower valued inductor and smaller output capacitors leading to the smallest solution size. A lower switching frequency will result in a larger solution size but better efficiency. The user will typically set the frequency for the minimum tolerable efficiency to avoid excessively large external components. A switching frequency of 600 kHz is a good trade-off between efficiency and solution size. The appropriate resistor value is found from the resistance versus frequency graph of Figure 5, or calculated using Equation 1. R4 is calculated to be 78.4 kΩ and the nearest standard value resistor of 78.7 kΩ is selected. A resistor must be placed from the FREQ pin to ground, even if an external oscillation is applied for synchronization. 12 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 DETERMINING THE DUTY CYCLE The input to output voltage conversion ratio of the TPS55340 is limited by the worst case maximum duty cycle of 89% and the minimum duty cycle which is determined by the minimum on-time of 77 ns and the switching frequency. The minimum duty cycle can be estimated with Equation 7. With a 600 kHz switching frequency the minimum duty cycle is 4%. DPS = TON min × ƒsw (7) The duty cycle at which the converter operates is dependent on the mode in which the converter is running. If the converter is running in discontinuous conduction mode (DCM), where the inductor current ramps to zero at the end of each cycle, the duty cycle varies with changes of the load much more than it does when running in continuous conduction mode (CCM). In continuous conduction mode, where the inductor maintains a minimum dc current, the duty cycle is related primarily to the input and output voltages as computed below. Assume a 0.5 V drop VD across the Schottky rectifier. At the minimum input of 5 V, the duty cycle will be 80%. At the maximum input of 12 V, the duty cycle is 51%. + VD - VIN V D = OUT VOUT + VD (8) At light loads the converter will operate in DCM. In this case the duty cycle is a function of the load, input and output voltages, inductance and switching frequency as computed below. This can be calculated only after an inductance is chosen in the following section. While operating in DCM with very light load conditions the duty cycle demand will force the TPS55340 to operate with the minimum on time. The converter will then begin pulse skipping which can increase the output ripple. D= 2 ´ (VOUT + VD - VIN ) ´ L ´ IOUT ´ ¦ SW VIN (9) All converters using a diode as the freewheeling or catch component have a load current level at which they transit from discontinuous conduction mode to continuous conduction mode. This is the point where the inductor current just falls to zero during the off-time of the power switch. At higher load currents, the inductor current does not fall to zero and diode and switch current assume a trapezoidal wave shape as opposed to a triangular wave shape. The load current boundary between discontinuous conduction and continuous conduction can be found for a set of converter parameters as follows. IOUT(crit) = (VOUT + VD - VIN )´ VIN2 2 2 ´ (VOUT + VD ) ´ ¦ SW ´ L (10) For loads higher than the result of the Equation 10, the duty cycle is given by Equation 8. For loads less than the results of Equation 10, the duty cycle is given Equation 9. For Equation 7 through Equation 10, the variable definitions are as follows. • VOUT is the output voltage of the converter in V • VD is the forward conduction voltage drop across the rectifier or catch diode in V • VIN is the input voltage to the converter in V • IOUT is the output current of the converter in A • L is the inductor value in H • ƒSW is the switching frequency in Hz Unless otherwise stated, the design equations that follow assume that the converter is running in continuous conduction mode, which typically results in a higher efficiency for the power levels of this converter. SELECTING THE INDUCTOR (L1) The selection of the inductor affects steady state operation as well as transient behavior and loop stability. These factors make it the most important component in power regulator design. There are three important inductor specifications: inductor value, DC resistance and saturation current. Considering inductor value alone is not enough. Inductor values can have ±20% tolerance with no current bias. When the inductor current approaches saturation level, the effective inductance can fall to a fraction of the zero current value. Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 13 TPS55340 SLVSBD4 – MAY 2012 www.ti.com The minimum value of the inductor should be able to meet inductor current ripple (ΔIL) requirement at worst case. In a boost converter, maximum inductor current ripple occurs at 50% duty cycle. For the applications where duty cycle is always smaller or larger than 50%, Equation 12 should be used with the duty cycle closest to 50% and corresponding input voltage to calculate the minimum inductance. For applications that need to operate with 50% duty cycle when input voltage is somewhere between the minimum and the maximum input voltage, Equation 13 should be used. KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum input current (IINDC = ILavg). The maximum input current can be estimated with Equation 11, with an estimated efficiency based on similar applications (ηEST). The inductor ripple current will be filtered by the output capacitor. Therefore, choosing high inductor ripple currents will impact the selection of the output capacitor since the output capacitor must have a ripple current rating equal to or greater than the inductor ripple current. In general, the inductor ripple value (KIND) is at the discretion of the designer. However, the following guidelines may be used. For CCM operation, it is recommended to use KIND values in the range of 0.2 to 0.4. Choosing KIND closer to 0.2 results in a larger inductance value, maximizes the converter’s potential output current and minimizes EMI. Choosing KIND closer to 0.4 results in a smaller inductance value, a physically smaller inductor, and improved transient response, but potentially worse EMI and lower efficiency. Using an inductor with a smaller inductance value may result in the converter operating in DCM. This reduces the boost converter’s maximum output current, causes larger input voltage and output voltage ripple and reduced efficiency. For this design, choose KIND = 0.3 and a conservative efficiency estimate of 85% with the minimum input voltage and maximum output current. Equation 12 is used with the maximum input voltage because this corresponds to duty cycle closest to 50%. The maximum input current is estimated at 4.52A and the minimum inductance is 7.53 µH. A standard value of 10 µH is chosen. VOUT ´ IOUT IINDC = hEST ´ VIN min (11) LO min ³ LO min ³ VIN D ´ IINDC ´ KIND ¦ SW (VOUT + VD ) IINDC ´ KIND ´ , D ≠ 50%, VIN with D closest to 50% 1 , D=50% 4 ´ ¦ SW (12) (13) After choosing the inductance, the required current ratings can be calculated. The inductor will be closest to its ratings with the minimum input voltage. The ripple with the chosen inductance is calculated with Equation 14. The RMS and peak inductor current can be found with Equation 15 and Equation 16. For this design the current ripple is 663mA, the RMS inductor current is 4.52 A, and the peak inductor current is 4.85 A. It is generally recommended for the peak inductor current rating of the selected inductor be 20% higher to account for transients during power up, faults or transient load conditions. The most conservative approach is to specify an inductor with a saturation current greater than the maximum peak current limit of the TPS55340. This helps to avoid saturation of the inductor. The chosen inductor is a Würth Elektronik 74437368100. It has a saturation current rating of 12.5 A, RMS current rating of 5.2 A, and typical DCR of 27.0mΩ. V min Dmax DIL = IN ´ LO ¦ SW (14) æ DI ö (IINDC )2 + ç L ÷ è 12 ø DI IL peak = IINDC + L 2 2 IL rms = (15) (16) The TPS55340 has built-in slope compensation to avoid sub-harmonic oscillation associated with current mode control. If the inductor value is too small, the slope compensation may not be adequate, and the loop can be unstable. 14 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 COMPUTING THE MAXIMUM OUTPUT CURRENT The over-current limit for the integrated power MOSFET limits the maximum input current and thus the maximum input power for a given input voltage. Maximum output power is less than maximum input power due to power conversion losses. Therefore, the current limit setting, input voltage, output voltage and efficiency can all change maximum current output (IOUTmax). The current limit clamps the peak inductor current, therefore the ripple has to be subtracted to derive maximum DC current. Decreasing the KIND or designing for a higher efficiency will increase the maximum output current. This can be evaluated with the chosen inductance or the chosen KIND. This should be evaluated with the minimum input voltage and minimum peak current limit (ILIM) of 5.25 A. DIL ö æ æ KIND ö VIN min ´ ç ILIM ÷ ´ hEST VIN min ´ ILIM ´ ç 1 - 2 ÷ ´ hEST 2 è ø è ø IOUT max = = VOUT VOUT (17) In this design with 5 V input boosted to 24 V output and a 10 μH inductor with an assumed the Schottky forward voltage of 0.5 V and estimated efficiency of 85%, the maximum output current is 871 mA. With the 12 V input and increased estimated efficiency of 90%, the maximum output current increases to 2.13 A. This circuit was evaluated to its maximum output currents with both the minimum and maximum input voltage. SELECTING THE OUTPUT CAPACITOR (C8-C10) At least 4.7 µF of ceramic type X5R or X7R capacitance is recommended at the output. The output capacitance is mainly selected to meet the requirements for the output ripple (VRIPPLE) and voltage change during a load transient. Then the loop is compensated for the output capacitor selected. The output capacitance should be chosen based on the most stringent of these criteria. The output ripple voltage is related to the capacitance and equivalent series resistance (ESR) of the output capacitor. Assuming a capacitor with zero ESR, the minimum capacitance needed for a given ripple can be calculated by Equation 18. If high ESR capacitors are used it will contribute additional ripple. The maximum ESR for a specified ripple is calculated with Equation 19. ESR ripple can be neglected for ceramic capacitors but must be considered if tantalum or electrolytic capacitors are used. The minimum ceramic output capacitance needed to meet a load transient requirement can be estimated by the Equation 20. Equation 21 can be used to calculate the RMS current that the output capacitor needs to support. Dmax ´ IOUT COUT ³ ¦ SW ´ VRIPPLE (18) æ Dmax ´ IOUT ö ç VRIPPLE ÷ ¦ SW ´ COUT ø ESR £ è DIL (19) DITRAN ³ 2 ´ p ´ ¦BW ´ DVTRAN (20) COUT ICOrms = IOUT Dmax (1 - Dmax ) (21) Using Equation 18 for this design, the minimum output capacitance for the specified 120 mV output ripple is 8.8 µF. For a maximum transient voltage change (ΔVTRAN) of 960 mV with a 400 mA load transient (ΔITRAN) and a 6 kHz control loop bandwidth (fBW) with Equation 20, the minimum output capacitance is 11.1 µF. The most stringent criteria is the 11.1 µF for the required load transient. Equation 21 gives a 1.58 A RMS current in the output capacitor. The capacitor should also be properly rated for the desired output voltage. Care must be taken when evaluating ceramic capacitors that derate under dc bias, aging and AC signal conditions. For example, larger form factor capacitors (in 1206 size) have self-resonant frequencies in the range of converter switching frequency. Self-resonance causes the effective capacitance to be significantly lower. The DC bias can also significantly reduce capacitance. Ceramic capacitors can lose as much as 50% of the capacitance whan operated at the rated voltage. Therefore, allow margin in selected capacitor voltage rating to ensure adequate capacitance at the required output voltage. For this example, three 4.7 µF, 50V 1210 X7R ceramic capacitors are used in parallel leading to a negligible ESR. Choosing 50 V capacitors instead of 35 V reduces the effects of DC bias and allows this example circuit to be rated for the maximum output voltage range of the TPS55340. Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 15 TPS55340 SLVSBD4 – MAY 2012 www.ti.com SELECTING THE INPUT CAPACITORS (C2, C7) At least 4.7µF of ceramic input capacitance is recommended. Additional input capacitance may be required to meet ripple and/or transient requirements. High quality ceramic, type X5R or X7R are recommended to minimize capacitance variations over temperature. The capacitor must also have an RMS current rating greater than the maximum RMS input current of the TPS55340 calculated with Equation 22. The input capacitor must also be rated greater than the maximum input voltage. The input voltage ripple can be calculated with Equation 23. DI ICIrms = L 12 (22) DIL Vripple = + DIL ´ RCIN I 4 ´ ¦ SW ´ CIN (23) In the design example, the input RMS current is calculated to be 191 mA. The chosen input capacitor is a 10 µF, 35V 1210 X7R with 3 mΩ ESR. Although one with a lower voltage rating can be used, a 35 V rated capacitor was chosen to limit the affects of dc bias and to allow it the circuit to be rated for the entire input range of the TPS55340. The input ripple is calculated to be 30 mV. An additional 0.1 µF, 50V 0603 X5R is located close to the VIN and GND pins for extra decoupling. SETTING OUTPUT VOLTAGE (R1, R2) To set the output voltage in either DCM or CCM, select the values of R1 and R2 according to the following equations. æ R1 ö VOUT = 1.229V ´ ç + 1÷ è R2 ø (24) æ V ö R1 = R2 ´ ç OUT - 1÷ è 1.229V ø (25) Considering the leakage current through the resistor divider and noise decoupling into FB pin, an optimum value for R2 is around 10 kΩ. The output voltage tolerance depends on the VFB accuracy and the tolerance of R1 and R2. In this example with a 24 V output using Equation 25, R1 is calculated to 185.3 kΩ. The nearest standard value of 187 kΩ is used. SETTING THE SOFT-START TIME (C7) Choose the appropriate capacitor to set soft-start time and avoid overshoot. Increasing the soft-start time reduces the overshoot during start-up. A 0.047 µF ceramic capacitor is used in this example. SELECTING THE SCHOTTKY DIODE (D1) The high switching frequency of the TPS55340 demands high-speed rectification for optimum efficiency. Ensure that the diode’s average and peak current rating exceed the average output current and peak inductor current. In addition, the diode’s reverse breakdown voltage must exceed the regulated output voltage. The diode must also be rated for the power dissipated which can be calculated with Equation 26. PD = VD × IOUT (26) In this conservative design example, the diode is chosen to be rated for the maximum output current of 2.13 A. During normal operation with 800 mA output current and assuming a Schottky diode drop of 0.5V, the diode must be capable of dissipating 400 mW. The recommended minimum ratings for this design are a 40 V, 3 A diode. However to improve the flexibility of this design, a Diodes Inc B540-13-F in an SMC package is used with voltage and current ratings of 40 V and 5A. COMPENSATING THE CONTROL LOOP (R3, C4, C5) The TPS55340 requires external compensation which allows the loop response to be optimized for each application. The COMP pin is the output of the internal error amplifier. An external resistor R3 and ceramic capacitor C4 are connected to the COMP pin to provide a pole and a zero, shown in the application circuit. This pole and zero, along with the inherent pole and zero of a boost converter, determine the closed loop frequency response. This is important for converter stability and transient response. Loop compensation should be designed for the minimum operating voltage. 16 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 The following equations summarize the loop equations for the TPS55340 configured as a CCM boost converter. They include the power stage output pole (ƒOUT) and the right-half-plane zero (ƒRHPZ) of a boost converter calculated with Equation 27 and Equation 28 respectively. When calculating ƒOUT it is important to include the derating of ceramic output capacitors. In the example with an estimated 10.2 µF capacitance, these frequencies are calculated to 980 kHz and 22.1 kHz respectively. The DC gain (A) of the power stage is calculated with Equation 27 and is 39.9 dB in this design. The compensation pole (ƒP) and zero (ƒZ) generated by R3, C4 and internal transconductance amplifier are calculated with Equation 30 and Equation 31 respectively. Most CCM boost converters will have a stable control loop if fZ is set slightly above ƒP through proper sizing of R3 and C4. A good starting point is C4 = 0.1 µF and R3 = 2kΩ. Increasing R3 or reducing C4 increases the closed loop bandwidth, and therefore improves the transient response. Adjusting R3 and C4 in opposite direction increases the phase and gain margin of the loop, which improves loop stability. It is generally recommended to limit the bandwidth of the loop to the lower of either 1/5 of the switching frequency ƒSW or 1/3 the RHPZ frequency, ƒRHPZ shown in Equation 28. The spreadsheet tool located in the TPS55340 product folder at www.ti.com can also be used to aid in compensation design. 2 ¦ OUT » 2p ´ ROUT ´ COUT (27) 2 ROUT æ VIN ö ´ç ÷ 2p ´ L è VOUT ø VIN 1.229 1 A= ´ Gea ´ 10MW ´ ´ ROUT ´ VOUT VOUT ´ RSENSE 2 ¦RHPZ » (28) (29) 1 ¦P = 2p ´ 10MW ´ C4 1 ¦Z = 2p ´ R3 ´ C4 ¦ ¦ co1 = SW 5 ¦ ¦ co2 = RHPZ 3 (30) (31) (32) (33) Where COUT is the equivalent output capacitor (COUT=C8+C9+C10) ROUT is the equivalent load resistance (VOUT/IOUT) Gea is the error amplifier transconductance located in the ELECTRICAL CHARACTERISTICS table RSENSE (15mΩ, typical) is the sense resistor in the current control loop ƒco1 and ƒco2 are possible bandwith. An additional capacitor from the COMP pin to GND (C5) can be used to place a high frequency pole in the control loop. This is not always necessary with ceramic output capacitors. If a non-ceramic output capacitor is used, there is an additional zero (fZESR) in the control loop which can be calculated with Equation 35. The value of C5 and the pole created by C5 can be calculated with Equation 36 and Equation 34 respectively. Finally if more phase margin is needed, an additional zero (fZFF) can be added by placing a capacitor (CFF) in parallel with the top feedback resistor R1. It is recommended to place the zero at the target cross-over frequency or higher. The feed forward capacitor also adds a pole at a higher frequency. The recommended value of CFF can be calculated with Equation 37. 1 ¦P2 = 2p ´ R3 ´ C5 (34) 1 ¦ ZESR » 2p ´ RESR ´ COUT (35) C5 = RESR ´ COUT R3 (36) Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 17 TPS55340 SLVSBD4 – MAY 2012 CFF = www.ti.com 1 2p ´ R1´ ¦ ZFF ´ VREF VOUT (37) where RESR is the ESR of the output capacitor. If a network measurement tool is available, the most accurate compensation design can be achieved following this procedure. The power stage frequency response is first measured using a network analyzer at the minimum 5 V input and maximum 800 mA load. This measurement is shown in Figure 16. In this design only one pole and one zero are used, so the maximum phase increase from the compensation will be 180 degrees. For a 60 degree phase margin, the power stage phase must be –120 degrees at its lowest point. Based on the target 6 kHz bandwidth, the measured power stage gain, KPS(fBW), is 24.84 dB and the phase is –110.3 degrees. 0 60 Gain Phase 40 −30 −60 0 −90 Gain (dB) 20 −20 −120 −40 −150 −60 100 1k 10k Frequency (Hz) −180 100k G016 Figure 16. Power Stage Gain and Phase of the Boost Converter R3 is then chosen to set the compensation gain to be the reciprocal of the power stage gain at the target bandwidth using Equation 38. C4 is then chosen to place a zero at 1/10 the target bandwidth with Equation 39. In this case R3 is calculated to be 2.56 kΩ, the nearest standard value 2.55 kΩ is used. C4 is calculated at 0.104 µF and the nearest standard value 0.100 µF is used. Although not necessary because this design uses all ceramic capacitors, a 100 pF capacitor is selected for C5 to add a high frequency pole at a frequency 100 times the target bandwidth. 1 R3 = KPS (¦BW ) æ ö 20 R1 ç Gea ´ ÷ ´ 10 çç ÷÷ (R1 + R2 ) è ø (38) 1 C4 = ¦ 2p ´ R3 ´ BW 10 (39) 18 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 CHARACTERISTICS OF THE BOOST CONVERTER 100 95 VOUT (ac coupled) = 500mV/div Efficiency (%) 90 85 80 75 IOUT = 200mA/div 70 65 60 VIN = 5 V VIN = 12 V 55 50 0 0.4 0.8 1.2 Output Current (A) 1.6 2 Time - 1ms/div G017 Figure 17. Efficiency vs Output Current IL = 1A/div Figure 18. Load Transient Response IL = 1A/div VOUT (ac coupled) = 100mV/div VOUT (ac coupled) = 10mV/div SW = 20V/div SW = 20V/div Time - 1ms/div Time - 1ms/div Figure 19. CCM PWM Operation Figure 20. DCM PWM operation VIN = 1V/div IL = 200mA/div EN = 2V/div VOUT (ac coupled) = 20mV/div SW = 10V/div SW = 20V/div VOUT = 10V/div Time - 50ms/div Time - 500ms/div Figure 21. Pulse Skipping Figure 22. Start Up Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 19 TPS55340 SLVSBD4 – MAY 2012 www.ti.com 180 60 40 120 20 60 0 0 −60 −20 −40 −60 100 Phase (°) Gain (dB) IOUT = 800 mA 5 V Gain 5 V Phase 15 V Gain 15 V Phase −120 1k 10k Frequency (Hz) −180 100k G023 Figure 23. Closed Loop Gain and Phase of the Boost Converter 20 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com SLVSBD4 – MAY 2012 DESIGN GUIDE-STEP-BY-STEP DESIGN PROCEDURE OF SEPIC CONVERTER Lb TP4 VIN La VIN 6-18V 12V, 1A 1 TP5 49.9 TP2 VIN 86.6k SYNC SYNC 1 TP3 Not Populated Figure 24. SEPIC Converter Application Schematic The following parameters are used for a SEPIC converter design. These calculations are performed only for CCM operation, and the use of a coupled inductor is assumed. Table 2. Key Parameters of SEPIC Converter Example PARAMETER VALUE Output Voltage 12 V Input Voltage 6 V to 18 V, 12 V nominal Maximum Output Current 1A Transient Response 50% load step (ΔVOUT = 4%) 480 mV Output Voltage Ripple (0.5% of VOUT) 60 mV SELECTING THE SWITCHING FREQUENCY (R4) A 500 kHz switching frequency (ƒSW) is selected for this design. Using Equation 1, R4 is calculated and the nearest standard value 95.3 kΩ is used. DUTY CYCLE The duty cycle of a SEPIC converter is calculated with Equation 40. With the 6 V minimum input the duty cycle is 68% and with the 18 V maximum input voltage the duty cycle is 41%. VOUT + VD D= VOUT + VD + VIN (40) Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 21 TPS55340 SLVSBD4 – MAY 2012 www.ti.com SELECTING THE INDUCTOR (L1) With an estimated 85% efficiency, the input current is calculated with Equation 9 to be 2.35 A. With KIND of 0.3 and the maximum 18 V input the minimum inductance is calculated to be 10.5µH using Equation 41. The nearest standard value of 12µH is used. As mentioned previously, this equation assumes a coupled inductor is used. VIN max´ Dmin L³ 2 ´ ¦ SW ´ IINDC ´ KIND (41) The inductor ripple current is recalculated to be 615mA with Equation 42. The peak current is calculated to 3.69 A. The typical current limit is used as the saturation rating for the inductor used. The RMS current for La is approximately the average input current 2.35 A. The RMS current for Lb is approximately the output current of 1 A. For this design a CoilCraft MSD1260-123 is used with 6.86 A saturation, 74 mΩ DCR and 3.12 A RMS current rating for one winding. V max´ Dmin DIL = IN 2 ´ ¦ SW ´ L (42) DI ö æ DI ö æ IL peak = ILa peak + ILb peak = ç IINDC + L ÷ + ç IOUT + L ÷ 2 ø è 2 ø è (43) CALCULATING THE MAXIMUM OUTPUT CURRENT The maximum output current with the minimum input voltage 6 V, chosen inductance 12µH, 5.25 A minimum current limit and estimated 85% efficiency is calculated to be 1.47 A using Equation 44. (ILIM - DIL ) (ILIM - IINDC ´ KIND ) IOUT max = = æ ö æ ö VOUT VOUT + 1÷ ç + 1÷ ç è VIN min´ hEST ø è VIN min´ hEST ø (44) SELECTING THE OUTPUT CAPACITOR (C8-C10) To meet the 60 mV ripple specification, the minimum output capacitance is calculated to be 22.5µF with Equation 45. This design uses ceramic output capacitors and the effects of ESR are ignored. To meet the transient response of 500mA with less than 480mV voltage change and a 7kHz control loop bandwidth, the minimum output capacitance is calculated to be 23.7µF using Equation 46. The RMS current is calculated with Equation 22 to be 1.44 A. The output capacitors used in this design is 3 x 22 µF, 25 V X7R 1210 ceramic capacitors. With voltage derating, the effective total output capacitance is estimated to be 30.4 µF. Dmax ´ IOUT COUT ³ ¦ SW ´ VRIPPLE (45) COUT ³ DITRAN 2p ´ ¦BW ´ DVTRAN (46) SELECTING THE SERIES CAPACITOR (C6) The series capacitor is chosen to limit the ripple current to 5% of the maximum input voltage. Using Equation 47 the minimum capacitance is 1.5 µF. Using Equation 48 the RMS current is calculated to be 1.63A. A 2.2 µF ceramic capacitor in a 1206 package is selected. IOUT ´ Dmax CP ³ 0.05 ´ VIN max ´ ƒSW (47) ICPrms = IINDC ´ (1 - Dmax) Dmax (48) SELECTING THE INPUT CAPACITOR (C2, C7) Based on the minimum 4.7µF ceramic recommended for the TPS55340, a 10µF X7R input capacitor is used with an additional 0.1µF placed close to the VIN and GND pins. With an estimated 6 µF capacitance after voltage derating, the input ripple voltage is calculated to be 39.9mV using Equation 49. The RMS current of the input capacitance is calculated to be 0.177 A with Equation 50. 22 Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 www.ti.com VIripple = ICIrms = SLVSBD4 – MAY 2012 DIL 4 ´ ¦ SW ´ CIN (49) DIL 12 (50) SELECTING THE SCHOTTKY DIODE (D1) The selected diode must have a minimum breakdown voltage (VBR) calculated with Equation 51 which is 30.5 V in this design. The average current rating is recommended to be greater than the maximum output current. With the maximum 18 V input, average current is calculated to be 2.6 A using Equation 17. The package must also be capable of handling the power dissipation. With an estimated 0.5V forward voltage, power dissipation is calculated with Equation 26 to be 500mW. Diodes Inc B340B is chosen with a 40 V, 3 A rating in a SMB package. VBR = VO + VINmax + VF (51) SETTING THE OUTPUT VOLTAGE (R1, R2) With R2 fixed at 10 kΩ using Equation 25 the nearest standard value of 86.6 kΩ is chosen for R1. SETTING THE SOFT-START TIME (C3) The recommended 0.047µF soft-start capacitor is used. MOSFET RATING CONSIDERATIONS In a SEPIC converter the MOSFET must be rated to handle the sum of the input and output voltages. In this design with the maximum input voltage of 18 V and output voltage of 12 V the FET will see approximately 30 V. A 10% tolerance is recommended to account for any ringing. The 40V rating of the TPS55340 power MOSFET comfortably satisfies this requirement. COMPENSATING THE CONTROL LOOP (R3, C4) This design was compensated by measuring the frequency response of the power stage at the lowest input voltage of 6 V and choosing the components for the desired bandwidth. The lowest right half plane zero (ƒRHPZ) is calculated to be 36.7kHz with Equation 52. Using the recommendation to limit the bandwidth to 1/3 of ƒRHPZ the maximum recommended is 12.2kHz. VOUT IOUT ¦RHPZ = 2 æ D ö 2´ p´L ´ç ÷ ç (1 - D ) ÷ è ø (52) This design also uses only one pole and one zero. In order to achieve approximately 60 degrees of phase margin, the power stage phase must be no lower than approximately –120 degrees at the desired bandwidth. To ensure a stable design, R3 was initially set to 1kΩ and C4 was 1µF. Figure 25 shows the measurement of the power stage At 7kHz the power stage has a gain of 19.52 dB and phase of –118.1 degrees. Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 23 TPS55340 SLVSBD4 – MAY 2012 www.ti.com 180 60 6 V Input Gain 6 V Input Phase 40 120 0 0 Phase (°) 60 Gain (dB) 20 −20 −60 −40 −120 −60 100 1k −180 100k 10k Frequency (Hz) G025 Figure 25. SEPIC Power Stage Gain and Phase As there are no changes in the transconductance amplifier, the equations used to calculate the external compensation components in a boost design can be used in the SEPIC design. Using the maximum Gea from the electrical specification of 440 µmho, Equation 38 calculates the nearest standard value of R3 to be 2.37 kΩ. Using Equation 39, C4 is calculated to the nearest standard value of 0.1 µF. CHARACTERISTICS OF THE SEPIC CONVERTER 100 95 IOUT = 500mA/div Efficiency (%) 90 85 VOUT (ac coupled) = 200mV/div 80 75 70 65 VIN = 6 V VIN = 12 V VIN = 18 V 60 55 50 0 0.2 0.4 0.6 0.8 Output Current (A) 1 1.2 Time - 500ms/div G026 Figure 26. Efficiency vs Output Current Figure 27. Load Transient Response VIN = 2V/div SW = 10V/div EN = 2V/div ILb = 1A/div SW = 20V/div ILa = 1A/div VOUT = 5V/div VOUT (ac coupled) = 50mV/div Time - 2ms/div Time - 1ms/div Figure 28. CCM PWM Operation 24 Figure 29. Output Voltage Soft-start Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 TPS55340 SLVSBD4 – MAY 2012 60 180 40 120 20 60 0 0 −60 −20 −40 −60 100 Phase (°) Gain (dB) www.ti.com 6 V Gain 6 V Phase 18 V Gain 18 V Phase −120 1k 10k Frequency (Hz) −180 100k G030 Figure 30. Closed Loop Gain and Phase of the SEPIC Converter Submit Documentation Feedback Copyright © 2012, Texas Instruments Incorporated Product Folder Link(s) :TPS55340 25 PACKAGE OPTION ADDENDUM www.ti.com 30-May-2012 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp (3) TPS55340RTER ACTIVE WQFN RTE 16 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS55340RTET ACTIVE WQFN RTE 16 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 13-Jun-2012 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant TPS55340RTER WQFN RTE 16 3000 330.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2 TPS55340RTET WQFN RTE 16 250 180.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 13-Jun-2012 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) TPS55340RTER WQFN RTE 16 3000 346.0 346.0 29.0 TPS55340RTET WQFN RTE 16 250 210.0 185.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. 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