ALD1000 ® ALD 100 0 Precision Programmable CURRENT/VOLTAGE TRANSMITTER FEATURES APPLICATIONS ● SWITCHABLE OUTPUT ±10V OR 4-20mA ● PROGRAMMABLE CONTROLLERS ● DRIVES 1000Ω || 1µF AT 20mA ● VOLTAGE AND CURRENT SENSE ● STANDARDIZED OUTPUTS FOR TERMINATION PANELS ● GROUND NOISE SUPPRESSION ● ERROR DETECTION FLAG ● OUTPUT DISABLE ● INDUSTRIAL PROCESS CONTROL ● PROGRAMMABLE CURRENT SOURCE ● MOTOR CONTROL SYSTEMS ● ACCURACY: 0.05% max ● WIDE SUPPLY RANGE: ±11V TO +24/–15V ● PC AND VME BASED INSTRUMENTATION ● CONDITIONER FOR STANDARD SENSOR OUTPUTS ● TEST EQUIPMENT PIN DRIVER DESCRIPTION Voltage Error Indication Phase Compensation This product is a monolithic programmable voltage-tocurrent or voltage-to-voltage analog line driver circuit. It can convert a ±10V input into either an output voltage or current with remote sensing. It provides drive for external transistors to boost output current to greater than ±25mA levels. Current and voltage sensing can be performed simultaneously. Current sensing is achieved through a single external sense resistor. Voltage sensing is performed directly across the load. The logic inputs provide for both output disable and switching between constant current or constant voltage output functions. An open collector output provides an error flag for open circuit loads. The output disable function allows full control of the output even during power-on and power-off sequencing. The instrumentation amplifiers are designed to insure that load noise is not circulated within the control loop. Phase V Sense Output Disable External PNP Drive Input Select 100Ω Internal Drive External NPN Drive VIN2 Open-Loop Gain Control VIN1 IA Gain Control IA1 IA2 IAs Provide Closed-Loop Gain Control Input Over-Voltage Protection International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111 Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 © 1996 Burr-Brown Corporation PDS-1292A Printed in U.S.A. October, 1996 SPECIFICATIONS At +VS = 24V, –VS = 15V, TAMB = 25°C, and 2N2222, 2N2907 external transistors, unless otherwise noted. ALD1000U PARAMETER CONDITIONS MIN SWOP INPUTS Linear Range Min Linear Range Max Input Bias Current Internal Drive Transistors 5mA Load 10 XTR OUTPUT Positive Overvoltage Sense Negative Overvoltage Sense Positive Overcurrent Sense Negative Overcurrent Sense Internal Drive Transistors TYP MAX UNITS –10 50 V V pA 19.5 –10.5 +25 –15 V V mA mA TRANSMITTER LOGIC INPUTS Logic Low Logic High 0.8 4.0 LOGIC OUTPUTS Logic High Logic Low 5V Logic Supply with 10k pull-up resistor OUTPUT—VOLTAGE MODE (Gain = 1 unless otherwise specified) Span Error Span Drift Linear Range Min 0.1% of FS Linear Range Max 0.1% of FS Output Current Min Internal Drive Transistors Output Current Max Internal Drive Transistors Short-Circuit Current Internal Drive Transistors Short-Circuit Current Internal Drive Transistors Non-Linearity Initial Offset Voltage—RTI Offset Voltage vs Temperature OUTPUT—CURRENT MODE (Gain = 5 with 50Ω shunt resistor unless otherwise specified) Span Error Span Drift Gain = 1(1) Output Current Min Internal Drive Transistors (2) Output Current Max Internal Drive Transistors (2) Compliance Min Compliance Max Offset Current Min Offset Current Min 2.6 4.0 0.8 0.5 50 1 V V V V % ppm/°C of FS –10 10 –5 5 25 –15 0.005 2 20 0.05 5 50 –5 5 –10 15 –25 25 mA mA mA mA % mV µV/°C % ppm/°C of FS mA mA V V µA µA INSTRUMENTATION AMPLIFIERS RLOAD = 10k IA INPUTS Linear Input Voltage Min Linear Input Voltage Max Common-Mode Input Voltage Min Common-Mode Input Voltage Max Input Bias Current Initial Offset Voltage CMRR –10 20 VIN = 0 VIN = 0 20 G=1 G = 10 –1 80 100 IA OUTPUTS (with 10k Load) Output Voltage Max Output Voltage Min + Short Circuit Current – Short Circuit Current 1 100 20 –10 5 –12 GAIN EQUATION (gain = 1+50k/RG) Gain Error, G = 1 G=5 G = 100 Non-Linearity, G = 1 G=5 G = 100 0.3 0.6 0.8 0.004 0.008 0.02 ® ALD1000 –10 2 V V V V nA mV V V mA mA %±FS %±FS %±FS %±FS %±FS %±FS SPECIFICATIONS (CONT) At +VS = 24V, –VS = 15V, TAMB = 25°C, and 2N2222, 2N2907 external transistors, unless otherwise noted. ALD1000U PARAMETER CONDITIONS FREQUENCY RESPONSE G=1 G=5 G = 100 Slew Rate MIN VO = ±10V, G = 10 SETTLING TIME, 0.01% G=1 G=5 G = 100 POWER SUPPLY Quiescent Current Internal Drive Transistors TEMPERATURE RANGE Operating Storage –40 –65 TYP MAX UNITS 700 400 50 4 kHz kHz kHz V/µS 20 20 30 µS µS µS 5 mA +85 +150 °C °C NOTES: (1) Gain drift depends on tempco of 50K factor on gain equation when gain is greater than 1. (2) External Drive capacity varies with configuration. See Application Note. ABSOLUTE MAXIMUM RATINGS ELECTROSTATIC DISCHARGE SENSITIVITY Supply Voltage (±VS) .............................................................. +25V, –18V IA Inputs ............................................................................................ ±40V SWOP Inputs ....................................................................................... ±VS Logic Inputs .................................................................... +VS, –VS + 0.5V Junction Temperature ...................................................................... 150°C Storage Temperature ..................................................... –65°C to +150°C Lead Temperature (soldering, 10s) ............................................... +300°C Output Short-to-Ground at 25°C ............................................. Continuous This integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER(1) ALD1000U 28-Pin SOIC 217 NOTE: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 ALD1000 PIN CONFIGURATION Top View 28-Lead SOIC XTR GND 1 28 VERR Disable 2 27 Bias Select 3 26 C CC2 4 25 XN CC1 5 24 XP P1 6 23 E N1 7 22 P2 ACOM 8 21 N2 +VS 9 20 –VS VOUT1 10 19 VOUT2 VINP1 11 18 VINP2 RGB1 12 17 RGB2 RGA1 13 16 RGA2 VINN1 14 15 VINN2 TA IA IA PIN ASSIGNMENTS PIN # NAME 1 XTR GND DESCRIPTION 2 Disable A 5V signal puts the internal drive in a high impedance state and limits the external drive capacity. 3 Select Selects the SWOP amp input. A 5V signal selects inputs N1 and P1. 4 CC2 CC1 and CC2 are for the external compensation capacitor. 5 CC1 CC1 and CC2 are for the external compensation capacitor. 6 P1 Non-inverting input to the XTR SWOP amp 1. 7 N1 Inverting input to the XTR SWOP amp 1. 8 ACOM Signal ground for the instrumentation amplifiers. 9 +VS Positive power supply voltage. 10 VOUT1 Output of the instrumentation amplifier 1. 11 VINP1 Non-inverting input to instrumentation amplifier 1. 12 RGB1 Gain set resistor for instrumentation amplifier 1. 13 RGA1 Gain set resistor for instrumentation amplifier 1. 14 VINN1 Inverting input of instrumentation amplifier 1. 15 VINN2 Inverting input of instrumentation amplifier 2. 16 RGA2 Gain set resistor for instrumentation amplifier 2. 17 RGB2 Gain set resistor for instrumentation amplifier 2. 18 VINP2 Non-inverting input to instrumentation amplifier 2. 19 VOUT2 Output of the instrumentation amplifier 2. 20 –VS Negative power supply voltage. 21 N2 Inverting input to the XTR SWOP amp 2. 22 P2 Non-inverting input to the XTR SWOP amp 2. 23 E Inverting input (emitter) of the output transconductance amplifier. 24 XP Base drive for an external, PNP, driver transistor (optional). 25 XN Base drive for an external, NPN, driver transistor (optional). 26 C Output (collector) of the output transconductance amplifier. 27 Bias Open collector output indicating an internal overcurrent condition. 28 VERR Open collector output indicating an overvoltage condition. Power ground pin. ® ALD1000 4 TYPICAL PERFORMANCE CURVES At TA = +25°C; +VS = 24V, –VS = –15V, unless otherwise noted. STEP RESPONSE IN VOLTAGE FEEDBACK STEP RESPONSE IN THE CURRENT FEEDBACK MODE 5 8 4 6 3 4 Output (V) Output (V) 2 1 0 Figure 8, RI = 400Ω, CI = 100nf No external comp capacitor –1 2 Figure 8, RI = 400Ω, CI = 100nf No External Comp Capacitor 0 –2 –2 –3 –4 –4 0 100 200 300 400 500 600 0 100 200 Time (µs) 300 400 500 600 Time (µs) STEP RESPONSE IN VOLTAGE FEEDBACK COMPENSATION CAPACITOR vs LOAD CAPACITANCE 5 1.0E-08 4 Compensation Capacitor 3 Output (V) 2 1 0 Figure 8, RI = 400Ω, CI = 100nf 420pf Compensation Capacitor –1 –2 1.0E-09 1.0E-10 Gain = 1 –3 –4 1.0E-11 0 100 200 300 400 500 600 1.0E-12 1.0E-11 1.0E-10 1.0E-09 1.0E-08 1.0E-07 1.0E-06 Time (µs) Load Capacitance STEP RESPONSE LOAD IN THE VOLTAGE FEEDBACK MODE 5 5 4 4 3 3 2 2 Output (V) Output (V) STEP RESPONSE IN CURRENT FEEDBACK MODE 1 0 –1 1 0 –1 Figure 8, RI = 400Ω, CI = 1µf No Comp Capacitor –2 Figure 8, RI = 400Ω, CI = 1µf Comp = 2.2nf –2 –3 –3 –4 –4 0 500 1000 1500 2000 2500 3000 3500 4000 4500 0 Time (µs) 100 200 300 400 500 600 700 800 900 Time (µs) ® 5 ALD1000 TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C; +VS = +24V, –VS = –15V, unless otherwise noted. STEP RESPONSE IN VOLTAGE FEEDBACK MODE STEP RESPONSE IN CURRENT FEEDBACK MODE 15 20 15 10 Output (V) Output (V) 10 5 0 5 0 –5 Figure 7, RI = 220Ω, CI = 1µf No Compensation Capacitor –5 Figure 7, RI = 400Ω, CI = 1µf –10 –15 –10 0 0 1000 2000 3000 4000 5000 6000 7000 8000 9000 10000 200 400 600 800 1000 1200 1400 1600 1800 Time (µs) Time (µs) STEP RESPONSE IN VOLTAGE FEEDBACK MODE 25 10 20 Output (V) Output (V) STEP RESPONSE IN VOLTAGE FEEDBACK MODE 15 5 0 Figure 7, RI = 220Ω, CI = 1µf 2.2nf Compensation Capacitor –5 15 10 Figure 9 5 –10 0 0 200 400 600 800 1000 1200 1400 1600 1800 0 50 100 150 200 250 300 350 Time (µs) Time (µs) STEP RESPONSE WITH 2kΩ LOAD CURRENT FEEDBACK STEP RESPONSE WITH 2kΩ LOAD VOLTAGE FEEDBACK 20 15 15 10 400 450 Output (V) Output (V) 10 5 Figure 8, CI = 15pf No Comp Capacitor, RE = 5.5kΩ 0 5 0 Figure 8, CI = 15pf No comp capacitor, RE = 5.5kΩ –5 –5 –10 –10 0 20 40 60 80 0 100 120 140 160 180 200 ® ALD1000 20 40 60 80 100 120 140 160 180 200 Time (µs) Time (µs) 6 TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C; +VS = +24V, –VS = –15V, unless otherwise noted. GAIN vs FREQUENCY COMMON-MODE REJECTION vs FREQUENCY 1000 120 Commom-Mode Rejection (dB) Gain = 100 Instrumentation Amplifiers Gain = 100 100 100 80 Gain Gain = 5 60 Gain = 5 Instrumentation Amplifiers with ±10V input 10 40 1 Gain = 1 20 Gain = 1 0.1 0 1 10 100 1k 10k 100k 1M 1 10M 10 INPUT BIAS CURRENT vs INPUT VOLTAGE 1000 10000 INPUT COMMON-MODE RANGE vs OUTPUT VOLTAGE 10 25 G = 10 8 G = 10 20 4 Gain = 100 Gain = 1 2 Common-Mode Voltage 6 Bias Current (mA) 100 Frequency (kHz) Frequency (Hz) 0 –2 Gain = 1 –4 –6 G=1 G=1 10 5 0 –5 All Gains –10 Gain = 100 –8 15 –10 –15 –40 –30 –20 –10 0 10 20 30 40 –15 50 Input Voltage (V) –10 –5 0 5 10 15 20 25 Output Voltage BASIC OPERATION SWOP amp has two pairs of inputs to provide flexibility of application. The SELECT logic input can switch between two input and feedback signals. Take care, however, to insure that the loop remains stable if switching between current and voltage feedback. ALD1000 FUNCTIONAL BLOCKS The typical ALD1000 control loop comprises three primary functional blocks (see Figure 1): the current transmitter (XTR), the load, and the instrumentation amplifier (IA). The XTR can be further viewed as divided into the switchable input operational amplifier (SWOP amp), and the voltage to current, transconductance amplifier (TA). Each of these blocks plays a role in the dynamic performance of the control loop, particularly in terms of loop stability with reactive loads. The ALD1000 handles a wide range of load conditions in either a voltage or current feedback application. The frequency characteristics of the potential load conditions vary widely. To accommodate these varying frequency characteristics the XTR includes a compensation network. It consists of a simple resistor divider network which forms a single pole, high pass, RC filter when a compensation capacitor is connected externally. The transconductance amplifier converts the output voltage of the SWOP amp into an output current to drive the load. Whether used in a current feedback or a voltage feedback loop, the ALD1000 transmitter should be viewed as a source of current not voltage. In a voltage loop, the output current is converted to a feedback voltage by the load. In a current loop the output current is converted to a feedback voltage by the shunt resistor. The external, XTR gain resistor, tied to E (Pin 23, Figure 1), sets the voltage to current ratio. THE CURRENT TRANSMITTER (XTR) The XTR produces the forward gain necessary for error amplification. It also controls the frequency response which must be adjusted to balance the trade-off between step response and stability when driving reactive loads. Within the XTR the SWOP amp serves as the input stage. It amplifies the error between the input and output signals to produce a precise signal to the TA to drive the load. The ® 7 ALD1000 +5V 10kΩ VERR XTR GND 1 10kΩ 28 27 +24V V Sense 9 4 Input Select Select Input Signal 5 Freq. Comp. 22 24 XP 21 26 C XTR 3 20Ω 2N2907 Local Shunt 100Ω 100Ω SWOP 25 6 7 TA 1kΩ 2N2222 20Ω 20 23 Loop Gain 2 XN Disable 14 VINN1 –15V 40kΩ 40kΩ 25kΩ 40kΩ 25kΩ 13 50kΩ 10 Monitor Current 12 VOUT1 40kΩ 8 11 VINP1 18 VINP2 IAs ACOM 40kΩ 25kΩ 40kΩ 19 Monitor Voltage 40kΩ 17 25kΩ 16 VOUT2 Remote Load RGB2 RGA2 40kΩ 15 VOUT VIN IOUT 0-2 0-10mA 1 0 0-10 0-50mA 1 0 DISABLE 0.8-4 4-20mA 1 0 2-10 10-50mA 1 0 0-5 0-5V 0 0 0-10 0-10V 0 0 ±5 ±5V 0 0 ±10 ±10V 0 0 OFF X 1 X OFF FIGURE 1. ® ALD1000 SELECT VINN2 8 400Ω THE INSTRUMENTATION AMPLIFIERS (IA) The ALD1000 includes two, general purpose, uncommitted, 3-op amp, instrumentation amplifiers (see Figure 1). The two IAs are connected to the common power supply and operate at full supply voltage. They share the same analog common reference. Otherwise they are configured independently for maximum flexibility. The instrumentation amplifier senses the feedback signal, reduces any common-mode component, and scales it to the level required. A more comprehensive discussion of the instrumentation amplifiers follows in a later section. VOLTAGE ERROR INDICATION The VERR error signal at pin 28 triggers when the voltage at C (pin 26) exceeds the Positive Overvoltage Sense or the Negative Overvoltage Sense (see SPECIFICATIONS) internal thresholds. When external transistors are used without connecting them to C, as shown in Figure 7, the load voltage cannot be detected. The logic signal will generally trigger to an error state (low). However, consider it indeterminate under these conditions. ALD1000 LOGIC SETTING THE GAIN INSTRUMENTATION AMPLIFIERS The IA gain is set by connecting a single, external resistor, Rg in Figure 3, between the gain set pins. The logic inputs used for the SWOP amp select and the XTR disable functions are simple, differential pair comparators as shown in Figure 2. The logic outputs are open collector NPN transistors with their emitters at ground (pin 1). G = 1+ INPUT PROTECTION The inputs of the ALD1000 instrumentation amplifiers are individually protected against over-voltage. Internal circuitry on each input provides low series impedance under normal signal conditions. If the input is overloaded, the protection circuitry limits the input current to a safe value of approximately 1ma to 5ma. Refer to the typical performance curve “Input Bias Current vs Input Voltage.” The inputs are protected even if the power supplies are disconnected or turned off. +VS Internal Logic Threshold Logic Input 50kΩ RG + Internal Logic IA INPUT OVERLOAD AND INPUT COMMON-MODE RANGE The linear voltage range of the input circuitry of the ALD1000 instrumentation amplifiers is from approximately 0.6V below the positive supply to 1V above the negative supply. However, the output swing of the input amplifiers –VS FIGURE 2. Simplified Diagram of the Internal ALD1000 Logic Input Circuitry. VIN– Over-Voltage Protection 40kΩ 40kΩ 40kΩ A3 A1 25kΩ RG VIN+ 25kΩ Over-Voltage Protection VOUT A2 40kΩ FIGURE 3. Simplified Schematic of the Instrumentation Amplifiers. Resistor RG Controls the Gain. ® 9 ALD1000 limit this range when a differential input voltage causes the output voltage to increase. Thus, the linear common-mode range relates to the output voltage of the complete amplifier. This behavior also depends in supply voltage—see performance curve “Input Common-Mode Range vs Output Voltage.” insure adequate transient response and loop stability: loop gain and phase. Together loop gain and phase set the phase margin which defines dynamic performance. Loop gain is the product of the forward voltage to current ratio, the load impedance, and the IA gain. The input error voltage is converted to an output current. The output current is converted to a feedback voltage by the load impedance. The feedback voltage is gained up by the feedback IA. All three blocks affect loop stability. The XTR gain resistor, which is connected to the E pin of the ALD1000, adjusts the voltage to current relationship. Increasing this resistor decreases loop gain. This, in turn, increases phase margin and slows step response. This resistor will typically be between 250Ω and 2500Ω. In a voltage feedback loop the frequency at which the loop gain starts to roll off decreases with increasing capacitance. It is necessary to compensate for the loss of bandwidth caused by load capacitance. The compensation network provides this capability. Typical performance curve “Compensation Capacitor vs Load Capacitance” illustrates typical compensation capacitor values for load capacitance varying from 1pf to 1µf. Exact capacitor values will vary with the load resistance, the XTR gain resistor value, IA gain, and variability of the open loop gain of the ALD1000 SWOP amp. This curve provides a starting point for empirical selection of the compensation capacitor value. The effect described above is much less significant with a current feedback loop since the shunt resistor’s capacitance can be easily controlled. The current feedback loop will be more robust when load conditions are unknown or varying. The combination of a significant differential signal and a high common-mode voltage as occurs in the current feedback configuration reduces the common-mode range. Exceeding the common-mode range results in a reduced IA output voltage. When this occurs the feedback loop can no longer balance. The forward gain of the ALD1000 amplifies this false error signal, the output voltage tries to increase, and this holds the IA in an overloaded condition. The ALD1000 applies two defenses against this problem. First, there is a 100Ω resistor in series with the transmitter output. This resistor, which primarily provides protection from over-voltage damage to the output terminal, acts to limit the output swing under high current conditions. Second, the ALD1000’s error detection circuitry signals when the transmitter output voltage exceeds rating. This serves to detect a potential lock condition. Limiting the transmitter’s output swing to within the instrumentation amplifier’s input range allows the loop to recover without reducing the input signal should a transient voltage level exceed the common-mode input range. However, the common-mode range of the instrumentation amplifiers varies with application specific factors. Lock-up can occur. The application designer must provide defenses against this condition where it is warranted. USING THE INSTRUMENTATION AMPLIFIERS WITH A FLOATING SIGNAL SOURCE The input impedance of the ALD1000 instrumentation amplifiers are very high—about 106Ω. Within a feedback loop, as shown in the examples, this characteristic acts to minimize errors caused by loading of the feedback signal. However, if used as an amplifier for a thermocouple, microphone, or other isolated signal source a path is needed for the input bias current. This current is nominally about 100nA. Without a return path the inputs will float to a potential that exceeds the common-mode range of the amplifier. See Figure 10. LOOP STABILITY AND THE INSTRUMENTATION AMPLIFIERS The frequency characteristics and gain of the instrumentation amplifiers affect loop stability when they are used in a feedback loop. There are two main contributions. First, the IA gain directly multiplies loop gain. As a result high IA gains reduce phase margin. Second, when the input exceeds the IA range the IA output can no longer provide the necessary feedback. This can result in a lock condition. Both of these situations are discussed further below. LOOP STABILITY The ALD1000 is designed for use in a feedback loop. When one of the instrumentation amplifiers is used as the feedback amplifier its gain directly contributes to loop gain. The loop can become unstable if the loop gain is too large. Conversely, it may be possible to stabilize a difficult loop by reducing the gain of the IA. Refer to Figure 4. In this circuit the ALD1000 is configured in a current loop with a 50Ω shunt resistor. A 20ma full scale current through the 50Ω shunt results in a 1V feedback signal. The IA must remove the common-mode level from the shunt voltage and scale the resulting differential signal up to the input signal level. LOOP GAIN AND THE INSTRUMENTATION AMPLIFIERS The stability of a closed loop system such as the intended application of the ALD1000 requires adequate phase margin. In contrast, excessive phase margin will reduce the circuit’s transient response to fast changing signals. It is the intent of this section to give an insight into how the ALD1000 circuits blocks affect dynamic performance. Selection of the loop architecture and compensation can then be done empirically. LOOP STABILITY AND THE XTR There are two critical parameters that must be controlled to ® ALD1000 10 USING THE ALD1000 WITH EXTERNAL DRIVE TRANSISTORS + 5V FS Signal Power dissipated by the internal driver stage affects the built-in instrumentation amplifiers compromising their accuracy. External transistors reduce the internal power dissapation. The external transistors are configured as current sources. A PNP transistor delivers positive current. An NPN supplies the negative drive. Either or both can be used. For example a 4ma to 20ma current loop may only require a PNP transistor since negative current drive is not required. See Figure 9. 50Ω C XTR E – Load 12.5kΩ IA ALD1000 Degeneration resistors are required (refer to R1 and R2 in Figure 6). The value of the degeneration resistors will affect stability, load sharing with the internal driver devices, and the current limit value. These issues are covered in more detail below. FIGURE 4. Simplified Block Diagram of a 20ma Current Loop. The IA is in a Gain of 5 to Match the 1V Full Scale Shunt Signal to the 5V Full Scale Input Signal. EXTERNAL TRANSISTORS AND THE CURRENT LIMIT VALUE The ALD1000 contains circuitry to prevent damage to the internal components due to excess current. When using the internal driver stage by itself, current to the load is limited to about 20ma at room temperature. When using external drivers, the current limit depends, approximately, on the load sharing ratio between the internal and external transistors. Figure 6 illustrates the circuit relationship between the current limit circuitry and external drive transistors. Here the input signal affects loop stability. A 10V full scale input signal would require an IA gain of 10. A lower input signal, 5V as shown in Figure 4, allows the IA gain to be reduced to 5. This results in a lower loop gain and increased phase margin. Note that it is possible to reduce the IA gain to less than 1 by using a voltage divider at the IA output. VOLTAGE FEEDBACK AND THE INSTRUMENTATION AMPLIFIERS The instrumentation amplifiers can be used for remote sensing in a voltage feedback loop as illustrated in Figure 5. Here the instrumentation amplifier tends to reject small ground potential differences between the source and load. The voltage loop, however, is more sensitive to reactive load impedance than the current loop. The ALD1000 emitter resistor and compensation capacitor need to be selected for the specific load conditions. Voltage feedback may not be appropriate for variable load conditions. 10V Signal + THE DEGENERATION RESISTORS The degeneration resistors, R1 and R2 in Figure 6, control the load sharing between the internal and external transistors. Choose the resistor values by measuring load current, current through the external transistor, and calculating the current being supplied by the internal drive. LOOP STABILITY AND THE DEGENERATION RESISTORS Loop stability depends on loop gain. Because the degeneration resistors affect the voltage to current ratio of the loop the value of these resistors also affect loop gain and thus stability. Smaller resistor values will increase loop gain. It may be necessary to compensate for this by adjusting the value of the XTR gain resistor connected to the E Pin, R4 in Figure 6. C XTR E – RCOMP CCOMP EXTERNAL TRANSISTORS AND OUTPUT VOLTAGE SWING The output voltage swing must be limited within the input range of the instrumentation amplifiers. The 100Ω resistor shown in Figure 6 limits output swing under high current conditions. Resistor R3 performs this function with external transistors. R3 must be sized to limit output swing, at the expected full load, within the input range of the instrumentation amplifiers. Refer to the section on the instrumentation amplifiers for further information. Load ALD1000 FIGURE 5. Simplified Block Diagram of the ALD1000 in Voltage Feedback. ® 11 ALD1000 +VS R1 To Positive Current Limit XP QP 24 100Ω C 26 To Load R3 Signal E 23 R4 To Negative Current Limit XN QN 25 R2 –VS ALD1000 OUTPUT STAGE FIGURE 6. Simplified Schematic Showing the Use of External Drive Transistors. R1 and R2 Provide Degeneration that Affects the Current Limit and Loop Stability. R4 Controls the Transconductance Amplifier Gain Affecting Loop Stability and Transient Response. See the Text. ® ALD1000 12 COMP 5V = I 0V = V 20Ω VIN 2N2907 50Ω 2N2222 20Ω 1kΩ 12.5kΩ RI CI ALD1000 FIGURE 7. Using External Transistors Without Internal Drive. Note that an Overvoltage Condition Can Not Be Detected. COMP 5V = I 0V = V VIN 50Ω Re 2.5kΩ 12.5kΩ R1 C1 ALD1000 FIGURE 8.Using Internal Drive Transistors. ® 13 ALD1000 100pf 10Ω VIN 2N2907 3.7kΩ Disable 220Ω 1µf ALD1000 FIGURE 9. Sharing Load Between Internal Drive Transistors and Positive External Drive Transistors to Increase Load Capacity. A Similiar Configuration is Possible with the Negative External Drive Transistor. COMP 5V = Off 0V = On Load Re CJ Sense Output 10kΩ Differential Input Signal ALD1000 FIGURE 10. Showing Flexibility in Application: Using Direct Voltage Feedback to Free Both IAs; Using an IA as a Differential Input; Using a Grounded Input to Provide an Off State; Providing a Ground Path for Bias Current With a Thermocouple. ® ALD1000 14