Evaluation board available. NX2154/2154A 300kHz SYNCHRONOUS PWM CONTROLLER PRELIMINARY DATA SHEET Pb Free Product DESCRIPTION The NX2154/2154A controller IC is a synchronous Buck controller IC designed for step down DC to DC converter applications. It is optimized to convert bus voltages from 2V to 40V to outputs as low as 0.8V voltage. The NX2154/2154A operates at fixed 300kHz. The NX2154/2154A employs fixed loss-less current limiting by sensing the Rdson of synchronous MOSFET followed by hiccup feature.NX2154A has higher current limit threshold than NX2154. Feedback under voltage also triggers hiccup. Other features of the device are: 5V gate drive, Adaptive deadband control, Internal digital soft start, Vcc undervoltage lock out and shutdown capability via the comp pin. FEATURES n n n n n n Bus voltage operation from 2V to 40V Fixed 300kHz voltage mode controller Internal Digital Soft Start Function Prebias Startup Less than 50 nS adaptive deadband Current limit triggers hiccup by sensing Rdson of Synchronous MOSFET n No negative spike at Vout during startup and shutdown n Pb-free and RoHS compliant APPLICATIONS n n n n n Graphic Card on board converters Memory Vddq Supply in mother board applications On board DC to DC such as 5V to 3.3V, 2.5V or 1.8V Hard Disk Drive Set Top Box TYPICAL APPLICATION Vin Vin +5V +33V MBR0530T1 7 HI=SD M3 1 BST Comp 12nF 82pF 13.3k 6 5 Vcc NX2154 1uF Hdrv 50ME180WX 0.1uF 2 M1A STM6920 15uH SW Ldrv Fb 1uF Vout +5V,3A 8 4 M1B 6ME1000WG(1000uF,30mohm) Gnd 3 1k 191 Figure1 - Typical application of 2154 ORDERING INFORMATION Device NX2154CSTR NX2154ACSTR Rev.1.2 02/26/07 Temperature 0 to 70oC 0 to 70o C Package SOIC-8L SOIC-8L Frequency 300kHz 300kHz OCP Threshold 360mV 540mV Pb-Free Yes Yes 1 NX2154/2154A ABSOLUTE MAXIMUM RATINGS (NOTE1) Vcc to GND & BST to SW voltage ................... 6.5V BST to GND Voltage ...................................... 50V Storage Temperature Range ............................. -65oC to 150oC Operating Junction Temperature Range ............. -40oC to 125oC NOTE1: Stresses above those listed in "ABSOLUTE MAXIMUM RATINGS", may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. PACKAGE INFORMATION 8-PIN PLASTIC SOIC (S) θJA ≈ 130o C/W BST 1 HDrv 2 8 SW 7 Comp Gnd 3 6 Fb LDrv 4 5 Vcc ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over Vcc = 5V, and TA = 0 to 70oC. Typical values refer to TA = 25oC. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER Reference Voltage Ref Voltage Ref Voltage line regulation Supply Voltage(Vcc) VCC Voltage Range VCC Supply Current (Static) VCC Supply Current (Dynamic) VCC ICC (Static) Outputs not switching ICC CLOAD=3300pF FS=300kHz (Dynamic) Supply Voltage(VBST) VBST Supply Current (Static) IBST (Static) Outputs not switching VBST Supply Current (Dynamic) CLOAD=3300pF IBST (Dynamic) Under Voltage Lockout VCC-Threshold VCC-Hysteresis VCC_UVLO VCC Rising VCC_Hyst VCC Falling Rev.1.2 02/26/07 SYM VREF Test Condition Min 4.5V<Vcc<5.5V TYP MAX 0.8 0.4 FS=300kHz 4.5 5 3 5 Units V % 5.5 V mA mA 0.15 mA 5 mA 4.2 0.22 V V 2 NX2154/2154A PARAMETER SS Soft Start time Oscillator (Rt) Frequency Ramp-Amplitude Voltage Max Duty Cycle Min Duty Cycle Error Amplifiers Transconductance Input Bias Current Comp SD Threshold FBUVLO Feedback UVLO threshold High Side Driver(C L=2200pF) Output Impedance , Sourcing Output Impedance , Sinking Sourcing Current Sinking Current Rise Time Fall Time Deadband Time SYM Tss Test Condition Min Fsw=300Khz FS VRAMP TYP MAX 3.4 1.7 mS 300 1.6 84 kHz V % % 0 2000 10 0.3 Ib percent of nominal 65 Units 70 umho nA V 75 % Rsource(Hdrv) I=200mA Rsink(Hdrv) I=200mA Isource(Hdrv) Isink(Hdrv) THdrv(Rise) THdrv(Fall) Tdead(L to Ldrv going Low to Hdrv H) going High, 10%-10% 1.9 1.7 1 1.2 14 17 30 ohm ohm A A ns ns ns Rsource(Ldrv) I=200mA 1.9 ohm Rsink(Ldrv) I=200mA 1 ohm Isource(Ldrv) 1 A Isink(Ldrv) 2 A 13 12 10 ns ns ns 360 540 mV Low Side Driver (C L=2200pF) Output Impedance, Sourcing Current Output Impedance, Sinking Current Sourcing Current Sinking Current Rise Time Fall Time Deadband Time OCP OCP voltage Rev.1.2 02/26/07 TLdrv(Rise) TLdrv(Fall) Tdead(H to SW going Low to Ldrv L) going High, 10% to 10% NX2154 NX2154A 3 NX2154/2154A PIN DESCRIPTIONS PIN # 1 BST PIN DESCRIPTION This pin supplies voltage to the high side driver. A high frequency ceramic capacitor of 0.1 to 1 uF must be connected from this pin to SW pin. 2 HDRV High side MOSFET gate driver. 3 GND Ground pin. 4 LDRV Low side MOSFET gate driver. For the high current application, a 4.7nF capacitor is recommend to placed on low side MOSFET's gate to ground. This is to prevent undesired Cdv/dt induced low side MOSFET's turn on to happen, which is caused by fast voltage change on the drain of low side MOSFET in synchronous buck converter and lower the system efficiency. 5 Vcc Voltage supply for the internal circuit as well as the low side MOSFET gate driver. A 1uF high frequency ceramic capacitor must be connected from this pin to GND pin. 6 FB This pin is the error amplifier inverting input. This pin is also connected to the output UVLO comparator. When this pin falls below 0.56V, both HDRV and LDRV outputs are in hiccup. 7 COMP This pin is the output of the error amplifier and together with FB pin is used to compensate the voltage control feedback loop. This pin is also used as a shut down pin. When this pin is pulled below 0.3V, both drivers are turned off and internal soft start is reset. 8 SW This pin is connected to the source of the high side MOSFET and provides return path for the high side driver. Also SW senses the low side MOSFETS current, when the pin voltage is lower than 360mV for NX2154, 540mV for NX2154A, hiccup will be triggered. Rev.1.2 02/26/07 PIN SYMBOL 4 NX2154/2154A BLOCK DIAGRAM VCC 70%Vp Hiccup Logic FB Bias Generator 1.25V OC 0.8V UVLO BST POR START HDRV COMP SW 0.3V OC Control Logic START 0.8V VCC PWM OSC Digital start Up ramp S R LDRV Q FB 0.6V CLAMP COMP START 360mV/540mV 1.3V CLAMP Hiccup Logic OCP comparator GND Figure 2 - Simplified block diagram of the NX2154/NX2154A Rev.1.2 02/26/07 5 NX2154/2154A APPLICATION INFORMATION Symbol Used In Application Information: VIN - Input voltage VOUT - Output voltage IOUT - Output current = VIN -VOUT VOUT 1 × × L OUT VIN FS ...(2) 33V-5V 5V 1 × × = 0.94A 15uH 33V 300kHz Output Capacitor Selection DVRIPPLE - Output voltage ripple FS ∆IRIPPLE = Output capacitor is basically decided by the - Working frequency amount of the output voltage ripple allowed during steady DIRIPPLE - Inductor current ripple state(DC) load condition as well as specification for the load transient. The optimum design may require a couple Design Example schematic is figure 1. of iterations to satisfy both condition. Based on DC Load Condition The amount of voltage ripple during the DC load VIN = 33V condition is determined by equation(3). The following is typical application for NX2154, the VOUT=5V ∆IRIPPLE 8 × FS × COUT ...(3) FS=300kHz ∆VRIPPLE = ESR × ∆IRIPPLE + IOUT=3A Where ESR is the output capacitors' equivalent DVRIPPLE <=50mV series resistance,COUT is the value of output capacitors. DVDROOP<=250mV @ 9A step Typically when large value capacitors are selected such as Aluminum Electrolytic,POSCAP and OSCON Output Inductor Selection The selection of inductor value is based on inductor ripple current, power rating, working frequency and efficiency. Larger inductor value normally means smaller ripple current. However if the inductance is chosen too large, it brings slow response and lower efficiency. Usually the ripple current ranges from 20% to 40% of the output current. This is a design freedom which can be types are used, the amount of the output voltage ripple is dominated by the first term in equation(3) and the second term can be neglected. For this example,electrolytic capacitors are chosen as output capacitors, the ESR and inductor current typically determines the output voltage ripple. ESR desire = decided by design engineer according to various application requirements. The inductor value can be calcu- VIN -VOUT VOUT 1 × × ∆IRIPPLE VIN FS IRIPPLE =k × IOUTPUT 6ME1000WG (1000uF,30mΩ) is chosen. ...(1) 33V-5V 5V 1 × × 0.3 × 3A 33V 300kHz L OUT =15.7uH L OUT = Choose inductor from COILCRAFT DO5022P-153 with L=15uH is a good choice. Rev.1.2 02/26/07 If low ESR is required, for most applications, multor. For example, SANYO electrolytic capacitor where k is between 0.2 to 0.4. Select k=0.3, then Current Ripple is recalculated as ...(4) tiple capacitors in parallel are better than a big capaci- lated by using the following equations: L OUT = ∆VRIPPLE 50mV = = 53m Ω ∆IRIPPLE 0.94A N = E S R E × ∆ IR I P P L E ∆ VR IPPLE ...(5) Number of Capacitor is calculated as N= 30mΩ× 0.94A 50mV N =0.566 The number of capacitor has to be round up to a integer. Choose N =1. If ceramic capacitors are chosen as output ca 6 NX2154/2154A pacitors, both terms in equation (3) need to be evalu- of output capacitor. For low frequency capacitor such ated to determine the overall ripple. Usually when this as electrolytic capacitor, the product of ESR and ca- type of capacitors are selected, the amount of capaci- pacitance is high and L ≤ L crit is true. In that case, the tance per single unit is not sufficient to meet the tran- transient spec is dependent on the ESR of capacitor. sient specification, which results in parallel configuration of multiple capacitors . capacitors in parallel. The number of capacitors can be For example, one 100uF, X5R ceramic capacitor with 2mΩ ESR is used. The amount of output ripple is ∆VRIPPLE In most cases, the output capacitors are multiple calculated by the following N= 0.94A = 2mΩ× 0.94A + 8 × 300kHz × 100uF = 5.4mV is specified as: ∆VDROOP <∆VTRAN @ step load DISTEP During the transient, the voltage droop during the transient is composed of two sections. One Section is dependent on the ESR of capacitor, the other section is a function of the inductor, output capacitance as well as input, output voltage. For example, for the overshoot, when load from high load to light load with a DISTEP transient load, if assuming the bandwidth of system is high enough, the overshoot can be estimated as the following equation. VOUT × τ2 2 × L × COUT ...(6) where τ is the a function of capacitor, etc. 0 if L ≤ L crit τ = L × ∆Istep − ESR × COUT V OUT if L ≥ L crit ESR × COUT × VOUT ESR E × C E × VOUT = ∆Istep ∆Istep VOUT × τ2 2 × L × C E × ∆Vtran 0 if L ≤ L crit τ = L × ∆Istep − ESR E × CE V OUT ...(9) ...(7) if L ≥ L crit ...(10) For example, assume voltage droop during transient is 250mV for 3A load step. If the SANYO electrolytic capaictor 6ME1000WG (1000uF, 30mΩ ) is used, the critical inductance is given as L crit = ESR E × C E × VOUT = ∆ I step 30m Ω × 1000µF × 5V = 50µH 3A The selected inductor is 15uH which is smaller than critical inductance. In that case, the output voltage transient only dependent on the ESR. number of capacitors is N= where L crit = ∆Vtran + where Although this meets DC ripple spec, however it needs to be studied for transient requirement. Based On Transient Requirement Typically, the output voltage droop during transient ∆Vovershoot = ESR × ∆Istep + ESR E × ∆Istep ESR E × ∆Istep ∆Vtran + VOUT × τ2 2 × L × CE × ∆Vtran 30mΩ × 3A + 250mV 5V × (0) 2 2 ×15µH ×1000µF × 250mV = 0.36 = ...(8) where ESRE and CE represents ESR and capaci- The number of capacitors has to satisfied both ripple and transient requirement. Overall, we can choose N=1. tance of each capacitor if multiple capacitors are used in parallel. The above equation shows that if the selected output inductor is smaller than the critical inductance, the voltage droop or overshoot is only dependent on the ESR Rev.1.2 02/26/07 7 NX2154/2154A It should be considered that the proposed equation is based on ideal case, in reality, the droop or overshoot is typically more than the calculation. The equation gives a good start. For more margin, more capacitors have to be chosen after the test. Typically, for high frequency capacitor such as high quality POSCAP especially ceramic capacitor, 20% to 100% (for ceramic) more capacitors have to be chosen since the ESR of capacitors is so low that the PCB parasitic can affect FZ1 = 1 2 × π × R 4 × C2 ...(11) FZ2 = 1 2 × π × (R 2 + R3 ) × C3 ...(12) FP1 = 1 2 × π × R3 × C3 ...(13) FP2 = the results tremendously. More capacitors have to be 1 ...(14) C × C2 2 × π × R4 × 1 C1 + C2 selected to compensate these parasitic parameters. where FZ1,FZ2,FP1 and FP2 are poles and zeros in Compensator Design Due to the double pole generated by LC filter of the the compensator. Their locations are shown in figure 4. The transfer function of type III compensator for power stage, the power system has 180o phase shift , transconductance amplifier is given by: and therefore, is unstable by itself. In order to achieve Ve 1 − gm × Z f = VOUT 1 + gm × Zin + Z in / R1 accurate output voltage and fast transient response,compensator is employed to provide highest possible bandwidth and enough phase margin.Ideally,the Bode plot of the closed loop system has crossover fre- For the voltage amplifier, the transfer function of compensator is phase margin greater than 50o and the gain crossing Ve −Z f = VOUT Zin 0dB with -20dB/decade. Power stage output capacitors To achieve the same effect as voltage amplifier, usually decide the compensator type. If electrolytic the compensator of transconductance amplifier must capacitors are chosen as output capacitors, type II com- satisfy this condition: R 4>>2/gm. And it would be desir- pensator can be used to compensate the system, be- able if R 1||R2||R3>>1/gm can be met at the same time. quency between1/10 and 1/5 of the switching frequency, cause the zero caused by output capacitor ESR is lower than crossover frequency. Otherwise type III compensator should be chosen. A. Type III compensator design Zin R3 R2 For low ESR output capacitors, typically such as Sanyo oscap and poscap, the frequency of ESR zero C3 sate the system with type III compensator. The following figures and equations show how to realize the type III C2 R4 Fb caused by output capacitors is higher than the crossover frequency. In this case, it is necessary to compen- Zf C1 Vout gm Ve R1 Vref compensator by transconductance amplifier. Figure 3 - Type III compensator using transconductance amplifier Rev.1.2 02/26/07 8 NX2154/2154A Case 1: 2. Set R2 equal to 10kΩ. FLC<FESR<FO Gain(db) R1= R2 × VREF 10kΩ× 0.8V = = 1.91kΩ VOUT -VREF 5V-0.8V Choose R1=1.91kΩ. 3. Set zero FZ2 = FLC and Fp1 =FESR . 4. Calculate C3 . power stage FLC 40dB/decade C3 = FESR 1 1 1 ×( ) 2 × π × R2 Fz2 Fp1 1 1 1 ×( ) 2 × π × 10kΩ 1.3kHz 5.3kHz =9.2nF = loop gain 20dB/decade Choose C3=10nF. 5. Calculate R3 . R3 = compensator 1 2 × π × FP1 × C 3 1 2 × π × 5.3kHz × 10nF = 3kΩ = FZ1 FZ2 FP1 FO FP2 Choose R3 =3kΩ. 6. Calculate R4 with FO=30kHz. R4 = Figure 4 - Bode plot of Type III compensator (FLC<FESR<FO) If electrolytic capacitors are used as output capacitors, typical design example of type III compensator in which the crossover frequency is selected as FLC<FESR<FO and F O<=1/10~1/5Fs is shown as the following steps. Here one SANYO 6ME1000WG with 30 mΩ is chosen as output capacitor. 1. Calculate the location of LC double pole F LC and ESR zero FESR. FLC = = 1 2 × π × L OUT × COUT 1 2 × π × 15uH × 1000uF = 1.3kHz FESR = 1 2 × π × ESR × COUT 1 2 × π × 30mΩ × 1000uF = 5.3kHz = Rev.1.2 02/26/07 VOSC 2 × π × FO × L R2 × R3 × × Vin ESR R2 + R3 1.5V 2 × π × 30kHz × 15uH 10kΩ × 3kΩ × × 33V 30mΩ 10kΩ + 3kΩ =9.9kΩ = Choose R4=10kΩ. 7. Calculate C2 with zero Fz1 at 75% of the LC double pole by equation (11). C2 = 1 2 × π × FZ1 × R 4 1 2 × π × 0.75 × 1.3kHz × 10k Ω = 12.2nF = Choose C2=12nF. 8. Calculate C 1 by equation (14) with pole F p2 at half the switching frequency. C1 = 1 2 × π × R 4 × FP2 1 2 × π × 10k Ω × 150kHz = 106pF = Choose C1=100pF. 9 NX2154/2154A Case 2: FLC<FO<FESR R 2 × VREF 10k Ω × 0.8V = = 8k Ω VOUT -VREF 1.8V-0.8V R1 = Gain(db) Choose R1=8kΩ. power stage 3. Set zero FZ2 = FLC and Fp1 =FESR . FLC 4. Calculate R4 and C3 with the crossover 40dB/decade frequency at 1/10~ 1/5 of the switching frequency. Set FO=30kHz. C3 = loop gain FESR 20dB/decade compensator 1 1 1 ×( ) 2 × π × R2 Fz2 Fp1 1 1 1 ×( ) 2 × π × 10kΩ 6.2kHz 60.3kHz =2.3nF = VOSC 2 × π × FO × L × × Cout Vin C3 R4 = 1.5V 2 × π × 30kHz × 1.5uH × × 440uF 5V 2.2nF =16.9kΩ = FZ1 FZ2 FO FP1 FP2 Choose C3=2.2nF, R 4=16.9kΩ. 5. Calculate C2 with zero Fz1 at 75% of the LC double pole by equation (11). Figure 5 - Bode plot of Type III compensator 1 2 × π × FZ1 × R 4 C2 = Design example for type III compensator are in order. The crossover frequency has to be selected as FLC<FO<FESR and FO<=1/10~1/5Fs. In this case, input voltage is 5V, output voltage is 1.8V, inductor is 1.5uH, two POSCAP 2R5TPE220MC(220uF,12 mΩ) are chosen as output capacitor. 1.Calculate the location of LC double pole F LC and ESR zero FESR. FLC = = 1 2 × π × L OUT × COUT 1 2 × π × 1.5uH × 440uF = 6.2kHz FESR 1 = 2 × π × ESR × C OUT 1 = 2 × π × 6m Ω × 440uF = 60.3kHz 2. Set R2 equal to 10kΩ. Rev.1.2 02/26/07 1 2 × π × 0.75 × 6.2kHz × 16.9kΩ = 2nF = Choose C2=2.2nF. 6. Calculate C 1 by equation (14) with pole F p2 at half the switching frequency. 1 2 × π × R 4 × FP2 C1 = 1 2 × π × 16.9kΩ × 150kHz = 63pF = Choose C1=68pF. 7. Calculate R 3 by equation (13). R3 = 1 2 × π × FP1 × C3 1 2 × π × 60.3kHz × 2.2nF = 1.2kΩ = Choose R3=1.2kΩ. 10 NX2154/2154A B. Type II compensator design If the electrolytic capacitors are chosen as power Vout stage output capacitors, usually the Type II compensator can be used to compensate the system. R2 Fb Type II compensator can be realized by simple RC circuit without feedback as shown in figure 6. R3 and C1 introduce a zero to cancel the double pole effect. C2 Ve gm R1 R3 Vref C2 introduces a pole to suppress the switching noise. The following equations show the compensator pole zero lo- C1 cation and constant gain. Gain=gm × R1 × R3 R1+R2 ... (15) Figure 7 - Type II compensator with 1 Fz = 2 × π × R3 × C1 Fp ≈ transconductance amplifier ... (16) 1 2 × π × R3 × C2 ... (17) For this type of compensator, FO has to satisfy FLC<FESR<<FO<=1/10~1/5Fs. The following is parameters for type II compensator design. Input voltage is 40V, output voltage is 5V, output inductor is 6uH, output capacitor is one 1000uF Gain(db) power stage with 30mΩ electrolytic capacitors. 40dB/decade 1.Calculate the location of LC double pole F LC and ESR zero FESR. FLC = loop gain = 20dB/decade compensator Gain 1 2 × π × L OUT × COUT 1 2 × π × 15uH × 1000uF = 1.3kHz FESR = 1 2 × π × ESR × COUT 1 2 × π × 30mΩ × 1000uF = 5.3kHz = FZ FLC FESR FO FP 2.Set R2 equal to 1kΩ. Figure 6 - Bode plot of Type II compensator R1 = R 2 × VREF 1kΩ × 0.8V = = 191Ω VOUT -VREF 5V-0.8V Choose R1=191Ω. 3. Set crossover frequency at 1/10~ 1/5 of the swithing frequency, here FO=30kHz. 4.Calculate R3 value by the following equation. Rev.1.2 02/26/07 11 NX2154/2154A 4.Calculate R3 value by the following equation. V 2 × π × FO × L 1 VOUT × × R3 = OSC × Vin RESR gm VREF 1.5V 2 × π × 30kHz × 15uH 1 × × 33V 30mΩ 2.0mA/V 5V × 0.8V =13.3kΩ Vout R2 Fb = R1 Vref Voltage divider Choose R 3 =13.3kΩ. 5. Calculate C1 by setting compensator zero FZ Figure 8 - Voltage divider at 75% of the LC double pole. C1= Input Capacitor Selection 1 2 × π × R 3 × Fz Input capacitors are usually a mix of high frequency 1 2 × π × 13.3kΩ × 0.75 × 1.3kHz =12.2nF = pacitors bypass the high frequency noise, and bulk capacitors supply switching current to the MOSFETs. Usually 1uF ceramic capacitor is chosen to decouple the Choose C1=12nF. 6. Calculate C 2 by setting compensator pole Fp at half the swithing frequency. C2= ceramic capacitors and bulk capacitors. Ceramic ca- high frequency noise.The bulk input capacitors are decided by voltage rating and RMS current rating. The RMS current in the input capacitors can be calculated as: 1 π × R 3 × Fs IRMS = IOUT × D × 1- D 1 π × 1 3 .3k Ω × 3 0 0 k H z =80pF = Choose C1=82pF. D= VOUT VIN ...(19) VIN = 33V, VOUT=5V, IOUT=3A, using equation (19), the result of input RMS current is 1.1A. For higher efficiency, low ESR capacitors are rec- Output Voltage Calculation Output voltage is set by reference voltage and external voltage divider. The reference voltage is fixed at 0.8V. The divider consists of two ratioed resistors so that the output voltage applied at the Fb pin is 0.8V when the output voltage is at the desired value. The following equation and picture show the relationship between VOUT , VREF and voltage divider.. R 2 × VR E F R 1= V O U T -V R E F ...(18) where R 2 is part of the compensator, and the value of R1 value can be set by voltage divider. See compensator design for R1 and R2 selection. ommended. One Sanyo electrolytic capacitor 50ME180WX 50V 180uF 46mΩ with 1.19A RMS rating is chosen as input bulk capacitors. Power MOSFETs Selection The power stage requires two N-Channel power MOSFETs. The selection of MOSFETs is based on maximum drain source voltage, gate source voltage, maximum current rating, MOSFET on resistance and power dissipation. The main consideration is the power loss contribution of MOSFETs to the overall converter efficiency. In this design example, two STM6920 are used. They have the following parameters: V DS=40V, I D =7A,RDSON =45mΩ,QGATE =8.7nC. There are two factors causing the MOSFET power loss:conduction loss, switching loss. Rev.1.2 02/26/07 12 NX2154/2154A Conduction loss is simply defined as: ISET = PHCON =IOUT 2 × D × RDS(ON) × K PLCON =IOUT 2 × (1 − D) × RDS(ON) × K PTOTAL =PHCON + PLCON ...(20) where the RDS(ON) will increases as MOSFET junc- 360mV K × RDSON If MOSFET R DSON=45mΩ, the worst case thermal consideration K=1.5, then ISET = 320mV 360mV = = 5.3A K × RDSON 1.5 × 45m Ω tion temperature increases, K is RDS(ON) temperature dependency. As a result, RDS(ON) should be selected for the worst case, in which K approximately equals to 1.5 at 125oC according to STM6920 datasheet. Conduction loss should not exceed package rating or overall system thermal budget. Switching loss is mainly caused by crossover conduction at the switching transition. The total switching loss can be approximated. The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results. There are two sets of components considered in the layout which are power components and small signal components. Power components usually consist of 1 PSW = × VIN × IOUT × TSW × FS ...(21) 2 where IOUT is output current, TSW is the sum of TR and TF which can be found in mosfet datasheet, and FS is switching frequency. Switching loss PSW is frequency dependent. Also MOSFET gate driver loss should be considered when choosing the proper power MOSFET. MOSFET gate driver loss is the loss generated by discharging the gate capacitor and is dissipated in driver circuits.It is proportional to frequency and is defined as: Pgate = (QHGATE × VHGS + QLGATE × VLGS ) × FS Layout Considerations ...(22) where QHGATE is the high side MOSFETs gate charge,QLGATE is the low side MOSFETs gate charge,VHGS input capacitors, high-side MOSFET, low-side MOSFET, inductor and output capacitors. A noisy environment is generated by the power components due to the switching power. Small signal components are connected to sensitive pins or nodes. A multilayer layout which includes power plane, ground plane and signal plane is recommended . Layout guidelines: 1. First put all the power components in the top layer connected by wide, copper filled areas. The input capacitor, inductor, output capacitor and the MOSFETs should be close to each other as possible. This helps to reduce the EMI radiated by the power loop due to the high switching currents through them. is the high side gate source voltage, and VLGS is the low 2. Low ESR capacitor which can handle input RMS side gate source voltage. This power dissipation should not exceed maximum power dissipation of the driver device. ripple current and a high frequency decoupling ceramic cap which usually is 1uF need to be practically touching the drain pin of the upper MOSFET, a plane connection is a must. Over Current Limit Protection Over current Limit for step down converter is achieved by sensing current through the low side 3. The output capacitors should be placed as close as to the load as possible and plane connection is required. MOSFET. For NX2154, the current limit is decided by 4. Drain of the low-side MOSFET and source of the RDSON of the low side mosfet. When synchronous the high-side MOSFET need to be connected thru a plane FET is on, and the voltage on SW pin is below 360mV, ans as close as possible. A snubber nedds to be placed the over current occurs. The over current limit can be as close to this junction as possible. calculated by the following equation. Rev.1.2 02/26/07 5. Source of the lower MOSFET needs to be con13 NX2154/2154A nected to the GND plane with multiple vias. One is not back to the resistor divider should not go through high enough. This is very important. The same applies to the frequency signals. output capacitors and input capacitors. 9. All GNDs need to go directly thru via to GND 6. Hdrv and Ldrv pins should be as close to plane. MOSFET gate as possible. The gate traces should be 10. The feedback part of the system should be wide and short. A place for gate drv resistors is needed kept away from the inductor and other noise sources, to fine tune noise if needed. and be placed close to the IC. 7. Vcc capacitor, BST capacitor or any other by- 11. In multilayer PCB, separate power ground and passing capacitor needs to be placed first around the IC analog ground. These two grounds must be connected and as close as possible. The capacitor on comp to together on the PC board layout at a single point. The GND or comp back to FB needs to be place as close to goal is to localize the high current path to a separate the pin as well as resistor divider. loop that does not interfere with the more sensitive ana- 8. The output sense line which is sensing output log control function. TYPICAL APPLICATION FOR HIGH CURRENT L2 1uH Vin +12V C3 33uF C5 1uF Cin 2 x 16SP180M D1 MBR0530T1 Vin 7 HI=SD M3 C1 220pF 1 BST Comp C2 15nF R4 5k 6 5 Vcc NX2154 C6 1uF +5V Hdrv Gnd 2 M1 IRF3706 L1 1uH SW Ldrv Fb C4 0.1uF 8 4 M2 2 x IRF3706 Vout +1.8V,20A Co 2 x (1500uF,13mohm) C7 4.7nF 3 R2 800 R1 1k Figure 9 - High output current application of 2154 Rev.1.2 02/26/07 14 NX2154/2154A TYPICAL APPLICATION FOR LED +5V Vin 78L05 +9V to 33V D1 MBR0530T1 1uF 10k 1N4148 1uF 7 1uF 5 1 Vcc BST Comp Hdrv 2 IRF7341 68nF 10uH SW 56pF NX2154 1.6k 6 220uF 0.1uF Fb Ldrv 8 Vout Co 1000uF,30mohm 4 Gnd 3 LUXEON III star LED LM358 0.1ohm POT 100k 1k Figure 10 - NX2154 LED application Waveforms for LED application 0.9 EFFICIENCY(%) 0.85 0.8 0.75 0.7 0.65 0.6 0.55 0 0.2 0.4 0.6 0.8 1 1.2 LED CURRENT(A) Figure 11 - LED application efficiency (One LUXEDN III star LED, VIN=12V) Rev.1.2 02/26/07 Figure 12 - Startup in NX2154 LED application 15 NX2154/2154A SOIC8 PACKAGE OUTLINE DIMENSIONS Rev.1.2 02/26/07 16 NX2154/2154A Rev.1.2 02/26/07 17