LM3429 N-Channel Controller for Constant Current LED Drivers General Description Features The LM3429 is a versatile high voltage N-channel MosFET controller for LED drivers . It can be easily configured in buck, boost, buck-boost and SEPIC topologies. This flexibility, along with an input voltage rating of 75V, makes the LM3429 ideal for illuminating LEDs in a very diverse, large family of applications. Adjustable high-side current sense voltage allows for tight regulation of the LED current with the highest efficiency possible. The LM3429 uses Predictive Off-time (PRO) control, which is a combination of peak current-mode control and a predictive off-timer. This method of control eases the design of loop compensation while providing inherent input voltage feed-forward compensation. The LM3429 includes a high-voltage startup regulator that operates over a wide input range of 4.5V to 75V. The internal PWM controller is designed for adjustable switching frequencies of up to 2.0 MHz, thus enabling compact solutions. Additional features include analog dimming, PWM dimming, over-voltage protection, under-voltage lock-out, cycle-by-cycle current limit, and thermal shutdown. The LM3429 comes in a low profile, thermally efficient TSSOP EP 14-lead package. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ VIN range from 4.5V to 75V Adjustable current sense voltage High-side current sensing 2Ω, 1A Peak MosFET gate driver Input under-voltage protection Over-voltage protection PWM dimming Analog dimming Cycle-by-cycle current limit Programmable switching frequency Thermal Shutdown TSSOP EP 14-lead package Applications ■ ■ ■ ■ ■ ■ LED Drivers Constant-Current Buck-Boost (or Flyback) Regulator Constant-Current Boost Regulator Constant-Current Buck Regulator Constant-Current SEPIC Regulator Thermo-Electric Cooler (Peltier) Driver Typical Application Circuit 30094422 Boost LED Driver © 2009 National Semiconductor Corporation 300944 www.national.com LM3429 N-Channel Controller for Constant Current LED Drivers August 3, 2009 LM3429 Connection Diagram Top View 30094404 14-Lead TSSOP EP NS Package Number MXA14A Ordering Information Order Number Spec. Package Type NSC Package Drawing LM3429MH NOPB TSSOP-14 EP MXA14A 94 Units, Rail LM3429MHX NOPB TSSOP-14 EP MXA14A 2500 Units, Tape and Reel Supplied As Pin Descriptions Pin Name Description Application Information 1 VIN Input Voltage Bypass with 100 nF capacitor to AGND as close to the device as possible in the circuit board layout. 2 COMP Compensation Connect a capacitor to AGND. 3 CSH Current Sense High Connect a resistor to AGND to set the signal current. For analog dimming, connect a controlled current source or a potentiometer to AGND as detailed in the Analog Dimming section. 4 RCT Resistor Capacitor Timing Connect a resistor from the switch node and a capacitor to AGND to set the switching frequency. 5 AGND Analog Ground Connect to PGND through the DAP copper circuit board pad to provide proper ground return for CSH, COMP, and RCT. 6 OVP Over-Voltage Protection Connect to a resistor divider from VO to program output over-voltage lockout (OVLO). Turn-off threshold is 1.24V and hysteresis for turn-on is provided by 20 µA current source. 7 nDIM Not DIM input Connect a PWM signal for dimming as detailed in the PWM Dimming section and/or a resistor divider from VIN to program input under-voltage lockout (UVLO). Turn-on threshold is 1.24V and hysteresis for turn-off is provided by 20 µA current source. 8 NC No Connection Leave open. 9 PGND Power Ground Connect to AGND through the DAP copper circuit board pad to provide proper ground return for GATE. 10 GATE Gate Drive Output 11 VCC Internal Regulator Output 12 IS Main Switch Current Sense 13 HSP High-Side LED Current Sense Connect through a series resistor to the positive side of the LED current sense Positive resistor. 14 HSN High-Side LED Current Sense Connect through a series resistor to the negative side of the LED current sense Negative resistor. DAP (15) DAP www.national.com Thermal pad on bottom of IC Connect to the gate of the external NFET. Bypass with a 2.2 µF–3.3 µF, ceramic capacitor to PGND. Connect to the drain of the main N-channel MosFET switch for RDS-ON sensing or to a sense resistor installed in the source of the same device. Star ground, connecting AGND and PGND. For thermal considerations please refer to (Note 4) of the Electrical Characteristics table. 2 PGND If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Junction Temperature Storage Temperature Range Maximum Lead Temperature (Reflow and Solder) (Note 5) Continuous Power Dissipation (Note 4) ESD Susceptibility (Note 6) Human Body Model VIN, nDIM -0.3V to 76.0V -1 mA continuous -0.3V to 76.0V -100 µA continuous -0.3V to 3.0V -1 mA to +5 mA continuous -0.3V to 76.0V -2V for 100 ns -1 mA continuous -0.3V to 8.0V -0.3V to 6.0V -200 µA to +200 µA Continuous -0.3V to VCC -2.5V for 100 ns VCC+2.5V for 100 ns -1 mA to +1 mA continuous OVP, HSP, HSN RCT IS VCC COMP, CSH GATE -0.3V to 0.3V -2.5V to 2.5V for 100 ns 150°C −65°C to +150°C 260°C Operating Conditions Operating Junction Temperature Range Input Voltage VIN Internally Limited 2 kV (Notes 1, 2) −40°C to +125°C 4.5V to 75V Electrical Characteristics (Note 2) Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40°C to +125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = +25°C, and are provided for reference purposes only. Unless otherwise stated the following condition applies: VIN = +14V. Symbol Parameter Conditions Min (Note 7) Typ (Note 8) Max (Note 7) Units STARTUP REGULATOR (VCC) VCC-REG VCC Regulation ICC = 0 mA ICC-LIM VCC Current Limit VCC = 0V IQ Quiescent Current Static 1.6 3.0 VCC-UVLO VCC UVLO Threshold VCC Increasing 4.17 4.50 VCC-HYS VCC UVLO Hysteresis VCC Decreasing 6.30 6.90 20 27 3.70 7.35 4.08 V mA V 0.1 OVER-VOLTAGE PROTECTION (OVP) VTH-OVP OVP OVLO Threshold OVP Increasing IHYS-OVP OVP Hysteresis Source Current OVP Active (high) 1.180 1.240 1.280 V 10 20 30 µA 1.210 1.235 1.260 V -0.6 0 0.6 26 40 ERROR AMPLIFIER VCSH CSH Reference Voltage With Respect to AGND Error Amplifier Input Bias Current COMP Sink / Source Current 10 Transconductance Linear Input Range (Note 9) Transconductance Bandwidth -6dB Unloaded Response (Note 9) 0.5 µA 100 µA/V ±125 mV 1.0 MHz OFF TIMER (RCT) tOFF-MIN Minimum Off-time RRCT RCT Reset Pull-down Resistance VRCT VIN/25 Reference Voltage RCT = 1V through 1 kΩ VIN = 14V 540 3 35 75 ns 36 120 Ω 565 585 mV www.national.com LM3429 Absolute Maximum Ratings (Notes 1, 2) LM3429 Symbol Parameter Conditions Min (Note 7) Typ (Note 8) Max (Note 7) Units PWM COMPARATOR COMP to PWM Offset 700 800 900 mV 215 245 275 mV 35 75 250 450 CURRENT LIMIT (IS) VLIM Current Limit Threshold VLIM Delay to Output tON-MIN Leading Edge Blanking Time 75 ns HIGH SIDE TRANSCONDUCTANCE AMPLIFIER Input Bias Current 10 µA Transconductance 20 119 mA/V Input Offset Current -1.5 0 1.5 µA Input Offset Voltage -7 0 7 mV 250 500 Transconductance Bandwidth ICSH = 100 µA (Note 9) kHz GATE DRIVER (GATE) RSRC(GATE) GATE Sourcing Resistance GATE = High 2.0 6.0 RSNK(GATE) GATE Sinking Resistance 1.3 4.5 GATE = Low Ω UNDER-VOLTAGE LOCKOUT and DIM INPUT (nDIM) VTH-nDIM nDIM / UVLO Threshold 1.180 1.240 1.280 V IHYS-nDIM nDIM Hysteresis Current 10 20 30 µA THERMAL SHUTDOWN TSD Thermal Shutdown Threshold (Notes 3, 9) THYS Thermal Shutdown Hysteresis (Notes 3, 9) 165 °C 25 THERMAL RESISTANCE θJA Junction to Ambient (Note 4) 14L TSSOP EP θJC Junction to Exposed Pad (DAP) 14L TSSOP EP 40 °C/W 5.5 °C/W Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in the Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be operated beyond such conditions. Note 2: All voltages are with respect to the potential at the AGND pin, unless otherwise specified. Note 3: Internal thermal shutdown circuitry protects the device from permanent damage. Thermal shutdown engages at TJ=165°C (typical) and disengages at TJ=140°C (typical). Note 4: Junction-to-ambient thermal resistance is highly board-layout dependent. The numbers listed in the table are given for a reference layout wherein the 14L TSSOP EP package has its DAP pad populated with 9 vias. In applications where high maximum power dissipation exists, namely driving a large MosFET at high switching frequency from a high input voltage, special care must be paid to thermal dissipation issues during board design. In high-power dissipation applications, the maximum ambient temperature may have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP = 125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to ambient thermal resistance of the package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP – (θJA × PD-MAX). In most applications there is little need for the full power dissipation capability of this advanced package. Under these circumstances, no vias would be required and the thermal resistances would be 104 °C/W for the 14L TSSOP EP. It is possible to conservatively interpolate between the full via count thermal resistance and the no via count thermal resistance with a straight line to get a thermal resistance for any number of vias in between these two limits. Note 5: Refer to National’s packaging website for more detailed information and mounting techniques. http://www.national.com/analog/packaging/ Note 6: Human Body Model, applicable std. JESD22-A114-C. Note 7: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% production tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL). Note 8: Typical numbers are at 25°C and represent the most likely norm. Note 9: These electrical parameters are guaranteed by design, and are not verified by test. Note 10: The measurements were made using the standard buck-boost evaluation board from AN-1985. Note 11: The measurements were made using the standard boost evaluation board from AN-1986. www.national.com 4 TA=+25°C and VIN = 14V unless otherwise specified Boost Efficiency vs. Input Voltage VO = 32V (9 LEDs) (Note 11) Buck-Boost Efficiency vs. Input Voltage VO = 20V (6 LEDs) (Note 10) 300944b5 300944b6 Boost LED Current vs. Input Voltage VO = 32V (9 LEDs) (Note 11) Buck-boost LED Current vs. Input Voltage VO = 20V (6 LEDs) (Note 10) 300944b8 300944b7 PWM Dimming VO = 20V (6 LEDs) (Note 10) Analog Dimming VO = 20V (6 LEDs) (Note 10) 300944c0 300944b9 5 www.national.com LM3429 Typical Performance Characteristics LM3429 VCSH vs. Junction Temperature VCC vs. Junction Temperature 300944b0 300944b1 VRCT vs. Junction Temperature VLIM vs. Junction Temperature 300944b3 300944b2 tON-MIN vs. Junction Temperature 300944b4 www.national.com 6 LM3429 Block Diagram 30094403 CURRENT REGULATORS Current regulators can be designed to accomplish three basic functions: buck, boost, and buck-boost. All three topologies in their most basic form contain a main switching MosFET, a recirculating diode, an inductor and capacitors. The LM3429 is designed to drive a ground referenced NFET which is perfect for a standard boost regulator. Buck and buck-boost regulators, on the other hand, usually have a high-side switch. When driving an LED load, a ground referenced load is often not necessary, therefore a ground referenced switch can be used to drive a floating load instead. The LM3429 can then be used to drive all three basic topologies as shown in the Basic Topology Schematics section. Looking at the buck-boost design, the basic operation of a current regulator can be analyzed. During the time that the NFET (Q1) is turned on (tON), the input voltage source stores energy in the inductor (L1) while the output capacitor (CO) provides energy to the LED load. When Q1 is turned off (tOFF), the re-circulating diode (D1) becomes forward biased and L1 provides energy to both CO and the LED load. Figure 1 shows the inductor current (iL(t)) waveform for a regulator operating in CCM. Theory of Operation The LM3429 is an N-channel MosFET (NFET) controller for buck, boost and buck-boost current regulators which are ideal for driving LED loads. The controller has wide input voltage range allowing for regulation of a variety of LED loads. The high-side differential current sense, with low adjustable threshold voltage, provides an excellent method for regulating output current while maintaining high system efficiency. The LM3429 uses a Predictive Off-time (PRO) control architecture that allows the regulator to be operated using minimal external control loop compensation, while providing an inherent cycle-by-cycle current limit. The adjustable current sense threshold provides the capability to amplitude (analog) dim the LED current and the output enable/disable function allows for PWM dimming using no external components. When designing, the maximum attainable LED current is not internally limited because the LM3429 is a controller. Instead it is a function of the system operating point, component choices, and switching frequency allowing the LM3429 to easily provide constant currents up to 5A. This simple controller contains all the features necessary to implement a high efficiency versatile LED driver. 7 www.national.com LM3429 30094498 FIGURE 1. Ideal CCM Regulator Inductor Current iL(t) The average output LED current (ILED) is proportional to the average inductor current (IL) , therefore if IL is tightly controlled, ILED will be well regulated. As the system changes input voltage or output voltage, the ideal duty cycle (D) is varied to regulate IL and ultimately ILED. For any current regulator, D is a function of the conversion ratio: PRO control was designed to mitigate “current mode instability” (also called “sub-harmonic oscillation”) found in standard peak current mode control when operating near or above 50% duty cycles. When using standard peak current mode control with a fixed switching frequency, this condition is present, regardless of the topology. However, using a constant off-time approach, current mode instability cannot occur, enabling easier design and control. Predictive off-time advantages: • There is no current mode instability at any duty cycle. • Higher duty cycles / voltage transformation ratios are possible, especially in the boost regulator. The only disadvantage is that synchronization to an external reference frequency is generally not available. Buck Boost SWITCHING FREQUENCY An external resistor (RT) connected between the RCT pin and the switch node (where D1, Q1, and L1 connect), in combination with a capacitor (CT) between the RCT and AGND pins, sets the off-time (tOFF) as shown in Figure 2. For boost and buck-boost topologies, the VIN proportionality ensures a virtually constant switching frequency (fSW). Buck-boost PREDICTIVE OFF-TIME (PRO) CONTROL PRO control is used by the LM3429 to control ILED. It is a combination of average peak current control and a one-shot off-timer that varies with input voltage. The LM3429 uses peak current control to regulate the average LED current through an array of HBLEDs. This method of control uses a series resistor in the LED path to sense LED current and can use either a series resistor in the MosFET path or the MosFET RDS-ON for both cycle-by-cycle current limit and input voltage feed forward. D is indirectly controlled by changes in both tOFF and tON, which vary depending on the operating point. Even though the off-time control is quasi-hysteretic, the input voltage proportionality in the off-timer creates an essentially constant switching frequency over the entire operating range for boost and buck-boost topologies. The buck topology can be designed to give constant ripple over either input voltage or output voltage, however switching frequency is only constant at a specific operating point . This type of control minimizes the control loop compensation necessary in many switching regulators, simplifying the design process. The averaging mechanism in the peak detection control loop provides extremely accurate LED current regulation over the entire operating range. www.national.com 30094499 FIGURE 2. Off-timer Circuitry for Boost and Buck-boost Regulators For a buck topology, RT and CT are also used to set tOFF, however the VIN proportionality will not ensure a constant switching frequency. Instead, constant ripple operation can be achieved. Changing the connection of RT in Figure 2 from VSW to VIN will provide a constant ripple over varying VIN. 8 AVERAGE LED CURRENT The LM3429 uses an external current sense resistor (RSNS) placed in series with the LED load to convert the LED current (ILED) into a voltage (VSNS) as shown in Figure 4. The HSP and HSN pins are the inputs to the high-side sense amplifier which are forced to be equal potential (VHSP=VHSN) through negative feedback. Because of this, the VSNS voltage is forced across RHSP to generate the signal current (ICSH) which flows out of the CSH pin and through the RCSH resistor. The error amplifier will regulate the CSH pin to 1.24V, therefore ICSH can be calculated: This means VSNS will be regulated as follows: 300944a0 ILED can then be calculated: FIGURE 3. Off-timer Circuitry for Buck Regulators The switching frequency is defined: Buck (Constant Ripple vs. VIN) The selection of the three resistors (RSNS, RCSH, and RHSP) is not arbitrary. For matching and noise performance, the suggested signal current ICSH is approximately 100 µA. This current does not flow in the LEDs and will not affect either the off state LED current or the regulated LED current. ICSH can be above or below this value, but the high-side amplifier offset characteristics may be affected slightly. In addition, to minimize the effect of the high-side amplifier voltage offset on LED current accuracy, the minimum VSNS is suggested to be 50 mV. Finally, a resistor (RHSN = RHSP) should be placed in series with the HSN pin to cancel out the effects of the input bias current (~10 µA) of both inputs of the high-side sense amplifier. Note that he CSH pin can also be used as a lowside current sense input regulated to the 1.24V. The high-side sense amplifier is disabled if HSP and HSN are tied to GND. Buck (Constant Ripple vs. VO) Boost and Buck-boost For all topologies, the CT capacitor is recommended to be 1 nF and should be located very close to the LM3429. 30094457 FIGURE 4. LED Current Sense Circuitry 9 www.national.com LM3429 Adding a PNP transistor as shown in Figure 3 will provide constant ripple over varying VO. LM3429 ANALOG DIMMING The CSH pin can be used to analog dim the LED current by adjusting the current sense voltage (VSNS). There are several different methods to adjust VSNS using the CSH pin: 1. External variable resistance : Adjust a potentiometer placed in series with RCSH to vary VSNS. 2. External variable current source: Source current (0 µA to ICSH) into the CSH pin to adjust VSNS. In general, analog dimming applications require a lower switching frequency to minimize the effect of the leading edge blanking circuit. As the LED current is reduced, the output voltage and the duty cycle decreases. Eventually, the minimum on-time is reached. The lower the switching frequency, the wider the linear dimming range. Figure 5 shows how both methods are physically implemented. Method 1 uses an external potentiometer in the CSH path which is a simple addition to the existing circuitry. However, the LEDs cannot dim completely because there is always some resistance causing signal current to flow. This method is also susceptible to noise coupling at the CSH pin since the potentiometer increases the size of the signal current loop. Method 2 provides a complete dimming range and better noise performance, though it is more complex. It consists of a PNP current mirror and a bias network consisting of an NPN, 2 resistors and a potentiometer (RADJ), where RADJ controls the amount of current sourced into the CSH pin. A higher resistance value will source more current into the CSH pin causing less regulated signal current through RHSP, effectively dimming the LEDs. VREF should be a precise external voltage reference, while Q7 and Q8 should be a dual pair PNP for best matching and performance. The additional current (IADD) sourced into the CSH pin can be calculated: CURRENT SENSE/CURRENT LIMIT The LM3429 achieves peak current mode control using a comparator that monitors the MosFET transistor current, comparing it with the COMP pin voltage as shown in Figure 6. Further, it incorporates a cycle-by-cycle over-current protection function. Current limit is accomplished by a redundant internal current sense comparator. If the voltage at the current sense comparator input (IS) exceeds 245 mV (typical), the on cycle is immediately terminated. The IS input pin has an internal N-channel MosFET which pulls it down at the conclusion of every cycle. The discharge device remains on an additional 250 ns (typical) after the beginning of a new cycle to blank the leading edge spike on the current sense signal. The leading edge blanking (LEB) determines the minimum achievable on-time (tON-MIN). 300944a2 FIGURE 6. Current Sense / Current Limit Circuitry There are two possible methods to sense the transistor current. The RDS-ON of the main power MosFET can be used as the current sense resistance because the IS pin was designed to withstand the high voltages present on the drain when the MosFET is in the off state. Alternatively, a sense resistor located in the source of the MosFET may be used for current sensing, however a low inductance (ESL) type is suggested. The cycle-by-cycle current limit (ILIM) can be calulated using either method as the limiting resistance (RLIM): The corresponding ILED for a specific IADD is: In general, the external series resistor allows for more design flexibility, however it is important to ensure all of the noise sensitive low power ground connections are connected together local to the controller and a single connection is made to the high current PGND (sense resistor ground point). 300944a1 FIGURE 5. Analog Dimming Circuitry www.national.com 10 LM3429 And the right half plane zero (ωZ1) is: CONTROL LOOP COMPENSATION The LM3429 control loop is modeled like any current mode controller. Using a first order approximation, the uncompensated loop can be modeled as a single pole created by the output capacitor and, in the boost and buck-boost topologies, a right half plane zero created by the inductor, where both have a dependence on the LED string dynamic resistance. There is also a high frequency pole in the model, however it is above the switching frequency and plays no part in the compensation design process therefore it will be neglected. Since ceramic capacitance is recommended for use with LED drivers due to long lifetimes and high ripple current rating, the ESR of the output capacitor can also be neglected in the loop analysis. Finally, there is a DC gain of the uncompensated loop which is dependent on internal controller gains and the external sensing network. A buck-boost regulator will be used as an example case. See the Design Guide section for compensation of all topologies. The uncompensated loop gain for a buck-boost regulator is given by the following equation: 300944a7 FIGURE 7. Uncompensated Loop Gain Frequency Response Figure 7 shows the uncompensated loop gain in a worst-case scenario when the RHP zero is below the output pole. This occurs at high duty cycles when the regulator is trying to boost the output voltage significantly. The RHP zero adds 20dB/ decade of gain while loosing 45°/decade of phase which places the crossover frequency (when the gain is zero dB) extremely high because the gain only starts falling again due to the high frequency pole (not modeled or shown in figure). The phase will be below -180° at the crossover frequency which means there is no phase margin (180° + phase at crossover frequency) causing system instability. Even if the output pole is below the RHP zero, the phase will still reach -180° before the crossover frequency in most cases yielding instability. Where the uncompensated DC loop gain of the system is described as: And the output pole (ωP1) is approximated: 300944a3 FIGURE 8. Compensation Circuitry 11 www.national.com LM3429 To mitigate this problem, a compensator should be designed to give adequate phase margin (above 45°) at the crossover frequency. A simple compensator using a single capacitor at the COMP pin (CCMP) will add a dominant pole to the system, which will ensure adequate phase margin if placed low enough. At high duty cycles (as shown in Figure 7), the RHP zero places extreme limits on the achievable bandwidth with this type of compensation. However, because an LED driver is essentially free of output transients (except catastrophic failures open or short), the dominant pole approach, even with reduced bandwidth, is usually the best approach. The dominant compensation pole (ωP2) is determined by CCMP and the output resistance (RO) of the error amplifier (typically 5 MΩ): OUTPUT OVER-VOLTAGE LOCKOUT (OVLO) 30094458 FIGURE 10. Over-Voltage Protection Circuitry The LM3429 can be configured to detect an output (or input) over-voltage condition via the OVP pin. The pin features a precision 1.24V threshold with 20 µA (typical) of hysteresis current as shown in Figure 10. When the OVLO threshold is exceeded, the GATE pin is immediately pulled low and a 20 µA current source provides hysteresis to the lower threshold of the OVLO hysteretic band. If the LEDs are referenced to a potential other than ground (floating), as in the buck-boost and buck configuration, the output voltage (VO) should be sensed and translated to ground by using a single PNP as shown in Figure 11. It may also be necessary to add one final pole at least one decade above the crossover frequency to attenuate switching noise and, in some cases, provide better gain margin. This pole can be placed across RSNS to filter the ESL of the sense resistor at the same time. Figure 8 shows how the compensation is physically implemented in the system. The high frequency pole (ωP3) can be calculated: The total system transfer function becomes: The resulting compensated loop gain frequency response shown in Figure 9 indicates that the system has adequate phase margin (above 45°) if the dominant compensation pole is placed low enough, ensuring stability: 30094459 FIGURE 11. Floating Output OVP Circuitry The over-voltage turn-off threshold (VTURN-OFF) is defined as follows: Ground Referenced Floating In the ground referenced configuration, the voltage across ROV2 is VO - 1.24V whereas in the floating configuration it is VO - 620 mV where 620 mV approximates the VBE of the PNP transistor. The over-voltage hysteresis (VHYSO) is defined as follows: 300944a4 FIGURE 9. Compensated Loop Gain Frequency Response www.national.com 12 LM3429 INPUT UNDER-VOLTAGE LOCKOUT (UVLO) The nDIM pin is a dual-function input that features an accurate 1.24V threshold with programmable hysteresis as shown in Figure 12. This pin functions as both the PWM dimming input for the LEDs and as a VIN UVLO. When the pin voltage rises and exceeds the 1.24V threshold, 20 µA (typical) of current is driven out of the nDIM pin into the resistor divider providing programmable hysteresis. 300944a6 FIGURE 13. PWM Dimming Circuit Figure 13 shows two ways the PWM signal can be applied to the nDIM pin: 1. Connect the dimming MosFET (QDIM) with the drain to the nDIM pin and the source to GND. Apply an external logic-level PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off QDIM if no signal is present. 2. Connect the anode of a Schottky diode (DDIM) to the nDIM pin. Apply an external inverted logic-level PWM signal to the cathode of the same diode. A minimum on-time must be maintained in order for PWM dimming to operate in the linear region of its transfer function. Because the controller is disabled during dimming, the PWM pulse must be long enough such that the energy intercepted from the input is greater than or equal to the energy being put into the LEDs. For boost and buck-boost regulators, the following condition must be maintained: 300944a5 FIGURE 12. UVLO Circuit When using the nDIM pin for UVLO and PWM dimming concurrently, the UVLO circuit can have an extra series resistor to set the hysteresis. This allows the standard resistor divider to have smaller resistor values minimizing PWM delays due to a pull-down MosFET at the nDIM pin (see PWM Dimming section). In general, at least 3V of hysteresis is necessary when PWM dimming if operating near the UVLO threshold. The turn-on threshold (VTURN-ON) is defined as follows: The hysteresis (VHYS) is defined as follows: UVLO only In the previous equation, tPULSE is the length of the PWM pulse in seconds. PWM dimming and UVLO STARTUP REGULATOR (VCC LDO) The LM3429 includes a high voltage, low dropout (LDO) bias regulator. When power is applied, the regulator is enabled and sources current into an external capacitor connected to the VCC pin. The VCC output voltage is 6.9V nominally and the supply is internally current limited to 20 mA (minimum). The recommended bypass capacitance range for the VCC regulator is 2.2 µF to 3.3 µF. The output of the VCC regulator is monitored by an internal UVLO circuit that protects the device during startup, normal operation, and shutdown from attempting to operate with insufficient supply voltage. PWM DIMMING The active low nDIM pin can be driven with a PWM signal which controls the main NFET (Q1). The brightness of the LEDs can be varied by modulating the duty cycle of this signal. LED brightness is approximately proportional to the PWM signal duty cycle, so 30% duty cycle equals approximately 30% LED brightness. This function can be ignored if PWM dimming is not required by using nDIM solely as a VIN UVLO input as described in the Input Under-Voltage Lockout section or by tying it directly to VCC or VIN (if less than 76VDC). THERMAL SHUTDOWN The LM3429 includes thermal shutdown. If the die temperature reaches approximately 165°C the device will shut down (GATE pin low), until it reaches approximately 140°C where it turns on again. 13 www.national.com LM3429 Obtaining rLED is accomplished by refering to the manufacturer's LED I-V characteristic. It can be calculated as the slope at the nominal operating point as shown in Figure 14. For any application with more than 2 series LEDs, RSNS can be neglected allowing rD to be approximated as the number of LEDs multiplied by rLED. Design Considerations This section describes the application level considerations when designing with the LM3429. For corresponding calculations, refer to the Design Guide section. INDUCTOR The inductor (L1) is the main energy storage device in a switching regulator. Depending on the topology, energy is stored in the inductor and transfered to the load in different ways (as an example, buck-boost operation is detailed in the Current Regulators section). The size of the inductor, the voltage across it, and the length of the switching subinterval (tON or tOFF) determines the inductor current ripple (ΔiL-PP ). In the design process, L1 is chosen to provide a desired ΔiL-PP. For a buck regulator the inductor has a direct connection to the load, which is good for a current regulator. This requires little to no output capacitance therefore ΔiL-PP is basically equal to the LED ripple current ΔiLED-PP. However, for boost and buck-boost regulators, there is always an output capacitor which reduces ΔiLED-PP, therefore the inductor ripple can be larger than in the buck regulator case where output capacitance is minimal or completely absent. In general, ΔiLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED). Therefore, for the buck regulator with no output capacitance, ΔiL-PP should also be less than 40% of ILED. For the boost and buckboost topologies, ΔiL-PP can be much higher depending on the output capacitance value. However, ΔiL-PP is suggested to be less than 100% of the average inductor current (IL) to limit the RMS inductor current. L1 is also suggested to have an RMS current rating at least 25% higher than the calculated minimum allowable RMS inductor current (IL-RMS). OUTPUT CAPACITOR For boost and buck-boost regulators, the output capacitor (CO) provides energy to the load when the recirculating diode (D1) is reverse biased during the first switching subinterval. An output capacitor in a buck topology will simply reduce the LED current ripple (ΔiLED-PP) below the inductor current ripple (ΔiL-PP). In all cases, CO is sized to provide a desired ΔiLEDPP. As mentioned in the Inductor section, ΔiLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED-PP). CO should be carefully chosen to account for derating due to temperature and operating voltage. It must also have the necessary RMS current rating. Ceramic capacitors are the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested. INPUT CAPACITORS The input capacitance (CIN) provides energy during the discontinuous portions of the switching period. For buck and buck-boost regulators, CIN provides energy during tON and during tOFF, the input voltage source charges up CIN with the average input current (IIN). For boost regulators, CIN only needs to provide the ripple current due to the direct connection to the inductor. CIN is selected given the maximum input voltage ripple (ΔvIN-PP) which can be tolerated. ΔvIN-PP is suggested to be less than 10% of the nominal input voltage (VIN). An input capacitance at least 100% greater than the calculated CIN value is recommended to account for derating due to temperature and operating voltage. It must also have the necessary RMS current rating. Ceramic capacitors are again the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested. LED DYNAMIC RESISTANCE (rD) When the load is a string of LEDs, the output load resistance is the LED string dynamic resistance plus RSNS. LEDs are PN junction diodes, and their dynamic resistance shifts as their forward current changes. Dividing the forward voltage of a single LED (VLED) by the forward current (ILED) leads to an incorrect calculation of the dynamic resistance of a single LED (rLED). The result can be 5 to 10 times higher than the true rLED value. N-CHANNEL MosFET (NFET) The LM3429 requires an external NFET (Q1) as the main power MosFET for the switching regulator. Q1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node. In practice, all switching regulators have some ringing at the switch node due to the diode parasitic capacitance and the lead inductance. The current rating is recommended to be at least 10% higher than the average transistor current. The power rating is then verified by calculating the power loss given the RMS transistor current and the NFET on-resistance (RDS-ON). In general, the NFET should be chosen to minimize total gate charge (Qg) whenever switching frequencies are high and minimize RDS-ON otherwise. This will minimize the dominant power losses in the system. Frequently, higher current NFETs in larger packages are chosen for better thermal performance. 30094474 FIGURE 14. Dynamic Resistance www.national.com 14 CIRCUIT LAYOUT The performance of any switching regulator depends as much upon the layout of the PCB as the component selection. Following a few simple guidelines will maximimize noise rejection and minimize the generation of EMI within the circuit. 15 www.national.com LM3429 Discontinuous currents are the most likely to generate EMI, therefore care should be taken when routing these paths. The main path for discontinuous current in the LM3429 buck regulator contains the input capacitor (CIN), the recirculating diode (D1), the N-channel MosFET (Q1), and the sense resistor (RLIM). In the LM3429 boost and buck-boost regulators, the discontinuous current flows through the output capacitor (CO), D1, Q1, and RLIM. In either case, this loop should be kept as small as possible and the connections between all the components should be short and thick to minimize parasitic inductance. In particular, the switch node (where L1, D1 and Q1 connect) should be just large enough to connect the components. To minimize excessive heating, large copper pours can be placed adjacent to the short current path of the switch node. The RCT, COMP, CSH, IS, HSP and HSN pins are all highimpedance inputs which couple external noise easily, therefore the loops containing these nodes should be minimized whenever possible. In some applications the LED or LED array can be far away (several inches or more) from the LM3429, or on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or separated from the rest of the regulator, the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. RE-CIRCULATING DIODE A re-circulating diode (D1) is required to carry the inductor current during tOFF. The most efficient choice for D1 is a Schottky diode due to low forward voltage drop and near-zero reverse recovery time. Similar to Q1, D1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node and a current rating at least 10% higher than the average diode current. The power rating is verified by calculating the power loss through the diode. This is accomplished by checking the typical diode forward voltage from the I-V curve on the product datasheet and multiplying by the average diode current. In general, higher current diodes have a lower forward voltage and come in better performing packages minimizing both power losses and temperature rise. LM3429 Basic Topology Schematics BOOST REGULATOR (VIN < VO) 30094422 BUCK REGULATOR (VIN > VO) 30094451 www.national.com 16 LM3429 BUCK-BOOST REGULATOR 30094450 17 www.national.com LM3429 Buck (Constant Ripple vs. VO) Design Guide Refer to Basic Topology Schematics section. SPECIFICATIONS Number of series LEDs: N Single LED forward voltage: VLED Single LED dynamic resistance: rLED Nominal input voltage: VIN Input voltage range: VIN-MAX, VIN-MIN Switching frequency: fSW Current sense voltage: VSNS Average LED current: ILED Inductor current ripple: ΔiL-PP LED current ripple: ΔiLED-PP Peak current limit: ILIM Input voltage ripple: ΔvIN-PP Output OVLO characteristics: VTURN-OFF, VHYSO Input UVLO characteristics: VTURN-ON, VHYS Boost and Buck-boost 3. AVERAGE LED CURRENT For all topologies, set the average LED current (ILED) knowing the desired current sense voltage (VSNS) and solving for RSNS: If the calculated RSNS is too far from a desired standard value, then VSNS will have to be adjusted to obtain a standard value. Setup the suggested signal current of 100 µA by assuming RCSH = 12.4 kΩ and solving for RHSP: 1. OPERATING POINT Given the number of series LEDs (N), the forward voltage (VLED) and dynamic resistance (rLED) for a single LED, solve for the nominal output voltage (VO) and the nominal LED string dynamic resistance (rD): If the calculated RHSP is too far from a desired standard value, then RCSH can be adjusted to obtain a standard value. Solve for the ideal nominal duty cycle (D): 4. INDUCTOR RIPPLE CURRENT Set the nominal inductor ripple current (ΔiL-PP) by solving for the appropriate inductor (L1): Buck Buck Boost Boost and Buck-boost Buck-boost To set the worst case inductor ripple current, use VIN-MAX and DMIN when solving for L1. The minimum allowable inductor RMS current rating (IL-RMS) can be calculated as: Using the same equations, find the minimum duty cycle (DMIN) using maximum input voltage (VIN-MAX) and the maximum duty cycle (DMAX) using the minimum input voltage (VINMIN). Also, remember that D' = 1 - D. Buck 2. SWITCHING FREQUENCY Set the switching frequency (fSW) by assuming a CT value of 1 nF and solving for RT: Boost and Buck-boost Buck (Constant Ripple vs. VIN) www.national.com 18 LM3429 Where the pole (ωP1) is approximated: 5. LED RIPPLE CURRENT Set the nominal LED ripple current (ΔiLED-PP), by solving for the output capacitance (CO): Buck Buck Boost Boost and Buck-boost Buck-boost To set the worst case LED ripple current, use DMAX when solving for CO. The minimum allowable RMS output capacitor current rating (ICO-RMS) can be approximated: And the RHP zero (ωZ1) is approximated: Boost Buck Buck-boost Boost and Buck-boost And the uncompensated DC loop gain (TU0) is approximated: 6. PEAK CURRENT LIMIT Set the peak current limit (ILIM) by solving for the transistor path sense resistor (RLIM): Buck Boost 7. LOOP COMPENSATION Using a simple first order peak current mode control model, neglecting any output capacitor ESR dynamics, the necessary loop compensation can be determined. First, the uncompensated loop gain (T U) of the regulator can be approximated: Buck-boost Buck For all topologies, the primary method of compensation is to place a low frequency dominant pole (ωP2) which will ensure that there is ample phase margin at the crossover frequency. This is accomplished by placing a capacitor (CCMP) from the COMP pin to GND, which is calculated according to the lower value of the pole and the RHP zero of the system (shown as a minimizing function): Boost and Buck-boost If analog dimming is used, CCMP should be approximately 4x larger to maintain stability as the LEDs are dimmed to zero. 19 www.national.com LM3429 A high frequency compensation pole (ωP3) can be used to attenuate switching noise and provide better gain margin. Assuming RFS = 10Ω, CFS is calculated according to the higher value of the pole and the RHP zero of the system (shown as a maximizing function): Buck-boost 9. NFET The NFET voltage rating should be at least 15% higher than the maximum NFET drain-to-source voltage (VT-MAX): Buck The total system loop gain (T) can then be written as: Boost Buck Buck-boost The current rating should be at least 10% higher than the maximum average NFET current (IT-MAX): Boost and Buck-boost Buck Boost and Buck-boost 8. INPUT CAPACITANCE Set the nominal input voltage ripple (ΔvIN-PP) by solving for the required capacitance (CIN): Approximate the nominal RMS transistor current (IT-RMS) : Buck Buck Boost Boost and Buck-boost Buck-boost Given an NFET with on-resistance (RDS-ON), solve for the nominal power dissipation (PT): 10. DIODE The Schottky diode voltage rating should be at least 15% higher than the maximum blocking voltage (VRD-MAX): Use DMAX to set the worst case input voltage ripple, when solving for CIN in a buck-boost regulator and DMID = 0.5 when solving for CIN in a buck regulator. The minimum allowable RMS input current rating (ICIN-RMS) can be approximated: Buck Buck Boost Boost Buck-boost www.national.com 20 12. INPUT UVLO For all topologies, input UVLO is programmed with the turnon threshold voltage (VTURN-ON) and the desired hysteresis (VHYS). Method #1: If no PWM dimming is required, a two resistor network can be used. To set VHYS, solve for RUV2: Buck Boost and Buck-boost Replace DMAX with D in the ID-MAX equation to solve for the average diode current (ID). Given a diode with forward voltage (VFD), solve for the nominal power dissipation (PD): To set VTURN-ON, solve for RUV1: Method #2: If PWM dimming is required, a three resistor network is suggested. To set VTURN-ON, assume RUV2 = 10 kΩ and solve for RUV1 as in Method #1. To set VHYS, solve for RUVH: 11. OUTPUT OVLO For boost and buck-boost regulators, output OVLO is programmed with the turn-off threshold voltage (VTURN-OFF) and the desired hysteresis (VHYSO). To set VHYSO, solve for ROV2: 13. PWM DIMMING METHOD PWM dimming can be performed several ways: Method #1: Connect the dimming MosFET (Q3) with the drain to the nDIM pin and the source to GND. Apply an external PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off Q3. Method #2: Connect the anode of a Schottky diode to the nDIM pin. Apply an external inverted PWM signal to the cathode of the same diode. To set VTURN-OFF, solve for ROV1: Boost Buck-boost 14. ANALOG DIMMING METHOD Analog dimming can be performed several ways: Method #1: Place a potentiometer in series with the RCSH resistor to dim the LED current from the nominal ILED to near zero. Method #2: Connect a controlled current source as detailed in the Analog Dimming section to the CSH pin. Increasing the current sourced into the CSH node will decrease the LEDs from the nominal ILED to zero current. A small filter capacitor (COVP = 47 nF) should be added from the OVP pin to ground to reduce coupled switching noise. 21 www.national.com LM3429 The current rating should be at least 10% higher than the maximum average diode current (ID-MAX): LM3429 Design Example #1 BUCK-BOOST APPLICATION - 6 LEDs at 1A 300944i1 Solve for D, D', DMAX, and DMIN: SPECIFICATIONS N=6 VLED = 3.5V rLED = 325 mΩ VIN = 24V VIN-MIN = 10V VIN-MAX = 70V fSW = 700 kHz VSNS = 100 mV ILED = 1A ΔiL-PP = 500 mA ΔiLED-PP = 50 mA ΔvIN-PP = 1V ILIM = 6A VTURN-ON = 10V VHYS = 3V VTURN-OFF = 40V VHYSO = 10V 2. SWITCHING FREQUENCY Assume CT = 1 nF and solve for RT: 1. OPERATING POINT Solve for VO and rD: The closest standard resistor is actually 35.7 kΩ therefore the fSW is: The chosen components from step 2 are: www.national.com 22 LM3429 5. OUTPUT CAPACITANCE Solve for CO: 3. AVERAGE LED CURRENT Solve for RSNS: Assume RCSH = 12.4 kΩ and solve for RHSP: The closest standard capacitor is 6.8 µF therefore the actual ΔiLED-PP is: The closest standard resistor for RSNS is actually 0.1Ω and for RHSP is actually 1 kΩ therefore ILED is: The chosen components from step 3 are: Determine minimum allowable RMS current rating: The chosen components from step 5 are: 4. INDUCTOR RIPPLE CURRENT Solve for L1: 6. PEAK CURRENT LIMIT Solve for RLIM: The closest standard inductor is 33 µH therefore the actual ΔiL-PP is: The closest standard resistor is 0.04 Ω therefore ILIM is: Determine minimum allowable RMS current rating: The chosen component from step 6 is: 7. LOOP COMPENSATION ωP1 is approximated: The chosen component from step 4 is: ωZ1 is approximated: 23 www.national.com LM3429 TU0 is approximated: 9. NFET Determine minimum Q1 voltage rating and current rating: To ensure stability, calculate ωP2: A 100V NFET is chosen with a current rating of 32A due to the low RDS-ON = 50 mΩ. Determine IT-RMS and PT: Solve for CCMP: The chosen component from step 9 is: To attenuate switching noise, calculate ωP3: 10. DIODE Determine minimum D1 voltage rating and current rating: Assume RFS = 10Ω and solve for CFS: A 100V diode is chosen with a current rating of 12A and VD = 600 mV. Determine PD: The chosen components from step 7 are: The chosen component from step 10 is: 8. INPUT CAPACITANCE Solve for the minimum CIN: 11. INPUT UVLO Solve for RUV2: To minimize power supply interaction a 200% larger capacitance of approximately 14 µF is used, therefore the actual ΔvIN-PP is much lower. Since high voltage ceramic capacitor selection is limited, three 4.7 µF X7R capacitors are chosen. Determine minimum allowable RMS current rating: The closest standard resistor is 150 kΩ therefore VHYS is: Solve for RUV1: The chosen components from step 8 are: www.national.com 24 The closest standard resistor is 499 kΩ therefore VHYSO is: Solve for ROV1: The chosen components from step 11 are: The closest standard resistor is 15.8 kΩ making VTURN-OFF: 12. OUTPUT OVLO Solve for ROV2: The chosen components from step 12 are: Design #1 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 0.22 µF X7R 10% 25V MURATA GRM21BR71E224KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 3 CIN 4.7 µF X7R 10% 100V TDK C5750X7R2A475K 1 CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100V 12A VISHAY 12CWQ10FNPBF 1 L1 33 µH 20% 6.3A COILCRAFT MSS1278-333MLB 1 Q1 NMOS 100V 32A FAIRCHILD FDD3682 1 Q2 PNP 150V 600 mA FAIRCHILD MMBT5401 1 RCSH 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0kΩ 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04Ω 1% 1W VISHAY WSL2512R0400FEA 1 ROV1 15.8 kΩ 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1Ω 1% 1W VISHAY WSL2512R1000FEA 1 RT 35.7 kΩ 1% VISHAY CRCW080535K7FKEA 1 RUV1 21 kΩ 1% VISHAY CRCW080521K0FKEA 1 RUV2 150 kΩ 1% VISHAY CRCW0805150KFKEA 25 www.national.com LM3429 The closest standard resistor is 21 kΩ making VTURN-ON: LM3429 Applications Information DESIGN #2: BOOST PWM DIMMING APPLICATION - 9 LEDs at 1A 300944h5 Design #2 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 2 CCMP, CFS 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 2, 1 CIN, CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60V 5A COMCHIP CDBC560-G 1 L1 33 µH 20% 6.3A COILCRAFT MSS1278-333MLB 1 Q1 NMOS 60V 8A VISHAY SI4436DY 1 Q2 NMOS 60V 115 mA ON SEMI 2N7002ET1G 2 RCSH, ROV1 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0 kΩ 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.06Ω 1% 1W VISHAY WSL2512R0600FEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1Ω 1% 1W VISHAY WSL2512R1000FEA 1 RT 35.7 kΩ 1% VISHAY CRCW080535K7FKEA 1 RUV1 1.82 kΩ 1% VISHAY CRCW08051K82FKEA 1 RUV2 10 kΩ 1% VISHAY CRCW080510KFKEA 1 RUVH 17.8 kΩ 1% VISHAY CRCW080517K8FKEA www.national.com 26 LM3429 DESIGN #3: BUCK-BOOST ANALOG DIMMING APPLICATION - 4 LEDs at 2A 300944h6 Design #3 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 1.0 µF X7R 10% 10V MURATA GRM21BR71A105KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 50V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60V 5A VISHAY CDBC560-G 1 L1 22 µH 20% 7.2A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 60V 8A VISHAY SI4436DY 1 Q2 PNP 150V 600 mA FAIRCHILD MMBT5401 1 RADJ 1.0 MΩ potentiometer BOURNS 3352P-1-105 1 RCSH 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0 kΩ 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04Ω 1% 1W VISHAY WSL2512R0400FEA 1 ROV1 18.2 kΩ 1% VISHAY CRCW080518K2FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.05Ω 1% 1W VISHAY WSL2512R0500FEA 1 RT 41.2 kΩ 1% VISHAY CRCW080541K2FKEA 1 RUV1 21 kΩ 1% VISHAY CRCW080521K0FKEA 1 RUV2 150 kΩ 1% VISHAY CRCW0805150KFKEA 27 www.national.com LM3429 DESIGN #4: BOOST ANALOG DIMMING APPLICATION - 12 LEDs at 700mA 300944h7 Design #4 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 1.0 µF X7R 10% 10V MURATA GRM21BR71A105KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 50V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100V 12A VISHAY 12CWQ10FNPBF 1 L1 47 µH 20% 5.3A COILCRAFT MSS1278-473MLB 1 Q1 NMOS 100V 32A FAIRCHILD FDD3682 1 Q2 NPN 40V 200 mA FAIRCHILD MMBT3904 1 Q3, Q4 (dual pack) Dual PNP 40V 200 mA FAIRCHILD FFB3906 1 RADJ 100 kΩ potentiometer BOURNS 3352P-1-104 1 RBIAS 40.2 kΩ 1% VISHAY CRCW080540K2FKEA 1 RCSH, ROV1, RUV1 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.05 kΩ 1% VISHAY CRCW08051K05FKEA 1 RLIM 0.06Ω 1% 1W VISHAY WSL2512R0600FEA 1 RMAX 4.99 kΩ 1% VISHAY CRCW08054K99FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.15Ω 1% 1W VISHAY WSL2512R1500FEA 1 RT 35.7 kΩ 1% VISHAY CRCW080535K7FKEA 1 RUV2 100 kΩ 1% VISHAY CRCW0805100KFKEA 1 VREF 5V precision reference NSC LM4040 www.national.com 28 LM3429 DESIGN #5: BUCK-BOOST PWM DIMMING APPLICATION - 6 LEDs at 500mA 300944h9 Design #5 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 0.68 µF X7R 10% 25V MURATA GRM21BR71E684KA88L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 3 CIN 4.7 µF X7R 10% 100V TDK C5750X7R2A475K 1 CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100V 12A VISHAY 12CWQ10FNPBF 1 D2 Schottky 30V 500 mA ON SEMI BAT54T1G 1 L1 68 µH 20% 4.3A COILCRAFT MSS1278-683MLB 1 Q1 NMOS 100V 32A VISHAY FDD3682 1 Q2 PNP 150V 600 mA FAIRCHILD MMBT5401 1 RCSH 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0 kΩ 1% VISHAY CRCW08051K00FKEA 1 ROV1 15.8 kΩ 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.2Ω 1% 1W VISHAY WSL2512R2000FEA 1 RT 35.7 kΩ 1% VISHAY CRCW080535K7FKEA 1 RUV1 1.43 kΩ 1% VISHAY CRCW08051K43FKEA 1 RUV2 10 kΩ 1% VISHAY CRCW080510K0FKEA 1 RUVH 17.4 kΩ 1% VISHAY CRCW080517K4FKEA 29 www.national.com LM3429 DESIGN #6: BUCK APPLICATION - 3 LEDS AT 1.25A 300944h8 Design #6 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 0.015 µF X7R 10% 50V MURATA GRM21BR71H153KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.01 µF X7R 10% 50V MURATA GRM21BR71H103KA01L 2 CIN 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 CO 1 µF X7R 10% 50V TDK C4532X7R1H105K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60V 5A COMCHIP CDBC560-G 1 L1 22 µH 20% 7.3A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 60V 8A VISHAY SI4436DY 1 Q2 PNP 150V 600 mA FAIRCHILD MMBT5401 1 RCSH 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RT 49.9 kΩ 1% VISHAY CRCW080549K9FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0 kΩ 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04Ω 1% 1W VISHAY WSL2512R0400FEA 1 ROV1 21.5 kΩ 1% VISHAY CRCW080521K5FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.08Ω 1% 1W VISHAY WSL2512R0800FEA 1 RUV1 11.5 kΩ 1% VISHAY CRCW080511K5FKEA 1 RUV2 100 kΩ 1% VISHAY CRCW0805100KFKEA www.national.com 30 LM3429 DESIGN #7: BUCK-BOOST THERMAL FOLDBACK APPLICATION - 8 LEDs at 2.5A 300944i0 Design #7 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 3 CIN 4.7 µF X7R 10% 100V TDK C5750X7R2A475K 1 CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100V 12A VISHAY 12CWQ10FNPBF 1 L1 22 µH 20% 7.2A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 100V 32A FAIRCHILD FDD3682 1 Q2 PNP 150V 600 mA FAIRCHILD MMBT5401 2 RCSH, ROV1 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.0 kΩ 1% VISHAY CRCW08051K00FKEA 2 RLIM, RSNS 0.04Ω 1% 1W VISHAY WSL2512R0400FEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RT 49.9 kΩ 1% VISHAY CRCW080549K9FKEA 1 RUV1 13.7 kΩ 1% VISHAY CRCW080513K7FKEA 1 RUV2 150 kΩ 1% VISHAY CRCW0805150KFKEA 31 www.national.com LM3429 DESIGN #8: SEPIC APPLICATION - 5 LEDs at 750mA 300944i8 Design #8 Bill of Materials Qty Part ID Part Value Manufacturer Part Number 1 LM3429 Boost controller NSC LM3429MH 1 CCMP 0.47 µF X7R 10% 25V MURATA GRM21BR71E474KA01L 1 CF 2.2 µF X7R 10% 16V MURATA GRM21BR71C225KA12L 1 CFS 0.1 µF X7R 10% 25V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 µF X7R 10% 50V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60V 5A COMCHIP CDBC560-G 1 L1, L2 68 µH 20% 4.3A COILCRAFT DO3340P-683 1 Q1 NMOS 60V 8A VISHAY SI4436DY 1 Q2 NMOS 60V 115 mA ON SEMI 2N7002ET1G 1 RCSH 12.4 kΩ 1% VISHAY CRCW080512K4FKEA 1 RFS 10Ω 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 750Ω 1% VISHAY CRCW0805750RFKEA 1 RLIM 0.04Ω 1% 1W VISHAY WSL2512R0400FEA 2 ROV1, RUV1 15.8 kΩ 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 kΩ 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1Ω 1% 1W VISHAY WSL2512R1000FEA 1 RT 49.9 kΩ 1% VISHAY CRCW080549K9FKEA 1 RUV2 100 kΩ 1% VISHAY CRCW0805100KFKEA www.national.com 32 LM3429 Physical Dimensions inches (millimeters) unless otherwise noted TSSOP-14 Pin EP Package (MXA) For Ordering, Refer to Ordering Information Table NS Package Number MXA14A 33 www.national.com LM3429 N-Channel Controller for Constant Current LED Drivers Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Design Support Amplifiers www.national.com/amplifiers WEBENCH® Tools www.national.com/webench Audio www.national.com/audio App Notes www.national.com/appnotes Clock and Timing www.national.com/timing Reference Designs www.national.com/refdesigns Data Converters www.national.com/adc Samples www.national.com/samples Interface www.national.com/interface Eval Boards www.national.com/evalboards LVDS www.national.com/lvds Packaging www.national.com/packaging Power Management www.national.com/power Green Compliance www.national.com/quality/green Switching Regulators www.national.com/switchers Distributors www.national.com/contacts LDOs www.national.com/ldo Quality and Reliability www.national.com/quality LED Lighting www.national.com/led Feedback/Support www.national.com/feedback Voltage Reference www.national.com/vref Design Made Easy www.national.com/easy www.national.com/powerwise Solutions www.national.com/solutions Mil/Aero www.national.com/milaero PowerWise® Solutions Serial Digital Interface (SDI) www.national.com/sdi Temperature Sensors www.national.com/tempsensors SolarMagic™ www.national.com/solarmagic Wireless (PLL/VCO) www.national.com/wireless www.national.com/training PowerWise® Design University THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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