LM3405A 1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in Tiny SOT23 and eMSOP Package General Description Features The LM3405A is a 1A constant current buck LED driver designed to provide a simple, high efficiency solution for driving high power LEDs. With a 0.205V reference voltage feedback control to minimize power dissipation, an external resistor sets the current as needed for driving various types of LEDs. Switching frequency is internally set to 1.6MHz, allowing small surface mount inductors and capacitors to be used. The LM3405A utilizes current-mode control and internal compensation offering ease of use and predictable, high performance regulation over a wide range of operating conditions. With a maximum input voltage of 22V, it can drive up to 5 HighBrightness LEDs in series at 1A forward current, with the single LED forward voltage of approximately 3.7V. Additional features include user accessible EN/DIM pin for enabling and PWM dimming of LEDs, thermal shutdown, cycle-by-cycle current limit and over-current protection. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ VIN operating range of 3V to 22V Drives up to 5 High-Brightness LEDs in series at 1A Thin SOT23-6 package & eMSOP8 with exposed DAP 1.6MHz switching frequency EN/DIM input for enabling and PWM dimming of LEDs 300mΩ NMOS switch 40nA shutdown current at VIN = 5V Internally compensated current-mode control Cycle-by-cycle current limit Input voltage UVLO Over-current protection Thermal shutdown Applications ■ ■ ■ ■ ■ LED Driver Constant Current Source Industrial Lighting LED Flashlights LED Lightbulbs Typical Application Circuit Efficiency vs LED Current (VIN = 12V) 30015201 30015273 © 2008 National Semiconductor Corporation 300152 www.national.com LM3405A 1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in Tiny SOT23 and eMSOP Package October 23, 2008 LM3405A Connection Diagrams 30015205 30015260 6-Lead SOT-23 Package NS Package Number MK06A 8-Lead eMSOP Package NS Package Number MUY08A Ordering Information Part Number Package Type NS Package Drawing Package Marking Supplied As LM3405AXMK TSOT-6 MK06A SSEB 1000 Units on Tape and Reel LM3405AXMKX TSOT-6 MK06A SSEB 3000 Units on Tape and Reel LM3405AXMY eMSOP8 MUY08A SVSA 1000 Units on Tape and Reel LM3405AXMYX eMSOP8 MUY08A SVSA 3500 Units on Tape and Reel SOT23-6 Pin Descriptions Pin(s) Name Application Information 1 BOOST 2 GND 3 FB 4 EN/DIM 5 VIN Input supply voltage. Connect a bypass capacitor locally from this pin to GND. 6 SW Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor. Boost voltage that drives the NMOS output switch. A bootstrap capacitor is connected between the BOOST and SW pins. Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin. Feedback pin. Connect FB to external resistor divider to set output voltage. Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of varying duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be greater than VIN + 0.3V. eMSOP8 Pin Descriptions Pin Name 1 FB 2, 7 GND 3 NC 4 BOOST Function Feedback pin. Connect FB to external resistor divider to set output voltage. Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin. Tie pins 2, 7 and DAP to one GND plane. No Connection Boost voltage that drives the NMOS output switch. A bootstrap capacitor is connected between the BOOST and SW pins. 5 SW Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor. 6 VIN Input supply voltage. Connect a bypass capacitor locally from this pin to GND. 8 EN/DIM DAP www.national.com Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of varying duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be greater than VIN + 0.3V. Attach to power ground pin 2 If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN SW Voltage Boost Voltage Boost to SW Voltage FB Voltage EN/DIM Voltage Junction Temperature ESD Susceptibility (Note 2) Storage Temperature Operating Ratings (Note 1) VIN -0.5V to 24V -0.5V to 24V -0.5V to 30V -0.5V to 6.0V -0.5V to 3.0V -0.5V to (VIN + 0.3V) 150°C 2kV -65°C to +150°C 220°C 3V to 22V -0.5V to (VIN + 0.3V) 2.5V to 5.5V -40°C to +125°C EN/DIM voltage Boost to SW Voltage Junction Temperature Range Thermal Resistance θJA (Note 3) SOT23 118°C/W Thermal Resistance θJA (Note 4) ILED SOT23 pkg ILED eMSOP pkg eMSOP8 73°C/W 400mA 1A Electrical Characteristics Unless otherwise specified, VIN = 12V. Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm, and are provided for reference purposes only. Symbol VFB Parameter Conditions Feedback Voltage Min Typ Max Units 0.188 0.205 0.220 V ΔVFB/(ΔVINxVFB) Feedback Voltage Line Regulation VIN = 3V to 22V IFB UVLO Feedback Input Bias Current Sink/Source Under-voltage Lockout VIN Rising Under-voltage Lockout VIN Falling 0.01 1.9 UVLO Hysteresis fSW Switching Frequency DMAX Maximum Duty Cycle RDS(ON) ICL IQ VEN/DIM_TH IEN/DIM ISW Switch ON Resistance %/V 10 250 nA 2.74 2.95 V 2.3 V 0.44 1.2 VFB = 0V 85 1.6 V 1.9 94 MHz % SOT23 (VBOOST - VSW = 3V) 300 600 eMSOP (VBOOST - VSW = 3V) 360 700 2.0 2.8 A 2.8 mA mΩ Switch Current Limit VBOOST - VSW = 3V, VIN = 3V Quiescent Current Switching, VFB = 0.195V 1.8 Quiescent Current (Shutdown) VEN/DIM = 0V 0.3 Enable Threshold Voltage VEN/DIM Rising Shutdown Threshold Voltage VEN/DIM Falling EN/DIM Pin Current Sink/Source 0.01 µA Switch Leakage VIN = 22V 0.1 µA 1.2 µA 1.8 V 0.4 V Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under which the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: Human body model, 1.5kΩ in series with 100pF. Note 3: Thermal shutdown will occur if the junction temperature (TJ) exceeds 165°C. The maximum allowable power dissipation (PD) at any ambient temperature (TA) is PD = (TJ(MAX) – TA)/θJA . This number applies to packages soldered directly onto a 3" x 3" PC board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θJA = 204°C/W. Note 4: Thermal shutdown will occur if the junction temperature (TJ) exceeds 165°C. The maximum allowable power dissipation (PD) at any ambient temperature (TA) is PD = (TJ(MAX) – TA)/θJA . This number applies to packages soldered directly onto a 1" x 0.75" PC board with 1oz. copper on 4 layers in still air. 3 www.national.com LM3405A Soldering Information Infrared/Convection Reflow (15sec) Absolute Maximum Ratings (Note 1) LM3405A Typical Performance Characteristics Unless otherwise specified, VIN = 12V, VBOOST - VSW = 5V and TA = 25°C. Efficiency vs LED Current (VIN=5V) Efficiency vs Input Voltage (IF = 1A) 30015271 30015231 Efficiency vs Input Voltage (IF = 0.7A) Efficiency vs Input Voltage (IF = 0.35A) 30015232 30015233 VFB vs Temperature Oscillator Frequency vs Temperature 30015236 30015227 www.national.com 4 SOT23 RDS(ON) vs Temperature (VBOOST - VSW = 3V) 30015272 30015230 Quiescent Current vs Temperature Startup Response to EN/DIM Signal (VIN = 15V, IF = 0.2A) 30015234 30015268 5 www.national.com LM3405A Current Limit vs Temperature LM3405A Block Diagram 30015252 FIGURE 1. Simplified Block Diagram Application Information THEORY OF OPERATION The LM3405A is a PWM, current-mode controled buck switching regulator designed to provide a simple, high efficiency solution for driving LEDs with a preset switching frequency of 1.6MHz. This high frequency allows the LM3405A to operate with small surface mount capacitors and inductors, resulting in LED drivers that need only a minimum amount of board space. The LM3405A is internally compensated, simple to use, and requires few external components. The following description of operation of the LM3405A will refer to the Simplified Block Diagram (Figure 1) and to the waveforms in Figure 2. The LM3405A supplies a regulated output current by switching the internal NMOS power switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the internal NMOS power switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and the inductor current (IL) increases with a linear slope. IL is measured by the current sense amplifier, which generates an output proportional to the switch current. The sense signal is summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the feedback voltage and VREF. When the PWM comparator output goes high, the internal power switch turns off until the next switching cycle begins. During the switch off-time, inductor current discharges through the catch diode D1, which forces the SW pin to swing below ground by the forward voltage (VD1) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output current (IF) through the LED, by forcing FB pin voltage to be equal to VREF (0.205V). www.national.com 30015207 FIGURE 2. SW Pin Voltage and Inductor Current Waveforms of LM3405A BOOST FUNCTION Capacitor C3 and diode D2 in Figure 1 are used to generate a voltage VBOOST. The voltage across C3, VBOOST - VSW, is the gate drive voltage to the internal NMOS power switch. To properly drive the internal NMOS switch during its on-time, VBOOST needs to be at least 2.5V greater than VSW. A large value of VBOOST - VSW is recommended to achieve better efficiency by minimizing both the internal switch ON resistance (RDS(ON)), and the switch rise and fall times. However, VBOOST - VSW should not exceed the maximum operating limit of 5.5V. 6 placing a zener diode D3 in series with D2 as shown in Figure 4. When using a series zener diode from the input, the gate drive voltage is VIN - VD3 - VD2 + VD1. VBOOST - VSW = VIN - VD2 + VD1 FIGURE 4. VBOOST derived from VIN through a Series Zener 30015299 When the NMOS switch turns on (TON), the switch pin rises to: An alternate method is to place the zener diode D3 in a shunt configuration as shown in Figure 5. A small 350mW to 500mW, 5.1V zener in a SOT-23 or SOD package can be used for this purpose. A small ceramic capacitor such as a 6.3V, 0.1µF capacitor (C5) should be placed in parallel with the zener diode. When the internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The 0.1µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time. Resistor R2 should be chosen to provide enough RMS current to the zener diode and to the BOOST pin. A recommended choice for the zener current (IZENER) is 1mA. The current IBOOST into the BOOST pin supplies the gate current of the NMOS power switch. It reaches a maximum of around 3.6mA at the highest gate drive voltage of 5.5V over the LM3405A operating range. For the worst case IBOOST, increase the current by 50%. In that case, the maximum boost current will be: VSW = VIN – (RDS(ON) x IL) Since the voltage across C3 remains unchanged, VBOOST is forced to rise thus reverse biasing D2. The voltage at VBOOST is then: VBOOST = 2VIN – (RDS(ON) x IL) – VD2 + VD1 Depending on the quality of the diodes D1 and D2, the gate drive voltage in this method can be slightly less or larger than the input voltage VIN. For best performance, ensure that the variation of the input supply does not cause the gate drive voltage to fall outside the recommended range: 2.5V < VIN - VD2 + VD1 < 5.5V The second method for deriving the boost voltage is to connect D2 to the output as shown in Figure 3. The gate drive voltage in this configuration is: VBOOST - VSW = VOUT – VD2 + VD1 IBOOST-MAX = 1.5 x 3.6mA = 5.4mA Since the gate drive voltage needs to be in the range of 2.5V to 5.5V, the output voltage VOUT should be limited to a certain range. For the calculation of VOUT, see OUTPUT VOLTAGE section. R2 will then be given by: R2 = (VIN - VZENER) / (IBOOST_MAX + IZENER) For example, let VIN = 12V, VZENER = 5V, IZENER = 1mA, then: R2 = (12V - 5V) / (5.4mA + 1mA) = 1.09kΩ 30015293 FIGURE 3. VBOOST derived from VOUT 30015294 The third method can be used in the applications where both VIN and VOUT are greater than 5.5V. In these cases, C3 cannot be charged directly from these voltages; instead C3 can be charged from VIN or VOUT minus a zener voltage (VD3) by FIGURE 5. VBOOST derived from VIN through a Shunt Zener 7 www.national.com LM3405A When the LM3405A starts up, internal circuitry from VIN supplies a 20mA current to the BOOST pin, flowing out of the BOOST pin into C3. This current charges C3 to a voltage sufficient to turn the switch on. The BOOST pin will continue to source current to C3 until the voltage at the feedback pin is greater than 123mV. There are various methods to derive VBOOST: 1. From the input voltage (VIN) 2. From the output voltage (VOUT) 3. From a shunt or series zener diode 4. From an external distributed voltage rail (VEXT) The first method is shown in the Simplified Block Diagram of Figure 1. Capacitor C3 is charged via diode D2 by VIN. During a normal switching cycle, when the internal NMOS power switch is off (TOFF) (refer to Figure 2), VBOOST equals VIN minus the forward voltage of D2 (VD2), during which the current in the inductor (L1) forward biases the catch diode D1 (VD1). Therefore the gate drive voltage stored across C3 is: LM3405A The fourth method can be used in an application which has an external low voltage rail, VEXT. C3 can be charged through D2 from VEXT, independent of VIN and VOUT voltage levels. Again for best performance, ensure that the gate drive voltage, VEXT - VD2 + VD1, falls in the range of 2.5V to 5.5V. with respect to VIN on the LED current is shown in Figure 7. For a fast rising input voltage (200µs for example), there is no need to delay the EN/DIM signal since soft-start can smoothly bring up the LED current as shown in Figure 8. SETTING THE LED CURRENT LM3405A is a constant current buck regulator. The LEDs are connected between VOUT and the FB pin as shown in the Typical Application Circuit. The FB pin is at 0.205V in regulation and therefore the LED current IF is set by VFB and resistor R1 from FB to ground by the following equation: IF = VFB / R1 IF should not exceed the 1A current capability of LM3405A and therefore R1 minimum must be approximately 0.2Ω. IF should also be kept above 200mA for stable operation, and therefore R1 maximum must be approximately 1Ω. If average LED currents less than 200mA are desired, the EN/DIM pin can be used for PWM dimming. See LED PWM DIMMING section. OUTPUT VOLTAGE The output voltage is primarily determined by the number of LEDs (n) connected from VOUT to FB pin and therefore VOUT can be written as : 30015276 FIGURE 6. Startup Response to VIN with 5ms rise time VOUT = ((n x VF) + VFB) where VF is the forward voltage of one LED at the set LED current level (see LED manufacturer datasheet for forward characteristics curve). ENABLE MODE / SHUTDOWN MODE The LM3405A has both enable and shutdown modes that are controlled by the EN/DIM pin. Connecting a voltage source greater than 1.8V to the EN/DIM pin enables the operation of LM3405A, while reducing this voltage below 0.4V places the part in a low quiescent current (0.3µA typical) shutdown mode. There is no internal pull-up on EN/DIM pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to 0.3V above VIN. It should be noted that when the EN/DIM pin voltage rises above 1.8V while the input voltage is greater than UVLO, there is a finite delay before switching starts. During this delay the LM3405A will go through a power on reset state after which the internal softstart process commences. The soft-start process limits the inrush current and brings up the LED current (IF) in a smooth and controlled fashion. The total combined duration of the power on reset delay, soft-start delay and the delay to fully establish the LED current is in the order of 100µs (refer to Figure 10). The simplest way to enable the operation of LM3405A is to connect the EN/DIM pin to VIN which allows self start-up of LM3405A whenever the input voltage is applied. However, when an input voltage of slow rise time is used to power the application and if both the input voltage and the output voltage are not fully established before the soft-start time elapses, the control circuit will command maximum duty cycle operation of the internal power switch to bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.205V, the duty cycle will have to reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this reduction of duty cycle and this will result in a spike in LED current for a short duration as shown in Figure 6. In applications where this LED current overshoot is undesirable, EN/ DIM pin voltage can be separately applied and delayed such that VIN is fully established before the EN/DIM pin voltage reaches the enable threshold. The effect of delaying EN/DIM www.national.com 30015277 FIGURE 7. Startup Response to VIN with EN/DIM delayed 30015275 FIGURE 8. Startup Response to VIN with 200µs rise time 8 LM3405A LED PWM DIMMING The LED brightness can be controlled by applying a periodic pulse signal to the EN/DIM pin and varying its frequency and/ or duty cycle. This so-called PWM dimming method controls the average light output by pulsing the LED current between the set value and zero. A logic high level at the EN/DIM pin turns on the LED current whereas a logic low level turns off the LED current. Figure 9 shows a typical LED current waveform in PWM dimming mode. As explained in the previous section, there is approximately a 100µs delay from the EN/ DIM signal going high to fully establishing the LED current as shown in Figure 10. This 100µs delay sets a maximum frequency limit for the driving signal that can be applied to the EN/DIM pin for PWM dimming. Figure 11 shows the average LED current versus duty cycle of PWM dimming signal for various frequencies. The applicable frequency range to drive LM3405A for PWM dimming is from 100Hz to 5kHz. The dimming ratio reduces drastically when the applied PWM dimming frequency is greater than 5kHz. 30015283 FIGURE 11. Average LED Current versus Duty Cycle of PWM Dimming Signal at EN/DIM Pin UNDER-VOLTAGE LOCKOUT Under-voltage lockout (UVLO) prevents the LM3405A from operating until the input voltage exceeds 2.74V (typical). The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until VIN drops below 2.3V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic. CURRENT LIMIT The LM3405A uses cycle-by-cycle current limit to protect the internal power switch. During each switching cycle, a current limit comparator detects if the power switch current exceeds 2.0A (typical), and turns off the switch until the next switching cycle begins. 30015266 OVER-CURRENT PROTECTION The LM3405A has a built in over-current comparator that compares the FB pin voltage to a threshold voltage that is 60% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level (typically 328mV), the internal NMOS power switch is turned off, which allows the feedback voltage to decrease towards regulation. This threshold provides an upper limit for the LED current. LED current overshoot is limited to 328mV/R1 by this comparator during transients. FIGURE 9. PWM Dimming of LEDs using the EN/DIM Pin THERMAL SHUTDOWN Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction temperature exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on until the junction temperature drops below approximately 150°C. Design Guide INDUCTOR (L1) The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN): 30015267 FIGURE 10. Startup Response to EN/DIM with IF = 1A 9 www.national.com LM3405A tor selection, refer to Circuit Examples and Recommended Inductance Range in Table 1. The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to calculate a more accurate duty cycle. Calculate D by using the following formula: TABLE 1. Recommended Inductance Range IF Inductance Range and Inductor Current Ripple 6.8µH-15µH 1.0A Inductance ΔiL / IF* VSW can be approximated by: 6.8µH 10µH 15µH 51% 36% 24% 10µH 15µH 22µH 58% 39% 26% 10µH-22µH VSW = IF x RDS(ON) 0.6A Inductance ΔiL / IF* The diode forward drop (VD1) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VD1 is, the higher the operating efficiency of the converter. The inductor value determines the output ripple current (ΔiL, as defined in Figure 2). Lower inductor values decrease the size of the inductor, but increases the output ripple current. An increase in the inductor value will decrease the output ripple current. The ratio of ripple current to LED current is optimized when it is set between 0.3 and 0.4 at 1A LED current. This ratio r is defined as: 15µH-27µH 0.2A Inductance 15µH 22µH 27µH ΔiL / IF* 116% 79% 65% *Maximum over full range of VIN and VOUT. INPUT CAPACITOR (C1) An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL (Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any recommended deratings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be greater than: One must also ensure that the minimum current limit (1.2A) is not exceeded, so the peak current in the inductor must be calculated. The peak current (ILPK) in the inductor is calculated as: ILPK = IF + ΔiL/2 When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.2A, r can be made as high as 0.7. The ripple ratio can be increased at lighter loads because the net ripple is actually quite low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for the maximum ripple ratio at any current below 2A is: It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the point where the duty cycle D, is closest to 0.5. The ESL of an input capacitor is usually determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL and an 0805 ceramic chip capacitor will have very low ESL. At the operating frequency of the LM3405A, certain capacitors may have an ESL so large that the resulting inductive impedance (2πfL) will be higher than that required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low ESR and low ESL. A 10µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at least a 1µF ceramic capacitor close to the VIN pin. For MLCCs it is recommended to use X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions. r = 0.387 x IOUT-0.3667 Note that this is just a guideline. The LM3405A operates at a high frequency allowing the use of ceramic output capacitors without compromising transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing LED current ripple. See the output capacitor and feed-forward capacitor sections for more details on LED current ripple. Now that the ripple current or ripple ratio is determined, the inductance is calculated by: OUTPUT CAPACITOR (C2) The output capacitor is selected based upon the desired reduction in LED current ripple. A 1µF ceramic capacitor results in very low LED current ripple for most applications. Due to the high switching frequency, the 1µF capacitor alone (without feed-forward capacitor C4) can filter more than 90% of the inductor current ripple for most applications where the sum of LED dynamic resistance and R1 is larger than 1Ω. Since the internal compensation is tailored for small output capacitance with very low ESR, it is strongly recommended to use a ceramic capacitor with capacitance less than 3.3µF. where fSW is the switching frequency and IF is the LED current. When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating correctly. Because of the operating frequency of the LM3405A, ferrite based inductors are preferred to minimize core losses. This presents little restriction since the variety of ferrite based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended inducwww.national.com 10 LM3405A Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3405A, there is really no need to review other capacitor technologies. A benefit of ceramic capacitors is their ability to bypass high frequency noise. A certain amount of switching edge noise will couple through the parasitic capacitances in the inductor to the output. A ceramic capacitor will bypass this noise. In cases where large capacitance is required, use Electrolytic or Tantalum capacitors with large ESR, and verify the loop performance on the bench. Like the input capacitor, multilayer ceramic capacitors are recommended X7R or X5R. Again, verify actual capacitance at the desired operating voltage and temperature. Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor is: 30015270 FIGURE 13. PWM Dimming with a 1µF Feed-Forward Capacitor One may select a 1206 size ceramic capacitor for C2 since its current rating is typically higher than 1A, more than enough for the requirement. CATCH DIODE (D1) The catch diode (D1) conducts during the switch off-time. A Schottky diode is required for its fast switching time and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than: FEED-FORWARD CAPACITOR (C4) The feed-forward capacitor (designated as C4) connected in parallel with the LED string is required to provide multiple benefits to the LED driver design. It greatly improves the large signal transient response and suppresses LED current overshoot that may otherwise occur during PWM dimming; it also helps to shape the rise and fall times of the LED current pulse during PWM dimming thus reducing EMI emission; it reduces LED current ripple by bypassing some of inductor ripple from flowing through the LED. For most applications, a 1µF ceramic capacitor is sufficient. In fact, the combination of a 1µF feed-forward ceramic capacitor and a 1µF output ceramic capacitor leads to less than 1% current ripple flowing through the LED. Lower and higher C4 values can be used, but bench validation is required to ensure the performance meets the application requirement. Figure 12 shows a typical LED current waveform during PWM dimming without feed-forward capacitor. At the beginning of each PWM cycle, overshoot can be seen in the LED current. Adding a 1µF feed-forward capacitor can totally remove the overshoot as shown in Figure 13. ID1 = IF x (1-D) The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin. To improve efficiency, choose a Schottky diode with a low forward voltage drop. BOOST DIODE (D2) A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than 3.3V, a small-signal Schottky diode is recommended for better efficiency. A good choice is the BAT54 small signal diode. BOOST CAPACITOR (C3) A 0.01µF ceramic capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide the best performance. POWER LOSS ESTIMATION The main power loss in LM3405A includes three basic types of loss in the internal power switch: conduction loss, switching loss, and gate charge loss. In addition, there is loss associated with the power required for the internal circuitry of IC. The conduction loss is calculated as: If the inductor ripple current is fairly small (for example, less than 40%) , the conduction loss can be simplified to: PCOND = IF2 x RDS(ON) x D The switching loss occurs during the switch on and off transition periods, where voltage and current overlap resulting in power loss. The simplest means to determine this loss is to empirically measure the rise and fall times (10% to 90%) of the voltage at the switch pin. Switching power loss is calculated as follows: 30015269 FIGURE 12. PWM Dimming without Feed-Forward Capacitor PSW = 0.5 x VIN x IF x fSW x ( TRISE + TFALL ) 11 www.national.com LM3405A The gate charge loss is associated with the gate charge QG required to drive the switch: PCB Layout Considerations When planning the layout there are a few things to consider when trying to achieve a clean, regulated output. The most important consideration when completing the layout is the close coupling of the GND connections of the input capacitor C1 and the catch diode D1. These ground ends should be close to one another and be connected to the GND plane with at least two through-holes. Place these components as close to the IC as possible. The next consideration is the location of the GND connection of the output capacitor C2, which should be near the GND connections of C1 and D1. There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node island. The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup that causes inaccurate regulation. The LED current setting resistor R1 should be placed as close as possible to the IC, with the GND of R1 placed as close as possible to the GND of the IC. The VOUT trace to LED anode should be routed away from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible. Radiated noise can be decreased by choosing a shielded inductor. The remaining components should also be placed as close as possible to the IC. Please see Application Note AN-1229 for further considerations and the LM3405A demo board as an example of a four-layer layout. PG = fSW x VIN x QG The power loss required for operation of the internal circuitry: PQ = IQ x VIN IQ is the quiescent operating current, and is typically around 1.8mA for the LM3405A. The total power loss in the IC is: PINTERNAL = PCOND + PSW + PG + PQ An example of power losses for a typical application is shown in Table 2: TABLE 2. Power Loss Tabulation Conditions Power loss VIN 12V VOUT 3.9V IOUT 1.0A VD1 0.45V RDS(ON) 300mΩ fSW 1.6MHz PCOND 108mW PSW 288mW TRISE 18ns TFALL 12ns IQ 1.8mA PQ 22mW QG 1.4nC PG 27mW D is calculated to be 0.36 Σ ( PCOND + PSW + PQ + PG ) = PINTERNAL PINTERNAL = 445mW www.national.com 12 LM3405A LM3405A Circuit Examples 30015242 FIGURE 14. VBOOST derived from VIN ( VIN = 5V, IF = 1A ) Bill of Materials for Figure 14 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 6.3V, X5R C3216X5R0J106M TDK C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.37V at 1A, VR = 10V MBRM110LT1G ON Semiconductor D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1.5A, White LED LXK2-PW14 Lumileds 13 www.national.com LM3405A 30015243 FIGURE 15. VBOOST derived from VOUT ( VIN = 12V, IF = 1A ) Bill of Materials for Figure 15 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1.5A, White LED LXK2-PW14 Lumileds www.national.com 14 LM3405A 30015244 FIGURE 16. VBOOST derived from VIN through a Shunt Zener Diode (D3) ( VIN = 18V, IF = 1A ) Bill of Materials for Figure 16 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C5, Shunt Cap 0.1µF, 16V, X7R GRM219R71C104KA01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 4.7V, 350mW, SOT-23 BZX84C4V7 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay R2 1.91kΩ, 1% CRCW08051K91FKEA Vishay LED1 1.5A, White LED LXK2-PW14 Lumileds 15 www.national.com LM3405A 30015249 FIGURE 17. VBOOST derived from VIN through a Series Zener Diode (D3) ( VIN = 15V, IF = 1A ) Bill of Materials for Figure 17 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 11V, 350mW, SOT-23 BZX84C11 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1.5A, White LED LXK2-PW14 Lumileds www.national.com 16 LM3405A 30015250 FIGURE 18. VBOOST derived from VOUT through a Series Zener Diode (D3) ( VIN = 18V, IF = 1A ) Bill of Materials for Figure 18 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 3.6V, 350mW, SOT-23 BZX84C3V6 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1.5A, White LED LXK2-PW14 Lumileds LED2 1.5A, White LED LXK2-PW14 Lumileds 17 www.national.com LM3405A 30015251 FIGURE 19. LED MR16 Lamp Application ( VIN = 12V AC, IF = 0.75A ) Bill of Materials for Figure 19 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405A National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C5, Input Cap 220µF, 25V, electrolytic ECE-A1EN221U Panasonic D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Rectifier Diode Schottky, 0.385V at 500mA CMHSH5-2L Central Semiconductor D4, Rectifier Diode Schottky, 0.385V at 500mA CMHSH5-2L Central Semiconductor D5, Rectifier Diode Schottky, 0.385V at 500mA CMHSH5-2L Central Semiconductor D6, Rectifier Diode Schottky, 0.385V at 500mA CMHSH5-2L Central Semiconductor L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.27Ω, 0.33W, 1% ERJ8BQFR27 Panasonic LED1 1A, White LED LXHL-PW09 Lumileds www.national.com 18 LM3405A Physical Dimensions inches (millimeters) unless otherwise noted 6-Lead TSOT Package NS Package Number MK06A 8-Lead eMSOP Package NS Package Number MUY08A 19 www.national.com LM3405A 1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in Tiny SOT23 and eMSOP Package Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Design Support Amplifiers www.national.com/amplifiers WEBENCH www.national.com/webench Audio www.national.com/audio Analog University www.national.com/AU Clock Conditioners www.national.com/timing App Notes www.national.com/appnotes Data Converters www.national.com/adc Distributors www.national.com/contacts Displays www.national.com/displays Green Compliance www.national.com/quality/green Ethernet www.national.com/ethernet Packaging www.national.com/packaging Interface www.national.com/interface Quality and Reliability www.national.com/quality LVDS www.national.com/lvds Reference Designs www.national.com/refdesigns Power Management www.national.com/power Feedback www.national.com/feedback Switching Regulators www.national.com/switchers LDOs www.national.com/ldo LED Lighting www.national.com/led PowerWise www.national.com/powerwise Serial Digital Interface (SDI) www.national.com/sdi Temperature Sensors www.national.com/tempsensors Wireless (PLL/VCO) www.national.com/wireless THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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