MIC2103/04 75V, Synchronous Buck Controllers featuring Adaptive On-Time Control Hyper Speed Control™ Family General Description Features The Micrel MIC2103/04 are constant-frequency, synchronous buck controllers featuring a unique adaptive ON-time control architecture. The MIC2103/04 operates over an input supply range from 4.5V to 75V and can be used to supply up to 15A of output current. The output voltage is adjustable down to 0.8V with a guaranteed accuracy of ±1%. The device operates with programmable switching frequency from 200kHz to 600kHz. Micrel’s Hyper Light Load™ architecture provides the same high-efficiency and ultra fast transient response as the Hyper Speed Control architecture under the medium to heavy loads, but also maintains high efficiency under light load conditions by transitioning to variable frequency, discontinuous-mode operation. The MIC2103/04 offers a full suite of protection features to ensure protection of the IC during fault conditions. These include under-voltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, fold-back current limit, “hiccup” mode shortcircuit protection and thermal shutdown. All support documentation can be found on Micrel’s web site at: www.micrel.com. Hyper Speed Control architecture enables - High delta V operation (VIN = 75V and VOUT = 1.2V) - Any CapacitorTM stable 4.5V to 75V input voltage 0.8V Reference Voltage with ±1% accuracy 200kHz to 600kHz, programmable switching frequency Hyper Light Load Control (MIC2103 only) Hyper Speed Control (MIC2104 only) Enable input, Power-Good output Built-in 5V regulator for single-supply operation Programmable current limit and fold-back “hiccup” mode short-circuit protection 5ms internal soft-start, internal compensation, and thermal shutdown Supports safe start-up into a pre-biased output –40C to +125C junction temperature range Available in 16-pin 3mm x 3mm MLF® package Applications Distributed power systems Networking/Telecom Infrastructure Printers, scanners, graphic cards and video cards _________________________________________________________________________________________________________________________ Typical Application Efficiency (VIN = 48V) vs. Output Current (MIC2103) 100 90 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V EFFICIENCY (%) 80 70 60 50 40 30 fSW = 200kHz (CCM) 20 10 0 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 OUTPUT CURRENT (A) Hyper Speed Control, Hyper Light Load and Any Capacitor are trademarks of Micrel, Inc. MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com August 2012 M9999-080712-A Micrel, Inc. MIC2103/04 Ordering Information Part Number Switching Frequency Features Package Junction Temperature Range Lead Finish MIC2103YML 200kHz to 600kHz Hyper Light Load 16-pin 3mm x 3mm MLF –40°C to +125°C Pb-Free MIC2104YML 200kHz to 600kHz Hyper Speed Control 16-pin 3mm x 3mm MLF –40°C to +125°C Pb-Free Pin Configuration 16-Pin 3mm x 3mm MLF (ML) (TOP VIEW) Pin Description Pin Number Pin Name 1 VDD Internal +5V linear regulator output. VDD is the internal supply bus for the device. A 1μF ceramic capacitor from VDD to AGND is required for decoupling. In the applications with VIN<+5.5V, VDD should be tied to VIN to by-pass the linear regulator. 2 PVDD 5V supply input for the low-side N-channel MOSFET driver, which can be tied to VDD externally. A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling. 3 ILIM Current Limit Setting. Connect a resistor from SW to ILIM to set the over-current threshold for the converter. DL Low-Side Drive output. High-current driver output for external low-side MOSFET of a buck converter. The DL driving voltage swings from ground to VDD. Adding a small resistor between DL pin and the gate of the low-side N-channel MOSFET can slow down the turn-on and turn-off speed of the MOSFET. 5 PGND Power Ground. PGND is the return path for the buck converter power stage. The PGND pin connects to the sources of low-side N-Channel external MOSFET, the negative terminals of input capacitors, and the negative terminals of output capacitors. The return path for the power ground should be as small as possible and separate from the Signal ground (AGND) return path. 6 FREQ Switching Frequency Adjust input. Tie this pin to VIN to operate at 600kHz and place a resistor divider to reduce the frequency. DH High-Side Drive output. High-current driver output for external high-side MOSFET of a buck converter. The DH driving voltage is floating on the switch node voltage (VSW). Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFET can slow down the turn-on and turn-off speed of the MOSFET. SW Switch Node and Current-Sense input. High current output driver return. The SW pin connects directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be routed away from sensitive nodes. The SW pin also senses the current by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to the SW pin using a Kelvin connection. 4 7 8 August 2012 Pin Function 2 M9999-080712-A Micrel, Inc. MIC2103/04 Pin Description (Continued) Pin Number Pin Name 9, 11 NC No connection. 10 BST Voltage Supply Pin input for the high-side N-channel MOSFET driver, which can be powered by a bootstrapped circuit connected between VDD and SW, using a Schottky diode and a 0.1μF ceramic capacitor. Adding a small resistor at BST pin can slow down the turn-on speed of the high-side MOSFET. 12 AGND Signal ground for VDD and the control circuitry, which is connected to Thermal Pad electronically. The signal ground return path should be separate from the power ground (PGND) return path. 13 FB Feedback input. Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to set the desired output voltage. 14 PG Power Good output. Open Drain Output, an external pull-up resistor to VDD or external power rails is required. 15 EN Enable input. A logic signal to enable or disable the buck converter operation. The EN pin is CMOS compatible. Logic high enables the device, logic low shutdowns the regulator. In the disable mode, the VDD supply current for the device is minimized to 0.7mA typically. 16 VIN Supply voltage. The VIN operating voltage range is from 4.5V to 75V. A 1μF ceramic capacitor from VIN to AGND is required for decoupling. EP ePad August 2012 Pin Function Exposed Pad. Connect the EPAD to PGND plain on the PCB to improve the thermal performance. 3 M9999-080712-A Micrel, Inc. MIC2103/04 Absolute Maximum Ratings(1) Operating Ratings(3) VIN ................................................................ –0.3V to +76V VDD, VPVDD ........................................................ –0.3V to +6V VSW, VFREQ, VILIM, VEN............................ 0.3V to (VIN +0.3V) VBST to VSW ........................................................ 0.3V to 6V VBST ................................................................ 0.3V to 82V VPG ..................................................... 0.3V to (VDD + 0.3V) VFB . .................................................... 0.3V to (VDD + 0.3V) PGND to AGND ........................................... 0.3V to +0.3V Junction Temperature .............................................. +150°C Storage Temperature (TS).........................65C to +150C Lead Temperature (soldering, 10sec)........................ 260°C ESD Rating(2) ................................................. ESD Sensitive Supply Voltage (VIN).......................................... 4.5V to 75V Enable Input (VEN) .................................................. 0V to VIN VSW, VFEQ, VILIM, VEN ............................................... 0V to VIN Junction Temperature (TJ) ........................ 40C to +125C Junction Thermal Resistance 3mm x 3mm MLF-16 (JA) ....................................50.8°C/W 3mm x 3mm MLF-16 (JC) ....................................25.3°C/W Electrical Characteristics(4) VIN = 48V, VOUT = 5V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C. Parameter Condition Min Typ Max Units Power Supply Input Input Voltage Range (VIN)(5) 4.5 75 V Quiescent Supply Current (MIC2103) VFB = 1.5V 400 750 µA Quiescent Supply Current (MIC2104) VFB = 1.5V 2.1 3 mA Shutdown Supply Current SW unconnected, VEN = 0V 0.1 10 µA VDD Supply VDD Output Voltage VIN = 7V to 75V, IDD = 10mA 4.8 5.2 5.4 V VDD UVLO Threshold VDD rising 3.8 4.2 4.6 V VDD UVLO Hysteresis Load Regulation 400 IDD = 0 to 40mA mV 0.6 2 3.6 TJ = 25°C (±1.0%) 0.792 0.8 0.808 -40°C ≤ TJ ≤ 125°C (±2%) 0.784 0.8 0.816 5 500 % Reference Feedback Reference Voltage FB Bias Current VFB = 0.8V V nA Enable Control EN Logic Level High V 1.8 EN Logic Level Low 0.6 EN Hysteresis EN Bias Current August 2012 200 VEN = 48V 23 4 V mV 40 µA M9999-080712-A Micrel, Inc. MIC2103/04 Electrical Characteristics(4) (Continued) VIN = 48V, VOUT = 5V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C. Oscillator Switching Frequency VFREQ = VIN 400 VFREQ = 50%VIN 750 300 Maximum Duty Cycle Minimum Duty Cycle 600 VFB > 0.8V Minimum Off-Time 140 kHz 85 % 0 % 200 260 ns Soft Start Soft-Start time 5 ms Short Circuit Protection Current-Limit Threshold VFB = 0.79V -30 -14 0 mV Short-Circuit Threshold VFB = 0V -23 -7 9 mV Current-Limit Source Current VFB = 0.79V 60 80 100 µA Short-Circuit Source Current VFB = 0V 27 36 47 µA 0.1 V FET Drivers DH, DL Output Low Voltage ISINK = 10mA VPVDD - 0.1V DH, DL Output High Voltage Or ISOURCE = 10mA V VBST - 0.1V DH On-Resistance, High State 2.1 3.3 Ω DH On-Resistance, Low State 1.8 3.3 Ω DL On-Resistance, High State 1.8 3.3 Ω DL On-Resistance, Low State 1.2 2.3 Ω 50 µA 95 %VOUT SW, BST Leakage Current Power Good Power Good Threshold Voltage Sweep VFB from Low to High Power Good Hysteresis Sweep VFB from High to Low 6 %VOUT Power Good Delay Time Sweep VFB from Low to High 100 µs Power Good Low Voltage VFB < 90% x VNOM, IPG = 1mA 70 TJ Rising 160 °C 4 °C 85 90 200 mV Thermal Protection Over-Temperature Shutdown Over-Temperature Shutdown Hysteresis Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF. 3. The device is not guaranteed to function outside operating range. 4. Specification for packaged product only. 5. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH. August 2012 5 M9999-080712-A Micrel, Inc. MIC2103/04 Typical Characteristics VIN Operating Supply Current vs. Input Voltage (MIC2103) 1.0% 1.60 1.20 0.80 0.6% 0.4% 0.2% 0.0% -0.2% -0.4% -0.6% VOUT = 5.0V VOUT = 5V -0.8% IOUT = 0A to 10A IOUT = 0A -1.0% 0.40 0.00 0.808 0.8% INPUT VOLTAGE (V) 10 15 20 25 30 35 40 45 50 55 60 65 70 75 INPUT VOLTAGE (V) 5.015 5.010 5.005 5.000 VOUT = 5V 1.60 1.20 0.80 V IN = 48V 0.40 4.990 -50 -25 0.3% 0.6% 0.5% 0.4% 0.2% 0.1% 0.0% VIN = 48V VOUT = 5.0V IOUT = 0A to 10A August 2012 0 25 50 75 100 0.792 -50 125 -25 0 100 125 25 50 75 100 125 TEMPERATURE (°C) Feedback Voltage vs. Output Current (MIC2103) 0.808 0.3% 0.2% 0.1% 0.0% -0.1% -0.2% -0.3% -0.4% V IN = 12V to 75V V OUT = 5.0V -0.5% -0.6% -0.3% TEMPERATURE (°C) VOUT = 5.0V FEEDBACK VOLTAGE (V) 0.8% 0.7% LINE REGULATION (%) 0.4% 75 VIN = 48V Line Regulation vs. Temperature (MIC2103) Load Regulation vs. Temperature (MIC2103) 50 0.796 TEMPERATURE (°C) INPUT VOLTAGE (V) 25 0.800 IOUT = 0A IOUT = 0A 0.00 10 15 20 25 30 35 40 45 50 55 60 65 70 75 0 0.804 V OUT = 5.0V IOUT = 0A -25 Feedback Voltage vs. Temperature (MIC2103) 0.808 FEEBACK VOLTAGE (V) SUPPLY CURRENT (mA) OUTPUT VOLTAGE (V) 5.020 -50 VOUT = 5.0V 0.792 2.00 -0.2% 0.796 VIN Operating Supply Current vs. Temperature (MIC2103) 5.025 -0.1% 0.800 INPUT VOLTAGE (V) Output Voltage vs. Input Voltage (MIC2103) 4.995 0.804 IOUT = 0A 10 15 20 25 30 35 40 45 50 55 60 65 70 75 10 15 20 25 30 35 40 45 50 55 60 65 70 75 LOAD REGULATION (%) FEEDBACK VOLTAGE (V) TOTAL REGULATION (%) SUPPLY CURRENT (mA) 2.00 Feedback Voltage vs. Input Voltage (MIC2103) Output Regulation vs. Input Voltage (MIC2103) IOUT = 0A -50 -25 0 25 50 75 TEMPERATURE (°C) 6 0.804 0.800 0.796 VIN = 48V VOUT = 5.0V fSW = 200kHz 0.792 100 125 0 1 2 3 4 5 6 7 8 9 OUTPUT CURRENT (A) M9999-080712-A 10 Micrel, Inc. MIC2103/04 Typical Characteristics (Continued) Line Regulation vs. Output Current (MIC2103) Efficiency (VIN =12V) vs. Output Current (MIC2103) 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 0.2% 80 EFFICIENCY (%) 0.1% 0.0% -0.1% 60 50 40 30 0 1 2 3 4 5 6 7 8 9 1 2 3 OUTPUT CURRENT (A) EFFICIENCY (%) 70 60 50 40 30 20 4 5 6 7 8 9 10 11 12 13 14 fSW = 200kHz (CCM) 0 1 2 3 4 5 6 7 8 fSW = 200kHz (CCM) 0 1 2 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 80 70 60 9 10 11 12 13 14 50 40 30 4 5 6 7 8 9 10 11 12 13 14 90 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 80 70 60 50 40 30 20 10 fSW = 200kHz (CCM) fSW = 200kHz (CCM) 10 0 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 OUTPUT CURRENT (A) OUTPUT CURRENT (A) 3 Efficiency (VIN = 48V) vs. Output Current (MIC2103) 100 90 0 0 30 OUTPUT CURRENT (A) 20 10 40 10 Efficiency (VIN = 38V) vs. Output Current (MIC2103) 100 EFFICIENCY (%) 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 80 50 OUTPUT CURRENT (A) Efficiency (VIN = 24V) vs. Output Current (MIC2103) 90 60 0 0 10 70 20 0 -0.3% 100 80 fSW = 200kHz (CCM) 10 VOUT = 5.0V 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 90 20 VIN = 12V to 75V -0.2% 70 100 EFFICIENCY (%) LINE REGULATION (%) 90 EFFICIENCY (%) 100 0.3% Efficiency (VIN = 18V) vs. Output Current (MIC2103) 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 OUTPUT CURRENT (A) Efficiency (VIN = 75V) vs. Output Current (MIC2103) 100 90 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V EFFICIENCY (%) 80 70 60 50 40 30 20 fSW = 200kHz (CCM) 10 0 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 OUTPUT CURRENT (A) August 2012 7 M9999-080712-A Micrel, Inc. MIC2103/04 Typical Characteristics (Continued) 0.812 FEEDBACK VOLTAGE (V) 40 30 20 VOUT = 5V 10 IOUT = 0A 0.808 0.804 0.800 VOUT = 5.0V 0.796 fSW = 200kHz 0.4% 0.2% 0.0% -0.2% -0.4% fSW = 200kHz -1.0% INPUT VOLTAGE (V) LOAD REGULATION (%) 32 28 24 20 16 VIN = 48V VOUT = 5.0V IOUT = 0A 4 0.4% 0.0% 0.3% -0.2% 0.2% 0.1% 0.0% VIN = 48V -0.1% 0 25 50 75 100 VOUT = 5.0V IOUT = 0A to 10A -0.2% fSW = 200kHz -0.4% -0.6% -0.8% -1.0% -1.2% V IN = 12V to 75V -1.4% V OUT = 5.0V -1.6% -50 0 -25 Line Regulation vs. Temperature (MIC2104) -0.3% fSW = 200kHz -50 INPUT VOLTAGE (V) Load Regulation vs. Temperature (MIC2104) 36 8 IOUT = 0A to 10A fSW = 200kHz 10 15 20 25 30 35 40 45 50 55 60 65 70 75 INPUT VOLTAGE (V) 12 VOUT = 5.0V -0.6% IOUT = 0A 10 15 20 25 30 35 40 45 50 55 60 65 70 75 40 SUPPLY CURRENT (mA) 0.6% -0.8% 10 15 20 25 30 35 40 45 50 55 60 65 70 75 VIN Operating Supply Current vs. Temperature (MIC2104) -25 0 25 50 75 100 125 IOUT = 0A -1.8% TEMPERATURE (°C) 125 -50 -25 0 25 50 75 100 125 TEMPERATURE (°C) TEMPERATURE (°C) Feedback Voltage vs. Output Current (MIC2104) Line Regulation vs. Output Current (MIC2104) 0.0% LINE REGULATION (%) 0.808 FEEDBACK VOLTAGE (V) 0.8% 0.792 0 Output Regulation vs. Input Voltage (MIC2104) 1.0% LINE REGULATION (%) SUPPLY CURRENT (mA) 50 Feedback Voltage vs. Input Voltage (MIC2104) TOTAL REGULATION (%) VIN Operating Supply Current vs. Input Voltage (MIC2104) 0.804 0.800 0.796 V IN = 48V V OUT = 5.0V -0.1% -0.2% -0.3% -0.4% V IN = 12V to 75V -0.5% V OUT = 5.0V fSW = 200kHz fSW = 200kHz 0.792 -0.6% 0 1 2 3 4 5 6 7 8 OUTPUT CURRENT (A) August 2012 9 10 0 1 2 3 4 5 6 7 8 9 10 OUTPUT CURRENT (A) 8 M9999-080712-A Micrel, Inc. MIC2103/04 Typical Characteristics (Continued) 60 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 80 50 40 30 20 70 60 50 40 30 1 2 3 4 5 6 7 8 9 10 11 12 13 14 40 30 1 2 3 4 5 6 7 8 0 9 10 11 12 13 14 Efficiency (VIN = 48V) vs. Output Current (MIC2104) 100 60 50 EFFICIENCY (%) 70 40 30 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 70 60 50 40 30 2 3 4 5 6 7 8 1 2 3 DIE TEMPERATURE (°C) 80 60 40 VIN = 12V VOUT = 5.0V 20 4 5 6 7 8 3 4 5 6 7 OUTPUT CURRENT (A) 40 30 fSW = 200kHz 0 1 2 8 9 10 3 4 5 6 7 8 9 10 11 12 13 14 OUTPUT CURRENT (A) Die Temperature* (VIN = 75V) vs. Output Current 120 100 80 60 40 V IN = 48V V OUT = 5.0V 20 fSW = 200kHz 0 2 50 140 fSW = 200kHz 0 1 60 9 10 11 12 13 14 140 100 9 10 11 12 13 14 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V Die Temperature* (VIN = 48V) vs. Output Current 120 8 70 OUTPUT CURRENT (A) OUTPUT CURRENT (A) Die Temperature* (VIN = 12V) vs. Output Current 7 0 0 9 10 11 12 13 14 6 10 fSW = 200kHz 0 0 5 20 10 fSW = 200kHz 4 80 20 20 3 90 80 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 1 2 Efficiency (VIN = 75V) vs. Output Current (MIC2104) 100 EFFICIENCY (%) 80 0 1 OUTPUT CURRENT (A) 90 10 fSW = 200kHz OUTPUT CURRENT (A) 90 EFFICIENCY (%) 50 0 0 Efficiency (VIN = 38V) vs. Output Current (MIC2104) 100 DIE TEMPERATURE (°C) 60 10 fsw = 200kHz OUTPUT CURRENT (A) 0 70 0 0 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 80 20 10 fSW = 200kHz 0 140 90 20 10 Efficiency (VIN = 24V) vs. Output Current (MIC2104) 100 EFFICIENCY (%) 70 90 DIE TEMPERATURE (°C) EFFICIENCY (%) 80 Efficiency (VIN = 18V) vs. Output Current (MIC2104) 100 5.0V 3.3V 2.5V 1.8V 1.2V 0.8V 90 EFFICIENCY (%) 100 Efficiency (VIN =12V) vs. Output Current (MIC2104) 120 100 80 60 40 VIN = 75V VOUT = 5.0V 20 fSW = 200kHz 0 0 1 2 3 4 5 6 7 8 OUTPUT CURRENT (A) 9 10 0 1 2 3 4 5 6 7 8 9 OUTPUT CURRENT (A) * Case Temperature: The temperature measurement was taken at the hottest point on the MIC2103 case mounted on a 5 square inch PCB, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. August 2012 9 M9999-080712-A 10 Micrel, Inc. MIC2103/04 Typical Characteristics (Continued) VIN Shutdown Current vs. Input Voltage Enable Threshold vs. Input Voltage 1.50 Rising 540 480 420 360 300 240 180 IDD = 10mA 6 4 120 IDD = 40mA V OUT = 5.0V 2 60 V EN = 0V 220 180 15 10 5 VOUT = 5.0V IOUT = 2A Feedback Voltage vs. Temperature Output Peak Current Limit vs. Temperature SHUTDOWN CURRENT (uA) CURRENT LIMIT (A) 15 12 9 6 V IN =48V V OUT = 5.0V 3 fSW = 200kHz 0 0 25 50 75 August 2012 100 125 6 8 10 400 IOUT = 0A TEMPERATURE (°C) 4 VIN Shutdown Current vs. Temperature 18 VOUT = 5.0V 2 OUTPUT CURRENT (A) 21 VIN = 48V VIN = 48V VOUT = 5.0V 0 INPUT VOLTAGE (V) 0.800 -40°C 125°C 150 10 15 20 25 30 35 40 45 50 55 60 65 70 75 INPUT VOLTAGE (V) 0.804 25°C 200 0 10 15 20 25 30 35 40 45 50 55 60 65 70 75 0.808 250 100 fSW = 200kHz 100 Switching Frequency vs. Output Current 300 20 VOUT = 5.0V -25 INPUT VOLTAGE (V) SWITCHING FREQUENCY (kHz) CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) 260 -50 10 15 20 25 30 35 40 45 50 55 60 65 70 75 Output Peak Current Limit vs. Input Voltage 25 300 0.792 Hyst 0.30 INPUT VOLTAGE (V) Switching Frequency vs. Input Voltage 0.796 0.60 10 15 20 25 30 35 40 45 50 55 60 65 70 75 INPUT VOLTAGE (V) 0.812 Falling 0.90 0.00 0 10 15 20 25 30 35 40 45 50 55 60 65 70 75 140 1.20 fSW = 200kHz 0 FEEBACK VOLTAGE (V) ENABLE THRESHOLD (V) 8 VDD VOLTAGE (V) SHUTDOWN CURRENT (uA) 600 VDD Voltage vs. Input Voltage 10 320 240 160 V IN =48V 80 V EN = 0V IOUT = 0A 0 -50 -25 0 25 50 75 TEMPERATURE (°C) 10 100 125 -50 -25 0 25 50 75 100 125 TEMPERATURE (°C) M9999-080712-A Micrel, Inc. MIC2103/04 Typical Characteristics (Continued) VDD UVLO Threshold vs. Temperature VDD Voltage vs. Temperature 4.3 5.0 4.2 VDD Voltage (V) 4.5 4.0 IDD = 40mA IDD = 10mA 3.5 3.0 2.5 2.0 1.5 1.21 Rising 1.01 4.1 0.91 4.0 Falling 3.9 0.81 3.8 0.71 3.7 0.61 3.6 1.0 VIN = 48V 3.5 VIN =48V 0.5 IOUT = 0A 3.4 IOUT = 0A 0.0 0.51 0.41 3.3 -50 -25 0 25 50 75 100 125 0.31 -50 -25 0 TEMPERATURE (°C) 25 50 75 100 125 TEMPERATURE (°C) EN Bias Current vs. Temperature -50 -25 0 25 50 75 100 125 TEMPERATURE (C) Enable Threshold vs. Temperature 1.5 100 Rising 1.4 80 60 40 20 VIN =48V VEN = 0V 0 -50 -25 0 25 50 75 TEMPERATURE (°C) 100 125 ENABLE THRESHOLD (V) EN BIAS CURRENT (µA) 1.11 PG THRESHOLD (V) 4.4 5.5 VDD THRESHOLD (V) 6.0 PG Threshold/VREF Ratio vs. Temperature 1.3 Falling 1.2 1.1 1.0 0.9 0.8 0.7 VIN = 48V 0.6 0.5 -50 -25 0 25 50 75 100 125 TEMPERATURE (°C) August 2012 11 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Characteristics August 2012 12 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Characteristics (Continued) August 2012 13 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Characteristics (Continued) August 2012 14 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Characteristics (Continued) August 2012 15 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Characteristics (Continued) August 2012 16 M9999-080712-A Micrel, Inc. MIC2103/04 Functional Diagram Figure 1. MIC2103/04 Functional Diagram August 2012 17 M9999-080712-A Micrel, Inc. MIC2103/04 The maximum duty cycle is obtained from the 200ns tOFF(min): Functional Description The MIC2103/04 are adaptive on-time synchronous buck controllers built for high-input voltage to low-output voltage conversion applications. They are designed to operate over a wide input voltage range, from 4.5V to 75V, and the output is adjustable with an external resistive divider. An adaptive on-time control scheme is employed to obtain a constant switching frequency and to simplify the control compensation. Over-current protection is implemented by sensing low-side MOSFET’s RDS(ON). The device features internal softstart, enable, UVLO, and thermal shutdown. DMAX VOUT VIN f SW tS 1 200ns tS (Eq. 2) where tS = 1/fSW. It is not recommended to use MIC2103/04 with a OFF-time close to tOFF(min) during steady-state operation. The adaptive ON-time control scheme results in a constant switching frequency in the MIC2103/04. The actual ON-time and resulting switching frequency will vary with the different rising and falling times of the external MOSFETs. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications. During load transients, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop operation, we will analyze both the steady-state and load transient scenarios. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the feedback voltage. Figure 2 shows the MIC2103/04 control loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple plus injected voltage ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. Theory of Operation Figure 1 illustrates the block diagram of the MIC2103/04. The output voltage is sensed by the MIC2103/04 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low-gain transconductance (gm) amplifier. If the feedback voltage decreases and the amplifier output is below 0.8V, thenthe error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the “Fixed tON Estimator” circuitry: t ON(estimated) t S t OFF(MIN) (Eq. 1) where VOUT is the output voltage, VIN is the power stage input voltage, and fSW is the switching frequency. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 200ns, the MIC2103/04 control logic will apply the tOFF(min) instead. TOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the highside MOSFET. Figure 2. MIC2103/04 Control Loop Timing August 2012 18 M9999-080712-A Micrel, Inc. MIC2103/04 Figure 3a shows the operation of the MIC2103/04 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This will cause the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time tOFF(min) is generated to charge CBST since the feedback voltage is still below VREF. Then, the next ON-time period is triggered due to the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC2103/04 converter. Discontinuous Mode (MIC2103 only) In continuous mode, the inductor current is always greater than zero; however, at light loads, the MIC2103 is able to force the inductor current to operate in discontinuous mode. Discontinuous mode is where the inductor current falls to zero, as indicated by trace (IL) shown in Figure 3b. During this period, the efficiency is optimized by shutting down all the non-essential circuits and minimizing the supply current. The MIC2103 wakes up and turns on the high-side MOSFET when the feedback voltage VFB drops below 0.8V. The MIC2103 has a zero crossing comparator (ZC Detection) that monitors the inductor current by sensing the voltage drop across the low-side MOSFET during its ON-time. If the VFB > 0.8V and the inductor current goes slightly negative, then the MIC2103 automatically powers down most of the IC circuitry and goes into a low-power mode. Once the MIC2103 goes into discontinuous mode, both LSD and HSD are low, which turns off the high-side and low-side MOSFETs. The load current is supplied by the output capacitors and VOUT drops. If the drop of VOUT causes VFB to go below VREF, then all the circuits will wake up into normal continuous mode. First, the bias currents of most circuits reduced during the discontinuous mode are restored, then a tON pulse is triggered before the drivers are turned on to avoid any possible glitches. Finally, the high-side driver is turned on. Figure 3b shows the control loop timing in discontinuous mode. Figure 3a. MIC2103/04 Load Transient Response Unlike true current-mode control, the MIC2103/04 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. In order to meet the stability requirements, the MIC2103/04 feedback voltage ripple should be in phase with the inductor current ripple and are large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV over full input voltage range. If a low ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to “Ripple Injection” subsection in Application Information for more details about the ripple injection technique. August 2012 Figure 3b. MIC2103 Control Loop Timing (Discontinuous Mode) 19 M9999-080712-A Micrel, Inc. MIC2103/04 During discontinuous mode, the bias current of most circuits are reduced. As a result, the total power supply current during discontinuous mode is only about 400μA, allowing the MIC2103 to achieve high efficiency in light load applications. created by RCL and CCL should be much less than the minimum off time. The VCL drop allows programming of short limit through the value of the resistor (RCL), If the absolute value of the voltage drop on the bottom FET is greater than VCL’ in that case the V(ILIM) is lower than PGND and a short circuit event is triggered. A hiccup cycle to treat the short event is generated. The hiccup sequence including the soft start reduces the stress on the switching FETs and protects the load and supply for severe short conditions. The short circuit current limit can be programmed by using the following formula. Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC2103/04 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with 9.7mV steps. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function correctly. Current Limit The MIC2103/04 uses the RDS(ON) and external resistor connected from ILIM pin to SW node to decide the current limit. RCL ( I CLIM PP 0.5) RDS ( ON ) VCL I CL (Eq. 3) Where ISH = Desired Current limit ΔPP = Inductor current peak to peak RDS (ON) = On resistance of low-side power MOSFET VCL = Current limit threshold, the typical value is 14mV in EC table ICL = Current Limit source current, the typical value is 80µA in EC table. In case of hard short, the short limit is folded down to allow an indefinite hard short on the output without any destructive effect. It is mandatory to make sure that the inductor current used to charge the output capacitance during soft start is under the folded short limit, otherwise the supply will go in hiccup mode and may not be finishing the soft start successfully. The MOSFET RDS(ON) varies 30 to 40% with temperature; therefore, it is recommended to add a 50% margin to ICL in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect SW pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON). Figure 4. MIC2103/04 Current Limiting Circuit In each switching cycle of the MIC2103/04 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. The sensed voltage V(ILIM) is compared with the power ground (PGND) after a blanking time of 150nS. In this way the drop voltage over the resistor RCL (VCL) is compared with the drop over the bottom FET generating the short current limit. The small capacitor (CCL) connected from ILIM pin to PGND filters the switching node ringing during the off time allowing a better short limit measurement. The time constant August 2012 20 M9999-080712-A Micrel, Inc. MIC2103/04 the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e., ΔBST = 10mA x 3.33μs/0.1μF = 333mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the VDD supply voltage. The nominal low-side gate drive voltage is VDD and the nominal high-side gate drive voltage is approximately VDD – VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. MOSFET Gate Drive The MIC2103/04 high-side drive circuit is designed to switch an N-Channel MOSFET. Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reverse biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF is sufficient to hold August 2012 21 M9999-080712-A Micrel, Inc. MIC2103/04 Application Information MOSFET Selection The MIC2103/04 controllers work from input voltages of 4.5V to 75V and have an internal 5V VDD LDO. This internal VDD LDO provides power to turn the external NChannel power MOSFETs for the high-side and low-side switches. For applications where VDD < 5V, it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VDD > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. There are different criteria for choosing the high-side and low-side MOSFETs. These differences are more significant at lower duty cycles. In such an application, the high-side MOSFET is then required to switch as quickly as possible in order to minimize transition losses, whereas the low-side MOSFET can switch slower, but must handle larger RMS currents. When the duty cycle approaches 50%, the current carrying capability of the high-side MOSFET starts to become critical. It is important to note that the on-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current limit. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2103/04 gate-drive circuit. At 200kHz switching frequency, the gate charge can be a significant source of power dissipation in the MIC2103/04. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is: Setting the Switching Frequency The MIC2103/04 are adjustable-frequency, synchronous buck controllers featuring a unique adaptive on-time control architecture. The switching frequency can be adjusted between 200kHz and 600kHz by changing the resistor divider network consisting of R19 and R20. Figure 5. Switching Frequency Adjustment The following formula gives the estimated switching frequency: f SW _ ADJ f O R 20 R19 R 20 (Eq. 4) Where fO = Switching Frequency when R19 is 100k and R20 being open, fO is typically 550kHz. For a more precise setting, it is recommended to use the following graph: Switching Frequency 600 500 I G[high -side] (avg) QG fSW R19 = 100k, IOUT =10A SW FREQ (kHz) VIN = 48V where: IG[high-side](avg) = Average high-side MOSFET gate current QG = Total gate charge for the high-side MOSFET taken from the manufacturer’s data sheet for VGS = VDD. fSW = Switching Frequency 400 VIN =75V 300 200 100 0 10.00 100.00 1000.00 The low-side MOSFET is turned on and off at VDS = 0 because an internal body diode or external freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the 10000.00 R20 (k Ohm) Figure 6. Switching Frequency vs. R20 August 2012 (Eq. 5) 22 M9999-080712-A Micrel, Inc. MIC2103/04 transition times are equal; the transition times can be approximated by: gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET: IG[low -side] (avg) C ISS VGS f SW tT (Eq. 6) The total high-side MOSFET switching loss is: PGATEDRIVE VDD (IG[high- side] (avg) IG[low -side] (avg)) PAC (VHSD VD ) IPK t T f SW A convenient figure of merit for switching MOSFETs is the on resistance multiplied by the total gate charge; RDS(ON) × QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2103/04. Also, the RDS(ON) of the low-side MOSFET will determine the current-limit value. Please refer to “Current Limit” subsection is Functional Description for more details. Parameters that are important to MOSFET switch selection are: Voltage rating On-resistance Total gate charge (Eq. 8) PCONDUCTION ISW(RMS) 2 RDS(ON) (Eq. 9) PAC PAC(off ) PAC(on) The high-side MOSFET switching losses increase with the switching frequency and the input voltage VHSD. The low-side MOSFET switching losses are negligible and can be ignored for these calculations. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. (Eq. 10) The inductance value is calculated by Equation 13: where: RDS(ON) = On-resistance of the MOSFET switch D = Duty Cycle = VOUT / VHSD L VOUT (VIN(max) VOUT ) VIN(max) fsw 20% IOUT(max) (Eq. 13) where: Making the assumption that the turn-on and turn-off August 2012 (Eq. 12) where: tT = Switching transition time VD = Body diode drop (0.5V) fSW = Switching Frequency The voltage ratings for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VHSD. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitic elements. The power dissipated in the MOSFETs is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses during the period of time when the MOSFETs turn on and off (PAC). PSW PCONDUCTION PAC (Eq. 11) where: CISS and COSS are measured at VDS = 0 IG = Gate-drive current Since the current from the gate drive comes from the VDD, the power dissipated in the MIC2103/04 due to gate drive is: (Eq. 7) C ISS VIN C OSS VHSD IG 23 M9999-080712-A Micrel, Inc. MIC2103/04 PWINDING(Ht) = RWINDING(20°C) (1 + 0.0042 × (TH – T20°C)) (Eq. 18) fSW = Switching frequency 20% = Ratio of AC ripple current to DC output current VIN(max) = Maximum power stage input voltage where: TH = temperature of wire under full load T20°C = ambient temperature RWINDING(20°C) = room temperature winding resistance (usually specified by the manufacturer) The peak-to-peak inductor current ripple is: IL(pp) VOUT (VIN(max) VOUT ) VIN(max) fsw L (Eq. 14) Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated: The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) =IOUT(max) 0.5 ΔIL(pp) (Eq. 15) The RMS inductor current is used to calculate the I2R losses in the inductor. 2 IL(RMS) IOUT(max) ΔIL(PP) 2 12 ESR COUT (Eq. 16) The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated in Equation 20: 2 ΔVOUT(pp) ΔIL(PP) 2 ΔIL(PP) ESR C OUT C f 8 OUT SW (Eq. 20) where: D = duty cycle COUT = output capacitance value fsw = switching frequency Copper loss in the inductor is calculated by Equation 17: (Eq. 17) As described in the “Theory of Operation” subsection in Functional Description, the MIC2103/04 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. August 2012 (Eq. 19) ΔIL(PP) where: ΔVOUT(pp) = peak-to-peak output voltage ripple ΔIL(PP) = peak-to-peak inductor current ripple Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2103/04 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetic vendor. PINDUCTOR(Cu) = IL(RMS)2 RWINDING ΔVOUT(pp) 24 M9999-080712-A Micrel, Inc. MIC2103/04 smaller than the ripple caused by the output capacitor ESR. If low ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide enough feedback voltage ripple. Please refer to the “Ripple Injection” subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 21: ICOUT (RMS) ΔIL(PP) Figure 7. Voltage-Divider Configuration (Eq. 21) 12 The output voltage is determined by the equation: The power dissipated in the output capacitor is: VOUT VFB (1 2 PDISS(COUT ) ICOUT (RMS) ESR COUT where, VFB = 0.8V. A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 (Eq. 23) (Eq. 24) The power dissipated in the input capacitor is: PDISS(CIN) = ICIN(RMS)2 × ESRCIN (Eq. 25) Voltage Setting Components The MIC2103 requires two resistors to set the output voltage as shown in Figure 7: August 2012 VFB R1 VOUT VFB (Eq. 27) Ripple Injection The VFB ripple required for proper operation of the MIC2103/04 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as a 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator cannot sense it, then the MIC2103/04 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1. Enough ripple at the feedback voltage due to the large ESR of the output capacitors. As shown in Figure 8a, the converter is stable without any ripple injection. The feedback voltage ripple is: The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN(RMS) IOUT(max) D (1 D) (Eq. 26) (Eq. 22) Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: ΔVIN = IL(pk) × ESRCIN R1 ) R2 25 M9999-080712-A Micrel, Inc. MIC2103/04 ΔVFB(pp) R2 ESR COUT ΔIL (pp) R1 R2 In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 8c. The injected ripple is: (Eq. 28) where ΔIL(pp) is the peak-to-peak value of the inductor current ripple. 2. Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feed-forward capacitor Cff in this situation, as shown in Figure 8b. The typical Cff value is between 1nF and 100nF. With the feed-forward capacitor, the feedback voltage ripple is very close to the output voltage ripple: ΔVFB(pp) ESR ΔIL (pp) ΔVFB(pp) VIN K div D (1 - D) K div R1//R2 R inj R1//R2 1 fSW (Eq. 30) (Eq. 31) where: VIN = Power stage input voltage D = Duty cycle fSW = Switching frequency (Eq. 29) τ = (R1//R2//Rinj) Cff 3. Virtually no ripple at the FB pin voltage due to the very-low ESR of the output capacitors: In Equations 30 and 32, it is assumed that the time constant associated with Cff must be much greater than the switching period: 1 T 1 fSW (Eq. 32) If the voltage divider resistors R1 and R2 are in the kΩ range, then a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 33: Figure 8a. Enough Ripple at FB Figure 8b. Inadequate Ripple at FB K div ΔVFB(pp) VIN fSW D (1 D) (Eq. 33) Then the value of Rinj is obtained as: R inj (R1//R2) ( Figure 8c. Invisible Ripple at FB August 2012 1 K div 1) (Eq. 34) Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. 26 M9999-080712-A Micrel, Inc. MIC2103/04 Inductor PCB Layout Guidelines Warning: To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2103 converter. IC The signal ground pin (GND) must be connected directly to the ground planes. Do not route the GND pin to the PGND pin on the top layer. Place the IC close to the point of load (POL). Use fat traces to route the input and output power lines. Signal and power grounds should be kept separate and connected at only one location. Keep the inductor connection to the switch node (SW) short. Do not route any digital lines underneath or close to the inductor. Keep the switch node (SW) away from the feedback (FB) pin. The SW pin should be connected directly to the drain of the low-side MOSFET to accurate sense the voltage across the low-side MOSFET. To minimize noise, place a ground plane underneath the inductor. Output Capacitor The 1µF ceramic capacitors, which are connected to the VDD and PVDD pins, must be located right at the IC. The VDD pin is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the VDD and PGND pins. Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. Input Capacitor MOSFETs Place the input capacitor next. Place the input capacitors on the same side of the board and as close to the MOSFETs as possible. Place several vias to the ground plane close to the input capacitor ground terminal. Low-side MOSFET gate drive trace (DL pin to MOSFET gate pin) must be short and routed over a ground plane. The ground plane should be the connection between the MOSFET source and PGND. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. Chose a low-side MOSFET with a high CGS/CGD ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. Do not put a resistor between the Low-side MOSFET gate drive output and the gate. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. Use a 4.5V VGS rated MOSFET. Its higher gate threshold voltage is more immune to glitches than a 2.5V or 3.3V rated MOSFET. MOSFETs that are rated for operation at less than 4.5V VGS should not be used. RC Snubber Place the RC snubber on the same side of the board and as close to the SW pin as possible. August 2012 27 M9999-080712-A Micrel, Inc. MIC2103/04 Evaluation Board Schematic Figure 9. Schematic of MIC2103/04 Evaluation Board (J1, J9, J12, R14, and R21 are for testing purposes) August 2012 28 M9999-080712-A Micrel, Inc. MIC2103/04 Bill of Materials Item C1 Part Number EEU-FC2A101 GRM32ER72A225K C2, C3, C4 C14 12101C225KAT2A (4) AVX GRM32ER60J107ME20L Murata 12106D107MAT2A AVX C3225X5ROJ107M TDK AVX C1608X7R1H104K TDK C10 AVX C1608X5R0J105K TDK GRM188R72A104KA35D C1608X7S2A104K GRM188R72A102KA01D C11 C12 Murata AVX Murata TDK 100µF Aluminum Capacitor, 100V 1 2.2µF/100V Ceramic Capacitor, X7R, Size 1210 3 100µF/6.3V Ceramic Capacitor, X5R, Size 1210 1 0.1µF/50V Ceramic Capacitor, X7R, Size 0603 2 1µF/6.3V Ceramic Capacitor, X7R, Size 0603 3 0.47µF/100V Ceramic Capacitor, X7R, Size 0805 1 0.1µF/100V Ceramic Capacitor, X7R, Size 0603 0.1µF/100V,X7S,0603 1 Murata 06031C102KAT2A AVX C1608X7R2A102K TDK GRM188R72A222KA01D Qty Murata 06036C105KAT2A 08051C474KAT2A Description Murata 06035C104KAT2A GRM21BR72A474KA73 C9 Murata (1) (2) TDK(3) GRM188R70J105KA01D C7, C8, C17 Panasonic C3225X7R2A225K GRM188R71H104KA93D C6, C16 Manufacturer 1nF/100V Cermiac Capacitor, X7R, Size 0603 1 2.2nF/100V Cermiac Capacitor, X7R, Size 0603 1 Murata 06031C222KAT2A AVX C1608X7R2A222K TDK Sanyo(5) 470µF/6.3V, 7m-ohms, OSCON 6SEPC470M Sanyo 470µF/6.3V, 7m-ohms, OSCON C15 (OPEN) 6TPB470M Sanyo 470µF/6.3V, POSCAP C5 (OPEN) GRM32ER60J107ME20L Murata 100µF/6.3V Ceramic Capacitor, X7R, Size 1210 C18 GCM1885C2A100JA16D 06031A100JAT2A Murata AVX 10pF, 100V, 0603, NPO D1 BAT46W-TP MCC(6) 100V Small Signal Schottky Diode, SOD123 1 6.1µH Inductor, 14.8A RMS Current 1 C13 L1 6SEPC470MX CDEP147NP-6R1MC-95 Sumida (7) 1 1 Notes: 1. Panasonic: www.panasonic.com. 2. Murata: www.murata.com. 3. TDK: www.tdk.com. 4. AVX: www.avx.com 5. Sanyo: www.sanyo.com. 6. MCC.: www.mccsemi.com. 7. Sumida: www.sumida.com. August 2012 29 M9999-080712-A Micrel, Inc. MIC2103/04 Bill of Materials (Continued) Item Part Number Q1 SIR878DP Q3 SIR882DP Manufacturer (8) Vishay Description Qty MOSFET, N-CH, Power SO-8 1 Vishay MOSFET, N-CH, Power SO-8 1 Q2, Q4 (OPEN) R1 CRCW060310K0FKEA Vishay Dale 10kΩ Resistor, Size 0603, 1% 1 R2, R23 CRCW08051R21FKEA Vishay Dale 1.21Ω Resistor, Size 0805, 5% 2 R3 CRCW060395K30FKEA Vishay Dale 95.3kΩ Resistor, Size 0603, 1% 1 R4 CRCW060380K6FKEA Vishay Dale 80.6kΩ Resistor, Size 0603, 1% 1 R5 CRCW060340K2FKEA Vishay Dale 40.2kΩ Resistor, Size 0603, 1% 1 R6 CRCW060320K0FKEA Vishay Dale 20kΩ Resistor, Size 0603, 1% 1 R7 CRCW060311K5FKEA Vishay Dale 11.5kΩ Resistor, Size 0603, 1% 1 R8 CRCW06038K06FKEA Vishay Dale 8.06kΩ Resistor, Size 0603, 1% 1 R9 CRCW06034K75FKEA Vishay Dale 4.75kΩ Resistor, Size 0603, 1% 1 R10 CRCW06033K24FKEA Vishay Dale 3.24kΩ Resistor, Size 0603, 1% 1 R11 CRCW06031K91FKEA Vishay Dale 1.91kΩ Resistor, Size 0603, 1% 1 R12 (OPEN) CRCW0603715R0FKEA Vishay Dale 715Ω Resistor, Size 0603, 1% R13 (OPEN) CRCW0603348R0FKEA Vishay Dale 348Ω Resistor, Size 0603, 1% R14, R15 CRCW06030000FKEA Vishay Dale 0Ω Resistor, Size 0603, 5% 2 R16 CRCW08052R0FKEA Vishay Dale 2Ω Resistor, Size 0805, 5% 1 R17 CRCW06032K21FKEA Vishay Dale 2.21kΩ Resistor, Size 0603, 1% 1 R18, R20 CRCW060349K9FKEA Vishay Dale 49.9kΩ Resistor, Size 0603, 1% 2 R19, R22 CRCW0603100K0FKEA Vishay Dale 100kΩ Resistor, Size 0603, 1% 2 R21 CRCW060349R9FKEA Vishay Dale 49.9Ω Resistor, Size 0603, 1% 1 75V Synchronous Buck DC-DC Controller 1 MIC2103YML U1 MIC2104YML Micrel. Inc.(9) Notes: 8. Vishay: www.vishay.com. 9. Micrel, Inc.: www.micrel.com. August 2012 30 M9999-080712-A Micrel, Inc. MIC2103/04 PCB Layout Figure 10. MIC2103/04 Evaluation Board Top Layer August 2012 31 M9999-080712-A Micrel, Inc. MIC2103/04 PCB Layout (Continued) Figure 11. MIC2103/04 Evaluation Board Mid-Layer 1 (Ground Plane) August 2012 32 M9999-080712-A Micrel, Inc. MIC2103/04 PCB Layout (Continued) Figure 12. MIC2103/04 Evaluation Board Mid-Layer 2 August 2012 33 M9999-080712-A Micrel, Inc. MIC2103/04 PCB Layout (Continued) Figure 13. MIC2103/04 Evaluation Board Bottom Layer August 2012 34 M9999-080712-A Micrel, Inc. MIC2103/04 Recommended Land Pattern Red circle indicates Thermal Via. Size should be .300mm .350mm in diameter and it should be connected to GND plane for maximum thermal performance. ALL UNITS ARE IN mm, TOLERANCE 0.05, IF NOT NOTED LP # MLF33D-16LD-LP-1 August 2012 35 M9999-080712-A Micrel, Inc. MIC2103/04 Package Information 16-Pin 3mm 3mm MLF (ML) MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2012 Micrel, Incorporated. August 2012 36 M9999-080712-A