Ultralow Distortion, Ultralow Noise Op Amp AD797 PIN CONFIGURATION Low noise 0.9 nV/√Hz typical (1.2 nV/√Hz maximum) input voltage noise at 1 kHz 50 nV p-p input voltage noise, 0.1 Hz to 10 Hz Low distortion −120 dB total harmonic distortion at 20 kHz Excellent ac characteristics 800 ns settling time to 16 bits (10 V step) 110 MHz gain bandwidth (G = 1000) 8 MHz bandwidth (G = 10) 280 kHz full power bandwidth at 20 V p-p 20 V/μs slew rate Excellent dc precision 80 μV maximum input offset voltage 1.0 μV/°C VOS drift Specified for ±5 V and ±15 V power supplies High output drive current of 50 mA +IN 3 6 OUTPUT –VS 4 5 OFFSET NULL AD797 TOP VIEW 8 Figure 1. 8-Lead Plastic Dual In-Line Package [PDIP] and 8-Lead Standard Small Outline Package [SOIC] GENERAL DESCRIPTION The AD797 is a very low noise, low distortion operational amplifier ideal for use as a preamplifier. The low noise of 0.9 nV/√Hz and low total harmonic distortion of −120 dB at audio bandwidths give the AD797 the wide dynamic range necessary for preamps in microphones and mixing consoles. Furthermore, the AD797’s excellent slew rate of 20 V/μs and 110 MHz gain bandwidth make it highly suitable for low frequency ultrasound applications. The AD797 is also useful in infrared (IR) and sonar imaging applications, where the widest dynamic range is necessary. The low distortion and 16-bit settling time of the AD797 make it ideal for buffering the inputs to Σ-Δ ADCs or the outputs of high resolution DACs, especially when the device is used in critical applications such as seismic detection or in spectrum analyzers. Key features such as a 50 mA output current drive and the specified power supply voltage range of ±5 V to ±15 V make the AD797 an excellent general-purpose amplifier. APPLICATIONS Professional audio preamplifiers IR, CCD, and sonar imaging systems Spectrum analyzers Ultrasound preamplifiers Seismic detectors Σ-Δ ADC/DAC buffers 4 THD (dB) 3 2 –100 0.001 –110 0.0003 –120 0.0001 THD (%) –90 5 1 0 10 100 1k 10k 100k 1M FREQUENCY (Hz) 10M –130 100 300 1k 3k 10k FREQUENCY (Hz) 30k 100k 300k 00846-003 MEASUREMENT LIMIT 00846-002 INPUT VOLTAGE NOISE (nV/√Hz) –IN 2 DECOMPENSATION AND DISTORTION NEUTRALIZATION 7 +VS OFFSET NULL 1 00846-001 FEATURES Figure 2. AD797 Voltage Noise Spectral Density Figure 3. THD vs. Frequency Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2008 Analog Devices, Inc. All rights reserved. Rev. F AD797 TABLE OF CONTENTS Features .............................................................................................. 1 Low Frequency Noise ................................................................ 12 Applications....................................................................................... 1 Wideband Noise ......................................................................... 12 Pin Configuration............................................................................. 1 Bypassing Considerations ......................................................... 12 General Description ......................................................................... 1 The Noninverting Configuration............................................. 13 Revision History ............................................................................... 2 The Inverting Configuration .................................................... 14 Specifications..................................................................................... 3 Driving Capacitive Loads.......................................................... 14 Absolute Maximum Ratings............................................................ 5 Settling Time............................................................................... 14 ESD Caution.................................................................................. 5 Distortion Reduction ................................................................. 15 Typical Performance Characteristics ............................................. 6 Outline Dimensions ....................................................................... 18 Theory of Operation ...................................................................... 11 Ordering Guide .......................................................................... 19 Noise and Source Impedance Considerations........................ 11 REVISION HISTORY 1/08—Rev. E to Rev. F Changes to Absolute Maximum Ratings ....................................... 5 Change to Equation 1..................................................................... 12 Changes to the Noninverting Configuration Section................ 13 Updated Outline Dimensions ....................................................... 19 Changes to Ordering Guide .......................................................... 20 7/05—Rev. D to Rev. E Updated Figure 1 Caption ............................................................... 1 Deleted Metallization Photo ........................................................... 6 Changes to Equation 1 ................................................................... 12 Updated Outline Dimensions ....................................................... 19 Changes to Ordering Guide .......................................................... 20 10/02—Rev. C to Rev. D Deleted 8-Lead CERDIP Package (Q-8)..........................Universal Edits to Specifications ...................................................................... 2 Edits to Absolute Maximum Ratings ............................................. 3 Edits to Ordering Guide .................................................................. 3 Edits to Table I .................................................................................. 9 Deleted Operational Amplifiers Graphic .................................... 15 Updated Outline Dimensions ....................................................... 15 Rev. F | Page 2 of 20 AD797 SPECIFICATIONS TA = 25°C and VS = ±15 V dc, unless otherwise noted. Table 1. Parameter INPUT OFFSET VOLTAGE Supply Voltage (V) ±5 V, ±15 V Conditions Min TMIN to TMAX Offset Voltage Drift INPUT BIAS CURRENT ±5 V, ±15 V ±5 V, ±15 V TMIN to TMAX INPUT OFFSET CURRENT OPEN-LOOP GAIN DYNAMIC PERFORMANCE Gain Bandwidth Product –3 dB Bandwidth Full Power Bandwidth1 Slew Rate Settling Time to 0.0015% COMMON-MODE REJECTION POWER SUPPLY REJECTION INPUT VOLTAGE NOISE INPUT CURRENT NOISE INPUT COMMON-MODE VOLTAGE RANGE OUTPUT VOLTAGE SWING Short-Circuit Current Output Current 3 TOTAL HARMONIC DISTORTION ±5 V, ±15 V TMIN to TMAX VOUT = ±10 V RLOAD = 2 kΩ TMIN to TMAX RLOAD = 600 Ω TMIN to TMAX @ 20 kHz 1 ±15 V G = 1000 G = 1000 2 G = 10 VOUT = 20 V p-p, RLOAD = 1 kΩ RLOAD = 1 kΩ 10 V step VCM = CMVR TMIN to TMAX VS = ±5 V to ±18 V TMIN to TMAX f = 0.1 Hz to 10 Hz f = 10 Hz f = 1 kHz f = 10 Hz to 1 MHz f = 1 kHz 1 1 1 1 14,000 ±15 V 15 V ±15 V ±15 V ±15 V ±15 V ±5 V, ±15 V RLOAD = 2 kΩ RLOAD = 600 Ω RLOAD = 600 Ω RLOAD = 1 kΩ, CN = 50 pF, f = 250 kHz, 3 V rms RLOAD = 1 kΩ, f = 20 kHz, 3 V rms ±15 V ±15 V ±15 V ±15 V ±15 V ±15 V ±5 V ±15 V ±15 V ±5 V ±5 V, ±15 V ±5 V, ±15 V ±15 V ±15 V INPUT CHARACTERISTICS Input Resistance Differential Common Mode Input Capacitance Differential 4 Common Mode Rev. F | Page 3 of 20 AD797A Typ 25 50 0.2 0.25 0.5 100 120 20 6 15 5 20,000 Max 80 125/180 1.0 1.5 3.0 400 600/700 110 450 8 280 12.5 114 110 114 110 ±11 ±2.5 ±12 ±11 ±2.5 30 20 800 130 120 130 120 50 1.7 0.9 1.0 2.0 ±12 ±3 ±13 ±13 ±3 80 50 −98 −120 AD797B Typ 10 30 0.2 0.25 0.25 80 120 2 20 2 10 2 15 2 7 14,000 20,000 Min Max 40 60 0.6 0.9 2.0 200 300 110 450 8 MHz MHz MHz kHz 280 12.5 −90 20 800 130 120 114 120 50 1.7 0.9 1.0 2.0 ±11 ±2.5 ±13 ±13 ±3 80 50 −98 −110 −120 1200 120 114 120 130 1.2 1.3 ±12 ±11 ±2.5 30 Unit μV μV μV/°C μA μA nA nA V/μV V/μV V/μV V/μV V/V −90 V/μs ns dB dB dB dB nV p-p nV/√Hz nV/√Hz μV rms pA/√Hz V V V V V mA mA dB −110 dB 1200 2.5 1.2 1.2 ±12 ±3 7.5 100 7.5 100 kΩ MΩ 20 5 20 5 pF pF AD797 Parameter OUTPUT RESISTANCE POWER SUPPLY Operating Range Quiescent Current Conditions AV = 1, f = 1 kHz Supply Voltage (V) Min AD797A Typ Max 3 ±5 ±5 V, ±15 V 1 8.2 Full power bandwidth = slew rate/2π VPEAK. Specified using external decompensation capacitor. 3 Output current for |VS – VOUT| > 4 V, AOL > 200 kΩ. 4 Differential input capacitance consists of 1.5 pF package capacitance and 18.5 pF from the input differential pair. 2 Rev. F | Page 4 of 20 ±18 10.5 Min AD797B Typ ±5 8.2 Max 3 Unit mΩ ±18 10.5 V mA AD797 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter Supply Voltage Internal Power Dissipation @ 25°C1 PDIP SOIC Input Voltage Differential Input Voltage2 Output Short-Circuit Duration Storage Temperature Range (N, R Suffix) Operating Temperature Range Lead Temperature Range (Soldering 60 sec) 1 2 Ratings ±18 V 1.3 W − (TA − 25°C)/θJA 0.9 W (TA − 25°C)/θJA ±VS ±0.7 V Indefinite within maximum internal power dissipation −65°C to +125°C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ESD CAUTION −40°C to +85°C 300°C θJA = 95°C/W for the 8-lead PDIP; 155°C/W for the 8-lead SOIC. The AD797 inputs are protected by back-to-back diodes. To achieve low noise, internal current-limiting resistors are not incorporated into the design of this amplifier. If the differential input voltage exceeds ±0.7 V, the input current should be limited to less than 25 mA by series protection resistors. Note, however, that this degrades the low noise performance of the device. Rev. F | Page 5 of 20 AD797 TYPICAL PERFORMANCE CHARACTERISTICS 10 5 0 5 10 15 20 SUPPLY VOLTAGE (±V) HORIZONTAL SCALE (5sec/DIV) Figure 7. 0.1 Hz to 10 Hz Noise Figure 4. Input Common-Mode Voltage Range vs. Supply Voltage 0 INPUT BIAS CURRENT (µA) 15 10 +VOUT –VOUT 5 0 0 5 10 15 20 SUPPLY VOLTAGE (±V) –0.5 –1.0 –1.5 –2.0 –60 00846-005 –20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) Figure 5. Output Voltage Swing vs. Supply Voltage Figure 8. Input Bias Current vs. Temperature 30 140 SHORT-CIRCUIT CURRENT (mA) VS = ± 15V 20 10 VS = ±5 120 100 SOURCE CURRENT SINK CURRENT 80 60 0 10 100 1k LOAD RESISTANCE (Ω) 10k 40 –60 00846-006 OUTPUT VOLTAGE SWING (V p-p) –40 –40 –20 0 20 40 60 80 100 120 TEMPERATURE (°C) Figure 9. Short-Circuit Current vs. Temperature Figure 6. Output Voltage Swing vs. Load Resistance Rev. F | Page 6 of 20 140 00846-009 OUTPUT VOLTAGE SWING (±V) 20 00846-008 0 00846-007 VERTICAL SCALE (0.01µV/DIV) 15 00846-004 INPUT COMMON-MODE RANGE (±V) 20 AD797 +125°C 9 +25°C 8 7 5 15 10 20 SUPPLY VOLTAGE (±V) Figure 10. Quiescent Supply Current vs. Supply Voltage 125 CMR 60 100 40 75 20 1 10 100 1k 10k 100k FREQUENCY (Hz) –60 f = 1kHz RL = 600Ω G = +10 RL = 600Ω G = +10 f = 10kHz NOISE BW = 100kHz THD + NOISE (dB) 9 6 –80 VS = ±5V –100 3 0 ±5 ±10 ±15 ±20 SUPPLY VOLTAGE (±V) –120 0.01 0.1 1 10 OUTPUT LEVEL (V) Figure 11. Output Voltage vs. Supply Voltage for 0.01% Distortion 00846-014 VS = ±15V 00846-011 0 50 1M Figure 13. Power Supply and Common-Mode Rejection vs. Frequency 12 OUTPUT VOLTAGE (V rms) 150 80 00846-010 0 PSR +SUPPLY PSR –SUPPLY 100 –55°C 6 175 120 00846-013 10 COMMON MODE REJECTION (dB) 200 140 POWER SUPPLY REJECTION (dB) QUIESCENT SUPPLY CURRENT (mA) 11 Figure 14. Total Harmonic Distortion (THD) + Noise vs. Output Level 1.0 30 OUTPUT VOLTAGE (V p-p) ±15V SUPPLIES 0.0015% 0.6 0.01% 0.4 RL = 600Ω 20 10 0 0 2 4 6 8 STEP SIZE (V) 10 0 10k 100k 1M FREQUENCY (Hz) Figure 12. Settling Time vs. Step Size (±) Figure 15. Large-Signal Frequency Response Rev. F | Page 7 of 20 10M 00846-015 ±5V SUPPLIES 0.2 00846-012 SETTLING TIME (µs) 0.8 AD797 110 SLEW RATE (V/µs) 30 3 2 SLEW RATE RISING EDGE 25 100 SLEW RATE FALLING EDGE 20 1 0 10 100 1k 10k 100k 1M 10M FREQUENCY (Hz) 90 15 –60 –40 –20 0 20 40 60 80 100 120 80 140 TEMPERATURE (°C) 00846-019 GAIN/BANDWIDTH PRODUCT 4 GAIN/BANDWIDTH PRODUCT (MHz (G = 1000)) 120 35 00846-016 INPUT VOLTAGE NOISE (nV/√Hz) 5 Figure 19. Slew Rate and Gain/Bandwidth Product vs. Temperature Figure 16. Input Voltage Noise Spectral Density 120 100 160 PHASE MARGIN 60 80 40 60 GAIN 20 40 *RS = 100 WITHOUT RS* 20 *SEE FIGURE 25. 120 0 WITH RS* 10k 100k 1M FREQUENCY (Hz) 10M 100M 100 100 Figure 17. Open-Loop Gain and Phase Margin vs. Frequency Figure 20. Open-Loop Gain vs. Load Resistance 100 MAGNITUDE OF OUTPUT IMPEDANCE (Ω) OVERCOMPENSATED 150 0 –150 UNDER COMPENSATED –40 –20 0 20 40 60 80 100 TEMPERATURE (°C) 120 140 10 1 WITHOUT CN* 0.1 WITH CN* *SEE FIGURE 32. 0.01 00846-018 INPUT OFFSET CURRENT (nA) 300 –300 –60 10k 1k LOAD RESISTANCE (Ω) 00846-020 1k 140 10 100 1k 10k 100k FREQUENCY (Hz) Figure 21. Magnitude of Output Impedance vs. Frequency Figure 18. Input Offset Current vs. Temperature Rev. F | Page 8 of 20 1M 00846-021 0 100 OPEN-LOOP GAIN (dB) 80 PHASE MARGIN (Degrees) WITHOUT RS* WITH RS* 00846-017 OPEN-LOOP GAIN (dB) 100 AD797 20pF 100Ω +VS 1kΩ 2 7 AD797 3 VIN VOUT 6 RS* AD797 3 VOUT 6 600Ω 4 ** 4 –VS * 00846-022 *VALUE OF SOURCE RESISTANCE (SEE THE NOISE AND SOURCE IMPEDANCE CONSIDERATIONS SECTION). **SEE FIGURE 35. –VS *SEE FIGURE 35. Figure 22. Inverter Connection Figure 25. Follower Connection 1µs 5V 100 100 90 90 10 0% 00846-023 10 0% 5V Figure 23. Inverter Large-Signal Pulse Response 50mV 1µs 00846-026 1kΩ 7 Figure 26. Follower Large-Signal Pulse Response 100ns 50mV 100 100 90 90 10 0% 00846-024 10 0% 100ns 00846-027 VIN 2 * ** 00846-025 +VS Figure 27. Follower Small-Signal Pulse Response Figure 24. Inverter Small-Signal Pulse Response Rev. F | Page 9 of 20 AD797 500ns 50mV 100 100 90 90 10 0% 00846-028 10 0% Figure 28. 16-Bit Settling Time Positive Input Pulse 500ns 00846-029 50mV Figure 29. 16-Bit Settling Time Negative Input Pulse Rev. F | Page 10 of 20 AD797 THEORY OF OPERATION The architecture of the AD797 was developed to overcome inherent limitations in previous amplifier designs. Previous precision amplifiers used three stages to ensure high open-loop gain (see Figure 30) at the expense of additional frequency compensation components. Slew rate and settling performance are usually compromised, and dynamic performance is not adequate beyond audio frequencies. As can be seen in Figure 30, the first stage gain is rolled off at high frequencies by the compensation network. Second stage noise and distortion then appears at the input and degrade performance. The AD797, on the other hand, uses a single ultrahigh gain stage to achieve dc as well as dynamic precision. As shown in the simplified schematic (Figure 31), Node A, Node B, and Node C track the input voltage, forcing the operating points of all pairs of devices in the signal path to match. By exploiting the inherent matching of devices fabricated on the same IC chip, high open-loop gain, CMRR, PSRR, and low VOS are guaranteed by pairwise device matching (that is, NPN to NPN and PNP to PNP), not by an absolute parameter such as beta and the early voltage. VOUT BUFFER gm RL C1 R1 benefit of making the low noise of the AD797 (<0.9 nV/√Hz) extend to beyond 1 MHz. This means new levels of performance for sampled data and imaging systems. All of this performance as well as load drive in excess of 30 mA are made possible by the Analog Devices, Inc., advanced complementary bipolar (CB) process. Another unique feature of this circuit is that the addition of a single capacitor, CN (see Figure 31), enables cancellation of distortion due to the output stage. This can best be explained by referring to a simplified representation of the AD797 using idealized blocks for the different circuit elements (Figure 32). A single equation yields the open-loop transfer function of this amplifier; solving it at Node B yields VOUT V IN a. where: gm is the transconductance of Q1 and Q2. A is the gain of the output stage (~1). VOUT is voltage at the output. VIN is differential input voltage. VOUT C2 V IN A2 R1 gm C CN jω − C N jω − C jω A A When CN is equal to CC, the ideal single-pole op amp response is attained: GAIN = gm × R1 × 5 × 106 gm = BUFFER A3 VOUT C1 00846-030 GAIN = gm × R1 × A2 × A3 b. gm jωC In Figure 32, the terms of Node A, which include the properties of the output stage, such as output impedance and distortion, cancel by simple subtraction. Therefore, the distortion cancellation does not affect the stability or frequency response of the amplifier. With only 500 μA of output stage bias, the AD797 delivers a 1 kHz sine wave into 60 Ω at 7 V rms with only 1 ppm of distortion. RL R2 = Figure 30. Model of AD797 vs. That of a Typical Three-Stage Amplifier VCC R2 R3 I1 CN R1 I2 CN I5 Q4 Q10 Q7 A Q9 Q1 Q2 I1 –IN Q5 C Q6 CC Q12 A VOUT +IN Q8 –IN Q1 Q11 CURRENT MIRROR Q2 VOUT CC 1 I6 I7 I4 VSS 00846-031 +IN B A B I3 C I4 00846-032 Q3 Figure 32. AD797 Block Diagram Figure 31. AD797 Simplified Schematic This matching benefits not just dc precision, but, because it holds up dynamically, both distortion and settling time are also reduced. This single stage has a voltage gain of >5 × 106 and VOS < 80 μV, while at the same time providing a THD + noise of less than −120 dB and true 16-bit settling in less than 800 ns. The elimination of second-stage noise effects has the additional NOISE AND SOURCE IMPEDANCE CONSIDERATIONS The AD797 ultralow voltage noise of 0.9 nV/√Hz is achieved with special input transistors running at nearly 1 mA of collector current. Therefore, it is important to consider the total input-referred noise (eNtotal), which includes contributions Rev. F | Page 11 of 20 AD797 from voltage noise (eN), current noise (iN), and resistor noise (√4 kTRS). e N total = [e N 2 + 4 kTR S + (i N × R S ) 2 ]1 / 2 (1) where RS is the total input source resistance. The plot in Figure 7 uses a slightly different technique: an FFT-based instrument (Figure 34) is used to generate a 10 Hz “brickwall” filter. A low frequency pole at 0.1 Hz is generated with an external ac coupling capacitor, which is also the instrument being dc coupled. This equation is plotted for the AD797 in Figure 33. Because optimum dc performance is obtained with matched source resistances, this case is considered even though it is clear from Equation 1 that eliminating the balancing source resistance lowers the total noise by reducing the total RS by a factor of 2. Several precautions are necessary to attain optimum low frequency noise performance: At very low source resistance (RS < 50 Ω), the voltage noise of the amplifier dominates. As source resistance increases, the Johnson noise of RS dominates until a higher resistance of RS > 2 kΩ is achieved; the current noise component is larger than the resistor noise. • • • 100 TOTAL NOISE • 100kΩ +VS RESISTOR NOISE ONLY 1 * 1Ω 2 7 AD797 3 10 100 1000 10000 SOURCE RESISTANCE (Ω) 4 HP 3465 DYNAMIC SIGNAL ANALYZER (10Hz) –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Figure 33. Noise vs. Source Resistance The AD797 is the optimum choice for low noise performance if the source resistance is kept <1 kΩ. At higher values of source resistance, optimum performance with respect to only noise is obtained with other amplifiers from Analog Devices (Table 3). Table 3. Recommended Amplifiers for Different Source Impedances RS (kΩ) 0 to <1 1 to <10 10 to <100 >100 VOUT * 00846-033 0.1 1.5µF 6 Recommended Amplifier AD797 AD743/AD745, OP27/OP37, OP07 AD743/AD745, OP07 AD548, AD549, AD711, AD743/AD745 00846-034 NOISE (nV/√Hz) 10 Care must be used to account for the effects of RS. Even a 10 Ω resistor has 0.4 nV/√Hz of noise (an error of 9% when root sum squared with 0.9 nV/√Hz). The test setup must be fully warmed up to prevent eOS drift from erroneously contributing to input noise. Circuitry must be shielded from air currents. Heat flow out of the package through its leads creates the opportunity for a thermoelectric potential at every junction of different metals. Selective heating and cooling of these by random air currents appears as 1/f noise and obscures the true device noise. The results must be interpreted using valid statistical techniques. Figure 34. Test Setup for Measuring 0.1 Hz to 10 Hz Noise WIDEBAND NOISE Due to its single-stage design, the noise of the AD797 is flat over frequencies from less than 10 Hz to beyond 1 MHz. This is not true of most dc precision amplifiers, where second-stage noise contributes to input-referred noise beyond the audio frequency range. The AD797 offers new levels of performance in wideband imaging applications. In sampled data systems, where aliasing of out-of-band noise into the signal band is a problem, the AD797 outperforms all previously available IC op amps. BYPASSING CONSIDERATIONS LOW FREQUENCY NOISE Analog Devices specifies low frequency noise as a peak-to-peak quantity in a 0.1 Hz to 10 Hz bandwidth. Several techniques can be used to make this measurement. The usual technique involves amplifying, filtering, and measuring the amplifier noise for a predetermined test time. The noise bandwidth of the filter is corrected for, and the test time is carefully controlled because the measurement time acts as an additional low frequency roll-off. Taking full advantage of the very wide bandwidth and dynamic range capabilities of the AD797 requires some precautions. First, multiple bypassing is recommended in any precision application. A 1.0 μF to 4.7 μF tantalum in parallel with 0.1 μF ceramic bypass capacitors are sufficient in most applications. When driving heavy loads, a larger demand is placed on the supply bypassing. In this case, selective use of larger values of tantalum capacitors and damping of their lead inductance with small-value (1.1 Ω to 4.7 Ω) carbon resistors can achieve an improvement. Figure 35 summarizes power supply bypassing recommendations. Rev. F | Page 12 of 20 AD797 VS OR 0.1µF Low noise preamplification is usually performed in the noninverting mode (Figure 38). For lowest noise, the equivalent resistance of the feedback network should be as low as possible. The 30 mA minimum drive current of the AD797 makes it easier to achieve this. The feedback resistors can be made as low as possible, with consideration to load drive and power consumption. 4.7µF TO 22.0µF 4.7µF 0.1µF 1.1Ω TO 4.7Ω KELVIN RETURN USE SHORT LEAD LENGTHS (<5mm) LOAD CURRENT LOAD CURRENT CL 00846-035 KELVIN RETURN USE SHORT LEAD LENGTHS (<5mm) R2 Figure 35. Recommended Power Supply Bypassing THE NONINVERTING CONFIGURATION +VS Ultralow noise requires very low values of the internal parasitic resistance (rBB) for the input transistors (≈6 Ω). This implies very little damping of input and output reactive interactions. With the AD797, additional input series damping is required for stability with direct output to input feedback. A 100 Ω resistor (R1) in the inverting input (Figure 36) is sufficient; the 100 Ω balancing resistor (R2) is recommended but is not required for stability. The noise penalty is minimal (eNtotal ≈ 2.1 nV/√Hz), which is usually insignificant. R1 100Ω +VS * 2 VIN R2 100Ω 7 AD797 3 6 RL 600Ω 4 VOUT –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 00846-036 * Figure 36. Voltage Follower Connection Best response flatness is obtained with the addition of a small capacitor (CL < 33 pF) in parallel with the 100 Ω resistor (Figure 37). The input source resistance and capacitance also affect the response slightly, and experimentation may be necessary for best results. CL +VS * RS AD797 3 600Ω 4 * CS 7 3 VOUT 6 RL 4 * –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Figure 38. Low Noise Preamplifier Table 4 provides some representative values for the AD797 when used as a low noise follower. Operation on 5 V supplies allows the use of a 100 Ω or less feedback network (R1 + R2). Because the AD797 shows no unusual behavior when operating near its maximum rated current, it is suitable for driving the AD600/AD602 (see Figure 50) while preserving low noise performance. Optimum flatness and stability at noise gains >1 sometimes require a small capacitor (CL) connected across the feedback resistor (R1 of Figure 38). Table 4 includes recommended values of CL for several gains. In general, when R2 is greater than 100 Ω and CL is greater than 33 pF, a 100 Ω resistor should be placed in series with CL. Source resistance matching is assumed, and the AD797 should not be operated with unbalanced source resistance >200 kΩ/G. Table 4. Values for Follower with Gain Circuit R1 1 kΩ 300 Ω 33.2 Ω 16.5 Ω 10 Ω R2 1 kΩ 300 Ω 300 Ω 316 Ω (G − 1) × 10 Ω CL ≈20 pF ≈10 pF ≈5 pF Noise (Excluding RS) 3.0 nV/√Hz 1.8 nV/√Hz 1.2 nV/√Hz 1.0 nV/√Hz 0.98 nV/√Hz VOUT 6 –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Figure 37. Alternative Voltage Follower Connection 00846-037 VIN 7 2 AD797 VIN Gain 2 2 10 20 >35 100Ω 2 * R1 00846-038 VS The I-to-V converter is a special case of the follower configuration. When the AD797 is used in an I-to-V converter, for example as a DAC buffer, the circuit shown in Figure 39 should be used. The value of CL depends on the DAC, and if CL is greater than 33 pF, a 100 Ω series resistor is required. A bypassed balancing resistor (RS and CS) can be included to minimize dc errors. Rev. F | Page 13 of 20 AD797 100Ω DRIVING CAPACITIVE LOADS R1 The capacitive load driving capabilities of the AD797 are displayed in Figure 41. At gains greater than 10, usually no special precautions are necessary. If more drive is desirable, however, the circuit shown in Figure 42 should be used. For example, this circuit allows a 5000 pF load to be driven cleanly at a noise gain ≥2. +VS * IIN 2 7 AD797 3 600Ω 4 100nF –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Figure 39. I-to-V Converter Connection THE INVERTING CONFIGURATION The inverting configuration (see Figure 40) presents a low input impedance, R1, to the source. For this reason, the goals of both low noise and input buffering are at odds with one another. Nonetheless, the excellent dynamics of the AD797 makes it the preferred choice in many inverting applications, and with careful selection of feedback resistors, the noise penalties are minimal. Some examples are presented in Table 5 and Figure 40. CAPACITIVE LOAD DRIVE CAPABILITY * RS 00846-039 CS VOUT 6 10nF 1nF 100pF 10pF 1pF 10 1 100 1k CLOSED-LOOP GAIN 00846-041 20pF TO 120pF Figure 41. Capacitive Load Drive Capability vs. Closed-Loop Gain CL 20pF R2 1kΩ +VS 200pF 100Ω * R1 +VS 7 AD797 3 * VOUT 6 1kΩ RL 4 2 VIN AD797 * –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 3 00846-040 RS 7 33Ω VOUT 6 C1 4 * –VS Figure 40. Inverting Amplifier Connection *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Table 5. Values for Inverting Circuit Gain −1 −1 −10 R1 1 kΩ 300 Ω 150 Ω R2 1 kΩ 300 Ω 1500 Ω CL ≈20 pF ≈10 pF ≈5 pF 00846-042 2 VIN Figure 42. Recommended Circuit for Driving a High Capacitance Load Noise (Excluding RS) 3.0 nV/√Hz 1.8 nV/√Hz 1.8 nV/√Hz SETTLING TIME The AD797 is unique among ultralow noise amplifiers in that it settles to 16 bits (<150 μV) in less than 800 ns. Measuring this performance presents a challenge. A special test circuit (see Figure 43) was developed for this purpose. The input signal was obtained from a resonant reed switch pulse generator, available from Tektronix as calibration Fixture No. 067-0608-00. When open, the switch is simply 50 Ω to ground and settling is purely a passive pulse decay and inherently flat. The low repetition rate signal was captured on a digital oscilloscope after being amplified and clamped twice. The selection of plug-in for the oscilloscope was made for minimum overload recovery. Rev. F | Page 14 of 20 AD797 TO TEKTRONIX 7A26 OSCILLOSCOPE 1MΩ PREAMP INPUT SECTION The benefits of adding C1 are evident for closed-loop gains of ≥100. A maximum value of ≈33 pF at gains of ≥1000 is recommended. At a gain of 1000, the bandwidth is 450 kHz. 20pF Table 6 and Figure 45 summarize the performance of the AD797 with distortion cancellation and decompensation. 4.26kΩ 226Ω (VIA LESS THAN 1FT 50Ω COAXIAL CABLE) 2 – A2 AD829 3 250Ω 6 7 + 50pF 2× HP2835 4 2× HP2835 R1 VERROR × 5 R2 0.47µF 2 0.47µF AD797 +VS VIN –VS 1kΩ 6 3 1kΩ a. 100Ω 1kΩ R1 20pF 1kΩ – C2 A1 AD797 3 6 7 + C1 R2 2 51pF 4 8 AD797 0.1µF +VS 0.1µF VIN NOTES USE CIRCUIT BOARD WITH GROUND PLANE. VOUT 6 3 –VS C1, SEE TABLE C2 = 50pF – C1 b. 00846-043 1µF 1µF Figure 44. Recommended Connections for Distortion Cancellation and Bandwidth Enhancement Figure 43. Settling Time Test Circuit Table 6. Recommended External Compensation for Distortion Cancellation and Bandwidth Enhancement The AD797 has distortion performance (THD < −120 dB, @ 20 kHz, 3 V rms, RL = 600 Ω) unequaled by most voltage feedback amplifiers. At higher gains and higher frequencies, THD increases due to a reduction in loop gain. However, in contrast to most conventional voltage feedback amplifiers, the AD797 provides two effective means of reducing distortion as gain and frequency are increased: cancellation of the output stage’s distortion and gain bandwidth enhancement by decompensation. By applying these techniques, gain bandwidth can be increased to 450 MHz at G = 1000, and distortion can be held to −100 dB at 20 kHz for G = 100. The unique design of the AD797 provides cancellation of the output stage’s distortion. To achieve this, a capacitance equal to the effective compensation capacitance, usually 50 pF, is connected between Pin 8 and the output (see C2 in Figure 44). Use of this feature improves distortion performance when the closed-loop gain is more than 10 or when frequencies of interest are greater than 30 kHz. Bandwidth enhancement via decompensation is achieved by connecting a capacitor from Pin 8 to ground (see C1 in Figure 44). Adding C1 results in subtracting from the value of the internal compensation capacitance (50 pF), yielding a smaller effective compensation capacitance and therefore a larger bandwidth. A/B R1 R2 (Ω) (Ω) 909 100 1k 10 10 k 10 Gain 10 100 1000 A C1 (pF) 0 0 0 C2 (pF) 50 50 50 B 3 dB BW 6 MHz 1 MHz 110 kHz C1 (pF) 0 15 33 C2 (pF) 50 33 15 3 dB BW 6 MHz 1.5 MHz 450 kHz 0.01 –80 G = +1000 RL = 600Ω –90 THD (dB) DISTORTION REDUCTION 0.003 NOISE LIMIT, G = +1000 G = +1000 RL = 10kΩ –100 0.001 G = +100 RL = 600Ω NOISE LIMIT, G = +100 0.0003 –110 G = +10 RL = 600Ω –120 100 300 1k THD (%) 2 3k 10k 30k 100k 0.0001 300k FREQUENCY (Hz) Figure 45. Total Harmonic Distortion (THD) vs. Frequency @ 3 V rms for Figure 44b Differential Line Receiver The differential receiver circuit of Figure 46 is useful for many applications, from audio to MRI imaging. The circuit allows Rev. F | Page 15 of 20 00846-045 VIN 00846-044 TEKTRONIX CALIBRATION FIXTURE 8 AD797 extraction of a low level signal in the presence of commonmode noise. As shown in Figure 47, the AD797 provides this function with only 9 nV/√Hz noise at the output. Figure 48 shows the AD797 20-bit THD performance over the audio band and the 16-bit accuracy to 250 kHz. 20pF 1kΩ 1kΩ +VS ** 50pF* DIFFERENTIAL INPUT 7 2 8 AD797 VOUT 6 4 3 A General-Purpose ATE/Instrumentation I/O Driver The ultralow noise and distortion of the AD797 can be combined with the wide bandwidth, slew rate, and load drive of a current feedback amplifier to yield a very wide dynamic range general-purpose driver. The circuit shown in Figure 49 combines the AD797 with the AD811 in just such an application. Using the component values shown, this circuit is capable of better than −90 dB THD with a ±5 V, 500 kHz output signal. The circuit is, therefore, suitable for driving a high resolution ADC as an output driver in automatic test equipment (ATE) systems. Using a 100 kHz sine wave, the circuit drives a 600 Ω load to a level of 7 V rms with less than −109 dB THD and a 10 kΩ load at less than −117 dB THD. 22pF ** 1kΩ –VS R2 1kΩ 2kΩ +VS * 20pF +VS 2 00846-046 * OPTIONAL ** USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 7 AD797 Figure 46. Differential Line Receiver VIN 1kΩ 3 6 3 7 AD811 4 * 16 2 4 649Ω VOUT * –VS 649Ω 12 *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 10 Figure 49. A General-Purpose ATE/Instrumentation I/O Driver Ultrasound/Sonar Imaging Preamp 8 The AD600 variable gain amplifier provides the time-controlled gain (TCG) function necessary for very wide dynamic range sonar and low frequency ultrasound applications. Under some circumstances, it is necessary to buffer the input of the AD600 to preserve its low noise performance. To optimize dynamic range, this buffer should have a maximum of 6 dB of gain. The combination of low noise and low gain is difficult to achieve. The input buffer circuit shown in Figure 50 provides 1 nV/√Hz noise performance at a gain of 2 (dc to 1 MHz) by using 26.1 Ω resistors in its feedback path. Distortion is only −50 dBc at 1 MHz for a 2 V p-p output level and drops rapidly to better than −70 dBc at an output level of 200 mV p-p. 10 100 1k 10k 100k 1M 10M FREQUENCY (Hz) 00846-047 6 Figure 47. Output Voltage Noise Spectral Density for Differential Line Receiver 0.003 WITHOUT OPTIONAL 50pF CN –100 MEASUREMENT LIMIT –110 0.001 0.0003 THD (%) –90 0.0001 –120 –130 100 WITH OPTIONAL 50pF CN 300 1k 3k 10k 30k 100k 300k FREQUENCY (Hz) 00846-048 THD (dB) 6 00846-049 OUTPUT VOLTAGE NOISE (nV/√Hz) –VS 14 * Figure 48. Total Harmonic Distortion (THD) vs. Frequency for Differential Line Receiver Rev. F | Page 16 of 20 AD797 26.1Ω Professional Audio Signal Processing—DAC Buffers The low noise and low distortion of the AD797 make it an ideal choice for professional audio signal processing. An ideal I-to-V converter for a current output DAC would simply be a resistor to ground, were it not for the fact that most DACs do not operate linearly with voltage on their output. Standard practice is to operate an op amp as an I-to-V converter, creating a virtual ground at its inverting input. Normally, clock energy and current steps must be absorbed by the op amp output stage. However, in the configuration shown in Figure 53, Capacitor CF shunts high frequency energy to ground while correctly reproducing the desired output with extremely low THD and IMD. +VS * * 26.1Ω 7 AD797 3 VOUT 4 * * –VS 00846-050 VIN AD600 6 VS = ±6Vdc *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. Figure 50. An Ultrasound Preamplifier Circuit Amorphous (Photodiode) Detector Large area photodiodes (CS ≥ 500 pF) and certain image detectors (amorphous Si) have optimum performance when used in conjunction with amplifiers with very low voltage (rather than very low current noise). Figure 51 shows the AD797 used with an amorphous Si (CS = 1000 pF) detector. The response is adjusted for flatness using capacitor CL, and the noise is dominated by voltage noise amplified by the ac noise gain. The AD797’s excellent input noise performance gives 27 μV rms total noise in a 1 MHz bandwidth, as shown by Figure 52. CF 82pF 100Ω 3kΩ +VS * AD1862 DAC 2 C1 2000pF 7 AD797 3 4 * CL 50pF 100Ω VOUT 6 –VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 10kΩ Figure 53. A Professional Audio DAC Buffer +VS +VS * IS CS 1000pF 7 AD797 3 –IN +IN –50 60 –60 40 –70 20 –80 100k 1M FREQUENCY (Hz) 10M 0 100M Figure 54. Offset Null Configuration 00846-052 VOUT (dB Re 1V/µA) 80 NOISE 20kΩ VOS ADJUST VOLTAGE NOISE (mV rms (0.1Hz FREQUENCY)) 100 10k VOUT –VS Figure 51. Amorphous Detector Preamp –30 1k 3 4 00846-051 *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35. 100 6 5 1 –VS VOUT 7 AD797 4 * –40 2 VOUT 6 Figure 52. Total Integrated Voltage Noise and VOUT of Amorphous Detector Preamp Rev. F | Page 17 of 20 00846-054 2 00846-053 2 AD797 OUTLINE DIMENSIONS 0.400 (10.16) 0.365 (9.27) 0.355 (9.02) 8 5 1 4 0.280 (7.11) 0.250 (6.35) 0.240 (6.10) 0.100 (2.54) BSC 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.060 (1.52) MAX 0.210 (5.33) MAX 0.015 (0.38) MIN 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) SEATING PLANE 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.195 (4.95) 0.130 (3.30) 0.115 (2.92) 0.015 (0.38) GAUGE PLANE 0.430 (10.92) MAX 0.005 (0.13) MIN 0.014 (0.36) 0.010 (0.25) 0.008 (0.20) 0.070 (1.78) 0.060 (1.52) 0.045 (1.14) 070606-A COMPLIANT TO JEDEC STANDARDS MS-001 CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS. Figure 55. 8-Lead Plastic Dual In-Line Package [PDIP] Narrow Body (N-8) Dimensions shown in inches and (millimeters) 5.00 (0.1968) 4.80 (0.1890) 8 1 5 6.20 (0.2441) 5.80 (0.2284) 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) 0.50 (0.0196) 0.25 (0.0099) 45° 8° 0° 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. Figure 56. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches) Rev. F | Page 18 of 20 012407-A 4.00 (0.1574) 3.80 (0.1497) AD797 ORDERING GUIDE Model AD797AN AD797ANZ 1 AD797AR AD797AR-REEL AD797AR-REEL7 AD797ARZ1 AD797ARZ-REEL1 AD797ARZ-REEL71 AD797BR AD797BR-REEL AD797BR-REEL7 AD797BRZ1 AD797BRZ-REEL1 AD797BRZ-REEL71 1 Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] Z = RoHS Compliant Part. Rev. F | Page 19 of 20 Package Option N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 AD797 NOTES ©2008 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D00846-0-1/08(F) Rev. F | Page 20 of 20