UCC2956 UCC3956 Switch Mode Lithium-Ion Battery Charger Controller PRELIMINARY FEATURES DESCRIPTION • Precision 4.1V Reference (1%) • High Efficiency Battery Charger Solution • Average Current Mode Control from Trickle to Over Charge • Resistor Programmable Charge Currents • Internal State Logic Provides Four Charge States • Programmable Over Charge Time • Fully Differential Switch Mode Current Sensing The UCC3956 family of Switch Mode Lithium-Ion Battery Charger Controllers accurately control lithium-ion battery charging with a highly efficient average current control loop. This chip is designed to work as a stand alone charger controller for a single cell or multiple cell battery pack. This chip combines charge state logic and average current PWM control circuitry with a 14 bit counter to program the over charge time. The charge state logic indicates current or voltage control depending on the charge state. The chip includes undervoltage lockout circuitry to insure sufficient supply voltage is present before output switching starts. Additional circuit blocks include a differential current sense amplifier, a 1% voltage reference, voltage and current error amplifiers, PWM latch, charge state decode bits, and a 500mA output driver. • CHG Pin Initiates Charging BLOCK DIAGRAM UDG-96197-1 1/98 Powered by ICminer.com Electronic-Library Service CopyRight 2003 UCC2956 UCC3956 CONNECTION DIAGRAM ABSOLUTE MAXIMUM RATINGS Supply Voltage VDD, OUT . . . . . . . . . . . . . . . . . . . . . . . . . 20V Output Current Sink Continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120mA Peak . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 600mA Output Current Source Continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120mA Peak . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 600mA CS+, CS– Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.5 to VDD Current with CS+, CS– less than –0.5. . . . . . . . . . . . . 50mA Remaining Pin Voltages . . . . . . . . . . . . . . . . . . . . . –0.3V to 6V Storage Temperature . . . . . . . . . . . . . . . . . . . –65°C to +150°C Junction Temperature . . . . . . . . . . . . . . . . . . . –55°C to +150°C Lead Temperature (Soldering, 10 sec.) . . . . . . . . . . . . . +300°C Currents are positive into, negative out of the specified terminal. Consult Packaging Section of Databook for thermal limitations and considerations of packages. DIP-20, SOIC-20 (Top View) J or N, DW Packages ELECTRICAL CHARACTERISTICS: Unless otherwise specified, TA = –40°C to +85 for UCC2956 and 0°C to +70°C for UCC3956, COSC = 500pF, RSET = 70k, CTO = 169nF, VDD = 12V, TA = TJ. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Current Sense Amplifier (CSA) DC Gain CS– = 0, CS+ = –50mV and CS+ = –250mV 4.9 5 5.1 V/V CS+ = 0, CS– = 50mV and CS– = 250mV 4.9 5 5.1 V/V CAO CS+ = CS– = 0V 1.99 2.05 2.11 mV 50 65 dB CMRR VCM = 1.1V to 18V, VDD = 18V VOL CS+ = –0.2V, CS– = 0.5V, IO = 1mA 0.3 V CS+ = 0.5V, CS– = –0.2V, IO = –500µA 3.7 4.1 4.4 V VOH Output Source Current IBAT = 3V, VID = 700mV –500 µA Output Sink Current IBAT = 1V, VID = –700mV 500 µA 0.1 3 MHz 3dB Bandwidth VCM = 0V, CS+ - CS– = 100mV (Note 2) Current Error Amplifier (CEA) 0.1 0.5 µA IB 8V < VDD< 18V, CHGENB = REF 1.99 2.05 2.11 V CA– Voltage 8V < VDD < 18V, CAO = CA– AVO 60 90 dB 1 3 MHz GBW TJ = 25°C, F = 100kHz IO = 250µA, CA– = 3V 0.5 V VOL VOH IO = –1mA, CA– = 2V 3.7 4.1 4.4 V VCHGENB = GND 8 10 12 µA ICA–, Itrck_control Voltage Error Amplifier (VEA) IB Total Bias Current; Regulating Level 0.5 3 µA 10 mV VIO 8V < VDD < 18V, –0.2 < VCM < 5V AVO 60 90 dB 0.75 3 MHz GBW TJ = 25°C, F = 100kHz VOL IO = 500µA, VA– = 3.8V 0.2 1 V IO = –500µA, VA– = 4.4V 3.8 4.1 4.3 V VOH –1 1 µA VAO Leakage VCHGENB = GND, STAT0 = 0 and STAT1 = 0, VAO = 2.05V Pulse Width Modulator Maximum Duty Cycle CAO = 0.5V 85 92 100 % Modulator Gain CAO = 1.7V, 2.1V 57 64 71 %/V Powered by ICminer.com Electronic-Library Service CopyRight 2003 2 UCC2956 UCC3956 ELECTRICAL CHARACTERISTICS: Unless otherwise specified, TA = –40°C to +85 for UCC2956 and 0°C to +70°C for UCC3956, COSC = 500pF, RSET = 70k, CTO = 169nF, VDD = 12V, TA = TJ. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT PWM Oscillator (OSC) 90 100 110 kHz Frequency 7V < VDD < 18V Over Charge Timer (OCT) 4.65 5 5.35 Hz Frequency 7V < VDD < 18V (Note 1) Reference 4.06 4.1 4.14 V Initial Accuracy TJ = 25°C 4.05 4.1 4.15 V Accuracy 0 < TJ < 70°C, VDD = 8V to 18V Load Regulation 0 < IO < 2mA 3 15 mV 4.03 4.1 4.17 V Accuracy –40°C < TJ < 85°C, VDD = 8V to 18V Short Circuit I REF = 0V 8 20 30 mA Charge Enable Comparator (CEC) Threshold Voltage 1.9 2.05 2.15 V Input Bias Current –0.5 –0.2 µA Voltage Sense Comparator (VSC) Threshold Voltage Volts below VA+ 50 125 200 mV Charge Current Comparator (CIC) Threshold Voltage CS+ = CS– = 0, Function of IBAT = 2.05V 2 2.05 2.1 V/V Input Bias Current Total Bias Current; Regulating Level –0.5 –0.2 µA Output Stage IO = 10mA 0.1 0.3 V VOL IO = –10mA 0.1 0.5 V VOH, Volts Below VDD Rise Time COUT = 1nF 30 70 ns 30 70 ns Fall Time COUT = 1nF STAT0 and STAT1 Open Drain Outputs 15 30 mA Maximum Sink Current VOUT = 12V IOUT = 1mA 0.1 0.2 V VOL Charge Control (CHG) Threshold Voltage 1.5 1.8 2.1 V Charge Pin Pull Down 3.0 5.0 kΩ Resistance UVLO Section Turn-on Threshold 6.0 6.5 6.75 V Hysteresis 100 150 400 mV IDD IDD (Run) 5 8 mA 0.25 0.75 mA IDD (UVLO) VDD = 5V is pulled low by an internal buffer. Another low to high transition is required to reset the timer and restart charging. PIN DESCRIPTIONS CA–: The inverting input to the current error amplifier. CAO: The output of the current error amplifier and inverting input of the PWM comparator. This pin is driven high during shutdown. CS–, CS+: The inverting and non-inverting inputs to the current sense amplifier. This amplifier has a fixed gain of 5. CHGENB: The input to a comparator that detects when the battery voltage is low and places the charger in trickle charge. The charge enable comparator forces the output of the voltage error amplifier to a high impedance state while forcing a fixed 10µA current into the CA– to set the trickle charge. CHG: A rising edge triggered input pin that indicates charging. Once the internal 14 bit timer has timed out the chip enters its shutdown charge state. At this point CHG COSC: The oscillator ramp pin which has a capacitor (COSC) to ground. The ramp oscillates between 0.8V to 3.2V and the frequency is determined by: Powered by ICminer.com Electronic-Library Service CopyRight 2003 3 UCC2956 UCC3956 PIN DESCRIPTIONS (cont.) Frequency = 3.475 RSET: This pin programs the charge current for the oscillator ramp. The oscillator charge current is determined by: (COSC + 20pF) RSET A rising edge on CHG initiates the oscillator. 1.37V CTO: The slow oscillator ramp pin which is used to generate a clock signal for the 14 bit timer to program the over charge time. A capacitor to ground is charged and discharged with equal currents at a frequency programmed between 0.75Hz to 5Hz. The ramp oscillates between 1.0V to 3.0V and the frequency is determined by: Frequency = RSET . The trickle control current (Itrck_control) is determined by: 0.68V RSET . STAT0, STAT1: CMOS open drain binary output decode pins indicating the four different charge states. The maximum high voltage sense comparator. 0.06 CTO RSET The oscillator operates only while in overcharge. VA–: The inverting input to the voltage error amplifier that is used as a battery sense input. It is also the input to the voltage sense comparator. The bulk charge state is completed and over charge state is initiated when VA– reaches 95% of VA+. GND: The reference point for the internal reference, all thresholds, and the return for the remainder of the device. IBAT: The output of the current sense amplifier. VA+: The non-inverting input to the voltage error amplifier that is used as the battery charge reference voltage. IMIN: The minimum charge current programming pin is provided to program an optional charge termination in addition to the programmable timer. VAO: The output of the voltage error amplifier. The upper output clamp of this amplifier is 4.1V. OUT: The output of the PWM driver. VDD: The input voltage of the chip. This chip is operational between 6V and 18V and should be bypassed with a 0.1µF capacitor. REF: The 4.1V precision reference which should be bypassed with a 0.1µF capacitor. CHARGE STATE DECODE CHART Trickle Charge Bulk Charge Over Charge Over Charge (Top Off) STAT1 0 0 1 1 STAT0 0 1 0 1 CHGENB < 2.05V VA– < 95% VA+ and CHGENB > 2.05V VA– > 95% VA+ and VIBAT< VIMIN VIBAT > VIMIN APPLICATION INFORMATION The UCC3956 contains all the necessary control functions for implementing an efficient switch mode LithiumIon battery charger. Lithium-Ion batteries are rapidly becoming the battery of choice for rechargeable portable and lap top products. When compared to NiCd, NiMH, and Lead Acid batteries, Lithium-Ion offer less weight and volume for the same energy. Lithium-Ion batteries do not suffer from the memory effect found in NiCd batteries. This effect, caused by not completely discharging and charging a battery, will reduce battery capacity over several charge cycles. Because Lithium-Ion batteries have a high average cell voltage of around 3.6V, they can often replace 2 to 3 Nickel based cells. the cell will typically have a voltage of 2.5V. A fully charged cell will typically have a voltage of 4.1V. Unlike many so called “smart” or “universal” chargers, the UCC3956 is optimized for Lithium-Ion characteristics. In order to restore capacity quickly, the chip features both constant current and constant voltage modes of operation. A programmable over charge time, provided by the UCC3956 timer, allows the charger to predictably restore 100% capacity to the battery. Charger Operation When CHG is transitioned from a low to high logic level, the chip will cycle through several charge states. If the battery voltage is severely depleted, the charger will begin in a low current trickle charge state. When the bat- The advantages that Lithium-Ion batteries offer come at the cost of a wide operating voltage. Near zero capacity, Powered by ICminer.com Electronic-Library Service CopyRight 2003 4 UCC2956 UCC3956 APPLICATION INFORMATION (cont.) tery voltage is above a user set threshold, the charger will initiate a constant current bulk charge state. Once the battery reaches 95% of it’s final voltage, the charger will enter an over charge state. During the over charge state, the converter will transition from a constant current to a constant voltage mode of operation. Figure 2 shows typical current, voltage, and capacity levels of a Lithium-Ion battery during a complete charge cycle. The UCC3956 is capable of operating at frequencies higher than 200kHz. However, the actual operating frequency of the buck converter will ultimately be determined by the usual tradeoffs of size, cost and efficiency. The application circuit frequency is set at 100kHz with COSC = 180pF and RSET = 162k. Trickle Charge State When the battery’s voltage is below a predetermined threshold, the battery is either deeply discharged or has shorted cells. The trickle charge state offers a low charging current to bring the battery up above zero capacity. In the case of shorted cells, the trickle charge state prevents the charger from delivering high currents during this fault condition. Stacking several cells makes the detection of a shorted cell more difficult. A Block Diagram of the UCC3956 is shown on the first page of the data sheet, while Figure 1 shows a typical application circuit for a Buck derived switch mode charger. The UCC3956 can be used for charging a single cell or multiple cells in series. If more than two cells are stacked in series, however, a level shifting gate drive will be needed to operate the buck switch. The application circuit charges a 1200mAh 2 cell stack at a 1C rate. For Lithium-Ion batteries, the trickle charge threshold is typically set to a value around 2.5V per cell (this corresponds to near zero capacity). When the cell voltage is below the threshold, only a trickle current will be applied to the battery. The threshold is established by programming CHGENB to 2.05V when the battery (or stack) voltage is at the threshold. Referring to the application circuit Setting the Oscillator Frequency The frequency of operation for the converter is set by picking values for RSET and COSC. fOSC = 3.475 (COSC + 20pF) RSET IBULK = 1.2A ITRICKLE = 90 mA fOSC = 100kHz Timeout = 120 minutes UDG-96198-1 Figure 1. Typical Application Circuit Powered by ICminer.com Electronic-Library Service CopyRight 2003 5 UCC2956 UCC3956 APPLICATION INFORMATION (cont.) reasonably small value of inductance to be used. The average current mode of the UCC3956 provides improved discontinuous duty cycle control, when compared to peak current mode implementations. of Figure 1, the trickle charge voltage threshold is determined by: VTRICKLE_ THRESHOLD = RS1+ RS2 + RS3 RS3 2.05 In Figure 2, the trickle charge state corresponds to the time interval between t0 (when CHG is transitioned from low to high) and t1. During the trickle charge state STAT0 and STAT1 are logic level lows. At time t1 the trickle threshold is met, and the charger transitions to the bulk charge state. In many instances, the battery voltage will initially be above the trickle threshold. In this case, the trickle charge state will not be needed. With a trickle threshold of 5V (for 2 cells) and setting RS3 to 10k, RS1+ RS2 should be approximately 14.4k. The applications circuit trickle charge current is set to about 7.5% of the bulk charge current. The current value is set by picking the appropriate value for RG1. Referring to the Block Diagram and Figure 1, during trickle charge a fixed current 0.68 Bulk Charge State RSET As the name implies, the bulk charge state is responsible for restoring a majority of the charge back into the battery. The bulk charge current is determined by the C rate and the capacity of the battery. In the application circuit, 2 stacked 1200mAH batteries are charged at a 1C rate. This will require 1.2A of current during bulk charge. In this case, a fully discharged battery will take about 60 minutes to reach approximately 80% capacity. Battery packs with a high ESR will typically have a shorter bulk period, due to the voltage drop generated by the bulk current and the ESR of the battery. flows out of the current amplifier’s inverting input and into RG1. The voltage amplifier output is disabled during trickle charge and acts as a high impedance node. The resulting voltage at the output of the current sense am- Both the voltage and current sense amplifiers are enabled during bulk charge. The voltage amplifier is saturated in this state as the battery voltage is slowly rising, but is not yet high enough to drive the voltage amplifier into regulation. The output of voltage amplifier is clamped at a nominal voltage of 4.1V. The current sense amplifier is configured such that its output voltage increases with decreasing RSENSE current. RSENSE should be sized such that the output voltage of the current sense amplifier VIBAT is within specification during bulk charge. VIBAT(BULK) = 2.05 - 5 IBULK RSENSE With 1.2A of bulk current and setting the current sense amplifier output at 1V, a sense resistor of 0.18Ω is required. As always, power dissipation and converter efficiency must be considered when choosing RSENSE. UDG-96262-1 Figure 2. Typical Charge Cycle Levels Referring to the Feedback Diagram of Figure 3, the output of the voltage and current sense amplifiers are summed together at the inverting input of the current amplifier. Assuming that the current amplifier is within regulation, the required value of RG2 can be calculated. The application circuit uses a value of 38.3k for RG2, setting the bulk current to 1.2A. plifier sets the trickle charge current. ITRICKLE = RG1 7.5 RSET RSENSE In the application circuit the sense resistor is 0.18Ω and RSET is 162k, for a trickle current of about 90mA a 20k resistor is selected for RG1. RG2 = The converter is typically designed to run in discontinuous conduction mode during trickle charge. This allows a Powered by ICminer.com Electronic-Library Service CopyRight 2003 2.05 RG1 5 IBULK RSENSE Referring to Figure 2, the bulk charge state corresponds 6 UCC2956 UCC3956 APPLICATION INFORMATION (cont.) Figure 3. Simplified Feedback Diagram out of saturation. Therefore, bulk current may continue in the battery during the initial portion of the over charge state (see Figure 2). When the voltage amplifier comes into regulation, the amplifier’s output voltage will begin to decrease. The current sense amplifier’s output voltage will need to increase, in order for the current amplifier’s inverting input to remain at 2.05V. This will translate into a decreasing battery current. The battery current will continue to decrease as the battery approaches 100% capacity. to the interval between t1 and t2. The step in voltage at time t1 is caused by bulk current flowing into the battery ESR and sense resistor. In the bulk charge state STAT0 is a logic level high and STAT1 is a logic level low. Over Charge State The over charge state of the converter starts when the battery reaches 95% of its final voltage (time t2 of Figure 2). The over charge state is initiated when the voltage at the inverting input of the voltage amplifier is 95% of the non-inverting input voltage. Using 95% rather than 100% of the final battery voltage assures that the over charge timer will always be set, before the battery current tapers off. At the beginning of over charge STAT0 indicates a logic level low and STAT1 indicates a logic level high. Although the bulk charge state restores a majority of the capacity to the battery, the over charge state will typically take a majority of the charge cycle time. The bulk charge state will usually take 1/3 of the total charge time, while the over charge state will take the remaining 2/3. Different methods are used to terminate the charge of Lithium-Ion batteries. Many chargers use a current threshold to terminate charge. While this method is simple to implement, the current tail near the end of charge is often quite flat (see Figure 2). To make matters worse, the current level versus battery capacity may differ from cell to cell. This makes it difficult to accurately terminate at 100% capacity. In order to avoid the possibility of over charging the battery, the design may require termination at a higher current level (before 100% capacity is reached). A more predictable method of charge termination is to use a fixed over charge time. In the application circuit of Figure 1, the voltage at which over charge is initiated is set by resistors RS1, RS2 and RS3. These resistors are also used to set the trickle charge threshold. A 0.1µF decoupling capacitor is added to this node as a filter. The battery (or stack) voltage that will initiate the over charge state is: VOC_ THRESHOLD = 0.95 RS1+ RS2 + RS3 RS2 + RS3 4.1 For a single cell stack, RS1 should be 0Ω. This results in a final battery voltage of 4.1V. It is important not to charge a Lithium-Ion battery above 4.2V. When charging a battery stack, RS1 should be selected to properly set the final stack voltage. In the application circuit, RS1 is selected to be 12.21k and RS2 is selected to be 2.21k. This sets the over charge level at 8.2V, while setting the trickle charge threshold to about 5V. The UCC3956 provides both a current level detection as well as a timer. In a typical design, the current level detection is used to give an indication of near full charge. In Figure 2 this occurs at time t4. This indication is useful since the time to charge from t4 to t5 may be quite long. Since Lithium-Ion batteries have no memory effect, there is little reason to have the user wait for the battery to be The battery voltage at the beginning of the over charge state may not correspond to the voltage amplifier coming Powered by ICminer.com Electronic-Library Service CopyRight 2003 UDG-96263-1 7 UCC2956 UCC3956 APPLICATION INFORMATION (cont.) 100% charged. If the battery is not taken from the charger at time t4, the charger will continue charging. The timer will expire and the charge cycle will terminate at time t5. L= Current Control Loop The UCC3956 features an outer voltage loop and an inner average current loop. The virtues of average current mode control are well documented in Reference [1]. A simplified block diagram of the feedback elements is provided in Figure 3. The network for the current amplifier could be as simple as a single capacitor, providing a dominant pole response, which may be adequate for a battery charger application. The current amplifier network of Figure 3 provides improved transient performance. The component values for CF3, CF4, and RF4 will be selected to give a constant gain from approximately fOSC/10 to fOSC. At frequencies below fOSC/10, the network gain will increase at 20dB/decade, giving a high DC gain. The network will attenuate at 20dB/decade above the switching frequency, giving noise immunity. RS5 RS4 + RS5 INEAR_ FULL = 2.05 - VIMIN 5 RSENSE A feedback design that optimizes transient response will have the amplified inductor current down-slope approach the PWM saw-tooth slope [1]. This occurs by designing the total loop gain to cross unity at 1/3 to 1/6 of the switching frequency. The applications circuit is designed to cross unity gain at 1/10 of the switching frequency (10kHz), with a 12V nominal input. The power stage small signal gain can be approximated by: The UCC3956 timer has a 14 bit counter that allows long over-charge times with reasonable component values. As stated above, the charger will continue charging the battery until the timer expires (unless the battery is pulled from the charger). Referring to Figure 2, the timer starts at time t2 and expires at time t5. The frequency of the timer can be determined as follows: fTIMER = IBULK fOSC A 150µH inductor is used in the application circuit. A typical value of current used to indicate near full charge is 1/10 of the bulk current value. This current level is established by setting the appropriate voltage on IMIN. IMIN is tied to an internal comparator along with the output of the current sense amplifier. When the current sense amplifier voltage becomes greater than the voltage on IMIN, the internal state machine indicates near full charge by setting STAT0 and STAT1 to logic level highs. In the application circuit of Figure 1, resistors RS4 and RS5 determine the voltage at IMIN. With RS4 at 11k and RS5 at 10k, near full charge is indicated at 120mA. VIMIN = 4.1 4 (VINPUT - VBAT) D 0.06 RSET CTO GPOWER_ STAGE = With a 14 bit counter the time-out period in minutes becomes: VIN RSENSE SL + RSENSE + ESR Referring to Figure 3, the current sense amplifier pro- TIMEOUT = 4550 CTO RSET In the application circuit, a value of 0.15µF is used for CTO to give 120 minutes of overcharge (more than twice the bulk charge time). When the timer expires, CHG is pulled low by an internal buffer and the charge cycle terminates. If tied to a bi-directional port, CHG can be read by a microprocessor. 80 60 40 GAIN (dB) FEEDBACK Inductor Sizing For good efficiency, the inductor should be sized to give continuous current in the bulk charge state. For a buck converter, duty cycle in continuous mode is given by: GAIN 20 POWER STAGE 0 -20 -40 D= VBATTERY + VSCHOTTKY -60 VINPUT + VSCHOTTKY 10 100 1000 10000 100000 FREQUENCY Allowing a 25% ripple in the bulk current will give a reasonable value of inductance. The inductor value can be calculated as follows: Powered by ICminer.com Electronic-Library Service CopyRight 2003 Figure 4a. Current Loop Power Stage and Feedback Gain 8 1000000 UCC2956 UCC3956 APPLICATION INFORMATION (cont.) 120 80 100 TOTAL PHASE 60 FEEDBACK GAIN 60 40 40 GAIN (dB) GAIN (dB) OR PHASE (degrees) 80 20 0 20 0 -20 -20 TOTAL LOOP GAIN -40 -40 -60 POWER STAGE GAIN -80 -60 10 100 1000 10000 100000 1000000 10 100 FREQUENCY GAIN (dB) OR PHASE (degrees) 50 RG1 RF4 is selected to be 15k, resulting in a 10kHz crossover frequency. Once RF4 is determined, CF3 and CF4 can be selected to give corner frequencies at fOSC/10 and fOSC respectively. CF4 = 0 TOTAL LOOP GAIN -50 -150 10 100 1000 10000 100000 1000000 FREQUENCY π fOSC RF4 10 Figure 5b. Voltage Loop Total Gain and Phase π fOSC RF4 plifier begins to decrease, demanding less current to the battery. With the current loop closed, the power stage gain of the voltage loop is equal to 1/(5*RSENSE) out to the crossover frequency (10kHz). In order to avoid interactions with the current loop, the voltage loop will cross unity at 2kHz. The voltage loop is attenuated by the divider RG1/(RG1+RG2). A single pole network is added to the voltage amplifier, giving a high gain at DC. Referring to Figure 3, the voltage amplifier gain is equal to the impedance of CF1 divided by RS1. A 2.2nF capacitor will give a total crossover frequency near 2kHz. Figure 5a shows the gain of the power and feedback stages for the voltage loop. Figure 5b shows the total gain and phase of the voltage loop. In the applications circuit, a value of 100pF is used for CF3 and 1.0nF is used for CF4. Figure 4a shows the power stage gain and feedback network gain for the current loop. Figure 4b shows the total open loop gain and phase. Adding the Voltage Control Loop The voltage loop comes into regulation during the overcharge period of operation. The output of the voltage am- UNITRODE CORPORATION 7 CONTINENTAL BLVD. • MERRIMACK, NH 03054 TEL. (603) 424-2410 • FAX (603) 424-3460 Powered by ICminer.com Electronic-Library Service CopyRight 2003 TOTAL PHASE -100 1 2 1000000 100 RF4 2 100000 Figure 5a. Voltage Loop Power Stage Gain vides a gain of 5, an inverting stage adds a gain of 1.5, and the modulator has a gain of 0.64; adding a fixed gain of 4.8 to the power stage. The current amplifier’s gain between fOSC/10 and fOSC is equal to RF4 divided by the parallel combination of RG1 and RG2 times the resistive divider RG2/(RG1+RG2), simplifying to: CF3 = 10000 FREQUENCY Figure 4b. Current Loop Total Gain and Phase GCA = 1000 9 IMPORTANT NOTICE Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. 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INCLUSION OF TI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO BE FULLY AT THE CUSTOMER’S RISK. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other intellectual property right of TI covering or relating to any combination, machine, or process in which such semiconductor products or services might be or are used. TI’s publication of information regarding any third party’s products or services does not constitute TI’s approval, warranty or endorsement thereof. Copyright 1999, Texas Instruments Incorporated Powered by ICminer.com Electronic-Library Service CopyRight 2003