ETC AB-093

APPLICATION BULLETIN
®
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ISOLATION AMPS HIKE ACCURACY AND RELIABILITY
Isolation Amplifiers Protect Critical Circuitry From Damage and Enhance Performance
by Tom Sommerville
Fault tolerance, transient protection, and interference rejection are valuable features that isolation amplifiers bring to
critical circuitry. Two useful examples of how designers can
employ isolation amplifiers to improve their systems’ performance and reliability are process temperature controllers
and electrocardiogram (ECG) amplifiers.
In the case of process control, isolation amplifiers galvanically isolate both the input channel and the current-loop
output driver from the controller hardware. Consequently,
neither accidental faults from line-powered manufacturing
plant equipment to the control system circuitry nor ground
loop voltages can compromise the process. In ECG amplifiers, a low-capacitance isolation barrier limits 60Hz leakage
current to safe levels, and a high barrier voltage rating
protects the monitoring equipment from defibrillator transients and electrosurgery (ESU) interference.
Process control loops, in particular, illustrate a number of
ways that isolation amplifiers improve performance and
reliability. In unisolated control systems, the long ground
lines can develop error potentials across the common impedance that can cause component failure and/or inaccurate
control. In contrast, isolating the distributed control systems
(DCS) inputs and outputs close to the controller interrupts
the dc path, replacing it with the large impedance of the
isolation amp’s high-voltage barrier (Figure 1).
Similarly, isolation protects the loop from interruption or
damage. In the example loop, which is a temperature controller, isolation protects the circuit if the resistance-temperature detector (RTD), a PT100, is accidentally shorted to
a grounded metal case or high-voltage conductor. Another
possible problem is a fault from the high-potential wire of a
twisted pair to earth ground. In addition, high current transients from motors and relays sharing an unisolated system
ground can create voltages that exceed the ±1V maximum
rating of the two-wire transmitter (XTR).
Isolated control loops also benefit from improved rejection
of 60Hz line interference, because low-level transducers are
especially susceptible to inductive-loop coupling to the
60Hz magnetic field and capacitive coupling to the electric
fields. Not only can the isolation amp prevent system damage, but also the component’s isolation-mode rejection (IMR)
can attenuate the effect on the output by over 1 million to
one.
High dv/dt from inductive current and from radiated electromagnetic interference (EMI) caused by relay arcing are
©
1994 Burr-Brown Corporation
Signal
Two-wire
Transmitter
Power Supply
4 to 20mA
Input
Distributed
Control
System
Isolation
Amplifiers
PT100
Transducer
Power Supply
Heat
Source
Signal
Valve
Output
20mA to 0
FIGURE 1. Isolating the Distributed Control System in a
Process Control Loop Interrupts the DC Path,
Blocking Possible Error Potentials that Could
Cause Component Failure and Control Inaccuracy.
more difficult to deal with. An isolation amp, however, can
reduce their effect on system accuracy, depending on the
amplifier’s transient immunity. This parameter is defined as
the greatest dv/dt that can appear between isolated and
unisolated ground before accuracy is lost at the amplifier’s
output. Few data sheets specify transient immunity, which
can vary from 0.1 to 10,000V/µs.
Choosing the best isolation amplifier from process-control
applications isn’t a simple matter. The common use of 4-to20mA current loop transmitters and receivers requires an
isolated power supply that can supply at least 25mA for the
loop and any signal conditioning circuitry. Another consideration is the input supply-voltage range. Some isolation
amps need a regulated 15V supply, while others can tolerate
sharing an unregulated system supply. The IMR needed
depends on the 60Hz line voltage encountered. Most isolation amps specify greater than 100dB rejection at 60Hz,
which is adequate for the majority of process-control applications.
AB-093
1
Printed in U.S.A. May, 1995
1mA 1mA
+
3
11
–
10
R4
1/2W
750
8
5
R2
119.4
R1
98.5
9
6
4
+
C5
1µF
D1
1
16
10
22
24
23
21
11
3
12
2N2222A
Q1
15
RCV420
14
13
2
7
C1
0.01µF
PT100
12
XTR101
+15V
+
C3
1µF
5
C2
0.01µF
4
R3
2.5kΩ
2
ISO
3
10
11
12
9
103
4
16
VOUT
15
C4
1µF
–
C6
1µF
–15V
FIGURE 2. In an Example Process Temperature Controller, the Isolated Input Channel Runs from the PT100 Transducer to the
XTR101 Two-Wire Transmitter.
TESTS MAY BE NEEDED
As noted, transient immunity is important in process control,
with the character of the ground noise determining the level
required. If the immunity isn’t specified, or is specified at a
low barrier voltage, designers must test the amplifier to
determine its suitability.
Reliability testing of the barrier integrity varies between
manufacturers and their products. For process-control applications, the UL1244 standard calls for 100%, 60Hz ac
breakdown testing at the rated voltage for 1 minute. By
comparing the isolation amp’s test-voltage condition with its
continuous rating voltage, designers can judge how conservative the rating is.
To minimize the temperature rise in the XTR101, the external 2N2222A conducts all but 4mA of the transmitter current. The 750Ω, 1/2W resistor limits the worst-case power
dissipation in the transistor to 19.6 x 16mA = 314mW,
where 19.6V is VCE and 16mA is determined by the transistor and XTR. Thus, the dissipation is below the TO-18 limit
of 400mW at room temperature. For applications at higher
ambient temperatures, a 2N6121 in a TO-220 package is a
more reliable substitute. This small, current-mode transmitter can be located close to the RTD, with signal and power
conducted through one twisted-wire pair from the remote
monitoring point to the central controller. At the controller,
a proportional analog voltage is reconstructed.
The RCV420 current-loop receiver has an output of 0 to 5V
for a 4-to-20mA output. The maximum voltage drop across
the receiver’s input is 75Ω (the internal resistance between
pins 2 and 3) times 20mA, or 1.5V.
Accuracy specifications, however, are relatively uniform. At
one time, only expensive discrete modules or in-house
custom designs offered better than 0.05% linearity. But
recently, small and inexpensive (less than $30) hybrids with
comparable performance became available. In some cases,
surface-mounting techniques have reduced size and cost.
Other designs include custom ICs and high-frequency ferrite
transformers that reduce complexity, as well as size and
cost.
The isolated input channel of the example control loop runs
from the PT100 transducer to the DCS input, beginning with
the XTR101 two-wire transmitter configured for the required process temperature of 50°C to 150°C (Figure 2). The
PT100 temperature resistance table indicates a resistance of
119.4Ω at 50°C and 157.31Ω at 150°C. One of the XTR101’s
two 1mA current sources flows through the PT100, so the
input amplifier voltage span is 37.9mV.
The span resistor, R1, is calculated from the input voltage
and output current span (4 to 20mA from 50°C to 150°C):
ERROR COMPENSATION
The isolation amp, an ISO103, creates both an isolated
signal buffer and an isolated dc-dc converter power supply that energizes the XTR101 and RCV420, as well as
the ISO103’s internal input amplifier. By adjusting R1
and R2 for input-temperature-to-voltage accuracy, designers can compensate for gain and offset-voltage errors in
the ISO103 and RCV420. The ISO103’s high continuousvoltage rating means that the circuit can tolerate line
voltages to 1500Vrms. The amplifier’s isolation-mode
rejection is 130dB at 60Hz, high enough to limit the
interference of a 1500Vrms fault to 0.5mVrms, or an
error of 0.014°C referred to the input.
Designers can verify the input-channel linearity by measuring the temperature-to-voltage error. The resulting 0.2%
error, which equates to a maximum 0.2°C error, is due
almost entirely to the temperature-to-resistance nonlinearity
of the PT100. If greater accuracy or a wider temperature
range is needed, designers can apply an RTD resistance
R1 = 40/[(20mA – 4mA)/37.91mV – 0.016mΩ] = 98.5Ω
The offset resistor, R2, is equal the PT100’s resistance at
50°C: 119.4Ω. The 5V common-mode bias needed for the
XTR101’s inputs is supplied by R3 (2mA x 2.5kΩ).
2
VIN
0
5V
Z1
6.2V
C3
1µF
23 22 21
16 15 14
ISO
+15V
1
C1
10µF
–15V
2
3
R2
20kΩ
IL = 0 to 20mA
RL ≤ 950
C4
1µF
13
113
9
4
Q1
VN2222
10 11 12
+
R1
250Ω
0.1%
C2
1µF
FIGURE 3. On the Output Side, the Process Control Loop Includes an Isolated 0-20mA Loop Driver.
correction factor in the controller, leaving only the 0.04%
error from the ISO103.
The output half of the process control loop consists of the
isolated 0-to-20mA loop driver (Figure 3). The circuit receives a 0-to-5V input from the controller in the DCS. The
driver output current energizes an actuator valve that controls the steam pressure, which regulates the process temperature. Thus, the process control loop is closed by an
isolated current-to-pressure converter built around the ISO113
internally powered output isolation amp.
The circuit operates by closing a voltage feedback loop, such
that VIN is developed across R1. This is done by connecting
the common pin to one end of R1 and the sense pin to the
other, with VOUT connected to the gate of the VN2222
MOSFET.
To use the maximum compliance voltage at the output, the
common pin voltage is referenced off the negative supply
with a 6.2V zener diode, Z1. To keep the zener active when
the output current is zero, the driver output current flows
through the diode to the negative supply, along with a
1.5mA current from the positive supply through R2. For the
30V supply in the example, 19V remains across the MOSFET
and load, allowing for a maximum load resistance of 950Ω.
The ISO113’s input offset voltage, which can be compensated by the controller’s transfer function, determines the
accuracy of the output offset current (IOS = VOS/R1). Gain
accuracy depends on the accuracy and stability of R1. If
adequate care isn’t taken, power dissipation in R1 can degrade the transfer function’s linearity.
As with the ISO103, the ISO113 is rated for 1500Vrms
continuous isolation voltage. Under this maximum interference condition at 60Hz, the 130dB isolation-mode rejection
results in a 2µA output-current error, only 0.01% of fullscale and negligible in this application.
Medical instrumentation, although a totally different class of
application from process control, illustrates an extreme example of isolation protection that may also be needed in
process control, automated test equipment, and data acquisition systems. An ECG amplifier, for instance, is designed to
accurately amplify the heart muscle’s action potential sensed
by surface skin electrodes. A specific design example is an
amplifier configured for measuring the difference between
the left arm (LA) electrode and the right arm (RA) electrode
while driving the right leg (RL) with a small current. This is
referred to as the “Lead I” configuration (Figure 4).
AVOIDING INTERFERENCE
Corrupting interference from nearby 60Hz line-operated
equipment is minimized by the RL drive amplifier (IC2A),
the high input impedance and common-mode rejection ratio
(CMRR) of the instrumentation amplifier (IC1), and the low
barrier capacitance of the isolation amp (IC3). The barrier
capacitance, 13pF, also ensures patient safety by limiting
60Hz leakage current from 240Vrms power-line coupling to
below 2µArms, which is one-tenth of the Underwriters
Laboratories’ standard for medical and dental equipment.
The electrode input-current limiting resistors (R1-R3), the
transistor clamps (Q1 and Q2, both 2N3904s), and the isolation amp’s internal insulation protect the amplifier from
defibrillator and ESU high-voltage transients.
The bandwidth needed to faithfully amplify the ECG waveform is 0.05Hz to 100Hz. The low-frequency limit is needed
to attenuate slowly varying potentials caused by chemical
reactions at the electrode-skin interface. Therefore, the gain
of the dc-coupled instrumentation amplifier is set at 10 to
prevent the amplified electrode potentials from creating an
For example, the temperature change in R1, for a resistor
thermal resistance of 300°C/W is 30°. Consequently, a
resistor temperature coefficient of 50ppm/°C causes a 0.15%
nonlinearity. To reduce this error, either the resistor thermal
resistance or temperature coefficient must be reduced. The
thermal resistance can be lowered by connecting four separate 1kΩ resistors in parallel to form the 250Ω resistance, or
by using a large wire-wound resistor.
3
+
R1
RA 330kΩ
1
11, 12
R 16
Q1
–
+
8
4
2210Ω
C1
2.2nF
R2
LA 330kΩ
R5
2210Ω
Q2
R8
2kΩ R
9
100kΩ
10
9
IC1
INA110
2
7
2 +
C3
0.33µF
R10
100kΩ
6
–
C2
10nF
5
IC2B
+
R3
RL 330kΩ
8
1
IC2A
14
31
29
15
IC3
ISO107
7
4
19
20
R12 2kΩ
VOUT
R13
10kΩ
13 16
Q4
R14
10kΩ
D1
R6
10kΩ
R7
10MΩ
18
30
6
3
–15V
–
Q3
–15V
R11
10kΩ
2
C4
0.1µF
3
4
–
NOTE: IC2A,B:OPA2111
FIGURE 4. In an ECG Amplifier, an Isolation Amplifier Protects the Circuitry and the Patient from Leakage Current from 60Hz
Line Operated Equipment and From High-Voltage Transients Caused By Other Equipment.
overload. Input low-pass filtering for differential inputs is
performed by R1, R2 and C1. The upper 3dB frequency:
= (1/2π)/[R1 + R2)(C1)]
= (0.159)/[(600kΩ)(2.2nF)] = 110Hz
back loop’s stability. Using the INA110’s gain equation, the
gain-setting resistance for a gain of G is:
R4 = R5 = [20kΩ/(G – 1) – 25]Ω.
The second amplifier stage (IC2b) is an ac-coupled, variable
gain, noninverting amplifier. The lower 3dB frequency is
0.05Hz. The unit used is an OPA2111 JFET input op amp
with a 1014Ω input resistance and a 3pA input bias current.
These values allow the use of a 10MΩ input resistor without
introducing dc errors. With the variable gain feature, designers can set a nominal 1V/mV gain or adjust the output
waveform amplitude for patient ECG variations.
An important consideration is the ECG amplifier’s common-mode input impedance. Because the LA and RA electrode impedances can be imbalanced by as much as 100kΩ,
the differential signal seen by IC1 may be in error if the
impedance is below 10MΩ. However, the common-mode
input impedance of the instrumentation amplifier used, an
INA110, is large enough, 2 x 1012Ω, so the cable shield’s
capacitance of 100pF dominates the ECG amplifier’s common-mode input impedance. The resulting reactance of
26MΩ at 60Hz is high enough to prevent errors due to
electrode impedance mismatching.
The RL amplifier reduces the 60Hz noise resulting from the
magnetic and electric fields surrounding line-voltage sources.
Typically, this noise voltage is 0.1 to 1Vp-p, about 100 to
1000 times the ECG signal. The RL amplifier helps reject
the noise by reducing the common-mode voltage with negative feedback through the op amp integrator (IC2A).
This stage includes an automatic-gain-control feedback loop
that ensures a rapid recovery from ESU interference. The
loop caps the amplifier’s output at 2V, a level determined by
the MOSFET’s threshold voltage (0.8 to 3V), the diode peak
detector, and value of R9. By limiting the output to 2V, the
input is kept below 20mV, so C2 can quickly discharge
during transient recovery.
TWO SOURCES OF TRANSIENTS
The ECG amplifier must deal with two sources of highvoltage transients. One is the ESU, essentially a highvoltage RF generator. The other is the defibrillator, which
generates electrical pulses.
The ESU has two modes: coagulate and cut. In the coagulate
mode, the unit applies a gated 1MHz damped sinusoid to
close small ruptured blood vessels with RF heating. In the
cut mode, a lower frequency (300kHz) sinusoid with less
damping is applied to tissue with a scalpel electrode, facilitating “bloodless” surgery (Figure 5).
The patient lies on a large electrode, supplying a return path
for the high-frequency current. Another current path is
created by the stray capacitance coupling the primary and
secondary windings of the power-supply transformer, which
The amount of loop gain available at 60Hz depends on the
compensation needed to stabilize the loop. The isolation
supplied by IC3 reduces the required compensation by adding series barrier capacitance between the patient and earth
ground, thereby maximizing the available 60Hz noise rejection.
A pair of gain-setting resistors (R4 and R5) create an ac
common-mode voltage sense point for the RL drive amplifier. The dc voltage at this point is 1V lower than the inputs,
so the driver amplifier floats the patient to +1V relative to
isolated ground as a consequence of forcing the commonmode point to zero. The design aims to maximize the RL
drive amplifier’s gain at 60Hz while maintaining the feed-
4
is 126Hz. The peak current and voltage turn out to be 69A
and 3460V. Some fraction of this transient appears at the
ECG electrodes and between the isolated patient ground and
the output (earth) ground.
During ESU and defibrillator operation, the two 2N3904
transistors clamp the INA110’s inputs at +8V and –0.7V.
With no overvoltages present, the transistor’s leakage current is less than 100pA. The 330kΩ input resistors limit the
peak defibrillator input current to less than 10mA, but they
must dissipate 41W of instantaneous power. Fortunately, the
defibrillator charges slowly enough to avoid significant
resistor heating and damage.
Larger ESU generators, however, can deliver 300W of RF
power to the patient. The result is a worst case of 300mW
dissipated in the 1/2W input resistors.
The ISO107 supplies an isolated gain of one for the ECG
signal and an isolated supply for the INA110 and OPA2111.
Besides limiting the 60Hz leakage current, the unit’s barrier
capacitance limits barrier RF current during ESU operation.
The path of the RF leakage current is through the 50pF stray
capacitance of the ESU line transformer, the amplifier’s
power-supply transformer capacitance, and the ISO107 isolation barrier (Figure 6).
The 13pF barrier capacitance conducts a transient current of
57mA peak. Though this level doesn’t damage the isolation
amp, it corrupts the output signal, because the slew rate of
the interference is 4400V/µs. As a result, an output latch is
used to ground the output during the interference. When the
output exceeds 1.4V, Q4 (also a 2N3904) conducts, forcing
the output to VSAT, which the output sense sees as positive
feedback. To release the clamp, the input voltage of the
isolation amp must go below ground.
The performance of the ECG amplifier is determined by the
frequency response, the common-mode rejection, and the
FIGURE 5. An Electrosurgery Unit Generates Two Damped
Sinusoids that Can Cause Transients in the ECG
Amplifier. The coagulate waveform is at 1MHz
(top), and the cut waveform is at 300kHz (bottom).
is connected to line voltage. The isolation amp completes
this second path through its power-supply winding capacitance and barrier capacitance.
The defibrillator, on the other hand, charges a 16µF highvoltage capacitor to an adjustable voltage level calibrated in
terms of energy (watt-seconds). The capacitor is discharged
against the patient’s chest through a 100mH inductor and
two large metal electrodes, or paddles. The result is an
underdamped transient known as the Lown waveform.
The value of this transient is derived by completing the LCR
circuit with a 50Ω resistor model for the human body. The
voltage that appears between the paddles is the voltage
across the 50Ω resistor. For a setting of 400Ws, the capacitor
is charged to 7kV, and the frequency of the damped sinusoid
CSTRAY 50pF
CSTRAY = 50pF
Electrosurgery +
Unit Generator
13pF
Scalpel
Line
Voltage
Patient
Return Pad
Right
Leg Drive
Isolation
Amplifier
Line
Voltage
FIGURE 6. Stray Capacitances Coupling the Windings of the ESU and Amplifier Power Transformers Supply Paths for RF
Leakage Current.
5
response to ESU interference and defibrillator pulses. Using
the isolation circuitry described, the amplifier’s 3dB bandwidth is 0.05 to 100Hz and CMRR ranges from 95dB at
0.05Hz to 80dB at 100Hz.
To simulate a patient load, a 50Ω resistor was placed across
the defibrillator output. With a 400Ws defibrillator pulse
applied directly between the amplifier’s isolated and output
grounds, the output clamps and briefly oscillates during the
transient, but settles back before the transient ends.
To test the ESU interference response, an ECG simulator
was used to apply a test input waveform of 1mV amplitude,
and 240beats/min. The isolation barrier RF interference
voltage and the ECG output were then displayed on an
oscilloscope. The output clamp effectively zeroes the output
during the transient, as expected, then releases during the
first negative-going input once the RF ceases (Figure 7).
FIGURE 7. With an Isolation Barrier RF Interference Voltage Applied (top), the Amplifier’s Output Clamp
Zeroes the Output (bottom).
Reprinted from Electronic Design, February 22, 1990.
The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes
no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change
without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant
any BURR-BROWN product for use in life support devices and/or systems.
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