LT3475/LT3475-1 Dual Step-Down 1.5A LED Driver U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ True Color PWMTM Delivers Constant Color with 3000:1 Dimming Range Wide Input Range: 4V to 36V Operating, 40V Maximum Accurate and Adjustable Control of LED Current from 50mA to 1.5A High Side Current Sense Allows Grounded Cathode LED Operation Open LED (LT3475) and Short Circuit Protection LT3475-1 Drives LED Strings Up to 25V Accurate and Adjustable 200kHz to 2MHz Switching Frequency Anti-Phase Switching Reduces Ripple Uses Small Inductors and Ceramic Capacitors Available in the Compact 20-Lead TSSOP Thermally Enhanced Surface Mount Package The LT®3475/LT3475-1 are dual step-down DC/DC converters designed to operate as a constant-current source. An internal sense resistor monitors the output current allowing accurate current regulation ideal for driving high current LEDs. The high side current sense allows grounded cathode LED operation. High output current accuracy is maintained over a wide current range, from 50mA to 1.5A, allowing a wide dimming range. Unique PWM circuitry allows a dimming range of 3000:1, avoiding the color shift normally associated with LED current dimming. The high switching frequency offers several advantages, permitting the use of small inductors and ceramic capacitors. Small inductors combined with the 20 lead TSSOP surface mount package save space and cost versus alternative solutions. The constant switching frequency combined with low-impedance ceramic capacitors result in low, predictable output ripple. U APPLICATIO S ■ ■ ■ ■ Automotive and Avionic Lighting Architectural Detail Lighting Display Backlighting Constant-Current Sources With its wide input range of 4V to 36V, the LT3475/LT3475-1 regulate a broad array of power sources. A current mode PWM architecture provides fast transient response and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patents Pending. U TYPICAL APPLICATIO Dual Step-Down 1.5A LED Driver Efficiency VIN 5V TO 36V 4.7μF BOOST2 SW1 *DIMMING CONTROL 2.2μF 0.1μF 1.5A LED CURRENT 10μH 85 SW2 OUT1 LED1 OUT2 LED2 PWM1 PWM2 VC1 VC2 REF RT VADJ1 DIMMING* CONTROL 24.3k 1.5A LED CURRENT SINGLE WHITE 1.5A LED 75 70 fSW = 600kHz 60 55 3475 TA01 *SEE APPLICATIONS SECTION FOR DETAILS 80 65 2.2μF 0.1μF VADJ2 GND TWO SERIES CONNECTED WHITE 1.5A LEDS 0.22μF LT3475 10μH VIN = 12V 90 EFFICIENCY (%) 0.22μF 95 SHDN VIN BOOST1 0 0.5 1 1.5 LED CURRENT (A) 3475 TA01b 3475fb 1 LT3475/LT3475-1 U W W W ABSOLUTE AXI U RATI GS PIN CONFIGURATION (Note 1) TOP VIEW VIN Pin .........................................................(-0.3V), 40V BOOST Pin Voltage ...................................................60V BOOST Above SW Pin ...............................................30V OUT, LED, Pins (LT3475) ...........................................15V OUT, LED Pins (LT3475-1).........................................25V PWM Pin ...................................................................15V VADJ Pin ......................................................................6V VC, RT, REF Pins ..........................................................3V SHDN Pin ...................................................................VIN Maximum Junction Temperature (Note 2)............. 125°C Operating Temperature Range (Note 3) LT3475E/LT3475E-1 ............................. –40°C to 85°C LT3475I/LT3475I-1 ............................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature Range (Soldering, 10 sec) ....... 300°C OUT1 1 20 PWM1 LED1 2 19 VADJ1 BOOST1 3 18 VC1 SW1 4 17 REF VIN 5 VIN 6 SW2 7 14 RT BOOST2 8 13 VC2 LED2 9 12 VADJ2 16 SHDN 21 OUT2 10 15 GND 11 PWM2 FE PACKAGE 20-LEAD PLASTIC TSSOP TJMAX = 125°C, θJA = 30°C/W, θJC = 8°C/W EXPOSED PAD (PIN 21) IS GROUND AND MUST BE ELECTRICALLY CONNECTED TO THE PCB. ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3475EFE#PBF LT3475EFE#TRPBF LT3475EFE 20-Lead Plastic TSSOP –40°C to 85°C LT3475IFE#PBF LT3475IFE#TRPBF LT3475IFE 20-Lead Plastic TSSOP –40°C to 125°C LT3475EFE-1#PBF LT3475EFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 85°C LT3475IFE-1#PBF LT3475IFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3) PARAMETER CONDITIONS MIN ● Minimum Input Voltage Input Quiescent Current Not Switching Shutdown Current SHDN = 0.3V, VBOOST = VOUT = 0V TYP MAX UNITS 3.7 4 V 6 8 mA 0.01 2 μA 3475fb 2 LT3475/LT3475-1 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3) PARAMETER CONDITIONS LED Pin Current VADJ Tied to VREF • 2/3 MIN TYP MAX UNITS 1.00 0.350 ● 0.97 0.94 0.336 0.325 0.31 1.03 1.04 0.364 0.375 0.385 A A A A A ● 1.22 1.25 1.27 V ● VADJ Tied to VREF • 7/30 LT3475E/LT3475E-1 0°C to 85°C REF Voltage Reference Voltage Line Regulation 4V < VIN < 40V Reference Voltage Load Regulation 0 < IREF < 500μA ● VADJ Pin Bias Current (Note 4) 0.05 %/V 0.0002 %/μA 40 400 nA Switching Frequency RT = 24.3k ● 530 600 640 kHz Maximum Duty Cycle RT = 24.3k RT = 4.32k RT = 100k ● 90 95 80 98 Switching Phase RT = 24.3k 150 180 Foldback Frequency RT = 24.3k, VOUT = 0V 210 80 SHDN Threshold (to Switch) SHDN Pin Current (Note 5) % % % VSHDN = 2.6V PWM Threshold Deg kHz 2.5 2.6 2.74 V 7 9 11 μA 0.3 0.8 1.2 V 0.8 V VC Source Current VC Switching Threshold VC = 1V 50 μA VC Sink Current VC = 1V 50 μA LED to VC Transresistance 500 LED to VC Current Gain V/A 1 mA/μA VC to Switch Current Gain 2.6 A/V VC Clamp Voltage 1.8 V VC Pin Current in PWM Mode VC = 1V, VPWM = 0.3V ● VOUT = 4V, VPWM = 0.3V ● OUT Pin Clamp Voltage (LT3475) OUT Pin Current in PWM Mode 13.5 Switch Current Limit (Note 6) 2.3 10 400 nA 14 14.5 V 25 50 μA 2.7 3.2 A Switch VCESAT ISW =1.5A 350 500 mV BOOST Pin Current ISW =1.5A 25 40 mA Switch Leakage Current 0.1 10 μA Minimum Boost Voltage Above SW 1.8 2.5 V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3475I and LT3475I-1 are guaranteed to meet performance specifications over the –40°C to 125°C operating temperature range. Note 4: Current flows out of pin. Note 5: Current flows into pin. Note 6: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. 3475fb 3 LT3475/LT3475-1 U W TYPICAL PERFOR A CE CHARACTERISTICS LED Current vs VADJ 1.50 LED Current vs Temperature TA = 25°C 600 VADJ = VREF • 2/3 1.25 0.75 0.50 SWITCH ON VOLTAGE (mV) 1.00 0.8 0.6 VADJ = VREF • 7/30 0.4 0 –50 –25 0 0 0.25 0.5 0.75 VADJ (V) 1.25 1 100 MINIMUM 1.5 1.0 TA = 25°C 0 20 40 60 DUTY CYCLE (%) Current Limit vs Output Voltage 3.5 3.0 3.0 2.5 2.5 2.0 1.5 1.0 50 25 75 0 TEMPERATURE (°C) 100 Oscillator Frequency vs Temperature 1.0 0 125 0 550 500 450 125 1.5 2.0 2.5 VOUT (V) 3.0 3.5 4.0 3475 G06 TA = 25°C RT = 24.3kΩ 600 TA = 25°C 1000 500 400 300 200 100 10 0 0.5 1.0 1.5 2.0 2.5 VOUT (V) 3475 G07 1.0 Oscillator Frequency vs RT 0 100 0.5 OSCILLATOR FREQUENCY (kHz) OSCILLATOR FREQUENCY (kHz) 600 50 25 75 0 TEMPERATURE (˚C) 1.5 Oscillator Frequency Foldback 700 RT = 24.3kΩ 400 –50 –25 2.0 3475 G05 3475 G04 650 TA = 25°C 0.5 0 –50 –25 100 80 2.0 3475 G03 0.5 0 1.5 0.5 1.0 SWITCH CURRENT (A) 0 CURRENT LIMIT (A) CURRENT LIMIT (A) CURRENT LIMIT (A) TYPICAL 0.5 OSCILLATOR FREQUENCY (kHz) 125 Switch Current Limit vs Temperature 3.0 700 200 3475 G02 Switch Current Limit vs Duty Cycle 2.0 300 0 50 25 75 0 TEMPERATURE (˚C) 3475 G01 2.5 400 100 0.2 0.25 TA = 25°C 500 1.0 LED CURRENT (A) LED CURRENT (A) Switch On Voltage 1.2 3475 G08 1 10 RT (kΩ) 100 3475 G09 3475fb 4 LT3475/LT3475-1 U W TYPICAL PERFOR A CE CHARACTERISTICS Boost Pin Current 35 7 TA = 25°C 50 TA = 25°C 14 INPUT CURRENT LT3475-1 12 20 15 10 OUTPUT VOLTAGE (V) 25 5 4 3 2 1.0 LT3475-1 10 20 OUTPUT VOLTAGE 15 0 40 30 2 0 5 1.26 4 VIN (V) 1.27 10 TA = 25°C TA = 25°C TO START 9 TO RUN LED VOLTAGE 8 TO START LED VOLTAGE 7 2 1.23 1 100 125 0 0 Minimum Input Voltage, Two Series Connected 1.5A White LEDs 3 1.24 40 3475 G12 VIN (V) 6 50 25 75 0 TEMPERATURE (˚C) 30 VIN (V) Minimum Input Voltage, Single 1.5A White LED 1.28 1.22 –50 –25 20 10 3475 G11 Reference Voltage 4 LT3475 VIN (V) 3475 G10 1.25 6 20 5 0 2.0 1.5 SWITCH CURRENT (A) 8 25 0 0.5 0 LT3475 30 10 1 0 10 35 TO RUN 6 5 0 0.5 1 LED CURRENT (A) 3475 G13 1.5 3475 G14 0 0.5 1 LED CURRENT (A) 1.5 3475 G15 U U U PI FU CTIO S OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the current sense resistor. Connect this pin to the inductor and the output capacitor. BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. LED1, LED2 (Pins 2, 9): The LED pin is the output of the current sense resistor. Connect the anode of the LED here. GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND pin and the exposed pad directly to the ground plane. The exposed pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation. Use a large ground plane and thermal vias to optimize thermal performance. VIN (Pins 5, 6): The VIN pins supply current to the internal circuitry and to the internal power switches and must be locally bypassed. SW1, SW2 (Pins 4, 7): The SW pin is the output of the internal power switch. Connect this pin to the inductor, switching diode and boost capacitor. 3475fb 5 INPUT CURRENT (mA) INPUT CURRENT (mA) 40 5 VREF (V) TA = 25°C 45 6 30 BOOST PIN CURRENT (mA) Open-Circuit Output Voltage and Input Current Quiescent Current LT3475/LT3475-1 U U U PI FU CTIO S RT (Pin 14): The RT pin is used to set the internal oscillator frequency. Tie a 24.3k resistor from RT to GND for a 600kHz switching frequency. VC1, VC2 (Pins 18, 13): The VC pin is the output of the internal error amp. The voltage on this pin controls the peak switch current. Use this pin to compensate the control loop. SHDN (Pin 16): The SHDN pin is used to shut down the switching regulator and the internal bias circuits. The 2.6V switching threshold can function as an accurate undervoltage lockout. Pull below 0.3V to shut down the LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/ LT3475-1. Tie to VIN if the SHDN function is unused. VADJ1, VADJ2 (Pins 19, 12): The VADJ pin is the input to the internal voltage-to-current amplifier. Connect the VADJ pin to the REF pin for a 1.5A output current. For lower output currents, program the VADJ pin using the following formula: ILED = 1.5A • VADJ/1.25V. REF (Pin 17): The REF pin is the buffered output of the internal reference. Either tie the REF pin to the VADJ pin for a 1.5A output current, or use a resistor divider to generate a lower voltage at the VADJ pin. Leave this pin unconnected if unused. PWM1, PWM2 (Pins 20, 11): The PWM pin controls the connection of the VC pin to the internal circuitry. When the PWM pin is low, the VC pin is disconnected from the internal circuitry and draws minimal current. If the PWM feature is unused, leave this pin unconnected. BLOCK DIAGRAM VIN RT CIN VIN SHDN RT INT REG AND UVLO D1 VIN MASTER OSC C1 C1 Q R ∑ SLOPE COMP SLOPE COMP MOSC 1 SW1 L1 Q Q1 SLAVE OSC D3 ∑ C2 C2 R Q S Q MOSC 2 S DRIVER Q2 SLAVE OSC FREQUENCY FOLDBACK L2 D4 FREQUENCY FOLDBACK OUT2 + – 0.067Ω 100Ω LED1 2V 2V – + 100Ω COUT2 0.067Ω LED2 gm1 DLED1 SW2 DRIVER OUT1 COUT1 D2 BOOST2 BOOST1 gm2 DLED 2 1.25V PWM 1 PWM2 Q3 Q4 VC1 VC2 1.25k 1.25k CC1 CC2 VADJ1 REF VADJ2 EXPOSED PAD GND 3475 BD 3475fb 6 LT3475/LT3475-1 OPERATION The LT3475 is a dual constant frequency, current mode regulator with internal power switches capable of generating constant 1.5A outputs. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3475 is shut down and draws minimal current from the input source tied to VIN. If the SHDN pin exceeds 1V, the internal bias circuits turn on, including the internal regulator, reference and oscillator. The switching regulators will only begin to operate when the SHDN pin exceeds 2.6V. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output current by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.8V, and an active clamp of 1.8V limits the output current. The voltage on the VADJ pin sets the current through the LED pin. The NPN, Q3, pulls a current proportional to the voltage on the VADJ pin through the 100Ω resistor. The gm amplifier servos the VC pin to set the current through the 0.067Ω resistor and the LED pin. When the voltage drop across the 0.067Ω resistor is equal to the voltage drop across the 100Ω resistor, the servo loop is balanced. Tying the REF pin to the VADJ pin sets the LED pin current to 1.5A. Tying a resistor divider to the REF pin allows the programming of LED pin currents of less than 1.5A. LED pin current can also be programmed by tying the VADJ pin directly to a voltage source. An LED can be dimmed with pulse width modulation using the PWM pin and an external NFET. If the PWM pin is unconnected or is pulled high, the part operates nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way, the VC pin and the output capacitor store the state of the LED pin current until the PWM is pulled high again. This leads to a highly linear relationship between pulse width and output light, allowing for a large and accurate dimming range. The RT pin allows programming of the switching frequency. For applications requiring the smallest external components possible, a fast switching frequency can be used. If low dropout or very high input voltages are required, a slower switching frequency can be programmed. During startup VOUT will be at a low voltage. The NPN, Q3, can only operate correctly with sufficient voltage of ≈1.7V at VOUT, A comparator senses VOUT and forces the VC pin high until VOUT rises above 2V, and Q3 is operating correctly. The switching regulator performs frequency foldback during overload conditions. An amplifier senses when VOUT is less than 2V and begins decreasing the oscillator frequency down from full frequency to 15% of the nominal frequency when VOUT = 0V. The OUT pin is less than 2V during startup, short circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions. The switch driver operates either from VIN or from the BOOST pin. An external capacitor and Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. 3475fb 7 LT3475/LT3475-1 APPLICATIONS INFORMATION Open Circuit Protection The LT3475 has internal open-circuit protection. If the LED is absent or is open circuit, the LT3475 clamps the voltage on the LED pin at 14V. The switching regulator then operates at a very low frequency to limit the input current. The LT3475-1 has no internal open circuit protection. With the LT3475-1, be careful not to violate the ABSMAX voltage of th BOOST pin; if VIN > 25V, external open circuit protection circuitry (as shown in Figure 1) may be necessary.The output voltage during an open LED condition is shown in the Typical Performance Characteristics section. OUT 10k 100k 3475 F01 Figure 1. External Overvoltage Protection Circuitry for the LT3475-1 LT3475 VIN VIN 2.6V Undervoltage Lockout Undervoltage lockout (UVLO) is typically used in situations where the input supply is current limited, or has high source resistance. A switching regulator draws constant power from the source, so the source current increases as the source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. An internal comparator will force the part into shutdown when VIN falls below 3.7V. If an adjustable UVLO threshold is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.6V. An internal resistor pulls 9μA to ground from the SHDN pin at the UVLO threshold. Choose resistors according to the following formula: R2 = 2.6V VTH – 2.6V – 9μA R1 VTH = UVLO Threshold Example: Switching should not start until the input is above 8V. VTH = 8V R1=100k 2.6V R2 = = 57.6k 8V – 2.6V – 9μA 100k VC 22V R1 VC SHDN 9μA C1 R2 GND 3475 F02 Figure 2. Undervoltage Lockout Keep the connections from the resistors to the SHDN pin short and make sure the coupling to the SW and BOOST pins is minimized. If high resistance values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from switching nodes. Setting the Switching Frequency The LT3475 uses a constant frequency architecture that can be programmed over a 200kHz to 2MHz range with a single external timing resistor from the RT pin to ground. A graph for selecting the value of RT for a given operating frequency is shown in the Typical Applications section. Table 1. Switching Frequencies SWITCHING FREQUENCY (MHz) RT (kΩ) 2 4.32 1.5 6.81 1.2 9.09 1 11.8 0.8 16.9 0.6 24.3 0.4 40.2 0.3 57.6 0.2 100 3475fb 8 LT3475/LT3475-1 APPLICATIONS INFORMATION Table 1 shows suggested RT selections for a variety of switching frequencies. Operating Frequency Selection The choice of operating frequency is determined by several factors. There is a tradeoff between efficiency and component size. A higher switching frequency allows the use of smaller inductors at the cost of increased switching losses and decreased efficiency. Another consideration is the maximum duty cycle. In certain applications, the converter needs to operate at a high duty cycle in order to work at the lowest input voltage possible. The LT3475 has a fixed oscillator off time and a variable on time. As a result, the maximum duty cycle increases as the switching frequency is decreased. Input Voltage Range The minimum operating voltage is determined either by the LT3475’s undervoltage lockout of 4V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: ( VOUT + VF ) DC = ( VIN – VSW + VF ) where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.4V at maximum load). This leads to a minimum input voltage of: V +V VIN(MIN ) = OUT F – VF + VSW DCMAX with DCMAX = 1–tOFF(MIN) • f where t0FF(MIN) is equal to 167ns and f is the switching frequency. The maximum operating voltage is determined by the absolute maximum ratings of the VIN and BOOST pins, and by the minimum duty cycle. V +V VIN(MAX ) = OUT F – VF + VSW DCMIN with DCMIN = tON(MIN) • f where tON(MIN) is equal to 140ns and f is the switching frequency. Example: f = 750kHz, VOUT = 3.4V DCMIN = 140ns • 750kHz = 0.105 3.4V + 0.4V VIN(MAX ) = – 0.4V + 0.4V = 36V 0.105 The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might allow a higher maximum operating voltage. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the Absolute Maximum Ratings of the VIN and BOOST pins. The input voltage should be limited to the VIN operating range (36V) during overload conditions (short circuit or start up). Minimum On Time The LT3475 will regulate the output current at input voltages greater than VIN(MAX). For example, an application with an output voltage of 3V and switching frequency of 1.2MHz has a VIN(MAX) of 20V, as shown in Figure 3. Figure 4 shows operation at 35V. Output ripple and peak inductor VOUT 500mV/DIV (AC COUPLED) IL 1A/DIV Example: f = 600kHz, VOUT = 4V DCMAX = 1− 167ns • 600kHz = 0.90 4V + 0.4V VIN(MIN ) = – 0.4V + 0.4V = 4.9V 0.9 VSW 20V/DIV 3475 F03 Figure 3. Operation at VIN(MAX) = 20V. VOUT = 3V and fSW = 1.2MHHz 3475fb 9 LT3475/LT3475-1 APPLICATIONS INFORMATION current have significantly increased. Exceeding VIN(MAX) is safe if the external components have adequate ratings to handle the peak conditions and if the peak inductor current does not exceed 3.2A. A saturating inductor may further reduce performance. VOUT 500mV/DIV (AC COUPLED) IL 1A/DIV VSW 20V/DIV 3475 F04 Figure 4. Operation above VIN(MAX). Output Ripple and Peak Inductor Current Increases Table 2. Inductors VALUE (μH) IRMS (A) DCR ( ) HEIGHT (mm) CR43-3R3 3.3 1.44 0.086 3.5 CR43-4R7 4.7 1.15 0.109 3.5 CDRH4D16-3R3 3.3 1.10 0.063 1.8 CDRH4D28-3R3 3.3 1.57 0.049 3.0 CDRH4D28-4R7 4.7 1.32 0.072 3.0 CDRH6D26-5R0 5.0 2.20 0.032 2.8 CDRH6D26-5R6 5.6 2.0 0.036 2.8 CDRH5D28-100 10 1.30 0.048 3.0 CDRH5D28-150 15 1.10 0.076 3.0 CDRH73-100 10 1.68 0.072 3.4 CDRH73-150 15 1.33 0.130 3.4 CDRH104R-150 15 3.1 0.050 4.0 DO1606T-332 3.3 1.30 0.100 2.0 DO1606T-472 4.7 1.10 0.120 2.0 DO1608C-332 3.3 2.00 0.080 2.9 DO1608C-472 4.7 1.50 0.090 2.9 MOS6020-332 3.3 1.80 0.046 2.0 MOS6020-472 10 1.50 0.050 2.0 DO3316P-103 10 3.9 0.038 5.2 DO3316P-153 15 3.1 0.046 5.2 PART NUMBER Sumida Coilcraft Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L = (VOUT + VF ) • 1.2MHz f where VF is the voltage drop of the catch diode (~0.4V), f is the switching frequency and L is in μH. With this value the maximum load current will be above 1.6A at all duty cycles. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.15Ω. Table 2 lists several vendors and types that are suitable. For robust operation at full load and high input voltages (VIN > 30V), use an inductor with a saturation current higher than 3.2A. The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations: L MIN = (VOUT + VF ) • 800kHz f 3475fb 10 LT3475/LT3475-1 APPLICATIONS INFORMATION The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3475 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3475 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor ΔIL = (1– DC)( VOUT + VF ) (L • f ) where f is the switching frequency of the LT3475 and L is the value of the inductor. The peak inductor and switch current is ΔI ISW (PK ) = IL (PK ) = IOUT + L 2 To maintain output regulation, this peak current must be less than the LT3475’s switch current limit ILIM. ILIM is at least 2.3A at low duty cycles and decreases linearly to 1.8A at DC = 0.9. The maximum output current is a function of the chosen inductor value: IOUT (MAX ) = ILIM – ΔIL 2 = 2.3A• (1–0.25•DC) – ΔIL 2 Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3475 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2. Input Capacitor Selection Bypass the input of the LT3475 circuit with a 4.7μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3475 input and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple. However, a conservative value is the RMS input current for the channel that is delivering most power (VOUT • IOUT): CINRMS = IOUT • VOUT (VIN – VOUT ) IOUT < VIN 2 and is largest when VIN = 2VOUT (50% duty cycle). As the second, lower power channel draws input current, the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single channel is ~1.5A, RMS ripple current will always be less than 0.75A. The high frequency of the LT3475 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10μF. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. 3475fb 11 LT3475/LT3475-1 APPLICATIONS INFORMATION An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10μF in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring, doubling the input voltage and damaging the LT3475. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work. The following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilizes the LT3475’s control loop. Because the LT3475 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equation: VRIPPLE = ΔIL / (8 • f • COUT) for ceramic capacitors where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ΔIL / 12 The low ESR and small size of ceramic capacitors make them the preferred type for LT3475 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Table 3 lists several capacitor vendors. Table 3. Low ESR Surface Mount Capacitors. VENDOR TYPE SERIES Taiyo-Yuden Ceramic X5R, X7R AVX Ceramic X5R, X7R TDK Ceramic X5R, X7R Diode Selection The catch diode (D3 from the Block Diagram) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT (VIN – VOUT)/VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch current limit. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 4 lists several Schottky diodes and their manufacturers. Diode reverse leakage can discharge the output capacitor during LED off times while PWM dimming. If operating at high ambient temperatures, use a low leakage Schottky for the widest PWM dimming range. 3475fb 12 LT3475/LT3475-1 APPLICATIONS INFORMATION Table 4. Schottky Diodes VF at 1A (mV) VR (V) IAVE(A) (A) MBR0540 40 0.5 620 MBRM120E 20 1 530 MBRM140 40 1 550 B120 20 1 500 B130 30 1 500 B140HB 40 1 530 DFLS140 40 1.1 510 B240 40 2 30 1 VF at 2A (mV) On Semiconductor Diodes Inc 500 International Rectifier 10BQ030 420 BOOST Pin Considerations The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22μF capacitor and fast switching diode (such as the CMDSH-3 or MMSD914LT1) will work well. Figure 5 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 3.3V and higher, the standard circuit (Figure 5a) is best. For outputs between 2.8V and 3.3V, use a small Schottky diode (such as the BAT-54). For lower output voltages, the boost diode can be tied to the input (Figure 5b). The circuit in Figure 5a is more efficient because the BOOST pin current comes from a lower voltage source. The anode of the boost diode can be tied to another source that is at least 3V. For example, if you are generating a 3.3V output, and the 3.3V output is on whenever the LED is on, the BOOST pin can be connected to the 3.3V output. For LT3475-1 applications with higher output voltages, an additional Zener diode may be necessary (Figure 5d) to maintain pin voltage below the absolute maximum. In any case, be sure that the maximum voltage at the BOOST pin is both less than 60V and the voltage difference between the BOOST and SW pins is less than 30V. The minimum operating voltage of an LT3475 application is limited by the undervoltage lockout (~3.7V) and by the maximum duty cycle. The boost circuit also limits the minimum input voltage for proper start up. If the input voltage ramps slowly, or the LT3475 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges D2 D2 C3 BOOST VIN VIN C3 BOOST LT3475 LT3475 VOUT SW VIN VIN VOUT SW GND GND VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT (5a) (5b) D2 D2 VIN2 > 3V BOOST BOOST C3 LT3475 VIN VIN C3 LT3475 SW VOUT VIN VIN GND VOUT SW GND 3475 F05 VBOOST – VSW ≅ VIN2 MAX VBOOST ≅ VIN2 + VIN MINIMUM VALUE FOR VIN2 = 3V VBOOST – VSW – VZ MAX VBOOST ≅ VIN + VOUT – VZ (5c) 3475 F05 (5d) Figure 5. Generating the Boost Voltage 3475fb 13 LT3475/LT3475-1 APPLICATIONS INFORMATION with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load current generally goes to zero once the circuit has started. The typical performance characteristics section shows a plot of minimum load to start and to run as a function of input voltage. Even without an output load current, in many cases the discharged output capacitor will present a load to the switcher that will allow it to start. The plots show the worst case, where VIN is ramping very slowly. Programming LED Current The LED current can be set by adjusting the voltage on the VADJ pin. For a 1.5A LED current, either tie VADJ to REF or to a 1.25V source. For lower output currents, program the VADJ using the following formula: ILED = 1.5A • VADJ/1.25V. Voltages less than 1.25V can be generated with a voltage divider from the REF pin, as shown in Figure 6. In order to have accurate LED current, precision resistors are preferred (1% or better is recommended). Note that the VADJ pin sources a small amount of bias current, so use the following formula to choose resistors: R2 = the voltage on the VADJ pin by tying a low on resistance FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation program an LED current of no less than 50mA. The maximum current dimming ratio (IRATIO) can be calculated from the maximum LED current (IMAX) and the minimum LED current (IMIN) as follows: IMAX/IMIN = IRATIO Another dimming control circuit (Figure 8) uses the PWM pin and an external NFET tied to the cathode of the LED. An external PWM signal is applied to the PWM pin and the gate of the NFET (For PWM dimming ratios of 20 to 1 or less, the NFET can be omitted). The average LED current is proportional to the duty cycle of the PWM signal. When the PWM signal goes low, the NFET turns off, turning off the LED and leaving the output capacitor charged. The PWM pin is pulled low as well, which disconnects the VC pin, storing the voltage in the capacitor tied there. Use the C-RC string shown in Figure 8 and Figure 9 tied to the VC pin for proper operation during startup. When the PWM pin goes high again, the LED current returns rapidly to its previous on state since the compensation and output capacitors are at the correct voltage. This fast settling time allows the VADJ 1.25V – VADJ + 50nA R1 REF R1 To minimize the error from variations in VADJ pin current, use resistors with a parallel resistance of less than 4k. Use resistor strings with a high enough series resistance so as not to exceed the 500μA current compliance of the REF pin. Dimming Control LT3475 VADJ GND R2 3475 F07 DIM Figure 7. Dimming with a MOSFET and Resistor Divider There are several different types of dimming control circuits. One dimming control circuit (Figure 7) changes PWM 100Hz TO 10kHz VC PWM 10k LT3475 REF 3.3nF LED R1 LT3475 VADJ R2 0.1μF GND 3475 F08 GND 3475 F06 Figure 6. Setting VADJ with a Resistor Divider Figure 8. Dimming Using PWM Signal 3475fb 14 LT3475/LT3475-1 APPLICATIONS INFORMATION LT3475 to maintain diode current regulation with PWM pulse widths as short as 7.5 switching cycles (12.5μs for fSW = 600kHz). Maximum PWM period is determined by the system and is unlikely to be longer than 12ms. Using PWM periods shorter than 100μs is not recommended. The maximum PWM dimming ratio (PWMRATIO) can be calculated from the maximum PWM period (tMAX) and minimum PWM pulse width (tMIN) as follows: tMAX/tMIN = PWMRATIO Total dimming ratio (DIMRATIO) is the product of the PWM dimming ratio and the current dimming ratio. Example: IMAX = 1A, IMIN = 0.1A, tMAX = 9.9ms tMIN = 3.3μs (fSW = 1.4MHz) IRATIO = 1A/0.1A =10:1 PWMRATIO = 9.9ms/3.3μs = 3000:1 DIMRATIO = 10 • 3000 = 30000:1 Layout Hints As with all switching regulators, careful attention must be paid to the PCB layout and component placement. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. The voltage signal of the SW and BOOST pins have sharp rise and fall edges. Minimize the area of all traces connected to the BOOST and SW pins and always use a ground plane under the switching regulator to minimize interplane coupling. In addition, the ground connection for frequency setting resistor RT and capacitors at VC1, VC2 pins (refer to the Block Diagram) should be tied directly to the GND pin and not shared with the power ground path, ensuring a clean, noise-free connection. 20 19 18 17 16 15 14 13 12 11 3 4 5 6 7 8 9 10 PWM2 2 To achieve the maximum PWM dimming ratio, use the circuit shown in Figure 9. This allows PWM pulse widths as short as 4.5 switching cycles (7.5μs for fSW = 600kHz). Note that if you use the circuit in Figure 9, the rising edge of the two PWM signals must align within 100ns. SHDN 1 PWM1 VIN 220pF RT VC 10k LT3475 1M 3.3nF 0.1μF PWM1 RT GND 3475 F09 3475 F10 VIA TO LOCAL GND PLANE Figure 9. Extending the PWM Dimming Range Figure 10. Recommended Component Placement 3475fb 15 LT3475/LT3475-1 TYPICAL APPLICATIONS Dual Step-Down 1A LED Driver VIN 5V TO 36V C1 4.7μF 50V D3 VIN SHDN BOOST1 C4 0.22μF 6.3V L2 10μH D4 BOOST2 C3 0.22μF 6.3V LT3475 SW1 L1 10μH SW2 D1 D2 C5 2.2μF 6.3V OUT1 OUT2 LED1 LED2 C2 2.2μF 6.3V LED 1 LED 2 C6 0.1μF R2 1k VC1 VC2 REF RT VADJ1 C7 0.1μF VADJ2 GND R3 2k R1 24.3k 3475 TA02 C1 TO C5: X5R OR X7R D1, D2: DFLS140 D3, D4: MBR0540 LED CURRENT = 1A fSW = 600kHz Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming VIN 6V TO 36V C1 4.7μF 50V D3 VIN SHDN BOOST1 C2 0.22μF 6.3V L2 10μH 1.5A LED CURRENT C3 0.22μF 6.3V LT3475 SW1 C4 2.2μF 6.3V D4 BOOST2 L1 10μH SW2 C5 2.2μF 6.3V D2 D1 OUT1 OUT2 LED1 LED2 PWM1 PWM2 LED 1 LED 2 R3 10k C6 3.3nF VC1 VC2 REF RT VADJ1 C7 3.3nF R4 10k VADJ2 M1 M2 GND C8 0.1μF C9 0.1μF M3 C8 220p 1M R2 R1 24.3k fSW = 600kHz PWM1 1.5A LED CURRENT 3475 TA03 PWM2 D1, D2: B260 D3, D4: MBR0540 C1 TO C5: X5R OR X7R M1, M2: Si2302ADS M3: 2n7002L 3475fb 16 LT3475/LT3475-1 TYPICAL APPLICATIONS Step-Down 3A LED Driver VIN 5V TO 36V C1 4.7μF 50V D3 VIN SHDN BOOST1 C2 0.22μF 6.3V L2 10μH D4 BOOST2 C3 0.22μF 6.3V LT3475 L1 10μH SW2 SW1 C5 2.2μF 6.3V D2 D1 OUT1 OUT2 C4 2.2μF 6.3V LED1 LED2 C6 0.1μF VC1 VC2 REF RT VADJ1 C7 0.1μF VADJ2 R1 24.3k GND 3A LED CURRENT LED 1 fSW = 600kHz D1, D2: B240A D3, D4: MBR0540 C1 TO C5: X5R OR X7R 3475 TA04 Dual Step-Down LED Driver with Series Connected LEDs VIN 10V TO 36V D3 C1 4.7μF 50V VIN SHDN BOOST1 C2 0.22μF 10V L2 15μH D4 BOOST2 C3 0.22μF 10V LT3475 SW2 SW1 D2 D1 C4 2.2μF 10V 1.5A LED CURRENT L1 15μH OUT1 OUT2 LED1 LED2 C5 2.2μF 10V LED 1 VC1 VC2 LED 2 C6 0.1μF REF RT C7 0.1μF LED 3 D1, D2: B240A D3, D4: MMSD4148T1 C1 TO C5: X5R OR X7R VADJ1 VADJ2 GND R1 24.3k 1.5A LED CURRENT LED 4 fSW = 600kHz 3475 TA05 3475fb 17 LT3475/LT3475-1 TYPICAL APPLICATIONS Dual Step-Down 1.5A Red LED Driver VIN 5V TO 28V C1 4.7μF 35V D3 VIN BOOST1 C2 0.22μF 35V L2 10μH D4 SHDN BOOST2 C3 0.22μF 35V LT3475 SW1 SW2 D2 D1 C4 2.2μF 6.3V C6 0.1μF 1.5A LED CURRENT LED 1 D1, D2: B240A D3, D4: MMSD4148T1 C1 TO C5: X5R OR X7R L1 10μH OUT1 OUT2 LED1 LED2 VC1 VC2 REF RT VADJ1 VADJ2 GND C5 2.2μF 6.3V C7 0.1μF R1 24.3k LED 2 1.5A LED CURRENT fSW = 600kHz 3475 TA06 3475fb 18 LT3475/LT3475-1 PACKAGE DESCRIPTION FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation CB 6.40 – 6.60* (.252 – .260) 3.86 (.152) 3.86 (.152) 20 1918 17 16 15 14 13 12 11 6.60 ±0.10 2.74 (.108) 4.50 ±0.10 6.40 2.74 (.252) (.108) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 9 10 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.25 REF 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 1.20 (.047) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE20 (CB) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3475fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3475/LT3475-1 TYPICAL APPLICATION Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output VIN 21V TO 36V D1 C1 4.7μF 50V D2 C2 0.22μF 16V L1 33μH SHDN VIN BOOST1 LT3475-1 SW1 R1 1k D7 C4 2.2μF 25V D5 D6 Q1 L2 33μH R2 1k 12V TO 18V LED VOLTAGE VC1 VC2 REF RT VADJ1 R6 100k D4 OUT2 LED2 12V TO 18V LED VOLTAGE C6 0.1μF C3 0.22μF 16V SW2 OUT1 LED1 R4 10k D3 BOOST2 VADJ2 GND 1.5A LED CURRENT* R5 10k C7 0.1μF R3 24.3k 1.5A LED CURRENT* C5 2.2μF 25V D8 R7 100k Q2 fSW = 600kHz 3475 TA08 D1, D4: 7.5V ZENER DIODE D2, D3: MMSD4148 D5, D6: B240A D7, D8: 22V ZENER DIODE R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL Q1, Q2: MMBT3904 C1 TO C5: X5R or X7R *DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C. RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1618 Constant-Current, 1.4MHz, 1.5A Boost Converter VIN(MIN) = 1.6V, VIN(MAX) = 18V, VOUT(MAX) = 35V, Analog/PWM, ISD < 1μA, MS10 Package LT3466 Dual Full Function Step-Up LED Driver Drivers Up to 20 LEDs, VIN: 2.7V to 24V, VOUT(MAX) = 40V, DFN, TSSOP16E Packages LT3474 36V, 1A (ILED), 2MHz Step-Down LED Driver VIN(MIN) = 4V, VIN(MAX) = 36V, 400:1 True Color PWM, ISD < 1μA, TSSOP16E Package LT3477 42V, 3A, 3.5MHz Boost, Buck-Boost, Buck LED Driver VIN(MIN) = 2.5V, VIN(MAX) = 25V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA, QFN, TSSOP20E Packages LT3479 3A, Full-Featured DC/DC Converter with VIN(MIN) = 2.5V, VIN(MAX) = 24V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA, Soft-Start and Inrush Current Protection DFN, TSSOP Packages LT3846 Dual 1.3A, 2MHz, LED Driver VIN: 2.5V to 24V, VOUT(MAX) = 36V, 1000:1 True Color PWMTM Dimmin, DFN, TSSOP16E Packages 3475fb 20 Linear Technology Corporation LT 1007 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2006