LTC3410-1.875 2.25MHz, 300mA Synchronous Step-Down Regulator in SC70 U FEATURES DESCRIPTIO ■ The LTC ®3410-1.875 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 26µA, dropping to <1µA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3410-1.875 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ High Efficiency: Up to 93% Very Low Quiescent Current: Only 26µA Low Output Voltage Ripple 300mA Output Current at VIN = 3V 380mA Minimum Peak Switch Current 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Stable with Ceramic Capacitors Shutdown Mode Draws < 1µA Supply Current ±2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Available in Low Profile SC70 Package Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors. The LTC3410-1.875 is specifically designed to work well with ceramic output capacitors, achieving very low output voltage ripple and a small PCB footprint. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. The LTC3410-1.875 is available in a tiny, low profile SC70 package. U APPLICATIO S ■ ■ ■ ■ Cellular Telephones Wireless and DSL Modems Digital Cameras MP3 Players Portable Instruments , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5994885, 6127815, 6304066, 6498466, 6580258, 6611131. U ■ TYPICAL APPLICATIO Efficiency and Power Loss vs Output Current 100 4.7µH VOUT 1.875V SW COUT 4.7µF CER LTC3410-1.875 RUN VOUT GND 80 4.2V 70 60 3.6V 0.01 50 40 20 0.1 3.6V 2.7V 30 34101875 TA01 1 4.2V POWER LOSS VIN POWER LOSS (W) CIN 4.7µF CER VIN EFFICIENCY (%) VIN 2.7V TO 5.5V 2.7V EFFICIENCY VIN 90 0.001 10 0 0.1 1 10 100 OUTPUT CURRENT (mA) 0.0001 1000 34101875 TA02 34101875f 1 LTC3410-1.875 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Supply Voltage .................................. – 0.3V to 6V RUN, VOUT Voltages................................... – 0.3V to VIN SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 500mA N-Channel Switch Sink Current (DC) ................. 500mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Notes 3, 5) ...................... 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW RUN 1 6 VOUT GND 2 5 GND SW 3 4 VIN LTC3410ESC6-1.875 SC6 PACKAGE 6-LEAD PLASTIC SC70 SC6 PART MARKING TJMAX = 125°C, θJA = 250°C/ W LCFQ Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER CONDITIONS IVOUT Output Voltage Feedback Current IPK Peak Inductor Current VOUT Regulated Output Voltage ∆VOUT Output Voltage Line Regulation VIN = 2.5V to 5.5V VLOADREG Output Voltage Load Regulation ILOAD = 50mA to 250mA VIN Input Voltage Range VUVLO Undervoltage Lockout Threshold VIN Rising VIN Falling IS Input DC Bias Current Burst Mode® Operation Shutdown (Note 4) VOUT = 1.945V, ILOAD = 0A VRUN = 0V fOSC Oscillator Frequency VOUT = 1.875V VOUT = 0V RPFET RDS(ON) of P-Channel FET ISW = 100mA RNFET RDS(ON) of N-Channel FET ISW = –100mA ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V VRUN RUN Threshold ● IRUN RUN Leakage Current ● MIN TYP MAX 3.3 6 380 500 1.837 1.875 1.913 0.04 0.4 ● VIN = 3V, VOUT = 1.64V, Duty Cycle < 35% ● ● ● 2.5 1.8 0.3 µA mA 0.5 ● UNITS V %/V % 5.5 V 2 1.94 2.3 V V 26 0.1 35 1 µA µA 2.25 310 2.7 MHz kHz 0.75 0.9 Ω 0.55 0.7 Ω ±0.01 ±1 µA 1 1.5 V ±0.01 ±1 µA Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3410E-1.875 is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3410-1.875: TJ = TA + (PD)(250°C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 34101875f 2 LTC3410-1.875 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1) Efficiency vs Input Voltage Efficiency vs Output Current 100 100 IOUT = 100mA 90 90 IOUT = 250mA 70 EFFICIENCY (%) EFFFICIENCY (%) 80 IOUT = 10mA 80 IOUT = 1mA 60 IOUT = 0.1mA 50 70 60 50 40 30 VIN = 2.7V VIN = 3.6V VIN = 4.2V 20 40 10 30 2.5 4.5 4 3.5 INPUT VOLTAGE (V) 3 5 0 0.1 5.5 1 10 100 OUTPUT CURRENT (mA) 34101875 G02 34101875 G01 Output Voltage vs Temperature Oscillator Frequency vs Temperature 1.911 2.7 VIN = 3.6V VIN = 3.6V 2.6 OSCILLATOR FREQUENCY (MHz) 1.899 OUTPUT VOLTAGE (V) 1000 1.887 1.875 1.863 1.851 2.5 2.4 2.3 2.2 2.1 2.0 1.9 1.839 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 1.8 –50 125 –25 0 25 50 75 TEMPERATURE (°C) 34101875 G03 125 34101875 G04 Oscillator Frequency vs Supply Voltage Output Voltage vs Load Current 2.7 1.900 2.6 1.895 1.890 2.5 OUTPUT VOLTAGE (V) OSCILLATOR FREQUENCY (MHz) 100 2.4 2.3 2.2 2.1 2.0 1.885 1.880 1.875 1.870 1.865 1.860 1.9 1.855 1.8 1.850 2 3 5 4 SUPPLY VOLTAGE (V) 6 34101875 G05 0 200 100 300 LOAD CURRENT (mA) 400 34101875 G06 34101875f 3 LTC3410-1.875 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1) RDS(ON) vs Input Voltage RDS(ON) vs Temperature 1.2 1.2 1.1 VIN = 4.2V 1.0 1.0 VIN = 2.7V MAIN SWITCH 0.8 0.8 RDS (ON) (Ω) RDS (ON) (Ω) 0.9 0.7 0.6 0.5 SYNCHRONOUS SWITCH 0.6 VIN = 4.2V 0.4 0.4 VIN = 2.7V 0.3 0.2 0.2 1 5 4 3 INPUT VOLTAGE (V) 2 6 VIN = 3.6V MAIN SWITCH SYNCHRONOUS SWITCH 0.1 0 VIN = 3.6V 0 –50 –30 –10 10 30 50 70 90 110 130 TEMPERATURE (°C) 7 34101875 G07 34101875 G08 Dynamic Supply Current vs Temperature Dynamic Supply Current vs VIN 50 50 DYNAMIC SUPPLY CURRENT (µA) DYNAMIC SUPPLY CURRENT (µA) ILOAD = 0A 40 30 20 10 0 2 1 4 3 5 40 30 20 10 0 –50 –25 6 VIN (V) 50 25 0 75 TEMPERATURE (°C) 34101875 G09 Switch Leakage vs Input Voltage Switch Leakage vs Temperature 600 VIN = 5.5V RUN = 0V 550 90 500 80 450 70 LEAKAGE CURRENT (pA) SWITCH LEAKAGE (nA) 125 34101875 G10 110 100 100 SYNCHRONOUS SWITCH 60 50 40 30 MAIN SWITCH 20 350 MAIN SWITCH 300 250 200 150 SYNCHRONOUS SWITCH 100 50 10 0 –50 400 0 –25 50 25 0 75 TEMPERATURE (°C) 100 125 34101875 G11 0 1 4 3 2 INPUT VOLTAGE (V) 5 6 34101875 G12 34101875f 4 LTC3410-1.875 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1) Burst Mode Operation Start-Up from Shutdown SW 5V/DIV RUN 2V/DIV VOUT 50mV/DIV AC COUPLED VOUT 1V/DIV IL 100mA/DIV IL 200mA/DIV VIN = 3.6V ILOAD = 10mA 2µs/DIV VIN = 3.6V ILOAD = 300mA 34101875 G13 Start-Up from Shutdown 200µs/DIV 34101875 G14 Load Step VOUT RUN 2V/DIV VOUT 1V/DIV 100mV/DIV AC-COUPLED IL 200mA/DIV IL 200mA/DIV ILOAD 200mA/DIV VIN = 3.6V ILOAD = 0A 200µs/DIV 10µs/DIV VIN = 3.6V ILOAD = 0mA TO 300mA 34101875 G15 34101875 G16 Load Step VOUT 100mV/DIV AC-COUPLED IL 200mA/DIV ILOAD 200mA/DIV 10µs/DIV VIN = 3.6V ILOAD = 15mA TO 300mA 34101875 G17 34101875f 5 LTC3410-1.875 U U U PI FU CTIO S RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1µA supply current. Do not leave RUN floating. GND (Pins 2, 5): Ground Pin. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2µF or greater ceramic capacitor. VOUT (Pin 6): Output Voltage Feedback. An internal resistive divider divides the output voltage down for comparison to the internal 0.8V reference voltage. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. W FU CTIO AL DIAGRA U U SLOPE COMP 0.65V OSC OSC 4 VIN FREQ SHIFT – 6 R1 322.5k + 0.8V VFB R2 240k 0.4V EN SLEEP – + – EA S Q R Q RUN 0.8V REF SHUTDOWN 5Ω + ICOMP BURST RS LATCH VIN 1 + – SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW + VOUT IRCMP 5 2 GND – 34101875 BD 34101875f 6 LTC3410-1.875 U OPERATIO (Refer to Functional Diagram) Main Control Loop Short-Circuit Protection The LTC3410-1.875 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VOUT pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from the internal resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. When the output is shorted to ground, the frequency of the oscillator is reduced to about 310kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 2.25MHz when VOUT rises above 0V. Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3410-1.875 uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. Burst Mode Operation The LTC3410-1.875 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 70mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 26µA. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier’s output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. 34101875f 7 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO The basic LTC3410-1.875 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. 4.7µH VIN 2.7V TO 5.5V CIN 4.7µF CER VIN VOUT 1.875V SW COUT 4.7µF CER LTC3410-1.875 RUN VOUT GND 34101875 F01 Figure 1. High Efficiency Step-Down Converter MANUFACTURER PART NUMBER Taiyo Yuden For most applications, the value of the inductor will fall in the range of 2.2µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 120mA (40% of 300mA). ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ ( f)(L) ⎝ VIN ⎠ Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3410-1.875 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3410-1.875 applications. Table 1. Representative Surface Mount Inductors Inductor Selection ∆IL = Inductor Core Selection (1) MAX DC VALUE CURRENT DCR HEIGHT CB2016T2R2M CB2012T2R2M LBC2016T3R3M 2.2µH 2.2µH 3.3µH 510mA 530mA 410mA 0.13Ω 1.6mm 0.33Ω 1.25mm 0.27Ω 1.6mm Panasonic ELT5KT4R7M 4.7µH 950mA 0.2Ω 1.2mm Sumida CDRH2D18/LD 4.7µH 630mA 0.086Ω 2mm Murata LQH32CN4R7M23 4.7µH 450mA Taiyo Yuden NR30102R2M NR30104R7M 2.2µH 4.7µH 1100mA 0.1Ω 1mm 750mA 0.19Ω 1mm FDK FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D 4.7µH 3.3µH 2.2µH 1100mA 0.11Ω 1mm 1200mA 0.1Ω 1mm 1300mA 0.08Ω 1mm 0.2Ω 2mm The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. 34101875f 8 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO CIN and COUT Selection Using Ceramic Input and Output Capacitors In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC34101.875’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. CIN required IRMS ≅ IOMAX [VOUT (VIN − VOUT )]1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ The recommended capacitance value to use is 4.7µF for both the input and output capacitors. where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. 34101875f 9 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3410-1.875 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 2. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. 1 VIN = 3.6V POWER LOSS (W) 0.1 0.01 0.001 0.0001 0.00001 0.1 10 100 1 LOAD CURRENT (mA) 1000 34101875 F02 Figure 2. Power Loss vs Load Current 34101875f 10 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO Thermal Considerations In most applications the LTC3410-1.875 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3410-1.875 is running at high ambient temperature with low supply voltage, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To prevent the LTC3410-1.875 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJAis the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3410-1.875 with an input voltage of 2.7V, a load current of 300mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the R DS(ON) of the P-channel switch at 70°C is approximately 1.05Ω and the RDS(ON) of the N-channel synchronous switch is approximately 0.75Ω. The series resistance looking into the SW pin is: RSW = 1.05Ω (0.69) + 0.75Ω (0.31) = 0.96Ω For the SC70 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.0864)(250) = 91.6°C which is well below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 86.4mW 34101875f 11 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3410-1.875. These items are also illustrated graphically in Figures 3 and 4. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 3. Keep the (–) plates of CIN and COUT as close as possible. Design Example As a design example, assume the LTC3410-1.875 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.3A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low 1 and high load currents is important. With this information we can calculate L using Equation (1), L= ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ ( f)(∆IL ) ⎝ VIN ⎠ Substituting VOUT = 1.875V, VIN = 4.2V, ∆IL = 100mA and f = 2.25MHz in Equation (3) gives: L= 1 . 875V ⎛ 1 . 875V ⎞ 1− = 4 .66µH 2 . 25MHz(100mA) ⎜⎝ 4 . 2V ⎟⎠ A 4.7µH inductor works well for this application. For best efficiency choose a 360mA or greater inductor with less than 0.3Ω series resistance. CIN will require an RMS current rating of at least 0.125A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.5Ω. In most cases, a ceramic capacitor will satisfy this requirement. Figure 5 shows the complete circuit along with its efficiency curve. VOUT RUN (3) VIN VIA TO VIN LTC3410-1.875 2 – VOUT GND 6 PIN 1 L1 COUT VOUT + 3 L1 SW VIN 5 LTC34101.875 4 SW CIN VIN COUT CIN 34101875 F03 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 3. LTC3410-1.875 Layout Diagram 34101875 F04 Figure 4. LTC3410-1.875 Suggested Layout 34101875f 12 LTC3410-1.875 U W U U APPLICATIO S I FOR ATIO VIN 2.7V TO 4.2V 4 † CIN 4.7µF CER VIN SW 3 4.7µH* COUT† 4.7µF CER LTC3410-1.875 1 RUN VOUT 6 VOUT 1.875V GND † TAIYO YUDEN JMK212BJ475 *MURATA LQH32CN4R7M23 2, 5 34101875 F05a Figure 5a 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 EFFICIENCY, VIN = 2.7V EFFICIENCY, VIN = 3.6V EFFICIENCY, VIN = 4.2V 1 10 LOAD (mA) 100 1000 34101875 F05b Figure 5b VOUT 100mV/DIV AC COUPLED IL 200mA/DIV ILOAD 200mA/DIV 20µs/DIV VIN = 3.6V ILOAD = 100mA TO 300mA 34101875 F05C Figure 5c 34101875f 13 LTC3410-1.875 U TYPICAL APPLICATIO Using Low Profile Components, <1mm Height VIN 2.7V TO 4.2V 4 CIN† 4.7µF VIN SW 3 4.7µH* COUT† 4.7µF CER LTC3410-1.875 1 RUN VOUT 6 VOUT 1.875V GND 2, 5 † TAIYO YUDEN JMK212BJ475 *FDK MIPF2520D 34101875 TA03 Low Profile Efficiency 100 VIN = 2.7V VIN = 3.6V VIN = 4.2V EFFICIENCY (%) 90 80 70 60 50 0.1 1 10 LOAD (mA) 100 1000 34101875 TA04 Load Step VOUT 100mV/DIV AC COUPLED IL 200mA/DIV ILOAD 200mA/DIV 20µs/DIV VIN = 3.6V ILOAD = 100mA TO 300mA 34101875 TA05 34101875f 14 LTC3410-1.875 U PACKAGE DESCRIPTIO SC6 Package 6-Lead Plastic SC70 (Reference LTC DWG # 05-08-1638) 0.47 MAX 0.65 REF 1.80 – 2.20 (NOTE 4) 1.00 REF INDEX AREA (NOTE 6) 1.80 – 2.40 1.15 – 1.35 (NOTE 4) 2.8 BSC 1.8 REF PIN 1 RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.10 – 0.40 0.65 BSC 0.15 – 0.30 6 PLCS (NOTE 3) 0.80 – 1.00 0.00 – 0.10 REF 1.00 MAX GAUGE PLANE 0.15 BSC 0.26 – 0.46 0.10 – 0.18 (NOTE 3) SC6 SC70 1205 REV B NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. DETAILS OF THE PIN 1 INDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE INDEX AREA 7. EIAJ PACKAGE REFERENCE IS EIAJ SC-70 8. JEDEC PACKAGE REFERENCE IS MO-203 VARIATION AB 34101875f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3410-1.875 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1616 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1µA, ThinSOT Package LT1676 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LTC1701/LTC1701B 750mA (IOUT), 1MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 2.5V to 5V, VOUT = 1.25V, IQ = 135µA, ISD = 1µA, ThinSOT Package LT1776 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 40V, VOUT = 1.24V, IQ = 3.2mA, ISD = 30µA, N8, S8 Packages LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 15µA, ISD = <1µA, TSSOP-16 Package LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable, IQ = 20µA, ISD = <1µA, DFN Package LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3406/LTC3406B 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3409 600mA (IOUT), 1.5MHz/2.25MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 1.6V to 5.5V, VOUT = 0.613V, IQ = 65µA, DD8 Package LTC3410/LTC3410B 300mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 26µA, SC70 Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, MS Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, TSSOP-16E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25µA, ISD = <1µA, MS Package 34101875f 16 Linear Technology Corporation LT 0306 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005