19-1321; Rev 1; 2/98 KIT ATION EVALU LE B A IL A AV Miniature, Low-Voltage, Precision Step-Down Controller ____________________________Features ♦ ±2% DC Accuracy Using synchronous rectification, the MAX1637 achieves up to 95% efficiency. Efficiency is greater than 80% over a 1000:1 load-current range, which extends battery life in system-suspend or standby mode. Excellent dynamic response corrects output load transients caused by the latest dynamic-clock CPUs within five 300kHz clock cycles. Powerful 1A on-board gate drivers ensure fast external N-channel MOSFET switching. ♦ 1µA Total Shutdown Current The MAX1637 features a logic-controlled and synchronizable, fixed-frequency, pulse-width-modulation (PWM) operating mode. This reduces noise and RF interference in sensitive mobile-communications and pen-entry applications. Asserting the SKIP pin enables fixed-frequency mode, for lowest noise under all load conditions. For a stand-alone device that includes a +5V VL linear regulator and low-dropout capabilities, refer to the MAX1636 data sheet. ________________________Applications Notebook Computers Subnotebook Computers ♦ 0.1% (typ) DC Load Regulation ♦ Adjustable Switching Frequency to 350kHz ♦ Idle Mode™ Pulse-Skipping Operation ♦ 1.10V to 5.5V Adjustable Output Voltage ♦ 3.15V Minimum IC Supply Voltage (at VCC pin) ♦ Internal Digital Soft-Start ♦ 1.1V ±2% Reference Output ♦ Output Overvoltage Crowbar Protection ♦ Output Undervoltage Shutdown (foldback) ♦ Tiny 16-Pin QSOP Package ______________Ordering Information PART MAX1637EEE TEMP. RANGE PIN-PACKAGE -40°C to +85°C 16 QSOP __________Typical Operating Circuit VBIAS VBATT VGG VCC SHDN Handy-Terminals, PDAs MAX1637 __________________Pin Configuration DH BST LX TOP VIEW CSH 1 16 SKIP CSL 2 15 LX CC FB 3 14 DH REF 13 BST SKIP CSH 12 PGND SYNC CSL SHDN 6 11 DL GND FB SYNC 7 10 VGG CC 4 MAX1637 REF 5 9 GND 8 OUTPUT DL PGND VCC QSOP Idle Mode is a trademark of Maxim Integrated Products. ________________________________________________________________ Maxim Integrated Products 1 For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800. For small orders, phone 408-737-7600 ext. 3468. MAX1637 General Description The MAX1637 synchronous, buck, switch-mode powersupply controller generates the CPU supply voltage in battery-powered systems. The MAX1637 is a strippeddown version of the MAX1636 in a smaller 16-pin QSOP package. The MAX1637 is intended to be powered separately from the battery by an external bias supply (typically the +5V system supply) in applications where the battery exceeds 5.5V. The MAX1637 achieves excellent DC and AC output voltage accuracy. This device can operate from a low input voltage (3.15V) and delivers the excellent load-transient response needed by upcoming generations of dynamic-clock CPUs. MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller ABSOLUTE MAXIMUM RATINGS GND to PGND .............................................................+2V to -2V LX, BST to GND......................................................-0.3V to +36V BST, DH to LX...........................................................-0.3V to +6V VCC, VGG, CSL, CSH, SHDN to GND.......................-0.3V to +6V DL to GND..................................................-0.3V to (VGG + 0.3V) REF, SKIP, SYNC, CC to GND ...................-0.3V to (VCC + 0.3V) REF Output Current.............................................................20mA REF Short-Circuit to GND ..............................................Indefinite Operating Temperature Range ...........................-40°C to +85°C Continuous Power Dissipation (TA = +70°C) QSOP (derate 8.3mW/°C above +70°C) ......................667mW Storage Temperature Range .............................-65°C to +160°C Junction Temperature ......................................................+150°C Lead Temperature (soldering, 10sec) .............................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (Circuit of Figure 1, VCC = VGG = 5V, SYNC = VCC, IREF = 0mA, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX UNITS 5.5 V 1.120 V SMPS CONTROLLER Input Voltage Range VCC, VGG 3.15 Output Voltage FB tied to VOUT, 0mV < (CSH - CSL) < 80mV, includes line and load regulation 1.080 VCC = 5V VREF 5.5 VCC = 3.3V VREF 3.6 Output Adjustment Range Current-Limit Threshold 1.100 V CSH > CSL 80 100 120 CSH < CSL -145 -100 -55 1.5 1 2.5 1.75 mW 0.5 3 µA VCC = VGG = 5V VCC = VGG = 3.3V Power Consumption Output not switching Shutdown Supply Current SHDN = GND, VCC = VGG FB Input Current VFB = VREF Soft-Start Ramp Time SHDN to full current limit, four levels Idle-Mode Switchover Threshold CSH - CSL AC Load Regulation CSH - CSL = 0mV to CSH - CSL = 100mV -50 50 512 20 30 mV nA clocks 40 2 mV % INTERNAL REFERENCE VCC Undervoltage Lockout Threshold Rising edge, hysteresis = 15mV 2.80 VGG Undervoltage Lockout Threshold Rising edge, hysteresis = 15mV 2.80 REF Output Voltage REF load = 0µA 1.080 REF Load Regulation REF Line Regulation 1.100 3.05 V 3.05 V 1.120 V REF load = 0µA to 50µA 10 mV VCC = 3.15V to 5.5V 3 mV 330 230 kHz OSCILLATOR Oscillator Frequency Maximum Duty Factor SYNC = VCC SYNC = GND SYNC = VCC SYNC = GND 270 170 89 93 300 200 92 96 % SYNC Input Pulse Width High 200 ns SYNC Input Pulse Width Low 200 ns SYNC Input Rise/Fall Time SYNC Input Frequency Range 2 (Note 1) 240 _______________________________________________________________________________________ 200 ns 340 kHz Miniature, Low-Voltage, Precision Step-Down Controller (Circuit of Figure 1, VCC = VGG = 5V, SYNC = VCC, IREF = 0mA, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX 7 10 UNITS OVERVOLTAGE PROTECTION Overvoltage Trip Threshold FB, with respect to regulation point Overvoltage Fault Propagation Delay FB to DL delay, 22mV overdrive, CGATE = 2000pF Output Undervoltage Lockout Threshold % of nominal output Output Undervoltage Lockout Delay 4 1.25 60 From shutdown or power-on-reset state 70 % µs 80 6144 % clocks INPUTS AND OUTPUTS Logic Input Voltage High SHDN, SKIP, SYNC Logic Input Voltage Low SHDN, SKIP, SYNC Logic Input Bias Current Pin at GND or VCC Current-Sense Input Leakage Current CSH = CSL = 5V, VCC = VGG = GND, either CSH or CSL input Gate Driver Sink/Source Current DH or DL forced to 2V Gate Driver On-Resistance High or low, DH or DL 2.4 V -1 0.8 V 1 µA 10 µA 7 Ω 1 A ELECTRICAL CHARACTERISTICS (Circuit of Figure 1, VCC = VGG = 5V, SYNC = VCC, IREF = 0mA, TA = -40°C to +85°C, unless otherwise noted.) (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS SMPS CONTROLLER Input Voltage Range VCC, VGG 3.15 5.5 V Output Voltage FB tied to VOUT, 0mV < (CSH - CSL) < 80mV, includes line and load regulation 1.080 1.120 V VCC = 5V VREF 5.5 VCC = 3.3V VREF 3.6 CSH > CSL 70 130 mV VCC = VGG = 5V, output not switching 2.5 mW VCC = VGG = 3.3V, output not switching 1.75 mW Output Adjustment Range Current-Limit Threshold Power Consumption V INTERNAL REFERENCE VCC Undervoltage Lockout Threshold Rising edge, hysteresis = 15mV 2.80 3.05 V VGG Undervoltage Lockout Threshold Rising edge, hysteresis = 15mV 2.80 3.05 V SYNC = VCC 262 338 SYNC = GND 170 230 OSCILLATOR Oscillator Frequency kHz SYNC Input Pulse Width High 200 ns SYNC Input Pulse Width Low 200 ns SYNC Input Rise/Fall Time SYNC Input Frequency Range 240 200 ns 340 kHz _______________________________________________________________________________________ 3 MAX1637 ELECTRICAL CHARACTERISTICS (continued) ELECTRICAL CHARACTERISTICS (continued) (Circuit of Figure 1, VCC = VGG = 5V, SYNC = VCC, IREF = 0mA, TA = -40°C to +85°C, unless otherwise noted.) (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS 4.0 10 % 60 80 % OVERVOLTAGE PROTECTION Overvoltage Trip Threshold FB, with respect to regulation point Output Undervoltage Lockout Threshold % of nominal output INPUTS AND OUTPUTS Logic Input Voltage High SHDN, SKIP, SYNC Logic Input Voltage Low SHDN, SKIP, SYNC 2.4 V 0.8 V Note 1: Guaranteed by design, not production tested. Note 2: Specifications from -40°C to 0°C are guaranteed by design and not production tested. __________________________________________Typical Operating Characteristics (VOUT = 3.3V, TA = +25°C, unless otherwise noted.) 90 EFFICIENCY (%) 80 VBATT = 15V VBATT = 22V 100 SKIP = LOW 90 VBATT = 15V 70 VBATT = 7V 60 50 40 MAX1637-03 VBATT = 22V 80 VBATT = 7V 70 SKIP = LOW 90 EFFICIENCY (%) SKIP = LOW EFFICIENCY (%) 100 MAX1637-01 100 EFFICIENCY vs. LOAD CURRENT (2.5V/2A CIRCUIT) EFFICIENCY vs. LOAD CURRENT (2.5V/3A CIRCUIT) MAX1637-02 EFFICIENCY vs. LOAD CURRENT (1.7V/7A CIRCUIT) VBATT = 7V 80 VBATT = 22V VBATT = 15V 70 30 60 60 20 10 50 0.1 1 10 0.1 1 0.1 1 EFFICIENCY vs. LOAD CURRENT (3.3V/3A CIRCUIT) SUPPLY CURRENT vs. SUPPLY VOLTAGE SUPPLY CURRENT vs. LOAD CURRENT VBATT = 30V 60 15 SKIP = HIGH 10 SKIP = LOW 5 0.1 1 LOAD CURRENT (A) 10 20 SKIP = LOW 15 SYNC = HIGH 10 SYNC = LOW 5 0 0 50 10 MAX1637-07 MAX1637-06 ILOAD = 1A VOUT = 3.3V VCC + VGG SUPPLY CURRENT (mA) VBATT = 5V 20 VCC + VGG SUPPLY CURRENT (mA) MAX1637-04 VBATT = 15V 0.01 0.01 10 LOAD CURRENT (A) 90 70 0.01 LOAD CURRENT (A) SKIP = LOW 80 0.001 LOAD CURRENT (A) 100 4 50 0 0.01 EFFICIENCY (%) MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller 3.0 3.5 4.0 4.5 5.0 SUPPLY VOLTAGE (V) 5.5 6.0 0 1 2 3 4 5 6 LOAD CURRENT (A) _______________________________________________________________________________________ 7 8 9 Miniature, Low-Voltage, Precision Step-Down Controller REF LOAD-REGULATION ERROR vs. REF LOAD CURRENT 4 2 0 -2 -4 -6 900 0.5 800 DROPOUT VOLTAGE (mV) 6 0.6 MAX1637-09 LOAD REGULATION ∆VOUT (mV) 8 REF LOAD REGULATION ∆V (mV) MAX1637-08 10 DROPOUT VOLTAGE vs. LOAD CURRENT 0.4 0.3 0.2 MAX1637-10 LOAD REGULATION vs. LOAD CURRENT VOUT FORCED TO 3.27V SYNC = VCC 700 MAX1637 ____________________________________Typical Operating Characteristics (continued) (VOUT = 3.3V, TA = +25°C, unless otherwise noted.) 600 500 400 300 200 0.1 100 -8 0 -10 0.01 0.1 1 0 0 10 10 20 30 40 50 60 70 80 90 100 0.01 0.1 REF LOAD CURRENT (µA) LOAD CURRENT (A) 1 10 LOAD CURRENT (A) LOAD-TRANSIENT RESPONSE (3.3V/3A, PWM MODE) LOAD-TRANSIENT RESPONSE (1.8V, PWM MODE) MAX1637 TOC12 MAX1637 TOC11 VOUT 50mV/div VOUT 50mV/div 4A LOAD 2A CURRENT 10A 0A 5A LOAD CURRENT 0A 100µs/div 100µs/div SWITCHING WAVEFORMS (PFM MODE) SWITCHING WAVEFORMS (PWM MODE) SWITCHING WAVEFORMS DROPOUT OPERATION MAX1637-15 MAX1637-14 MAX1637-13 SYNC = VCC VOUT = 1.7V VOUT 20mV/div VOUT 50mV/div 5V 5V VLX 0V 0V 1A 0A 1A INDUCTOR CURRENT 0A VOUT 10mV/div VLX 2V/div VLX 1A INDUCTOR CURRENT 0A INDUCTOR CURRENT VOUT FORCED TO 3.27V 1µs/div 20µs/div 1µs/div _______________________________________________________________________________________ 5 MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller ____________________________________Typical Operating Characteristics (continued) (VOUT = 3.3V, TA = +25°C, unless otherwise noted.) OVERVOLTAGE-PROTECTION WAVEFORMS (VIN SHORTED TO VOUT THROUGH A 0.5Ω RESISTOR) TIME EXITING SHUTDOWN (VOUT = 3.3V, ILOAD = 7A) MAX1637-16 MAX1637-17 VOUT 100mV/div VOUT 1V/div 5V VDL 0V 0A -5A VSHDN 5V/div INDUCTOR CURRENT -10A 500µs/div 10µs/div ______________________________________________________________Pin Description 6 PIN NAME FUNCTION 1 CSH High-Side Current-Sense Input 2 CSL Low-Side Current-Sense Input 3 FB Feedback Input. Connect to center of resistor divider. 4 CC Compensation Pin. Connect a small capacitor to GND to set the integration time constant. 5 REF 1.100V Reference Output. Capable of sourcing 50µA for external loads. Bypass with 0.22µF minimum. 6 SHDN Shutdown Control Input. Turns off entire IC. When low, reduces supply current below 0.5µA (typ). Drive with logic input or connect to RC network between GND and VCC for automatic start-up. 7 SYNC Oscillator Frequency Select and Synchronization Input. Tie to VCC for 300kHz operation; tie to GND for 200kHz operation. 8 GND Analog Ground 9 VCC Main Analog Supply-Voltage Input to the Chip. VCC powers the PWM controller, logic, and reference. Input range is 3.15V to 5.5V. Bypass to GND with a 0.1µF capacitor close to the pin. 10 VGG Gate-Drive and Boost-Circuit Power Supply. Can be driven from a supply other than VCC. If the same supply is used by both VCC and VGG, isolate VCC from VGG with a 20Ω resistor. Bypass to PGND with a 4.7µF capacitor. VGG current = (QG1 + QG2) x f, where QG is the MOSFET gate charge at VGS = VGG. 11 DL 12 PGND Low-Side Gate-Driver Output 13 BST Boost Capacitor Connection 14 DH High-Side Gate-Driver Output 15 LX Inductor Connection 16 SKIP Power Ground Low-Noise Mode Control. Forces fixed-frequency PWM operation when high. _______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller MAX1637 VBIAS +5V NOMINAL VBATT 20Ω VCC 4.7µF SYNC VGG 0.1µF C1 MAX1637 470pF CC CMPSH-3 BST DH Q1 0.1µF REF L1 R1 OUTPUT LX 1µF SKIP DL GND PGND Q2 C2 * CSH CSL 1M** R2 ON/OFF SHDN FB 0.01µF** R3 *SEE RECTIFIER CLAMP DIODE SECTION **OPTIONAL RC NETWORK FOR POWER-ON-RESET Figure 1. Standard Application Circuit ______Standard Application Circuit _______________Detailed Description The basic MAX1637 buck converter (Figure 1) is easily adapted to meet a wide range of applications where a 5V or lower supply is available. The components listed in Table 1 represent a good set of trade-offs among cost, size, and efficiency, while staying within the worstcase specification limits for stress-related parameters such as capacitor ripple current. Do not change the circuit’s switching frequency without first recalculating component values (particularly inductance value at maximum battery voltage). The power Schottky diode across the synchronous rectifier is optional because the MOSFETs chosen incorporate a high-speed silicon diode. However, installing the Schottky will generally improve efficiency by about 1%. If used, the Schottky diode DC current must be rated to at least one-third of the maximum load current. The MAX1637 is a BiCMOS, switch-mode power-supply (SMPS) controller designed primarily for buck-topology regulators in battery-powered applications where high efficiency and low quiescent supply current are critical. Light-load efficiency is enhanced by automatic idlemode operation—a variable-frequency, pulse-skipping mode that reduces transition and gate-charge losses. The step-down, power-switching circuit consists of two N-channel MOSFETs, a rectifier, and an LC output filter. Output voltage for this device is the average AC voltage at the switching node, which is regulated by changing the duty cycle of the MOSFET switches. The gate-drive signal to the high-side N-channel MOSFET, which must exceed the battery voltage, is provided by a flying-capacitor boost circuit that uses a 100nF capacitor between BST and LX. Figure 2 shows the major circuit blocks. _______________________________________________________________________________________ 7 MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller Table 1. Component Selection for Standard Applications LOAD CURRENT LOAD CURRENT COMPONENT 2A 3A (EV KIT) 7A (EV KIT) 3A Input Voltage Range 7V to 22V 7V to 22V 7V to 22V 4.75V to 30V Output Voltage Range 2.5V 2.5V 1.7V 3.3V Application Chipset Supply Chipset Supply CPU Core General Purpose Frequency 300kHz 300kHz 300kHz 300kHz Q1 High-Side MOSFET 1/2 Si4902DY or 1/2 MMDF3NO3HD International Rectifier IRF7403 or Siliconix Si4412 Fairchild FDS9412 or International Rectifier IRF7403 International Rectifier IRF7403 or Siliconix Si4412 Q2 Low-Side MOSFET 1/2 Si4902DY or 1/2 MMDF3NO3HD International Rectifier IRF7413 or Siliconix Si4410DY Fairchild FDS6680 or Siliconix Si4420DY International Rectifier IRF7413 or Siliconix Si4410DY C1 Input Capacitor 10µF, 25V ceramic Tokin C34Y5U1E106Z or Marcon/United Chemicon THCR40E1E106ZT 10µF, 25V ceramic Tokin C34Y5U1E106Z or Marcon/United Chemicon THCR40E1E106ZT 4 x 10µF, 25V ceramic Tokin C34Y5U1E106Z or Marcon/United Chemicon THCR40E1E106ZT 10µF, 30V Sanyo OS-CON C2 Output Capacitor 220µF, 6.3V tantalum Sprague 595D227X96R3C2 470µF, 6.3V tantalum Kemet T510X477(1)006AS or 470µF, 4V tantalum Sprague 594D477X0004R2T 3 x 470µF, 6.3V tantalum Kemet T510X477(1)006AS or 470µF, 4V tantalum Sprague 594D477X0004R2T 470µF, 6.3V tantalum Kemet T510X477(1)006AS or 470µF, 4V tantalum Sprague 594D477X0004R2T R1 Resistor 0.033Ω, 1% (2010) Dale WSL-2010-R033F 0.020Ω, 1% (2010) Dale WSL-2010-R020F 0.010Ω, 1% (2512) Dale WSL-2512-R010F 0.020Ω, 1% (2010) Dale WSL-2010-R020F L1 Inductor 10µH Coilcraft DO3316P-103 or Coiltronics UP2-100 10µH Sumida CDRH125-100 2.2µH Panasonic P1F2R0HL or Coiltronics UP4-2R2 or Coilcraft DO5022P-222HC 10µH Sumida CDRH125-100 Table 2. Component Suppliers COMPANY 8 FACTORY FAX (COUNTRY CODE) USA PHONE COMPANY FACTORY FAX (COUNTRY CODE) USA PHONE AVX (1) 803-626-3123 (803) 946-0690 (1) 847-696-9278 (847) 696-2000 Central Semiconductor Marcon/United Chemi-Con (1) 516-435-1824 (516) 435-1110 Coilcraft (1) 847-639-1469 (847) 639-6400 Matsuo Motorola (1) 714-960-6492 (1) 602-994-6430 (714) 969-2491 (602) 303-5454 Coiltronics (1) 561-241-9339 (561) 241-7876 Panasonic (1) 714-373-7183 (714) 373-7939 Sanyo (81) 7-2070-1174 (619) 661-6835 Dale (1) 605-665-1627 (605) 668-4131 Siliconix (1) 408-970-3950 (408) 988-8000 Fairchild (1) 408-721-1635 (408) 721-2181 Sprague (1) 603-224-1430 (603) 224-1961 (310) 322-3331 Sumida (81) 3-3607-5144 (847) 956-0666 TDK (1) 847-390-4428 (847) 390-4373 Tokin (1) 408-434-0375 (408) 432-8020 International Rectifier (IR) (1) 310-322-3332 IRC (1) 512-992-3377 (512) 992-7900 _______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller MAX1637 VBATT VBIAS 3.15V TO 5.5V + VCC SKIP SYNC MAX1637 IC POWER VGG BST 200kHz TO 300kHz OSC DH LX PWM LOGIC DL PGND REF SHDN VOUT + 1.1V REF. SHUTDOWN CONTROL OFF + - REF UNDERVOLTAGE FAULT SLOPE COMPENSATION - CC REF + VREF -30% gm - + + OVERVOLTAGE FAULT ERROR INTEGRATOR CSH CSL VREF +7% 60kHz LP FILTER FB GND Figure 2. Functional Diagram The pulse-width-modulation (PWM) controller consists of a multi-input PWM comparator, high-side and lowside gate drivers, and logic. It uses a 200kHz/300kHz synchronizable oscillator. The MAX1637 contains faultprotection circuits that monitor the PWM output for undervoltage and overvoltage. It includes a 1.100V pre- cision reference. The circuit blocks are powered from an internal IC power rail that receives power from VCC. VGG provides direct power to the synchronous-switch gate driver, but provides indirect power to the highside-switch gate driver via an external diode-capacitor boost circuit. _______________________________________________________________________________________ 9 MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller CSH 1X CSL FB 2X REF REF gm CC BST R LEVEL SHIFT Q S DH LX OSC SLOPE COMPENSATION 30mV SKIP CURRENT LIMIT DAC SHOOTTHROUGH CONTROL CK SHDN COUNTER SOFT-START SYNCHRONOUS RECTIFIER CONTROL R -100mV VGG Q LEVEL SHIFT S DL PGND Figure 3. PWM Controller Functional Diagram PWM Controller Block The heart of the current-mode PWM controller is a multi-input, open-loop comparator that sums four signals: the output voltage error signal with respect to the reference voltage, the current-sense signal, the integrated voltage-feedback signal, and the slope10 compensation ramp (Figure 3). The PWM controller is a direct-summing type, lacking a traditional error amplifier and the phase shift associated with it. This direct-summing configuration approaches ideal cycle-by-cycle control over the output voltage. ______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller Fixed-Frequency Mode When SKIP is high, the controller always operates in fixed-frequency PWM mode for lowest noise. Each pulse from the oscillator sets the main PWM latch that turns on the high-side switch for a period determined by the duty factor (approximately VOUT / VIN). As the high-side switch turns off, the synchronous rectifier latch is set; 60ns later, the low-side switch turns on. The low-side switch stays on until the beginning of the next clock cycle. SKIP LOAD CURRENT MODE DESCRIPTION Low Light Idle Pulse-skipping, discontinuous inductor current Low Heavy PWM Constant frequency PWM, continuous inductor current High Light PWM Constant frequency PWM, continuous inductor current High Heavy PWM Constant frequency PWM, continuous inductor current The current-mode feedback system regulates the peak inductor-current value as a function of the output voltage error signal. In continuous-conduction mode, the average inductor current is nearly the same as the peak current, so the circuit acts as a switch-mode transconductance amplifier. This pushes the second output LC filter pole, normally found in a duty-factorcontrolled (voltage-mode) PWM, to a higher frequency. To preserve inner-loop stability and eliminate regenerative inductor-current “staircasing,” a slope-compensation ramp is summed into the main PWM comparator to make the apparent duty factor less than 50%. In PWM mode, the controller operates as a fixed-frequency, current-mode controller in which the duty factor is set by the input/output voltage ratio. PWM mode (SKIP = high) forces two changes on the PWM controller. First, it disables the minimum-current comparator, ensuring fixed-frequency operation. Second, it changes the detection threshold for reverse-current limit from 0mV to -100mV, allowing the inductor current to reverse at light loads. This results in fixed-frequency operation and continuous inductor-current flow. PWM mode eliminates discontinuous-mode inductor ringing and improves cross-regulation of transformer-coupled, multiple-output supplies. The relative gains of the voltage-sense and currentsense inputs are weighted by the values of the current sources that bias four differential input stages in the main PWM comparator (Figure 4). The voltage sense into the PWM has been conditioned by an integrated component of the feedback voltage, yielding excellent DC output voltage accuracy. See the Output Voltage Accuracy section for details. VCC R1 R2 TO PWM LOGIC UNCOMPENSATED HIGH-SPEED LEVEL TRANSLATOR AND BUFFER FB CC I1 I2 I3 I4 OUTPUT DRIVER VBIAS REF CSH CSL SLOPE COMPENSATION Figure 4. Main PWM Comparator Functional Diagram ______________________________________________________________________________________ 11 MAX1637 Table 3. SKIP PWM Table Idle Mode When SKIP is low, idle-mode circuitry automatically optimizes efficiency throughout the load-current range. Idle mode dramatically improves light-load efficiency by reducing the effective frequency, subsequently reducing switching losses. It forces the peak inductor current to ramp to 30% of the full current limit, delivering extra energy to the output and allowing subsequent cycles to be skipped. Idle mode transitions seamlessly to fixed-frequency PWM operation as load current increases (Table 3). MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller REF, VCC, and VGG Supplies The 1.100V reference (REF) is accurate to ±2% over temperature, making REF useful as a precision system reference. Bypass REF to GND with a 0.22µF (min) capacitor. REF can supply up to 50µA for external loads. Loading REF reduces the main output voltage slightly because of the reference load-regulation error. The MAX1637 has two independent supply pins, VCC and VGG. VCC powers the sensitive analog circuitry of the SMPS, while VGG powers the high-current MOSFET drivers. No protection diodes or sequencing requirements exist between the two supplies. Isolate VGG from VCC with a 20Ω resistor if they are powered from the same supply. Bypass VCC to GND with a 0.1µF capacitor located directly adjacent to the pin. Use only smallsignal diodes for the boost circuit (10mA to 100mA Schottky or 1N4148 diodes are preferred), and bypass VGG to PGND with a 4.7µF capacitor directly at the package pins. The VCC and VGG input range is 3.15V to 5.5V. High-Side Boost Gate Drive (BST) Gate-drive voltage for the high-side N-channel switch is generated by a flying-capacitor boost circuit (Figure 2). The capacitor between BST and LX is alternately charged from the VGG supply and placed parallel to the high-side MOSFET’s gate-source terminals. On start-up, the synchronous rectifier (low-side MOSFET) forces LX to 0V and charges the boost capacitor to VGG. On the second half-cycle, the SMPS turns on the high-side MOSFET by closing an internal switch between BST and DH. This provides the necessary enhancement voltage to turn on the high-side switch, an action that boosts the gate-drive signal above the battery voltage. Ringing at the high-side MOSFET gate (DH) in discontinuous-conduction mode (light loads) is a natural operating condition. It is caused by residual energy in the tank circuit, formed by the inductor and stray capacitance at the switching node, LX. The gate-drive negative rail is referred to LX, so any ringing there is directly coupled to the gate-drive output. If the circuit is operating in continuous-conduction mode, the DL drive waveform is simply the complement of the DH high-side-drive waveform (with controlled dead time to prevent cross-conduction or “shootthrough”). In discontinuous (light-load) mode, the synchronous switch is turned off as the inductor current falls through zero. Shutdown Mode and Power-On Reset SHDN is a logic input with a threshold of about 1.5V that, when held low, places the IC in its 0.5µA shutdown mode. The MAX1637 has no power-on-reset circuitry, and the state of the device is not known on initial power-up. In applications that use logic to drive SHDN, it may be necessary to toggle SHDN to initialize the part once VCC is stable. In applications that require automatic start-up, drive SHDN through an external RC network (Figure 5). The network will hold SHDN low until VCC stabilizes. Typical values for R and C are 1MΩ and 0.01µF. For slow-rising VCC, use a larger capacitor. When cycling VCC, VCC must stay low long enough to discharge the 0.01µF capacitor, otherwise the circuit may not start. A diode may be added in parallel with the resistor to speed up the discharge. Current-Limiting and CurrentSense Inputs (CSH and CSL) The current-limit circuit resets the main PWM latch and turns off the high-side MOSFET switch whenever the voltage difference between CSH and CSL exceeds 100mV. This limiting is effective for both current flow directions, putting the threshold limit at ±100mV. The tolerance on the positive current limit is ±20%, so the external low-value sense resistor (R1) must be sized for 80mV / IPEAK, where IPEAK is the peak inductor current required to support the full load current. Components must be designed to withstand continuous current stresses of 120mV / R1. VIN R VGG VCC Synchronous-Rectifier Driver (DL) Synchronous rectification reduces conduction losses in the rectifier by shunting the normal Schottky catch diode with a low-resistance MOSFET switch. Also, the synchronous rectifier ensures proper start-up of the boost gate-driver circuit. If the synchronous power MOSFET is omitted for cost or other reasons, replace it with a small-signal MOSFET, such as a 2N7002. SHDN C MAX1637 R = 1MΩ C = 0.01µF Figure 5. Power-On Reset RC Network for Automatic Start-Up 12 ______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller Oscillator Frequency and Synchronization (SYNC) The SYNC input controls the oscillator frequency as follows: low selects 200kHz, high selects 300kHz. SYNC can also be used to synchronize with an external 5V CMOS or TTL clock generator. It has a guaranteed 240kHz to 340kHz capture range. A high-to-low transition on SYNC initiates a new cycle. Operation at 300kHz optimizes the application circuit for component size and cost. Operation at 200kHz increases efficiency, reduces dropout, and improves load-transient response at low input-output voltage differences (see the Low-Voltage Operation section). Output Voltage Accuracy (CC) Output voltage error is guaranteed to be within ±2% over all conditions of line, load, and temperature. The MAX1637’s DC load regulation is typically better than 0.1%, due to its integrator amplifier. The device optimizes transient response by providing a feedback signal with a direct path from the output to the main summing PWM comparator. The integrated feedback signal from the CC transconductance amplifier is also 50 Figure 6 shows the output voltage response to a 0A to 3A load transient with and without the integrator. With the integrator, the output voltage returns to within 0.1% of its no-load value with only a small AC excursion. Without the integrator, load regulation is degraded (Figure 6b). Asymmetrical clamping at the integrator output prevents worsening of load transients during pulse-skipping mode. Output Undervoltage Lockout The output undervoltage-lockout circuit protects against heavy overloads and short-circuits at the main SMPS output. This scheme employs a timer rather than a foldback current limit. The SMPS has an undervoltage-protection circuit, which is activated 6144 clock cycles after the SMPS is enabled. If the SMPS output is under 70% of the nominal value, it is latched off and does not restart until SHDN is toggled. Applications that use the recommended RC power-on-reset circuit will also clear the fault condition when VCC falls below 0.5V (typical). Note that undervoltage protection can INTEGRATOR ACTIVE CC = 470pF VOUT = 3.3V VOUT (mV) IOUT (A) summed into the PWM comparator, with the gain weighted so that the signal has only enough gain to correct the DC inaccuracies. The integrator’s response time is determined by the time constant set by the capacitor placed on the CC pin. The time constant should neither be so fast that the integrator responds to the normal VOUT ripple, nor too slow to negate the integrator’s effect. A 470pF to 1500pF CC capacitor is sufficient for 200kHz to 300kHz frequencies. 50 INTEGRATOR DEACTIVATED CC = REF VOUT = 3.3V VOUT (mV) -50 -50 4 4 IOUT (A) 2 0 2 0 (100µs/div) Figure 6a. Load-Transient Response with Integrator Active (100µs/div) Figure 6b. Load-Transient Response with Integrator Deactivated ______________________________________________________________________________________ 13 MAX1637 For prototyping or for very high-current applications, it may be useful to wire the current-sense inputs with a twisted pair rather than PC traces (two pieces of wrapped wire twisted together are sufficient). This reduces the noise picked up at CSH and CSL, which can cause unstable switching and reduced output current. MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller make prototype troubleshooting difficult since only 20ms or 30ms elapse before the SMPS is latched off. The overvoltage crowbar protection is disabled in output undervoltage mode. The exact time of the output rise depends on output capacitance and load current, but it is typically 1ms with a 300kHz oscillator. Output Overvoltage Protection The output voltage is set via a resistor divider connected to FB (Figure 1). Calculate the output voltage with the following formula: The overvoltage crowbar-protection circuit is intended to blow a fuse in series with the battery if the main SMPS output rises significantly higher than its standard level (Table 4). In normal operation, the output is compared to the internal precision reference voltage. If the output goes 7% above nominal, the synchronous-rectifier MOSFET turns on 100% (the high-side MOSFET is simultaneously forced off) in order to draw massive amounts of battery current to blow the fuse. This safety feature does not protect the system against a failure of the controller IC itself, but is intended primarily to guard against a short across the high-side MOSFET. A crowbar event is latched and can only be reset by a rising edge on SHDN (or by removal of the VCC supply voltage). The overvoltage-detection decision is made relative to the regulation point. Internal Digital Soft-Start Circuit Soft-start allows a gradual increase of the internal current-limit level at start-up to reduce input surge currents. The SMPS contains an internal digital soft-start circuit controlled by a counter, a digital-to-analog converter (DAC), and a current-limit comparator. In shutdown, the soft-start counter is reset to zero. When the SMPS is enabled, its counter starts counting oscillator pulses, and the DAC begins incrementing the comparison voltage applied to the current-limit comparator. The DAC output increases from 0mV to 100mV in five equal steps as the count increases to 512 clocks. As a result, the main output capacitor charges up relatively slowly. Setting the Output Voltage VOUT = VREF (1 + R2 / R3) where VREF = 1.1V nominal. Recommended normal values for R3 range from 5kΩ to 100kΩ. To achieve a 1.1V nominal output, connect FB directly to CSL. Remote output voltage sensing is possible by using the top of the external resistor divider as the remote sense point. __________________Design Procedure The standard application circuit (Figure 1) contains a ready-to-use solution for common application needs. Use the following design procedure to optimize the basic schematic for different voltage or current requirements. But before beginning a design, firmly establish the following: • Maximum input (battery) voltage, V IN(MAX) . This value should include the worst-case conditions, such as no-load operation when a battery charger or AC adapter is connected but no battery is installed. VIN(MAX) must not exceed 30V. • Minimum input (battery) voltage, VIN(MIN). This value should be taken at full load under the lowest battery conditions. If the minimum input-output difference is less than 1.5V, the filter capacitance required to maintain good AC load regulation increases (see Low-Voltage Operation section). Table 4. Operating Modes 14 MODE SHDN CONDITIONS VOUT in regulation STATUS NOTES Run High All circuit blocks active Normal operation Shutdown Low — All circuit blocks off Lowest current consumption Overvoltage (Crowbar) High VOUT greater than 7% above regulation point REF = off, DL = high Rising edge on SHDN exits crowbar Output Undervoltage Lockout High VOUT below 70% of nominal after 20ms to 30ms timeout expires REF = off, DL = low Rising edge on SHDN exits UVLO ______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller Three key inductor parameters must be specified: inductance value (L), peak current (IPEAK), and DC resistance (RDC). The following equation includes a constant, LIR, which is the ratio of inductor peak-topeak AC current to DC load current. A higher LIR value allows lower inductance, but results in higher losses and ripple. A good compromise is a 30% ripple-current to load-current ratio (LIR = 0.3), which corresponds to a peak inductor current 1.15 times higher than the DC load current. L = VOUT(VIN(MAX) - VOUT) / (VIN(MIN) x ƒ x IOUT x LIR) where ƒ = switching frequency (normally 200kHz or 300kHz), and IOUT = maximum DC load current. The peak current can be calculated as follows: IPEAK = ILOAD + [VOUT(VIN(MAX) - VOUT) / (2 x ƒ x L x VIN(MAX))] The inductor’s DC resistance should be low enough that RDC x IPEAK < 100mV, as it is a key parameter for efficiency performance. If a standard, off-the-shelf inductor is not available, choose a core with an LI2 rating greater than L x IPEAK2 and wind it with the largest diameter wire that fits the winding area. For 300kHz applications, ferrite-core material is strongly preferred; for 200kHz applications, Kool-Mu® (aluminum alloy) or even powdered iron is acceptable. If light-load efficiency is unimportant (in desktop PC applications, for example), then low-permeability iron-powder cores can be acceptable, even at 300kHz. For high-current applications, shielded-core geometries (such as toroidal or pot core) help keep noise, EMI, and switchingwaveform jitter low. Current-Sense Resistor Value The current-sense resistor value is calculated according to the worst-case, low-current limit threshold voltage (from the Electrical Characteristics) and the peak inductor current: RSENSE = 80mV / IPEAK Use IPEAK from the second equation in the Inductor Value section. Use the calculated value of RSENSE to size the MOSFET switches and specify inductor saturation-current ratings according to the worst-case highcurrent-limit threshold voltage: IPEAK = 120mV / RSENSE Low-inductance resistors, such as surface-mount metal film, are recommended. Input Capacitor Value Connect low-ESR bulk capacitors directly to the drain on the high-side MOSFET. The bulk input filter capacitor is usually selected according to input ripple current requirements and voltage rating, rather than capacitor value. Electrolytic capacitors with low enough equivalent series resistance (ESR) to meet the ripple-current requirement invariably have sufficient capacitance values. Aluminum electrolytic capacitors, such as Sanyo OS-CON or Nichicon PL, are superior to tantalum types, which risk power-up surge-current failure, especially when connecting to robust AC adapters or lowimpedance batteries. RMS input ripple current (IRMS) is determined by the input voltage and load current, with the worst case occurring at VIN = 2 x VOUT. Therefore, when VIN is 2 x VOUT: IRMS = ILOAD / 2 VCC and VGG should be isolated from each other with a 20Ω resistor and bypassed to ground independently. Place a 0.1µF capacitor between VCC and GND, as close to the supply pin as possible. A 4.7µF capacitor is recommended between VGG and PGND. Output Filter Capacitor Value The output filter capacitor values are generally determined by the ESR and voltage-rating requirements, rather than by actual capacitance requirements for loop stability. In other words, the low-ESR electrolytic capacitor that meets the ESR requirement usually has more output capacitance than is required for AC stability. Kool-Mu is a trademark of Magnetics, Inc. ______________________________________________________________________________________ 15 MAX1637 Inductor Value The exact inductor value is not critical and can be freely adjusted to allow trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but reduce efficiency due to higher peakcurrent levels. The smallest inductor value is obtained by lowering the inductance until the circuit operates at the border between continuous and discontinuous mode. Further reducing the inductor value below this crossover point results in discontinuous-conduction operation, even at full load. This helps lower output filter capacitance requirements, but efficiency suffers under these conditions, due to high I2R losses. On the other hand, higher inductor values produce greater efficiency, but also result in resistive losses due to extra wire turns—a consequence that eventually overshadows the benefits gained from lower peak current levels. High inductor values can also affect load-transient response (see the VSAG equation in the Low-Voltage Operation section). The equations in this section are for continuous-conduction operation. MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller Use only specialized low-ESR capacitors intended for switching-regulator applications, such as AVX TPS, Sprague 595D, Sanyo OS-CON, or Nichicon PL series. To ensure stability, the capacitor must meet both minimum capacitance and maximum ESR values as given in the following equations: COUT > VREF(1 + VOUT / VIN(MIN)) / VOUT x RSENSE x ƒ RESR < RSENSE x VOUT / VREF where RESR can be multiplied by 1.5, as discussed below. These equations are worst case, with 45 degrees of phase margin to ensure jitter-free, fixed-frequency operation, and provide a nicely damped output response for zero to full-load step changes. Some costconscious designers may wish to bend these rules with less-expensive capacitors, particularly if the load lacks large step changes. This practice is tolerable if some bench testing over temperature is done to verify acceptable noise and transient response. No well-defined boundary exists between stable and unstable operation. As phase margin is reduced, the first symptom is timing jitter, which shows up as blurred edges in the switching waveforms where the scope does not quite sync up. Technically speaking, this jitter (usually harmless) is unstable operation since the duty factor varies slightly. As capacitors with higher ESRs are used, the jitter becomes more pronounced, and the load-transient output voltage waveform starts looking ragged at the edges. Eventually, the load-transient waveform has enough ringing on it that the peak noise levels exceed the allowable output voltage tolerance. Note that even with zero phase margin and gross instability, the output voltage seldom declines beyond IPEAK x RESR (under constant loads). Designers of RF communicators or other noise-sensitive analog equipment should be conservative and stay within the guidelines. Designers of notebook computers and similar commercial-temperature-range digital systems can multiply the RESR value by a factor of 1.5 without affecting stability or transient response. The output voltage ripple, which is usually dominated by the filter capacitor’s ESR, can be approximated as IRIPPLE x RESR. There is also a capacitive term, so the full equation for ripple in continuous-conduction mode is VRIPPLE(p-p) = IRIPPLE x [RESR + 1 / (2πƒ x COUT)]. In idle mode, the inductor current becomes discontinuous, with high peaks and widely spaced pulses, so the noise can actually be higher at light load (compared to full load). In idle mode, calculate the output ripple as follows: VRIPPLE(p-p) = (0.02 x RESR / RSENSE) + [0.0003 x L x (1 / VOUT + 1 / (VIN - VOUT)) / RSENSE2 x CF ] 16 Selecting Other Components MOSFET Switches The high-current N-channel MOSFETs must be logiclevel types with guaranteed on-resistance specifications at VGS = 4.5V. Lower gate-threshold specifications are better (i.e., 2V max rather than 3V max). Drain-source breakdown voltage ratings must at least equal the maximum input voltage, preferably with a 20% margin. The best MOSFETs have the lowest on-resistance per nanocoulomb of gate charge. Multiplying RDS(ON) by Qg provides a good figure of merit for comparing various MOSFETs. Newer MOSFET process technologies with dense cell structures generally perform best. The internal gate drivers tolerate >100nC total gate charge, but 70nC is a more practical upper limit to maintain best switching times. In high-current applications, MOSFET package power dissipation often becomes a dominant design factor. I2R power losses are the greatest heat contributor for both high-side and low-side MOSFETs. I2R losses are distributed between Q1 and Q2 according to duty factor, as shown in the following equations. Generally, switching losses affect only the upper MOSFET since the Schottky rectifier usually clamps the switching node before the synchronous rectifier turns on. Gate-charge losses are dissipated by the driver and do not heat the MOSFET. Calculate the temperature rise according to package thermal-resistance specifications to ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature. The worstcase dissipation for the high-side MOSFET occurs at both extremes of input voltage, and the worst-case dissipation for the low-side MOSFET occurs at maximum input voltage. Duty = (VOUT + VQ2) / (VIN - VQ1) PD (UPPER FET) = ILOAD2 x RDS(ON) x duty + VIN x ILOAD x ƒ x [(VIN x CRSS) / IGATE + 20ns] PD (LOWER FET) = ILOAD2 x RDS(ON) x (1 - duty) where V Q = the on-state voltage drop (I LOAD x RDS(ON)), CRSS = the MOSFET reverse transfer capacitance, IGATE = the DH driver peak output current capability (1A typ), and the DH driver inherent rise/fall time is 20ns. The MAX1637’s output undervoltage shutdown function protects the synchronous rectifier under output short-circuit conditions. To reduce EMI, add a 0.1µF ceramic capacitor from the high-side switch drain to the low-side switch source. ______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller added capacitance can be supplied by a low-cost bulk capacitor in parallel with the normal low-ESR capacitor. __________Applications Information Heavy-Load Efficiency Considerations The major efficiency-loss mechanisms under loads are as follows, in the usual order of importance: • P(I2R) = I2R losses • P(tran) = transition losses • P(gate) = gate-charge losses • P(diode) = diode-conduction losses • P(cap) = capacitor ESR losses • P(IC) = losses due to the IC’s operating supply current Inductor core losses are fairly low at heavy loads because the inductor’s AC current component is small. Therefore, these losses are not considered in this analysis. Ferrite cores are preferred, especially at 300kHz, but powdered cores, such as Kool-Mu, can also work well. Efficiency = POUT / PIN x 100% = POUT / (POUT + PTOTAL) x 100% Boost-Supply Diode D2 A signal diode such as a 1N4148 works well in most applications. Do not use large power diodes, such as 1N5817 or 1N4001. Low-Voltage Operation Low input voltages and low input-output differential voltages each require extra care in their design. Low VIN-VOUT differentials can cause the output voltage to sag when the load current changes abruptly. The sag’s amplitude is a function of inductor value and maximum duty factor (DMAX, an Electrical Characteristics parameter, 93% guaranteed over temperature at f = 200kHz) as follows: VSAG = [(ISTEP)2 x L] / [2CF x (VIN(MIN) x DMAX VOUT)] Table 5 is a low-voltage troubleshooting guide. The cure for low-voltage sag is to increase the output capacitor’s value. For example, at VIN = 5.5V, VOUT = 5V, L = 10µH, ƒ = 200kHz, and ISTEP = 3A, a total capacitance of 660µF keeps the sag below 200mV. Note that only the capacitance requirement increases; the ESR requirements do not change. Therefore, the PTOTAL = P(I2R) + P(tran) + P(gate) + P(diode) + P(cap) + P(IC) 2R) = I 2 P = (I LOAD x (RDC + RDS(ON) +RSENSE) where RDC is the DC resistance of the coil, RDS(ON) is the MOSFET on-resistance, and RSENSE is the currentsense resistor value. The RDS(ON) term assumes identical MOSFETs for the high-side and low-side switches because they time-share the inductor current. If the MOSFETs are not identical, their losses can be estimated by averaging the losses according to duty factor. PD(tran) = transition loss = V IN x I LOAD x ƒ x [(VIN CRSS / IGATE ) + 20ns] where CRSS is the reverse transfer capacitance of the high-side MOSFET (a data sheet parameter), IGATE is the DH gate-driver peak output current (1.5A typ), and the rise/fall time of the DH driver is typically 20ns. Table 5. Low-Voltage Troubleshooting Guide SYMPTOM CONDITION ROOT CAUSE SOLUTION Sag or droop in VOUT under step-load change Low VIN-VOUT differential, under 1.5V Limited inductor-current slew rate per cycle Increase bulk output capacitance per formula (see Low-Voltage Operation section). Reduce inductor value. Dropout voltage is too high Low VIN-VOUT differential, under 1V Maximum duty-cycle limits exceeded Reduce operation to 200kHz. Reduce MOSFET on-resistance and coil DC resistance. ______________________________________________________________________________________ 17 MAX1637 Rectifier Clamp Diode The rectifier is a clamp across the low-side MOSFET that catches the negative inductor swing during the 60ns dead time between turning one MOSFET off and turning each low-side MOSFET on. The latest generations of MOSFETs incorporate a high-speed silicon body diode, which serves as an adequate clamp diode if efficiency is not of primary importance. A Schottky diode can be placed in parallel with the body diode to reduce the forward voltage drop, typically improving efficiency 1% to 2%. Use a diode with a DC current rating equal to one-third of the load current; for example, use an MBR0530 (500mA-rated) type for loads up to 1.5A, a 1N5819 type for loads up to 3A, or a 1N5822 type for loads up to 10A. The rectifier’s rated reversebreakdown voltage must be at least equal to the maximum input voltage, preferably with a 20% margin. MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller P(gate) = Qg x ƒ x VGG where Qg is the sum of the gate-charge values for lowside and high-side switches. For matched MOSFETs, Q g is twice the data-sheet value of an individual MOSFET. Efficiency can usually be optimized by connecting VGG to the most efficient 5V source, such as the system +5V supply. P(diode) = diode conduction losses = ILOAD x VFWD x tD x ƒ where tD is the diode conduction time (120ns typ), and VFWD is the diode forward voltage. This power is dissipated in the MOSFET body diode if no external Schottky diode is used. P(cap) = input capacitor ESR loss = IRMS2 x RESR where IRMS is the input ripple current as calculated in the Input Capacitor Value section. Light-Load Efficiency Considerations Under light loads, the PWM operates in discontinuous mode. The inductor current discharges to zero at some point during the charging cycle. This makes the inductor current’s AC component high compared to the load current, which increases core losses and I2R losses in the input-output filter capacitors. For best light-load efficiency, use MOSFETs with moderate gate-charge levels and use ferrite MPP or other low-loss core material. Avoid powdered-iron cores; even Kool-Mu (aluminum alloy) is not as desirable as ferrite. Low-Noise Operation Noise-sensitive applications such as hi-fidelity multimedia-equipped systems, cellular phones, RF communicating computers, and electromagnetic pen-entry systems should operate the controller in PWM mode (SKIP = high). This mode forces a constant switching frequency, reducing interference due to switching noise by concentrating the radiated EM fields at a known frequency outside the system audio or IF bands. Choose an oscillator frequency for which switchingfrequency harmonics do not overlap a sensitive frequency band. If necessary, synchronize the oscillator to a tight-tolerance external clock generator. Powering From a Single Low-Voltage Supply The circuit of Figure 7 is powered from a single 3.3V to 5.5V source and delivers 4A at 2.5V. At input voltages of 3.15V, this circuit typically achieves efficiencies of 90% at 3.5A load currents. When using a single supply to power both VBATT and VBIAS, be sure that it does not exceed the 5.5V rating (6V absolute maximum) for VGG 18 and VCC. Also, heavy current surges from the input may cause transient dips on VCC. To prevent this, the decoupling capacitor on V CC may need to be increased to 2µF or greater. This circuit uses lowthreshold (specified at VGS = 2.7V) IRF7401 MOSFETs which allow a typical startup of 3.15V at above 4A. Low input voltages demand the use of larger input capacitors. Sanyo OS-CONs are recommended for their high capacity and low ESR. PC Board Layout Considerations Good PC board layout is required to achieve specified noise, efficiency, and stable performance. The PC board layout artist must be given explicit instructions, preferably a pencil sketch showing the placement of power-switching components and high-current routing. See the PC board layout in the MAX1637 evaluation kit manual for examples. A ground plane is essential for optimum performance. In most applications, the circuit will be located on a multi-layer board, and full use of the four or more copper layers is recommended. Use the top layer for high-current connections, the bottom layer for quiet connections (REF, CC, GND), and the inner layers for an uninterrupted ground plane. Use the following step-by-step guide: 1) Place the high-power components (C1, C2, Q1, Q2, D1, L1, and R1) first, with their grounds adjacent. • Minimize current-sense resistor trace lengths and ensure accurate current sensing with Kelvin connections (Figure 8). • Minimize ground trace lengths in the high-current paths. • Minimize other trace lengths in the high-current paths. — Use >5mm-wide traces. — CIN to high-side MOSFET drain: 10mm max length — Rectifier diode cathode to low side — MOSFET: 5mm max length — LX node (MOSFETs, rectifier cathode, inductor): 15mm max length Ideally, surface-mount power components are butted up to one another with their ground terminals almost touching. These high-current grounds are then connected to each other with a wide, filled zone of top-layer copper so they do not go through vias. The resulting top-layer subground plane is connected to the normal inner-layer ground plane at the output ground terminals, which ensures that the IC’s analog ground is ______________________________________________________________________________________ Miniature, Low-Voltage, Precision Step-Down Controller MAX1637 VBIAS 3.15V TO 5.5V 4.7µF TANTALUM 20Ω 1µF VCC 220µF OS-CON VGG CMPSH-3 SYNC SKIP BST 0.1µF IRF7401 MAX1637 DH 10µH 20mΩ 1% OUTPUT = 2.5V AT 4A LX CDHR125-100 DL IRF7401 SHDN MBRS130 ON/OFF 130k 1% PGND 470µF LOW ESR TANTALUM CSH REF 1µF GND CSL CC FB 470pF 100k 1% Figure 7. 3.15V to 5.5V Single-Supply Application Circuit sensing at the supply’s output terminals without interference from IR drops and ground noise. Other high-current paths should also be minimized, but focusing primarily on short ground and current-sense connections eliminates about 90% of all PC board layout problems (see the PC board layouts in the MAX1637 evaluation kit manual for examples). 2) Place the IC and signal components. Keep the main switching nodes (LX nodes) away from sensitive analog components (current-sense traces and REF capacitor). Place the IC and analog components on the opposite side of the board from the powerswitching node. Important: The IC must be no further than 10mm from the current-sense resistors. Keep the gate-drive traces (DH, DL, and BST) shorter than 20mm and route them away from CSH, CSL, and REF. Place ceramic bypass capacitors close to the IC. The bulk capacitors can be placed further away. 3) Use a single-point star ground where the input ground trace, power ground (subground plane), and HIGH-CURRENT PATH SENSE RESISTOR MAX1637 Figure 8. Kelvin Connections for the Current-Sense Resistors normal ground plane meet at the supply's output ground terminal. Connect both IC ground pins and all IC bypass capacitors to the normal ground plane. ______________________________________________________________________________________ 19 ___________________Chip Information TRANSISTOR COUNT: 2164 ________________________________________________________Package Information QSOP.EPS MAX1637 Miniature, Low-Voltage, Precision Step-Down Controller Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 1998 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.