19-2622; Rev 2; 6/09 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators The MAX1951/MAX1952 high-efficiency, DC-to-DC step-down switching regulators deliver up to 2A of output current. The devices operate from an input voltage range of 2.6V to 5.5V and provide an output voltage from 0.8V to VIN, making the MAX1951/MAX1952 ideal for on-board postregulation applications. The MAX1951 total output error is less than 1% over load, line, and temperature. The MAX1951/MAX1952 operate at a fixed frequency of 1MHz with an efficiency of up to 94%. The high operating frequency minimizes the size of external components. Internal soft-start control circuitry reduces inrush current. Short-circuit and thermal-overload protection improve design reliability. The MAX1951 provides an adjustable output from 0.8V to VIN, whereas the MAX1952 has a preset output of 1.8V. Both devices are available in a space-saving 8-pin SO package. Features ♦ Compact 0.385in2 Circuit Footprint ♦ 10µF Ceramic Input and Output Capacitors, 2µH Inductor for 1.5A Output ♦ Efficiency Up to 94% ♦ 1% Output Accuracy Over Load, Line, and Temperature (MAX1951, Up to 1.5A) ♦ Guaranteed 2A Output Current ♦ Operate from 2.6V to 5.5V Supply ♦ Adjustable Output from 0.8V to VIN (MAX1951) ♦ Preset Output of 1.8V (1.5% Accuracy) (MAX1952) ♦ Internal Digital Soft-Soft ♦ Short-Circuit and Thermal-Overload Protection Ordering Information/ Selector Guide Applications ASIC/DSP/µP/FPGA Core and I/O Voltages Set-Top Boxes PART PINPACKAGE TEMP RANGE Cellular Base Stations Networking and Telecommunications OUTPUT MAX1951ESA+ -40°C to +85°C 8 SO Adj 0.8V to VIN MAX1952ESA+ -40°C to +85°C 8 SO Fixed 1.8V +Denotes a lead(Pb)-free/RoHS-compliant package. Pin Configuration Typical Operating Circuit OUTPUT 0.8V TO VIN, UP TO 2A INPUT 2.6V TO 5.5V TOP VIEW IN VCC 8 1 LX MAX1951 IN VCC REF 2 GND 3 MAX1951 MAX1952 FB 4 7 LX 6 PGND 5 COMP COMP PGND OFF FB REF GND ON SO OPTIONAL ________________________________________________________________ Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. MAX1951/MAX1952 General Description MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators ABSOLUTE MAXIMUM RATINGS IN, VCC to GND ........................................................-0.3V to +6V COMP, FB, REF to GND .............................-0.3V to (VCC + 0.3V) LX to Current (Note 1).........................................................±4.5A PGND to GND .............................................Internally Connected Continuous Power Dissipation (TA = +85°C) 8-Pin SO (derate 12.2mW/°C above +70°C)................976mW Package Junction-to-Ambient Thermal Resistance (θJA) (Note 2)...............................82°C/W Package Junction-to-Case Thermal Resistance (θJC) (Note 2) ..............................32°C/W Operating Temperature Range MAX195_ ESA..................................................-40°C to +85°C Junction Temperature Range ............................-40°C to +150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed the IC’s package power dissipation limits. Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX UNITS IN AND VCC IN Voltage Range 2.6 Supply Current Switching with no load, LX unconnected Shutdown Current COMP = GND VCC Undervoltage Lockout Threshold When LX starts/stops switching 5.5 V VIN = 5.5V 6 10 mA 1.0 2.5 mA VCC rising 0.5 2.35 VCC falling V 2 2.25 1.96 2 0.01 2.03 0.2 V % 0.01 0.4 % 12 22 Ω REF REF Voltage REF Load Regulation IREF = 0µA, VIN = 2.6V to 5.5V IREF = 0 to 40µA, VIN = 2.6V to 5.5V REF Line Regulation IREF = 20µA, VIN = 2.6V to 5.5V REF Shutdown Resistance From REF to GND, COMP = GND COMP MAX1951 40 60 80 MAX1952 26.7 40 53.3 COMP Transconductance From FB to COMP, VCOMP = 1.25V COMP Clamp Voltage, Low VIN = 2.6V to 5.5V, VFB = 1.3V 0.6 1 1.2 V COMP Clamp Voltage, High COMP Shutdown Resistance VIN = 2.6V to 5.5V, VFB = 1.1V From COMP to GND, VIN = 2V 1.97 2.15 15 2.28 30 V Ω COMP Shutdown Threshold When LX starts/stops switching 0.6 1 COMP Startup Current FB COMP rising COMP falling 0.17 0.4 COMP = GND 15 25 Output Voltage Range (MAX1951) When using external feedback resistors to drive FB 0.8 FB Regulation Voltage (Error Amp Only) VCOMP = 1V to 2V, IOUT = 0 to 1.5A VIN V µA V VIN = 2.6V to 5.5V MAX1951 0.787 0.795 0.803 VIN = 2.8V to 5.5V MAX1952 1.773 1.8 1.827 18 28 kΩ +0.1 µA FB Input Resistance MAX1952 13 FB Input Bias Current MAX1951 -0.1 2 40 µS _______________________________________________________________________________________ V 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators (VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX UNITS 266 mΩ 206 mΩ 0.24 0.35 Ω 3.1 4.5 LX LX On-Resistance, PMOS LX On-Resistance, NMOS VIN = 5V 116 VIN = 3.3V 140 VIN = 2.6V 163 VIN = 5V 93 VIN = 3.3V 106 VIN = 2.6V 116 LX Current-Sense Transimpedance From LX to COMP, VIN = 2.6V to 5.5V LX Current-Limit Threshold Duty cycle = 100%, VIN = 2.6V to 5.5V LX Leakage Current VIN = 5.5V LX Switching Frequency VIN = 2.6V to 5.5V 0.85 LX Maximum Duty Cycle VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V 100 LX Minimum Duty Cycle VCOMP = 1V, VIN = 2.6V to 5.5V 0.16 High side 2.4 Low side -0.6 VLX = 5.5V LX = GND 10 -10 1 1.1 A µA MHz % 15 % THERMAL CHARACTERISTICS Thermal-Shutdown Threshold When LX starts/stops switching TJ rising 160 TJ falling 145 °C ELECTRICAL CHARACTERISTICS (VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS IN AND VCC IN Voltage Range 2.6 Supply Current Switching with no load, VIN = 5.5V Shutdown Current COMP = GND VCC Undervoltage Lockout Threshold When LX starts/stops switching VCC rising VCC falling 5.5 V 10 mA 1 mA 2.5 1.95 V REF REF Voltage IREF = 0µA, VIN = 2.6V to 5.5V 2.03 V REF Load Regulation IREF = 0 to 40µA, VIN = 2.6V to 5.5V 1.95 0.2 % REF Line Regulation IREF = 20µA, VIN = 2.6V to 5.5V 0.4 % REF Shutdown Resistance From REF to GND, COMP = GND 22 Ω _______________________________________________________________________________________ 3 MAX1951/MAX1952 ELECTRICAL CHARACTERISTICS (continued) MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators ELECTRICAL CHARACTERISTICS (continued) (VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS COMP MAX1951 40 80 MAX1952 26.7 53.3 COMP Transconductance From FB to COMP, VCOMP = 1.25V COMP Clamp Voltage, Low VIN = 2.6V to 5.5V, VFB = 1.3V 0.6 1.2 V COMP Clamp Voltage, High VIN = 2.6V to 5.5V, VFB = 1.1V 1.97 2.28 V COMP Shutdown Resistance From COMP to GND, VIN = 2V 30 Ω COMP rising 1.2 µS COMP Shutdown Threshold When LX starts/stops switching COMP Startup Current COMP = GND 14 40 µA Output Voltage Range (MAX1951) When using external feedback resistors to drive FB 0.8 VIN V FB Regulation Voltage (Error Amp Only) VCOMP = 1V to 2V, VIN = 2.6V to 5.5V MAX1951 0.783 0.807 MAX1952 1.764 1.836 FB Input Resistance From FB to GND COMP falling 0.17 V FB V MAX1952 10 30 kΩ MAX1951 -0.1 +0.1 µA LX On-Resistance, PMOS 266 mΩ LX On-Resistance, NMOS LX Current Sense 206 mΩ From LX to COMP, VIN = 2.6V to 5.5V 0.16 0.35 Ω LX Current-Limit Threshold Duty cycle = 100%, VIN = 2.6V to 5.5V, high side 2.4 4.5 A FB Input Bias Current LX (Note 4) VLX = 5.5V 10 LX Leakage Current VIN = 5.5V LX Switching Frequency VIN = 2.6V to 5.5V 0.8 LX Maximum Duty Cycle VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V 100 LX = GND -10 Note 3: Specifications to -40°C are guaranteed by design and not production tested. Note 4: The LX output is designed to provide 2.4A RMS current. 4 _______________________________________________________________________________________ 1.1 µA MHz % 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators 70 VOUT = 1.5V 50 40 30 20 1.994 VOUT = 1.8V 70 60 VOUT = 1.5V 50 40 30 VOUT = 0.8V 1.995 VOUT = 0.8V 20 10 MAX1951 toc03 80 EFFICIENCY (%) VOUT = 2.5V 60 TA = +85°C 1.993 TA = +25°C 1.992 1.991 TA = -40°C 1.990 10 0 0 10 100 10,000 1000 1.989 10 100 LOAD CURRENT (mA) 10,000 1000 1.05 1.00 TA = +25°C 0.95 TA = -40°C 0.85 0.80 2.6 3.1 3.6 10 4.1 4.6 5.1 VOUT = 2.5V 2 1 0 -1 -2 -3 25 30 35 40 VOUT = 0.8V VOUT = 3.3V VOUT = 1.8V -4 -5 -6 5.6 0 0.4 0.8 1.6 1.2 LOAD CURRENT (A) LOAD-TRANSIENT RESPONSE LOAD-TRANSIENT RESPONSE MAX1951 toc06 MAX1951 toc07 OUTPUT VOLTAGE: 100mV/div, AC-COUPLED OUTPUT VOLTAGE: 100mV/div, AC-COUPLED OUTPUT CURRENT: 0.5A/div OUTPUT CURRENT: 0.5A/div VIN = 5V VOUT = 2.5V IOUT = 0.5 TO 1A 40μs/div 20 6 5 4 3 INPUT VOLTAGE (V) 0A 15 REF OUTPUT CURRENT (μA) MAX1951 toc05 TA = +85°C OUTPUT VOLTAGE DEVIATION (mV) MAX1951 toc04 1.15 0.90 5 OUTPUT VOLTAGE DEVIATION vs. LOAD CURRENT 1.20 1.10 0 LOAD CURRENT (mA) SWITCHING FREQUENCY vs. INPUT VOLTAGE SWITCHING FREQUENCY (MHz) EFFICIENCY (%) 80 VOUT = 2.5V 90 REF VOLTAGE vs. REF OUTPUT CURRENT REF VOLTAGE (V) VOUT = 3.3V 90 100 MAX 1951 toc01 100 EFFICIENCY vs. LOAD CURRENT (VCC = VIN = 3.3V) MAX 1951 toc02 EFFICIENCY vs. LOAD CURRENT (VCC = VIN = 5V) VIN = 3.3V VOUT = 1.5V IOUT = 0.5 TO 1A 0A 40μs/div _______________________________________________________________________________________ 5 MAX1951/MAX1952 Typical Operating Characteristics (Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25°C, unless otherwise noted. See Figure 2.) Typical Operating Characteristics (continued) (Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25°C, unless otherwise noted. See Figure 2.) SWITCHING WAVEFORMS SOFT-START WAVEFORMS MAX1951 toc08 MAX1951 toc09 INDUCTOR CURRENT 1A/div VCOMP 2V/div 0A VLX 5V/div 0V OUTPUT VOLTAGE 1V/div OUTPUT VOLTAGE 10mV/div, AC-COUPLED VIN = 3.3V VOUT = 1.8V ILOAD = 1.5A VIN = VCC = 3.3V VOUT = 2.5V ILOAD = 1.5A 200ns/div 1ms/div SOFT-START WAVEFORMS SHUTDOWN WAVEFORMS MAX1951 toc10 MAX1951 toc11 0V VCOMP 2V/div 0V VLX 5V/div VCOMP 2V/div OUTPUT VOLTAGE 0.5V/div VIN = VCC = 3.3V VOUT = 2.5V ILOAD = 1.5A VIN = VCC = 3.3V VOUT = 0.8V 0V 1ms/div 20μs/div SHUTDOWN CURRENT vs. INPUT VOLTAGE MAX1951 toc12 1.0 0.9 SHUTDOWN CURRENT (mA) MAX1951/MAX1952 1MHz, All Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) 6 _______________________________________________________________________________________ OUTPUT VOLTAGE 1V/div 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators PIN NAME FUNCTION 1 VCC Supply Voltage. Bypass VCC with 0.1µF capacitor to ground and 10Ω resistor to IN. 2 REF Reference Bypass. Bypass REF with 0.1µF capacitor to ground. 3 GND Ground 4 FB Feedback Input. Connect FB to the output to regulate using the internal feedback resistor string (MAX1952). Connect an external resistordivider from the output to FB and GND to set the output to a voltage between 0.8V and VIN (MAX1951). 5 Regulator Compensation. Connect series RC network from COMP to GND. Pull COMP below COMP 0.17V to shut down the regulator. COMP = GND when VIN is less than 2.25V (see the Compensation and Shutdown Mode section) 6 Power Ground. Internally connected to GND. PGND Keep power ground and signal ground planes separate. 7 8 LX Inductor Connection. Connect an inductor between LX and the regulator output. IN Power-Supply Voltage. Input voltage range from 2.6V to 5.5V. Bypass IN with a 10µF (min) ceramic capacitor to GND and a 10Ω resistor to VCC. Detailed Description The MAX1951/MAX1952 high-efficiency switching regulators are small, simple, DC-to-DC step-down converters capable of delivering up to 2A of output current. The devices operate in pulse-width modulation (PWM) at a fixed frequency of 1MHz from a 2.6V to 5.5V input voltage and provide an output voltage from 0.8V to VIN, making the MAX1951/MAX1952 ideal for on-board postregulation applications. The high switching frequency allows for the use of smaller external components, and internal synchronous rectifiers improve efficiency and eliminate the typical Schottky free-wheeling diode. Using the onresistance of the internal high-side MOSFET to sense switching currents eliminates current-sense resistors, further improving efficiency and cost. The MAX1951 total output error over load, line, and temperature (0°C to +85°C) is less than 1%. Controller Block Function The MAX1951/MAX1952 step-down converters use a PWM current-mode control scheme. An open-loop comparator compares the integrated voltage-feedback signal against the sum of the amplified current-sense signal and the slope compensation ramp. At each rising edge of the internal clock, the internal high-side MOSFET turns on until the PWM comparator trips. During this on-time, current ramps up through the inductor, sourcing current to the output and storing energy in the inductor. The currentmode feedback system regulates the peak inductor current as a function of the output voltage error signal. Since the average inductor current is nearly the same as the peak inductor current (<30% ripple current), the circuit acts as a switch-mode transconductance amplifier. To preserve inner-loop stability and eliminate inductor staircasing, a slope-compensation ramp is summed into the main PWM comparator. During the second half of the cycle, the internal high-side p-channel MOSFET turns off, and the internal low-side n-channel MOSFET turns on. The inductor releases the stored energy as its current ramps down while still providing current to the output. The output capacitor stores charge when the inductor current exceeds the load current, and discharges when the inductor current is lower, smoothing the voltage across the load. Under overload conditions, when the inductor current exceeds the current limit (see the Current Limit section), the high-side MOSFET does not turn on at the rising edge of the clock and the low-side MOSFET remains on to let the inductor current ramp down. Current Sense An internal current-sense amplifier produces a current signal proportional to the voltage generated by the high-side MOSFET on-resistance and the inductor current (RDS(ON) x ILX). The amplified current-sense signal and the internal slope compensation signal are summed together into the comparator’s inverting input. The PWM comparator turns off the internal high-side MOSFET when this sum exceeds the output from the voltage-error amplifier. Current Limit The internal high-side MOSFET has a current limit of 3.1A (typ). If the current flowing out of LX exceeds this limit, the high-side MOSFET turns off and the synchronous rectifier turns on. This lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. A synchronous rectifier current limit of -0.6A (typ) protects the device from current flowing into LX. If the negative current limit is exceeded, the synchronous rectifier turns off, forcing the inductor current to flow _______________________________________________________________________________________ 7 MAX1951/MAX1952 Pin Description MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators POSITIVE AND NEGATIVE CURRENT LIMITS VCC OSC IN CLOCK CURRENT SENSE PWM CONTROL LX RAMP GEN SLOPE COMP CLAMP ERROR SIGNAL THERMAL SHUTDOWN gm COMP SOFT-START/ UVLO PGND FB DAC REF 2V REF BANDGAP REF 1.25V MAX1951 GND Figure 1. Functional Diagram through the high-side MOSFET body diode, back to the input, until the beginning of the next cycle or until the inductor current drops to zero. The MAX1951/MAX1952 utilize a pulse-skip mode to prevent overheating during short-circuit output conditions. The device enters pulseskip mode when the FB voltage drops below 300mV, limiting the current to 3A (typ) and reducing power dissipation. Normal operation resumes upon removal of the short-circuit condition. VCC Decoupling Due to the high switching frequency and tight output tolerance (1%), decouple VCC with a 0.1µF capacitor connected from VCC to GND, and a 10Ω resistor connected from VCC to IN. Place the capacitor as close to VCC as possible. Soft-Start The MAX1951/MAX1952 employ digital soft-start circuitry to reduce supply inrush current during startup conditions. When the device exits undervoltage lockout (UVLO), shutdown mode, or restarts following a thermal-overload event, or the external pulldown on COMP is released, the digital soft-start circuitry slowly ramps up the voltages at REF and FB (see the Soft-Start Waveforms in the Typical Operating Characteristics). 8 Undervoltage Lockout If VCC drops below 2.25V, the UVLO circuit inhibits switching. Once V CC rises above 2.35V, the UVLO clears, and the soft-start sequence activates. Compensation and Shutdown Mode The output of the internal transconductance voltage error amplifier connects to COMP. The normal operation voltage for COMP is 1V to 2.2V. To shut down the MAX1951/MAX1952, use an NPN bipolar junction transistor or a very low output capacitance open-drain MOSFET to pull COMP to GND. Shutdown mode causes the internal MOSFETs to stop switching, forces LX to a high-impedance state, and shorts REF to GND. Release COMP to exit shutdown and initiate the softstart sequence. Thermal-Overload Protection Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds TJ = +160°C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 15°C, resulting in a pulsed output during continuous overload conditions. Following a thermal-shutdown condition, the soft-start sequence begins. _______________________________________________________________________________________ 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators Output Voltage Selection: Adjustable (MAX1951) or Preset (MAX1952) The MAX1951 provides an adjustable output voltage between 0.8V and VIN. Connect FB to output for 0.8V output. To set the output voltage of the MAX1951 to a voltage greater than VFB (0.8V typ), connect the output to FB and GND using a resistive divider, as shown in Figure 2a. Choose R2 between 2kΩ and 20kΩ, and set R3 according to the following equation: R3 = R2 x [(VOUT/VFB) – 1] The MAX1951 PWM circuitry is capable of a stable minimum duty cycle of 18%. This limits the minimum output voltage that can be generated to 0.18 ✕ VIN. Instability may result for VIN/VOUT ratios below 0.18. The MAX1952 provides a preset output voltage. Connect the output to FB, as shown in Figure 2b. Output Inductor Design Use a 2µH inductor with a minimum 2A-rated DC current for most applications. For best efficiency, use an inductor with a DC resistance of less than 20mΩ and a saturation current greater than 3A (min). See Table 2 for recommended inductors and manufacturers. For most designs, derive a reasonable inductor value (LINIT) from the following equation: LINIT = VOUT x (VIN - VOUT)/(VIN x LIR x IOUT(MAX) x fSW) where fSW is the switching frequency (1MHz typ) of the oscillator. Keep the inductor current ripple percentage LIR between 20% and 40% of the maximum load current for the best compromise of cost, size, and performance. Calculate the maximum inductor current as: IL(MAX) = (1 + LIR/2) x IOUT(MAX) Check the final values of the inductor with the output ripple voltage requirement. The output ripple voltage is given by: VRIPPLE = VOUT x (VIN - VOUT) x ESR / (VIN x LFINAL x fSW) where ESR is the equivalent series resistance of the output capacitors. Input Capacitor Design The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit’s switching. The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = (1/ VIN ) × (IOUT2 × VOUT × (VIN − VOUT )) For duty ratios less than 0.5, the input capacitor RMS current is higher than the calculated current. Therefore, use a +20% margin when calculating the RMS current at lower duty cycles. Use ceramic capacitors for their low ESR, equivalent series inductance (ESL), and lower cost. Choose a capacitor that exhibits less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability. After determining the input capacitor, check the input ripple voltage due to capacitor discharge when the high-side MOSFET turns on. Calculate the input ripple voltage as follows: VIN_RIPPLE = (IOUT x VOUT)/(fSW x VIN x CIN) Keep the input ripple voltage less than 3% of the input voltage. Output Capacitor Design The key selection parameters for the output capacitor are capacitance, ESR, ESL, and the voltage rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-to-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor’s ESR, and the voltage drop due to the capacitor’s ESL. Calculate the output voltage ripple due to the output capacitance, ESR, and ESL as: VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL) where the output ripple due to output capacitance, ESR, and ESL is: VRIPPLE(C) = IP-P/(8 x COUT x fSW) VRIPPLE(ESR) = IP-P x ESR VRIPPLE(ESL) = (IP-P/tON) x ESL or (IP-P/tOFF) x ESL, whichever is greater and IP-P the peak-to-peak inductor current is: IP-P = [ (VIN - VOUT )/fSW x L) ] x VOUT/VIN Use these equations for initial capacitor selection, but determine final values by testing a prototype or evaluation circuit. As a rule, a smaller ripple current results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output voltage ripple decreases with larger inductance. Use ceramic capacitors for their low ESR and ESL at the switching frequency of the converter. The low ESL of ceramic capacitors makes ripple voltages negligible. Load transient response depends on the selected output capacitor. During a load transient, the output instantly changes by ESR x ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time (see the Load Transient Response graph in the _______________________________________________________________________________________ 9 MAX1951/MAX1952 Design Procedure MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on the closed-loop bandwidth. A higher bandwidth yields a faster response time, thus preventing the output from deviating further from its regulating value. Compensation Design The double pole formed by the inductor and output capacitor of most voltage-mode controllers introduces a large phase shift, that requires an elaborate compensation network to stabilize the control loop. The MAX1951/ MAX1952 utilize a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, eliminating the double pole caused by the inductor and output capacitor, and greatly simplifying the compensation network. A simple type 1 compensation with single compensation resistor (R1) and compensation capacitor (C2) creates a stable and highbandwidth loop. An internal transconductance error amplifier compensates the control loop. Connect a series resistor and capacitor between COMP (the output of the error amplifier) and GND to form a pole-zero pair. The external inductor, internal current-sensing circuitry, output capacitor, and the external compensation circuit determine the loop system stability. Choose the inductor and output capacitor based on performance, size, and cost. Additionally, select the compensation resistor and capacitor to optimize control-loop stability. The component values shown in the typical application circuit (Figure 2) yield stable operation over a broad range of input-to-output voltages. The basic regulator loop consists of a power modulator, an output feedback divider, and an error amplifier. The power modulator has DC gain set by gmc x RLOAD, with a pole-zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The following equations define the power modulator: Modulator gain: GMOD = ΔVOUT/ΔVCOMP = gmc x RLOAD Modulator pole frequency: fpMOD = 1 / (2 x π x COUT x (RLOAD+ESR)) Modulator zero frequency: fzESR = 1 /(2 x π x COUT x ESR) where, RLOAD = VOUT/IOUT(MAX), and gmc = 4.2S. The feedback divider has a gain of GFB = VFB / VOUT, where VFB is equal to 0.8V. The transconductance error amplifier has a DC gain, GEA(DC), of 70dB. The compensation capacitor, C2, and the output resistance of the error amplifier, R OEA (20MΩ), set the dominant 10 pole. C2 and R1 set a compensation zero. Calculate the dominant pole frequency as: fpEA = 1/(2πx CC x ROEA) Determine the compensation zero frequency is: fzEA = 1/(2π x CC x RC) For best stability and response performance, set the closed-loop unity-gain frequency much higher than the modulator pole frequency. In addition, set the closedloop crossover unity-gain frequency less than, or equal to, 1/5 of the switching frequency. However, set the maximum zero crossing frequency to less than 1/3 of the zero frequency set by the output capacitance and its ESR when using POSCAP, SPCAP, OSCON, or other electrolytic capacitors.The loop-gain equation at the unity-gain frequency is: GEA(fc) x GMOD(fc) x VFB/VOUT = 1 where GEA(fc) = gmEA x R1, and GMOD(fc) = gmc x RLOAD x fpMOD/fC, where gmEA = 60µS. R1 calculated as: R1 = VOUT x K/(gmEA x VFB x GMOD(fc)) where K is the correction factor due to the extra phase introduced by the current loop at high frequencies (>100kHz). K is related to the value of the output capacitance (see Table 1 for values of K vs. C). Set the error-amplifier compensation zero formed by R1 and C2 at the modulator pole frequency at maximum load. C2 is calculated as follows: C2 = (VOUT x COUT/(R1 x IOUT(MAX)) As the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly, resulting in a constant closed-loop unitygain frequency. Use the following numerical example to calculate R1 and C2 values of the typical application circuit of Figure 2a. Table 1. K Value DESCRIPTION COUT (µF) 10 22 K Values are for output inductance from 1.2µH 0.55 to 2.2µH. Do not use output inductors larger 0.47 than 2.2µH. Use fC = 200kHz to calculate R1. VOUT = 1.5V IOUT(MAX) = 1.5A COUT = 10µF RESR = 0.010Ω gmEA = 60µS ______________________________________________________________________________________ 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators RLOAD = VOUT/IOUT(MAX) = 1.5V/1.5 A = 1Ω fpMOD = [1/(2π x COUT x (RLOAD + RESR)] = [1/(2 x π ×10 ×10-6 x (1 + 0.01)] = 15.76kHz. fzESR = [1/(2π xCOUT RESR)] = [1/(2 x π × 10 ×10-6 × 0.01)] = 1.59MHz. For 2µH output inductor, pick the closed-loop unity-gain crossover frequency (f C ) at 200kHz. Determine the power modulator gain at fC: GMOD(fc) = gmc × RLOAD × fpMOD/fC = 4.2 × 1 × 15.76kHz/200kHz = 0.33 then: R1 = VO x K/(gmEA x VFB x GMOD(fc)) = (1.5 x 0.55)/ (60 ×10-6 × 0.8 × 0.33) ≈ 51.1kΩ (1%) C2 = (VOUT × COUT)/(R × IOUT(max) ) = (1.5 × 10 × 10-6)/(51.1k × 1.5) ≈ 196pF, choose 220pF, 10% Applications Information PCB Layout Considerations Careful PCB layout is critical to achieve clean and stable operation. The switching power stage requires particular attention. Follow these guidelines for good PCB layout: 1) Place decoupling capacitors as close to the IC as possible. Keep power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. 2) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current (C1 to IN and C1 to PGND) short. Avoid vias in the switching paths. 4) If possible, connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP). Thermal Considerations The MAX1951 uses a fused-lead 8-pin SO package with a RTHJC rating of 32°C/W. The MAX1951 EV kit layout is optimized for 1.5A. The typical application circuit shown in Figure 2c was tested with the existing MAX1951 EV kit layout at +85°C ambient temperature, and GND lead temperature was measured at +113°C for a typical device. The estimated junction temperature was +138°C. Thermal performance can be further improved with one of the following options: 1) Increase the copper areas connected to GND, LX, and IN. 2) Provide thermal vias next to GND and IN, to the ground plane and power plane on the back side of PCB, with openings in the solder mask next to the vias to provide better thermal conduction. 3) Provide forced-air cooling to further reduce case temperature. ______________________________________________________________________________________ 11 MAX1951/MAX1952 gmc = 4.2S fSWITCH = 1MHz MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators L1 2μH 2.6V TO 5.5V IN R4 10Ω C5 0.1μF MAX1951ESA REF COMP C1 10μF R3 14.7kΩ 1% FB VCC R1 51.1kΩ GND PGND R2 16.9kΩ 1% OFF C2 220pF ON 1.5V AT 1.5A LX C3 0.1μF Q1 R5 10kΩ C4 10μF GND OUTPUT COMPONENT VALUES VOLTAGE (V) R1 (kΩ) R2 (kΩ) R3 (kΩ) C2 (pF) 220 SHORT OPEN 0.8 33.2 220 14.7 16.9 1.5 51.1 220 30 14 2.5 82.5 220 75 24 3.3 110 OPTIONAL SHUTDOWN CONTROL Figure 2a. MAX1951 Adjustable Output Typical Application Circuit L1 2μH 2.6V TO 5.5V IN R4 10Ω C5 0.1μF 1.8V AT 1.5A LX MAX1952ESA-18 FB VCC R1 68kΩ COMP C1 10μF PGND REF C4 10μF GND OFF C2 220pF ON Q1 R5 10kΩ C3 0.1μF GND OPTIONAL SHUTDOWN CONTROL Figure 2b. MAX1952 Fixed-Output Typical Application Circuit 12 ______________________________________________________________________________________ 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators IN R4 10Ω C5 0.1μF MAX1951ESA GND REF PGND OFF C2 100pF ON Q1 R5 10kΩ R3 12.7kΩ 1% FB COMP C1 10μF 1.8V, 2A LX VCC R1 100kΩ MAX1951/MAX1952 L1 1.1μH 3.3V ±5% C3 0.1μF R2 10kΩ 1% C4 22μF GND L1: TOKO A915AY-1R1M C1: TAIYO YUDEN JMK316BJ106ML C4: TAIYO YUDEN JMK325BJ226MM OPTIONAL SHUTDOWN CONTROL Figure 2c. MAX1951 Typical Application Circuit with 2A Output ______________________________________________________________________________________ 13 MAX1951/MAX1952 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators Table 2. External Components List COMPONENT (FIGURE 2) FUNCTION DESCRIPTION L1 Output inductor C1 Input filtering capacitor 10µF ±20%, 6.3V X5R capacitor Taiyo Yuden JMK316BJ106ML or TDK C3216X5R0J106MT C2 Compensation capacitor 220pF ±10%, 50V capacitor Murata GRM39X7R221K050AD or Taiyo Yuden UMK107CH221KZ 2µH ±20% inductor Sumida CDRH4D28-1R8 or Toko A915AY-2R0M C3 Reference bypass capacitor 0.1µF ±20%, 16V X7R capacitor Taiyo Yuden EMK107BJ104MA, TDK C1608X7R1C104K, or Murata GRM 39X7R104K016AD C4 Output filtering capacitor 10µF ±20%, 6.3V X5R capacitor Taiyo Yuden JMK316BJ106ML or TDK C3216X5R0J106MT C5 VCC bypass capacitor 0.1µF ±20%, 16V X7R capacitor Taiyo Yuden EMK107BJ104MA, TDK C1608X7R1C104K, or Murata GRM 39X7R104K016AD R1 Loop compensation resistor Figure 2a R2 Feedback resistor Figure 2a R3 Feedback resistor Figure 2a R4 Bypass resistor 10Ω ±5% resistor R5 Shutdown transistor base current bias (optional) 10kΩ ±5% resistor Q1 Shutdown transistor (optional) NPN bipolar junction transistor Fairchild MMBT3904 Zetex FMMT413 Table 3. Component Suppliers MANUFACTURER Murata Chip Information PHONE FAX 650-964-6321 650-964-8165 PROCESS: BiCMOS Sumida 847-545-6700 847-545-6720 Taiyo Yuden 800-348-2496 847-925-0899 Package Information TDK 847-803-6100 847-803-6296 Toko 1-800-PIK-TOKO 408-943-9790 For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. Note that a “+”, “#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. 14 PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 8 SO S8-F6 21-0041 ______________________________________________________________________________________ 1MHz, All-Ceramic, 2.6V to 5.5V Input, 2A PWM Step-Down DC-to-DC Regulators REVISION NUMBER REVISION DATE 0 10/02 Initial release 1 8/03 Updated data sheet title, Features, Typical Operating Circuit, Detailed Description and added Thermal Considerations section. 2 6/09 Revised Ordering Information, Electrical Characteristics, Typical Operating Characteristics, Pin Description, and the Compensation Design section. DESCRIPTION PAGES CHANGED — 1–15 1–7, 10, 11, 14 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15 © 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc. MAX1951/MAX1952 Revision History