February 1999 PRELIMINARY ML4803 8-Pin PFC and PWM Controller Combo GENERAL DESCRIPTION FEATURES The ML4803 is a space-saving controller for power factor corrected, switched mode power supplies that offers very low start-up and operating currents. ■ Internally synchronized PFC and PWM in one 8-pin IC ■ Patented one-pin voltage error amplifier with advanced input current shaping technique ■ Peak or average current, continuous boost, leading edge PFC (Input Current Shaping Technology) ■ High efficiency trailing-edge current mode PWM ■ Low supply currents; start-up: 150µA typ., operating: 2mA typ. ■ Synchronized leading and trailing edge modulation ■ Reduces ripple current in the storage capacitor between the PFC and PWM sections ■ Overvoltage, UVLO, and brownout protection ■ PFC VCCOVP with PFC Soft Start Power Factor Correction (PFC) offers the use of smaller, lower cost bulk capacitors, reduces power line loading and stress on the switching FETs, and results in a power supply fully compliant to IEC1000-3-2 specifications. The ML4803 includes circuits for the implementation of a leading edge, average current “boost” type PFC and a trailing edge, PWM. The ML4803-1’s PFC and PWM operate at the same frequency, 67kHz. The PFC frequency of the ML4803-2 is automatically set at half that of the 134kHz PWM. This higher frequency allows the user to design with smaller PWM components while maintaining the optimum operating frequency for the PFC. An overvoltage comparator shuts down the PFC section in the event of a sudden decrease in load. The PFC section also includes peak current limiting for enhanced system reliability. BLOCK DIAGRAM 7 VEAO 7V 4 + VCC PFC OFF COMP – 17.5V 16.2V 35µA REF + VCC OVP VREF GND COMP – 2 M3 –1 M4 M1 PFC CONTROL LOGIC M2 R1 C1 30pF PFC OUT 1 M7 LEADING EDGE PFC ONE PIN ERROR AMPLIFIER 3 ISENSE + + VCC OSCILLATOR PFC – 67kHz PWM – 134kHz 26k – TRAILING EDGE PWM DUTY CYCLE LIMIT PWM COMPARATOR COMP 40k + 1.2V 6 SOFT START PFC/PWM UVLO VREF VDC – PFC ILIMIT – –1V 5 COMP –4 ILIMIT PWM CONTROL LOGIC – COMP + PWM OUT 8 M6 1.5V – DC ILIMIT + 1 ML4803 PIN CONFIGURATION ML4803 8-Pin PDIP (P08) 8-Pin SOIC (S08) PFC OUT 1 8 PWM OUT GND 2 7 VCC ISENSE 3 6 ILIMIT VEAO 4 5 VDC TOP VIEW PIN DESCRIPTION PIN NAME FUNCTION 1 PFC OUT PFC driver output 2 GND 3 4 2 PIN NAME FUNCTION 5 VDC PWM voltage feedback input Ground 6 I LIMIT PWM current limit comparator input ISENSE Current sense input to the PFC current limit comparator 7 VCC Positive supply (may require an external shunt regulator) VEAO PFC one-pin error amplifier input 8 PWM OUT PWM driver output February 1999 ML4803 ABSOLUTE MAXIMUM RATINGS Absolute maximum ratings are those values beyond which the device could be permanently damaged. Absolute maximum ratings are stress ratings only and functional device operation is not implied. ICC Current (average) ............................................. 40mA VCC MAX ............................................................... 18.3V ISENSE Voltage .................................................. -5V to 1V Voltage on Any Other Pin ...... GND - 0.3V to VCC + 0.3V Peak PFC OUT Current, Source or Sink ....................... 1A Peak PWM OUT Current, Source or Sink ..................... 1A PFC OUT, PWM OUT Energy Per Cycle .................. 1.5µJ Junction Temperature .............................................. 150°C Storage Temperature Range ..................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) ..................... 260°C Thermal Resistance (qJA) Plastic DIP ..................................................... 110°C/W Plastic SOIC ................................................... 160°C/W OPERATING CONDITIONS Temperature Range ML4803CX-X ............................................. 0°C to 70°C ML4803IX-X ............................................-40°C to 85°C ELECTRICAL CHARACTERISTICS Unless otherwise specified, VCC = 15V, TA = Operating Temperature Range (Note 1) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 33.5 35.0 36.5 µA 0.1 0.3 µA 15.5 16.0 16.5 V -0.9 -1 -1.15 V 150 300 ns 1.5 1.6 V 150 300 ns 67 74 kHz ONE-PIN ERROR AMPLIFIER VEAO Output Current TA = 25ºC, VEAO = 6V Line Regulation 10V < VCC < 15V, VEAO = 6V VCC OVP COMPARATOR Threshold Voltage TA = 0ºC to 70ºC PFC ILIMIT COMPARATOR Threshold Voltage Delay to Output DC ILIMIT COMPARATOR Threshold Voltage 1.4 Delay to Output OSCILLATOR Initial Accuracy TA = 25°C Voltage Stability 10V < VCC < 15V 62 Temperature Stability 1 % 2 % Total Variation Over Line and Temp 60 67 74.5 kHz Dead Time PFC Only 0.3 0.45 0.65 µs Minimum Duty Cycle VEAO > 7.0V,ISENSE = -0.2V 0 % Maximum Duty Cycle VEAO < 4.0V,ISENSE = 0V PFC 95 % 8 15 W IOUT = -100mA 0.8 1.5 V IOUT = –10mA, VCC = 8V 0.7 1.5 V Output Low Impedance Output Low Voltage 90 February 1999 3 ML4803 ELECTRICAL CHARACTERISTICS (Continued) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 8 15 W PFC (Continued) Output High Impedance Output High Voltage IOUT = 100mA, VCC = 15V 13.5 Rise/Fall Time CL = 1000pF Duty Cycle Range TA = 0ºC to 70ºC, ML4803-2 0-43 TA = 0ºC to 70ºC, ML4803-1 0-49.5 14.2 V 50 ns PWM 0-50 % 0-50 % 8 15 W IOUT = –100mA 0.8 1.5 V IOUT = –10mA, VCC = 8V 0.7 1.5 V 8 15 W Output Low Impedance Output Low Voltage Output High Impedance Output High Voltage IOUT = 100mA, VCC = 15V Rise/Fall Time CL = 1000pF VCC Clamp Voltage (VCCZ) ICC = 10mA Start-up Current Operating Current 13.5 0-47 14.2 V 50 ns SUPPLY Note 1: 4 16.7 17.5 18.3 V VCC = 11V, CL = 0 0.2 0.4 mA VCC = 15V, CL = 0 2.5 4 mA Undervoltage Lockout Threshold 11.5 12 12.5 V Undervoltage Lockout Hysteresis 2.4 2.9 3.4 V Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions. February 1999 ML4803 FUNCTIONAL DESCRIPTION The ML4803 consists of an average current mode boost Power Factor Corrector (PFC) front end followed by a synchronized Pulse Width Modulation (PWM) controller. It is distinguished from earlier combo controllers by its low pin count, innovative input current shaping technique, and very low start-up and operating currents. The PWM section is dedicated to peak current mode operation. It uses conventional trailing-edge modulation, while the PFC uses leading-edge modulation. This patented Leading Edge/ Trailing Edge (LETE) modulation technique helps to minimize ripple current in the PFC DC buss capacitor. The ML4803 is offered in two versions. The ML4803-1 operates both PFC and PWM sections at 67kHz, while the ML4803-2 operates the PWM section at twice the frequency (134kHz) of the PFC. This allows the use of smaller PWM magnetics and output filter components, while minimizing switching losses in the PFC stage. In addition to power factor correction, several protection features have been built into the ML4803. These include soft start, redundant PFC over-voltage protection, peak current limiting, duty cycle limit, and under voltage lockout (UVLO). See Figure 12 for a typical application. DETAILED PIN DESCRIPTIONS VEAO This pin provides the feedback path which forces the PFC output to regulate at the programmed value. It connects to programming resistors tied to the PFC output voltage and is shunted by the feedback compensation network. ISENSE This pin ties to a resistor or current sense transformer which senses the PFC input current. This signal should be negative with respect to the IC ground. It internally feeds the pulse-by-pulse current limit comparator and the current sense feedback signal. The ILIMIT trip level is –1V. The ISENSE feedback is internally multiplied by a gain of four and compared against the internal programmed ramp to set the PFC duty cycle. The intersection of the boost inductor current downslope with the internal programming ramp determines the boost off-time. is offset internally by 1.2V and then compared against the opto feedback voltage to set the PWM duty cycle. PFC OUT and PWM OUT PFC OUT and PWM OUT are the high-current power drivers capable of directly driving the gate of a power MOSFET with peak currents up to ±1A. Both outputs are actively held low when VCC is below the UVLO threshold level. VCC VCC is the power input connection to the IC. The VCC startup current is 150µA . The no-load ICC current is 2mA. VCC quiescent current will include both the IC biasing currents and the PFC and PWM output currents. Given the operating frequency and the MOSFET gate charge (Qg), average PFC and PWM output currents can be calculated as IOUT = Qg x F. The average magnetizing current required for any gate drive transformers must also be included. The VCC pin is also assumed to be proportional to the PFC output voltage. Internally it is tied to the VCCOVP comparator (16.2V) providing redundant highspeed over-voltage protection (OVP) of the PFC stage. VCC also ties internally to the UVLO circuitry, enabling the IC at 12V and disabling it at 9.1V. VCC must be bypassed with a high quality ceramic bypass capacitor placed as close as possible to the IC. Good bypassing is critical to the proper operation of the ML4803. VCC is typically produced by an additional winding off the boost inductor or PFC Choke, providing a voltage that is proportional to the PFC output voltage. Since the VCCOVP max voltage is 16.2V, an internal shunt limits VCC overvoltage to an acceptable value. An external clamp, such as shown in Figure 1, is desirable but not necessary. VCC is internally clamped to 16.7V minimum, 18.3V maximum. This limits the maximum VCC that can be applied to the IC while allowing a VCC which is high VCC VDC This pin is typically tied to the feedback opto-collector. It is tied to the internal 5V reference through a 26kW resistor and to GND through a 40kW resistor. ILIMIT 1N4148 1N4148 1N5246B This pin is tied to the primary side PWM current sense resistor or transformer. It provides the internal pulse-by pulse-current limit for the PWM stage (which occurs at 1.5V) and the peak current mode feedback path for the current mode control of the PWM stage. The current ramp February 1999 GND Figure 1. Optional VCC Clamp 5 ML4803 FUNCTIONAL DESCRIPTION (Continued) enough to trip the VCCOVP. The max current through this zener is 10mA. External series resistance is required in order to limit the current through this Zener in the case where the VCC voltage exceeds the zener clamp level. GND GND is the return point for all circuits associated with this part. Note: a high-quality, low impedance ground is critical to the proper operation of the IC. High frequency grounding techniques should be used. POWER FACTOR CORRECTION Power factor correction makes a nonlinear load look like a resistive load to the AC line. For a resistor, the current drawn from the line is in phase with, and proportional to, the line voltage. This is defined as a unity power factor is (one). A common class of nonlinear load is the input of a most power supplies, which use a bridge rectifier and capacitive input filter fed from the line. Peak-charging effect, which occurs on the input filter capacitor in such a supply, causes brief high-amplitude pulses of current to flow from the power line, rather than a sinusoidal current in phase with the line voltage. Such a supply presents a power factor to the line of less than one (another way to state this is that it causes significant current harmonics to appear at its input). If the input current drawn by such a supply (or any other nonlinear load) can be made to follow the input voltage in instantaneous amplitude, it will appear resistive to the AC line and a unity power factor will be achieved. To hold the input current draw of a device drawing power from the AC line in phase with, and proportional to, the SW2 L1 + I2 I1 input voltage, a way must be found to prevent that device from loading the line except in proportion to the instantaneous line voltage. The PFC section of the ML4803 uses a boost-mode DC-DC converter to accomplish this. The input to the converter is the full wave rectified AC line voltage. No filtering is applied following the bridge rectifier, so the input voltage to the boost converter ranges, at twice line frequency, from zero volts to the peak value of the AC input and back to zero. By forcing the boost converter to meet two simultaneous conditions, it is possible to ensure that the current that the converter draws from the power line matches the instantaneous line voltage. One of these conditions is that the output voltage of the boost converter must be set higher than the peak value of the line voltage. A commonly used value is 385VDC, to allow for a high line of 270VACRMS. The other condition is that the current that the converter is allowed to draw from the line at any given instant must be proportional to the line voltage. Since the boost converter topology in the ML4803 PFC is of the current-averaging type, no slope compensation is required. LEADING/TRAILING MODULATION Conventional Pulse Width Modulation (PWM) techniques employ trailing edge modulation in which the switch will turn ON right after the trailing edge of the system clock. The error amplifier output voltage is then compared with the modulating ramp. When the modulating ramp reaches the level of the error amplifier output voltage, the switch will be turned OFF. When the switch is ON, the inductor I3 I4 VIN RL SW1 DC C1 RAMP VEAO REF U3 + EA – TIME DFF RAMP OSC U4 CLK + – U1 R Q D U2 Q CLK VSW1 TIME Figure 2. Typical Trailing Edge Control Scheme. 6 February 1999 ML4803 LEADING/TRAILING MODULATION (Continued) programming resistor. The nominal voltage at the VEAO pin is 5V. The VEAO voltage range is 4 to 6V. For a 11.3MW resistor chain to the boost output voltage and 5V steady state at the VEAO, the boost output voltage would be 400V. current will ramp up. The effective duty cycle of the trailing edge modulation is determined during the ON time of the switch. Figure 2 shows a typical trailing edge control scheme. In the case of leading edge modulation, the switch is turned OFF right at the leading edge of the system clock. When the modulating ramp reaches the level of the error amplifier output voltage, the switch will be turned ON. The effective duty-cycle of the leading edge modulation is determined during the OFF time of the switch. Figure 3 shows a leading edge control scheme. PROGRAMMING RESISTOR VALUE Equation 1 calculates the required programming resistor value. Rp = VBOOST − VEAO 400V − 50 . V = = 113 . MΩ IPGM 35µA PFC VOLTAGE LOOP COMPENSATION One of the advantages of this control technique is that it requires only one system clock. Switch 1 (SW1) turns OFF and Switch 2 (SW2) turns ON at the same instant to minimize the momentary “no-load” period, thus lowering ripple voltage generated by the switching action. With such synchronized switching, the ripple voltage of the first stage is reduced. Calculation and evaluation have shown that the 120Hz component of the PFC’s output ripple voltage can be reduced by as much as 30% using this method, substantially reducing dissipation in the highvoltage PFC capacitor. The voltage-loop bandwidth must be set to less than 120Hz to limit the amount of line current harmonic distortion. A typical crossover frequency is 30Hz. Equation 1, for simplicity, assumes that the pole capacitor dominates the error amplifier gain at the loop unity-gain frequency. Equation 2 places a pole at the crossover frequency, providing 45 degrees of phase margin. Equation 3 places a zero one decade prior to the pole. Bode plots showing the overall gain and phase are shown in Figures 5 and 6. Figure 4 displays a simplified model of the voltage loop. TYPICAL APPLICATIONS C COMP = Pin C COMP = The ML4803 utilizes a one pin voltage error amplifier in the PFC section (VEAO). The error amplifier is in reality a current sink which forces 35µA through the output SW2 L1 + I2 1 Rp × VBOOST × ∆VEAO × C OUT × 2 × π × f ONE PIN ERROR AMP I1 (1) 300W 0 6 2 (2) 5 113 . MW ´ 400V ´ 0.5V ´ 220mF ´ 2 ´ p ´ 30Hz 2 C COMP = 16nF I3 I4 VIN RL SW1 DC C1 RAMP VEAO REF U3 + –EA RAMP OSC U4 CLK VEAO + – CMP U1 TIME DFF R Q D U2 Q CLK VSW1 TIME Figure 3. Typical Leading Edge Control Scheme. February 1999 7 ML4803 TYPICAL APPLICATIONS (Continued) R COMP = R COMP = C ZERO = C ZERO = 1 (3) 2 ´ p ´ f ´ C COMP 1 = 330kW 6.28 ´ 30Hz ´ 16nF PFC CURRENT SENSE FILTERING 1 2´ p ´ to develop the internal ramp by charging the internal 30pF +12/–10% capacitor. See Figures 10 and 11. The frequency of the internal programming ramp is set internally to 67kHz. f ´ R COMP 10 (4) 1 = 0.16mF 6.28 ´ 3Hz ´ 330kW INTERNAL VOLTAGE RAMP The internal ramp current source is programmed by way of the VEAO pin voltage. Figure 7 displays the internal ramp current vs. the VEAO voltage. This current source is used In DCM, the input current wave shaping technique used by the ML4803 could cause the input current to run away. In order for this technique to be able to operate properly under DCM, the programming ramp must meet the boost inductor current down-slope at zero amps. Assuming the programming ramp is zero under light load, the OFF-time will be terminated once the inductor current reaches zero. Subsequently the PFC gate drive is initiated, eliminating the necessary dead time needed for the DCM mode. This forces the output to run away until the VCC OVP shuts down the PFC. This situation is corrected by adding an 60 Power Stage Overall Gain Compensation Network Gain 40 VO 11.3MΩ VEAO IOUT ML4803 220µF ∆VEAO + RLOAD 667Ω ML4803 GAIN (dB) 20 0 –20 330kΩ IVEAO 35µA –40 15nF – 0.15µF POWER STAGE –60 0.1 10 1 COMPENSATION Figure 4. Voltage Control Loop 0 1000 100 FREQUENCY (Hz) Figure 5. Voltage Loop Gain 50 Power Stage Overall Compensation Network FF @ –55ºC TYP @ –55ºC 40 50 IRAMP (µA) PHASE (º) TYP @ ROOM TEMP 100 150 200 0.1 TYP @ 155ºC 30 SS @ 155ºC 20 10 0 1 10 100 1000 0 8 2 3 4 5 6 7 VEAO (V) FREQUENCY (Hz) Figure 6. Voltage Loop Phase 1 Figure 7. Internal Ramp Current vs. VEAO February 1999 ML4803 TYPICAL APPLICATIONS (Continued) offset voltage to the current sense signal, which forces the duty cycle to zero at light loads. This offset prevents the PFC from operating in the DCM and forces pulse-skipping from CCM to no-duty, avoiding DMC operation. External filtering to the current sense signal helps to smooth out the sense signal, expanding the operating range slightly into the DCM range, but this should be done carefully, as this filtering also reduces the bandwidth of the signal feeding the pulse-by-pulse current limit signal. Figure 9 displays a typical circuit for adding offset to ISENSE at light loads. VEAO current sink and discharging the VEAO compensation components until the steady state operating point is reached. It should be noted that, as shown in Figure 8, once the VEAO pin exceeds 6.5V, the internal ramp is defeated. Because of this, an external Zener can be installed to reduce the maximum voltage to which the VEAO pin may rise in a shutdown condition. Clamping the VEAO pin externally to 7.4V will reduce the time required for the VEAO pin to recover to its steady state value. UVLO PFC Start-Up and Soft Start During steady state operation VEAO draws 35µA. At startup the internal current mirror which sinks this current is defeated until VCC reaches 12V. This forces the PFC error voltage to VCC at the time that the IC is enabled. With leading edge modulation VCC on the VEAO pin forces zero duty on the PFC output. When selecting external compensation components and VCC supply circuits VEAO must not be prevented from reaching 6V prior to VCC reaching 12V in the turn-on sequence. This will guarantee that the PFC stage will enter soft-start. Once VCC reaches 12V the 35µA VEAO current sink is enabled. VEAO compensation components are then discharged by way of the 35µA current sink until the steady state operating point is reached. See Figure 8. PFC SOFT RECOVERY FOLLOWING VCC OVP The ML4803 assumes that VCC is generated from a source that is proportional to the PFC output voltage. Once that source reaches 16.2V the internal current sink tied to the VEAO pin is disabled just as in the soft start turn-on sequence. Once disabled, the VEAO pin charges HIGH by way of the external components until the PFC duty cycle goes to zero, disabling the PFC. The VCC OVP resets once the VCC discharges below 16.2V, enabling the Once VCC reaches 12V both the PFC and PWM are enabled. The UVLO threshold is 9.1V providing 2.9V of hysteresis. GENERATING VCC An internal clamp limits overvoltage to VCC. This clamp circuit ensures that the VCC OVP circuitry of the ML4803 will function properly over tolerance and temperature while protecting the part from voltage transients. This circuit allows the ML4803 to deliver 15V nominal gate drive at PWM OUT and PFC OUT, sufficient to drive lowcost IGBTs. It is important to limit the current through the Zener to avoid overheating or destroying it. This can be done with a single resistor in series with the VCC pin, returned to a bias supply of typically 14V to 18V. The resistor value must be chosen to meet the operating current requirement of the ML4803 itself (4.0mA max) plus the current required by the two gate driver outputs. VCC OVP VCC is assumed to be a voltage proportional to the PFC output voltage, typically a bootstrap winding off the boost 10V/div. VCC C23 0.01µF 0 VEAO 10V/div. 0 VOUT PFC GATE 10V/div. 200V/div. 0 CR16 1N4148 R28 20kΩ C16 1µF R4 1kΩ R19 10kΩ ISENSE 0 VBOOST R29 20kΩ C5 0.0082µF R3 0.015Ω 3W VCC RTN 200ms/Div. Figure 8. PFC Soft Start Figure 9. ISENSE Offset for Light Load Conditions February 1999 9 ML4803 TYPICAL APPLICATIONS (Continued) inductor. The VCC OVP comparator senses when this voltage exceeds 16V, and terminates the PFC output drive while disabling the VEAO current sink. Once the VEAO current sink is disabled, the VEAO voltage will charge unabated, except for a diode clamp to VCC, reducing the PFC pulse width. Once the VCC rail has decreased to below 16.2V the VEAO sink will be enabled, discharging external VEAO compensation components until the steady state voltage is reached. Given that 15V on VCC corresponds to 400V on the PFC output, 16V on VCC corresponds to an OVP level of 426V. COMPONENT REDUCTION Components associated with the VRMS and IRMS pins of a typical PFC controller such as the ML4824 have been eliminated. The PFC power limit and bandwidth does vary with line voltage. Double the power can be delivered from a 220 V AC line versus a 110 V AC line. Since this is a combination PFC/PWM, the power to the load is limited by the PWM stage. VISENSE VC1 RAMP GATE DRIVE OUTPUT Figure 10. Typical Peak Current Mode Waveforms VOUT = 400V RP VC1 VEAO 4 RCOMP + C1 30pF CCOMP 35µA 5V R1 CZERO 3 ISENSE –4 VI SENSE Figure 11. ML4803 PFC Control 10 February 1999 COMP – GATE OUTPUT ML4803 LINE F1 5A 250V J1-1 R24 470kΩ 0.5W C19 4.7nF 250VAC 102T L2 TH1 10Ω 5A C4 0.47µF 250VAC Q5 R1 BR1 600V 4A 36Ω Q2 NEUTRAL J1-2 CR1 8A, 600V 1000µH R2 L3 C20 4.7nF 250VAC C1 220µF 450V 36Ω R22 10kΩ C16 0.01µF CR5 16V 0.5W R3 0.15Ω 3W Q4 CR18 51V R8 36Ω CR7 CR3 R30 200Ω T2 R13 5.62MΩ 3 10 4 C7 0.1µF R28 20kΩ 1 2 3 CR12 4 R25 390kΩ 7.0V C8 0.15µF PFC GND ISENSE VEAO PWM VCC ILIMIT VDC C6 1µF C28 1µF 12VRET J2-2 4T CR15 R6 1.2kΩ 5 6 R11 150Ω 5 R32 100Ω CR11 C27 0.01µF CR9 R21 10kΩ U3 1 R37 330Ω C17 0.1µF 2 4 R10 0.75Ω 3W C14 4.7µF R9 1.5kΩ Q3 R5 36Ω C10 2.2nF R14 150Ω 2W C26 0.01µF 500V 7 C5 8.2nF C15 0.015µF C9 1µF C21 1µF CR10 8 C2 2200µF L2 R31 10Ω ML4803 C3 1µF CR8 C22 1µF R19 10kΩ J2-1 CR2 30A, 60V L1 25µH R26 20kΩ 3W R29 20kΩ R4 1kΩ 12V CR2 30A 60V R27 20kΩ 3W CR16 IN4148 C25 0.01µF 500V R36 220Ω T1 C11 1000µF R12 5.62MΩ C18 4.7nF R23 10kΩ C29 0.01µF R7 10Ω C23 0.01µF Q1 R38 22Ω 1 R15 9.09kΩ C13 1nF R17 3.3kΩ CR4 R20 510Ω 3 1 U2 C12 0.1µF 2 R18 1kΩ R16 2.37kΩ Figure 12. Typical Application Circuit. Universal Input 240W 12V DC Output February 1999 11 ML4803 PHYSICAL DIMENSIONS inches (millimeters) Package: P08 8-Pin PDIP 0.365 - 0.385 (9.27 - 9.77) 0.055 - 0.065 (1.39 - 1.65) 8 0.240 - 0.260 0.299 - 0.335 (6.09 - 6.60) (7.59 - 8.50) PIN 1 ID 1 0.020 MIN (0.51 MIN) (4 PLACES) 0.100 BSC (2.54 BSC) 0.015 MIN (0.38 MIN) 0.170 MAX (4.32 MAX) 0.125 MIN (3.18 MIN) 0.016 - 0.020 (0.40 - 0.51) 0º - 15º 0.008 - 0.012 (0.20 - 0.31) SEATING PLANE Package: S08 8-Pin SOIC 0.189 - 0.199 (4.80 - 5.06) 8 PIN 1 ID 0.148 - 0.158 0.228 - 0.244 (3.76 - 4.01) (5.79 - 6.20) 1 0.017 - 0.027 (0.43 - 0.69) (4 PLACES) 0.050 BSC (1.27 BSC) 0.059 - 0.069 (1.49 - 1.75) 0º - 8º 0.055 - 0.061 (1.40 - 1.55) 0.012 - 0.020 (0.30 - 0.51) 0.004 - 0.010 (0.10 - 0.26) 0.015 - 0.035 (0.38 - 0.89) SEATING PLANE 12 February 1999 0.006 - 0.010 (0.15 - 0.26) ML4803 ORDERING INFORMATION PART NUMBER PFC/PWM FREQUENCY TEMPERATURE RANGE PACKAGE ML4803CP-1 ML4803CS-1 67kHz / 67kHz 67kHz / 67kHz 0°C to 70°C 0°C to 70°C 8-Pin PDIP (P08) 8-Pin SOIC (S08) ML4803IP-1 ML4803IS-1 67kHz / 67kHz 67kHz / 67kHz -40°C to 85°C -40°C to 85°C 8-Pin PDIP (P08) 8-Pin SOIC (S08) ML4803CP-2 ML4803CS-2 67kHz / 134kHz 67kHz / 134kHz 0°C to 70°C 0°C to 70°C 8-Pin PDIP (P08) 8-Pin SOIC (S08) ML4803IP-2 ML4803IS-2 67kHz / 134kHz 67kHz / 134kHz -40°C to 85°C -40°C to 85°C 8-Pin PDIP (P08) 8-Pin SOIC (S08) Micro Linear Corporation 2092 Concourse Drive San Jose, CA 95131 Tel: (408) 433-5200 Fax: (408) 432-0295 www.microlinear.com © Micro Linear 1999. property of their respective owners. is a registered trademark of Micro Linear Corporation. All other trademarks are the Products described herein may be covered by one or more of the following U.S. patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940; 5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; 5,652,479; 5,661,427; 5,663,874; 5,672,959; 5,689,167; 5,714,897; 5,717,798; 5,742,151; 5,747,977; 5,754,012; 5,757,174; 5,767,653; 5,777,514; 5,793,168; 5,798,635; 5,804,950; 5,808,455; 5,811,999; 5,818,207; 5,818,669; 5,825,165; 5,825,223; 5,838,723; 5.844,378; 5,844,941. Japan: 2,598,946; 2,619,299; 2,704,176; 2,821,714. Other patents are pending. Micro Linear makes no representations or warranties with respect to the accuracy, utility, or completeness of the contents of this publication and reserves the right to make changes to specifications and product descriptions at any time without notice. No license, express or implied, by estoppel or otherwise, to any patents or other intellectual property rights is granted by this document. The circuits contained in this document are offered as possible applications only. Particular uses or applications may invalidate some of the specifications and/or product descriptions contained herein. The customer is urged to perform its own engineering review before deciding on a particular application. Micro Linear assumes no liability whatsoever, and disclaims any express or implied warranty, relating to sale and/or use of Micro Linear products including liability or warranties relating to merchantability, fitness for a particular purpose, or infringement of any intellectual property right. Micro Linear products are not designed for use in medical, life saving, or life sustaining applications. DS4803-01 February 1999 13