PBL
385
73
June
1999
PBL 385 73
Speech Circuit for constant current
feeding systems
Description.
Key features.
PBL 38573 is a monolithic integrated speech transmission circuit for use in
electronic telephones. It is intended specially for telephone lines with constant current
feed. Maximum line current less than 130 mA in DIL package (100 mA in SO package).
It is designed to accomodate either a low impedance dynamic or an electret microphone.
A signal summing point at the transmitter input is available for DTMF dialler- and
possible monitor or handsfree signals. An available internally preset line length
compensation can be adjusted or shut off in low gain mode with external resistors.
Application dependent parameters such as line balance, side tone level, transmitter and
receiver gains and frequency responces are set independently by external components
which means an easy adaption to various market needs. The setting of the parameters
if carried out in certain order will counteract the interaction between the settings. A DCsupply is provided to feed microphones and diallers.
•
•
•
•
•
•
•
•
•
•
•
1
MIC.
13
PBL 385 73
9
AT
AM
10
REC
AR
14
DC-supply
7
Minimum number of external
components for function,with one filtered DC-supply. 6 capacitors and 10
resistors.
Easy adaption to various market
needs.
Mute control input for operation with
DTMF - generator.
Transmitter and receiver gain
regulation for automatic loop loss
compensation. Disconnectable.
Extended current and voltage range
5 - 130mA (DIP), down to 2 V.
Differential microphone input for good
balance to ground.
Balanced receiver output stage.
Stabilized DC - supply for low current
CMOS diallers and electret
microphones.
Short start up time.
Excellent RFI performance.
In 14- pin DIP and SO packages.
6
5
3
8
11
12
2
Telephone
line
+4
PB
L
38
5
73
Mute
(active low)
DTMF - input
4
5
DC supply for
external devices
2
+
3
+
1
B
P
1. Impedance to the line and radio interference suppression
2. Transmitter gain and frequency responce network
3. Receiver gain and frequency responce network
4. Sidetone balance network
5. DC supply components
L
38
5
73
14-pin plastic SO
14-pin plastic DIP
Figure 1. Functional diagram.
1
PBL 385 73
Maximum Ratings
Parameter
Symbol
Min
Max
Unit
Line voltage, tp = 2 s
Line current, continuous DIP package
Line current, continuous SO package (depending on mounting)
Operating temperature range
Storage temperature range
VL
IL
IL
TAmb
TStg
0
0
0
-40
-55
18
130
100
+70
+125
V
mA
mA
°C
°C
No input should be set on higher level than pin 4 (+C).
MUTE
VM
R = 0-4kΩ
L
0 ohm when artificial
line is used
5H+5H
feed = 400Ω+400Ω
+ LINE
Z Mic = 350Ω
MIC
C
V2
+
IL
ARTIFICIAL
LINE
+
R
IM
V3
PBL 38573
with external
components
See fig. 4
I DC
600Ω
V
E = 48.5V
L
VDC
V1
Z Rec= 350Ω
V4
REC
- LINE
C = 1µF when artificial line is used
470µF when no artificial line
Figure 2. Test set up without rectifier
bridge.
MUTE
VM
5H+5H
Uz= 15-16V
RL = 0 - 4kΩ
+
R
feed = 400Ω+400Ω
+
V
1µF
V2
IL
IM
+ LINE
Z Mic = 350Ω
MIC
L
600Ω
V3
PBL 38573
with external
components
See fig. 4
I DC
E = 50.0V
VDC
V1
Z Rec = 350Ω
V4
REC
- LINE
Figure 3. Test set up with rectifier
bridge.
+ Line
1
13
2.7k
Mic.
350
Ω
PBL 385 73
9
310
Ω
AT
AM
AR
10
REC.
350
Ω
14
DC-supply
7
6
5
R4
18k
8
3
C3
100nF
R7
910
Ω
R8
560
Mute
11
12
2
R10
C6
47nF
R11
6.2k
62k
C11
47µF
+
RG
R5
22k
R6
75
Ω
C5
100nF
R3
910
Figure 4. Circuit with external components for test set up.
Ω
Ω
DC supply
+4
R9
11k
R12
11k
+ C1
47µF
C2
15nF
-Line
2
DIP package pinning.
PBL 385 73
Electrical Characterisics
At TAmb = + 25° C. No cable and line rectifier unless otherwise specified.
Parameter
Line voltage, VL
Transmitting gain, note 1
No gain regulation
Transmitting gain, note 1
With gain regulation
Ref.
fig.
Conditions
Min
Typ
Max
Unit
3.3
11
3.7
13
4.1
15
V
V
41
43
45
dB
2
IL = 15 mA
IL = 100 mA
20 •10 log (V2 / V3); 1 kHz
Rg = 20kΩ
20 •10 log (V2 / V3); 1 kHz
RL = 0, RG = ∞
RL = 400 Ω
RL = 900 Ω - 2.2 kΩ
1 kHz, RL = 0 to 900 Ω
41
43.5
46
3
43
45.5
48
5
45
47.5
50
7
dB
dB
dB
dB
2
200 Hz to 3.4 kHz
-1
1
dB
2
2
1 kHz
13.5
2
2
2
2
Transmitting range of
regulation
Transmitting frequency
response
Microphone input impedance
Transmitter input impedance
pin 3
Transmitter dynamic output
2
Transmitter max output
2
Transmitter output noise
Receiving gain, note 1
No gain regulation
Receiving gain, note 1
With gain regulation
2
2
Receiving range of regulation
Receiving frequency response
Receiver input impedance
Receiver output impedance
Receiver dynamic output
note 2
Receiver max. output
2
2
2
2
2
3
Receiver output noise
2
Mute input voltage
at mute (active low)
DC-supply voltage, note 4
Pin 7
2
DC-supply voltage, pin 7
leakage current (no supply)
2
2
4
200 Hz - 3.4 kHz
≤ 2% distortion, IL = 20 - 100 mA
200 Hz - 3.4 kHz
IL = 0 - 100 mA, V3 = 0 - 1 V
Psoph-weighting, Rel 1 Vrms, RL = 0
20 • 10 log (V4 / V1); 1 kHz
Rg = 20kΩ
20 • 10 log (V4 / V1); 1 kHz
RL = 0, RG = ∞
RL = 400 Ω
RL = 900 Ω - 2.2 kΩ
1 kHz, RL = 0 to 900 Ω
200 Hz to 3.4 kHz
1 kHz,
1 kHz,
200 Hz - 3.4 kHz
≤ 2% distortion, IL = 20 - 100 mA
Measured with line rectifier
200 Hz - 3.4 kHz, IL = 0 - 100 mA,
V1= 0 - 50 V
A-weighting, Rel 1Vrms, with cable
0 - 3 km, Ø = 0.4 mm
0 - 5 km, Ø = 0.5 mm,
IL = 20 - 100 mA
IDC = 0 mA
IDC = 2 mA
VDC = 2.35 V
1.7//(2.7) note 3
17
20.5
kΩ
kΩ
1.5
Vp
3
Vp
-75
dBPsoph
-18.5
-16.5
-14.5
dB
-18.5
-16
-13.5
3
-1
-16.5
-14
-11.5
5
-14.5
-12
-9.5
7
1
38
3(+310)note 3
0.5
dB
dB
dB
dB
dB
kΩ
Ω
Vp
0.9
Vp
-85
dB A
0.3
V
2.1
1.95
2.35
2.2
0.1
2.6
2.6
V
V
µA
Notes
1. Adjustable to both higher and lower values with external components.
2. The dynamic output can be doubled, see applications information.
3. External resistor in the test set up.
4. The DC output voltage is reduced at low line voltage (see fig. 8).
3
PBL 385 73
+L 1
14
RE 2
TO 2
13
RE 1
TI 3
12
RI
TI 3
12 RI
+C 4
11
-L
+C 4
17 -L
11
Mute 5
10
MI 2
Mute 5
GR 6
9
MI 1
GR 6
9
MI 1
DCO 7
8
MO
DCO 7
8
MO
+L 1
14 RE 2
TO 2
13 RE 1
14-pin DIP
16 MI 2
10
14-pin SO
Figure 5. Pin configurations.
Pin Descriptions:
Refer to figure 5.
Pin Name
Function
1
+L
Output of the DC-regulator and transmit amplifier. This pin is connected to the line through a polarity
guard diode bridge.
2
TO
Output of the transmit amplifier. This pin is connected through a resistor of 47 to 100 ohm to -L,
sets the DC-resistance of the circuit. The output has a low AC output impedance and the
signal is used to drive a side tone balancing network.
3
TI
Input of transmit amplifier. Input impedance 17 kohm ± 20 %.
4
+C
Positive power supply terminal for most of the circuitry inside the PBL 385 73 ( current consumption
about 1 mA). The +C pin shall be connected to a decoupling capacitor of 47 µF to 150 µF.
5
6
7
Mute
GR
DCO
Maximum voltage (to mute) is 0.3 V, current sink requirement of external driver is 100 µA.
Input of the gain regulation with line length.
Output of the auxiliary DC-supply.
8
MO
Output of the microphone amplifier or DTMF-amplifier.
9
MI 1
10
MI 2
11
-L
The negative power terminal, connected to the line through a polarity guard diode bridge.
12
RI
Input of receiver amplifier. Input impedance 38 kohm ± 20 %.
13
RE 1
14
RE 2
4
}
}
Microphone amplifier Inputs. Input impedance 1.7 kohm ± 20 %.
Receiver amplifier outputs. Output impedance is approximately 3 ohm.
PBL 385 73
Functional description
Design procedure; ref. to fig.4.
+Line
The design is made easier through that all
settable parameters are returned to ground (-line), this feature differs it from bridge
type solutions.To set the parameters in the
following order will result in that the
interaction between the same is minimized.
1. Set the circuit impedance to the line,
either resistive (600Ω) or complex. (R3
and C1). C1 should be big enough to give
low impedance compared with R3 in the
telephone speech frequency band.Too
large C1 will make the start-up slow. See
fig. 6.
2. Set the DC-characteristic that is
required in the PTT specification or in case
of a system telephone,in the PBX
specification (R6).There are also internal
circuit dependent requirements like supply
voltages etc.
3. Set the attac point where the line
length regulation is supposed to cut in
(R15 and R16). Note that in some countries
the line length regulation is not allowed. In
most cases the end result is better and
more readily achieved by using the line
length regulation (line loss compensation)
than without. See fig. 13.
4. Set the transmitter gain and
frequency response.
5. Set the receiver gain and frequency
response. See text how to limit the max.
swing to the earphone.
6. Adjust the side tone balancing
network.
7. Set the RFI suppression
components in case necessary. In two
piece telephones the often ”helically”
wound cord acts as an aerial. The
microphone input with its high gain is
especially sensitive.
8. Circuit protection. Apart from any
other protection devices used in the design a good practice is to connect a 15V
1W zener diode across the circuit , from
pin 1 to -Line.
a)
1
PBL 38 573
4
R3
Rs
≈1Ω
+
C1
R6
C2
Figure 6. AC-impedance.
Impedance to the line
The AC- impedance to the line is
set by R3, C1 and C2. Fig.4. The circuits
relatively high parallel impedance will not
influence it to any noticeable extent. At low
frequencies the influence of C1 can not be
neglected. Series resistance of C1 that is
dependent on the temperature and the
quality of the component will cause some
of the line signal to enter pin 4. This
generates a closed loop in the transmitter
amplifier that in it´s turn will create an
active impedance thus lowering the
impedance to the line. The impedance at
high frequencies is set by C2 that also
acts as a RFI suppressor.
In many specifications the
impedance towards the line is specified as
a complex network. See fig. 6. In case a).
the error signal entering pin 4 is set by the
ratio ≈Rs/R3 (910Ω), where in case b). the
ratio at high frequencies will be Rs/220Ω
because the 820Ω resistor is bypassed by
3
AR
AT
4
Transmitter summing
input
- Line
Mute
Example:
How to connect a
complex network.
220Ω+820Ω//Cx
-Line
1
2
Cx
2
3
a capacitor. To help up this situation the
complex network capacitor is connected
directly to ground, case c). making the ratio
Rs/220Ω+820Ω and thus lessening the
error signal. Conclusion: Connect like in
case c) when complex impedance is
specified.
DC - characteristic
The DC - characteristic that a
telephone set has to fulfill is mainly given
by the network administrator. Following
parameters are useful to know when the
DC behaviour of the telephone is to be set:
•
•
•
•
•
•
AM
c)
220Ω
820Ω
+ Line
+
b)
The voltage of the feeding system
The line feeding resistance 2 x.......
ohms.
The maximum current from the line at
zero line length.
The min. current at which the telephone
has to work (basic function).
The lowest and highest voltage
permissible across the telephone set.
The highest voltage that the
telephone may have at different line
currents. Normally set by the
network owners specification.The
lowest voltage for the telephone is
normally set by the voltages that are
needed for the different parts of the
telephone to function. For ex. for
transmitter output amplifier, receiver output amplifier, dialler, speech
switching.
Figure 7. Block connections.
5
PBL 385 73
V
16
V telephone line
14
V line
V pin 4
12
10
V pin 2
8
6
4
V pin 7
(DC supply)
2
I
20
40
60
80
100
120
L
mA
Figure 8. DC - Characteristics. (R6 =75Ω)
R6 will set the slope of the DC-char. and
the rest of the level is set by some constants
in the circuit as shown in the equation
below. The slope of the DC-char. will also
influence the line length regulation (when
used ) and thus the gain of both transmitter
and receiver. See the table under gain
regulation. R6 also acts as power protection
for the circuit, this must be kept in mind
when low values of R6 are considered.
V Line ≈ 2 + 1.5 ⋅ R 6 ⋅ I line
V telephoneline ≈ 1.5 V + V line
Microphone amplifier
The microphone amplifier in PBL38573 is
divided into two stages. The first stage is a
true differential amplifier providing high
CMRR (-55 to -65 dB typical) with voltage
gain of 19 dB. This stage is followed by a
gain regulated amplifier with a regulation
range of 5 ± 2 dB. The input of the
6
microphone amplifier can be used for
dynamic or electret transducers. See fig.
10. An electret microphone with a built in
FET amplifier is to be seen from outside as
a high impedance constant current generator and is normally specified with a load
resistance of ≈ 2k. This is to be considered
as max. value and by using it will render
the max. gain from the microphone. This
level of input signal that is unnecessary
high will result in clipping in the microphone
amplifier and could in mute condition
permeate through the input to the circuits
reference and this way to all functions,
resulting among other things in a bad mute.
Hence it is better regarding noise
perfomance and mute to rather use the
gain of the microphone amplifier than the
gain of the microphone itself (in case of
electret) flat out. A more suitable level of
gain from the microphone is achieved by
using a load resistance of 330 - 820Ω. A
low microphone impedance will also
improve RFI suppression. Gain setting to
the line is done at the input of the transmitter. The microphone amplifier has its own
temperature stable reference to prevent
overhearing to other parts and functions
on the chip.It is possible to use the
microphone amplifier as a limiter ( added)
to the limiter in the transmitter output stage
of the transmitted signal. See fig.9. The
positive output swing is then limited by the
peak output current of the microphone
amplifier. The negative swing is limited by
the saturation voltage of the output
amplifier. The output of the amplifier is DCvice at internal reference level (1.2V). The
lowest negative level for the signal is
reference minus one diode and saturated
tranistor drop. (1.2-0.6-0.1 = 0.5V) The
correct clipping level is found by
determining the composite AC- and DCload that gives a maximum symmetrical
unclipped signal at the output. This signal
is then fed into the transmitter amplifier at
a level that renders a symmetrical signal
clipping on the line. (adjust with ratio R4,R5)
The total transmitter gain when an electret
microphone is used can then be adjusted
with the load resistor of the electret
microphones buffer amplifier.
PBL 385 73
PBL 38573
(a)
8
PBL
38573
DC
( ref. ≈ 1.2V )
(b)
4
9
9
M
constant
current
generator
10
10
+
Dynamic
microphone
ref. minus
a diode ≈ 0.5V
8
4
PBL 38573
(c)
ACload
8
Unbalanced electret
mic. with balanced
signal, DC-supply from
pin 4.
pin 7.
(d)
PBL 38573
8
9
9
M
C +
DC-load = R4+R5
AC-load = R4+R5//ZTI
Figure 9. Microphone amplifier output clipping.
M
+
DC
R
DCload
PBL 38573
8
10 +
Balanced electret microphone.
An additional RC filterlink is
recommended if pin 4 is used
as a supply.
M
10
+
Balanced electret
microphone
Figure 10. Microphone solutions.
Transmitter amplifier
The transmitter amplifier in PBL38573 consists of three stages. The first stage is an amplitude limiter for the input signal at TI, in
order to prevent the transmitted signal to exceed a certain set level and cause distortion. The second stage further amplifies the signal
from the first and adds it to a DC level from an internal DC-regulation loop in order to give the required DC characteristic to the telephone
set. The output of this stage is TO. The third stage is a current generator that presents a high impedance towards the line and has its
gain from TO to +L. The gain of this amplifier is ZL/R6 where ZL is the impedance across the telephone line. Hence, the absolute
maximum signal amplitude that can be transmitted to the line undistorted is dependent of R6. (amplitude limiting)
The transmitter gain and frequency response are set by the RC-network between the pins 11 and 3. See fig.11. The capacitor
for cutting the high end of frequency band is best to be placed directly at the microphone where it also will act as a RFI suppressor.
The input signal source impedance to the transmitter amplifier input TI should be reasonably low in order to keep the gain spread down,
saying that R4//R5 (see fig. 4) must be at least a factor 5 lower than the ZTin. Observe that the capacitor C1 should have a reasonably
good temperature behaviour in order to keep the impedance rather constant. The V+C´s influence on the transmitter DC-characteristic
is shown in the fig.7 (DC-characteristic), therefore the transmitter gain would change if the transmitted signal gives reason to an acvoltage leak signal across C1 since this is a feedback point. If the transmitter has an unacceptable low sving to the line at low line
currents <≈10mA, the first step should be to examine if the circuits DC characteristic can be adjusted upwards.
How to calculate the gains in the transmitter channel.
See fig. 2 and 4.
Microphone amplifiers first stage 19 dB.
Microphone amplifiers regulated second stage 10.5 dB - 15.5 dB
Regulation interval 10.5 - 15.5 dB
low gain 19.0 + 10.5 dB = 29.5 dB
high gain 19.0 + 15.5 dB = 34.5 dB
V2
RM
R5
R load
=
⋅ GM ⋅
⋅ G TX ⋅
V 3 Z mic + R M
R4 +R5
R6
RM = Microphone amplifier input resistance
Rload = Rline // Rtelephone
ex. calculate the gain of the transmitter stage GTX at 0 - line length:
43 = 20 log(
(1.7 / /2.7)k
(17 / /22)k
600Ω / /910Ω
) + 29.5 + 20 log(
) + G TX + 20 log(
)
350Ω + (1.7 / /2.7)k
18k + (17 / /22)k
75Ω
43 = −2.51+ 29.5 − 9.17 + GTX + 13.66
GTX = 11.52 dB
7
PBL 385 73
11
3
11
3
11
(b)
(a)
RA
CA
3
(c)
RA
RA
CA
(a),(c), (d)
CA
(a and b)
attn. = RTI//(RTI+RA)
RB
CB
attenuation
no attn. = RA = 0
11
CC
RA
11
3
(d)
3
11
3
(e)
(b),(e)
(f)
CC
RA
CA
CA
RB
attn.without dc.
RA
big CA
CA
RB
small CA
CB
RB
attenuation
CB
(f)
attn.without dc.
Figure 11. Possible network types between microphone amplifier and transmitter.
Receiver amplifier
The receiver amplifier consists of three
stages, the first stage being an input buffer
that renders the input a high impedance.
The second stage is a gain regulated differential amplifier and the third stage a
balanced power amplifier. The power
amplifier has a differential output with low
DC- offset voltage, therefore a series
capacitor with the load is normally not
necessary. The receiver amplifier uses at
max. swing 4-6 mA peak. This current is
drawn from the +Line. The driving capacity of the power stage can be optimized by
connecting a resistor in series with the
earphone itself fig.12 b.). The gain and
frequency response is set at the input RI
with a RC-network. The receiver gain can
be regulated. The range of regulation
from the input to the output is 5 ± 2 dB (19
to 24dB). The balanced earphone amplifier
can not be loaded to full (both current and
Gain regulation.
Both the receiver and transmitter
are gain regulated (line loss
compensated).There is a fixed default
compensation on the chip that can be
adjusted or or set to constant, low gain
mode.The input impedance at the gain
regulation pin 6 is 5.5k ± 20%. The default
regulation pattern is valid when the input
is left open. Fig.13 shows a typical trans8
signal level ) single ended.The signal would
be distorded when returned to ground. A
methode is shown in fig.12 d. how to
connect a light load (5k ac. or DC wise) to
the output. It is preferred that both outputs
are loaded the same. The receiver has, as
a principal protection, two series diodes
anti parallel across its output to limit the
signal to the earphone and thus preventing an acoustical shock. A resistor in
(b)
(a)
PBL 38 573
13
(c)
PBL 38 573
+
+
+
Rx
series with the output can very well be
used to increase the protection level. Note,
that the noise in the receiver is allways
transmitter noise that has been more or
less well balanced out by the side tone
network.
The RC - network (optional) at the
output is to stabilize against the inductive
load that an earphone represents.
(d)
PBL 38 573
13
+
13
+
Rx
(C)
+
Z
Rx
-
14
14
14
Z
(C)
Z > 5k
The capacitor C is optional
Figure 12. Receiver arrangements.
mitter or receiver gain pattern versus line
length. The following will show, what to
alter to change the look of the curve.
Fig.13.
a). Adjustable with the divider R4,R5
for the transmitter and with R13 for the
receiver. (fig. 19)
b). The attack point of the regulatorcan be adjusted with the divider R15,R16
to either direction, up or down, on the line
current axis. (fig. 19)
c). The angle of elevation of the curve
is mainly set by the value of R6 but is also
adjustable with R15. If the DC-characteristics is set according to the line parameters
and a correct value for R6 is chosen the
angle is mostly correct but it can be adjusted
with R6. The adjustement will affect the
DC-characteristics as well as most of the
other parameters. This is why the DCcharacteristic is set early in the design
phase.
PBL 385 73
Battery feed
R15
Regulation:
R16
dB
R6
c.
{
48V, 2 • 200Ω
48V, 2 • 400Ω
48V, 2 • 800Ω
∞
75Ω
∞
∞
47Ω
∞
∞
75Ω
450k
b.
a.
High limit
Sweden apply for spec. application
No regulation:
Low limit
All feedings:
Set for low gain
∞
22k
47 - 75Ω
Set for high gain
75k
∞
47 - 75Ω
I
L
Figure 13. Gain regulation principle.
where no balancing has been done is in
the order of 6 - 12 dB.
To understand that the side tone is
influenced by other factors like, the
impedance of the line and the signal that
enters the ear acoustically directly from the
mouth and from the mouth through the
material in the handset. The signal that
enters the microphone from the earphone
acoustically will also influence the return
loss factor to the telephone line.
To understand that the side tone network
can be trimmed to form a veritable
”distortion analyser”, so that the distortion
that is present from the microphone, will be
the only signal entering the earphone and
this signal even being small will sound very
bad. It is better to induce some of the
fundamental frequency back by making
What is balancing the side
tone?
To understand that side tone balancing
is to counteract the signal, that is
transmitted via the microphone and transmitter to the line, returning to the earphone
via the receiver.
That presence of a strong side tone
signal is disturbing in a way that one quite
instictively lowers ones own voice level
thus lowering the signal level for the other
party. But again, if the balance is too good
(seldom the case) the earphone will feel
”dead”. In practical terms what is expected
is the same amplitude of ones own voice
in the ear as when not talking in a telephone.
The need to lower the side tone level
Telephone
set side
Line side
a.)
1
Tx
Rx
2
14
12
Z2
c.)
R7
R10
C6
R11
R8
R6
Zbal
C5
A short guidance for understanding the side tone
principle. (See fig.14.)
13
PBL 38 573
b.)
the balance less perfect at that frequency.
This is valid for a network that is trimmed to
only one frequency. It is to strive to trim the
network such that it will attenuate the fundamental and the harmonic frequencies
alike throughout the different line
combinations.
To understand that if one of the two
signals entering the balancing system
from either direction, direct from
microphone or via the line, is clipped, will
result in a very distorted signal entering the
receiver amplifier and thus the earphone.
Further , to remember that side tone is a
small signal that is the difference of two
large signals and that the amplitude of the
distortion can be up to ten times the
amplitude of the fundamental frequency.
R12
R9
Z1
Assuming the line impedance to be 600Ω.
( theorethical value )
Z1 = Line impedance
Z2 = The telephone set impedance 600Ω
Z1//Z2 = 300Ω
R6 will have a certain value 39 - 100Ω to
give the telephone a specified DCcharacteristic and overcurrent protection.
Assuming that this DC-characteristic
requires R6=60Ω, hence it will be 1/5 of
the Z1//Z2. This will in transmitting mode
result that 1/5 of the ac-signal that is on the
line appear across R6.
Figure 14. The side tone suppression principle.
9
PBL 385 73
Note that the signals at points a. and b. are
180 degrees off phase.
10 x R6 ≈ R7 + Zbal
Note #1
R7 ≈ Zbal
Note#2
The ac-signal at point c. is now 1/10 of the
signal on the line because it is further
divided by two from point b. (R7≈Zbal).
Hence 10 x R10 ≈ R11 to satisfy the
balancing criteria. R12 is to set the receiver
gain. ( can also be a volume control potentiometer).
Note #1 These values ensure that the
frequency behaviour of the transmitter is
not influenced. With the ratio 1/10 the
influence is 1 dB, and with ratio 1/20 it´s 0.5
dB.
Note #2 If the R7 is made low ohmic
compared with Zbal, it will load the latter
and result in a bad side tone
perfomannce, again if the R7 is made high
ohmic compared with Zbal will result in a
low signal to balance the side tone with and
make the balancing difficult. Making any of
the impedances unnecessary high will
make the circuit sensitive to RFI. All values
given here are approximate and serve as
starting entities only. The final trimming
of side tone network is a cut and try proposition because a part of the balance lies
in the acoustical path between the
microphone and earphone.
Reverse side tone network.
This type of side tone balancing will
help when for some reason there is a need
to make the R6 low < 47Ω and thus the
signal for balancing gets small across R6.
By placing the balancing network like shown
in fig.15 the possible signal level is 6 dB
higher than in the first case and it will also
help in case when a volume control is
added to the receiver.
PBL 38573
2
12
R10
C6
R11
+Line
R6
R12
Z bal.
C*
* To give receiver flat
frequency response
Figure 15. Reverse side tone network
with complex R11.
10
a)
Mute IMute
c)
b)
PBL
38573
5
Mute IMute
PBL
38573
5
d)
PBL
38573
11
PBL
38573
5
-L
VMute
11
-L
VMute
13
Rx
11
Mute
14
-L
12
Muting
points
The diode has to be low
voltage drop type.
Receiver mute only.
Figure 16. Mute input.
Mute function.
The circuit has a mute function at pin 5.
Sinking current from this pin will cut off the
gain in the microphone amplifier
(attenuation min. 60dB) and decrease the
gain in the receiver amplifier to reach the
confidence tone level at DTMF-dialling.
The receiver mute is ≈ 40dB down from the
unmuted value to satisfy those who keep
the handset close to the ear at dialling.
Optional conditions:
For users who keep the handset from
the ear the confidence tone level is too low.
To alter the level, a signal can be taken
from DTMF generator output to receiver
input before the capacitor C6. The added
impedance to this point will hardly disturbe
the signal condition in active speech mode.
The microphone amplifier only, can be
muted by sinking current from the output
pin 11. See fig. 4 or 9.
Figure 16 b.) If the system mute signal
is used to other tasks than muting the
speech circuit it has to be isolated. If a
diode is used it has to be a low voltage
drop type. The input at mute has to be
below 300mV. If the mute signal has
reverse polarity out of the system it can be
phase changed like in e.) In case it is
required to mute the receiver only, d.) it
can be done by shorting the receiver input
to ground before or after the input capacitor.
Shorting the input pin to ground (does not
have to be absolute ground) actuates a
mute by driving the amplifier into saturation
thus blocking the signal path and rendering a mute with high attenuation but will
cause a DC-level shift at output which in
its turn will cause a ”click ” in the earphone.
This can be softened with a slower mute
signal flank. If the second approach,
grounding before the input capacitor is
chosen, the grounding has to be low ohmic
in order to render a high attenuating mute.
Start up circuit
The circuit contains a start up device
which function is to fast charge capacitor
C1 when the circuit goes into hook- off
condition. The fast charge circuit is a
thyristor function between pins 1 and 4 that
will stop conducting when the current drain
at pin 4 is lower than ≈ 700 µA + the internal
current consumption ( about 1 mA). Care
must be taken when connecting external
load to pin 4 in order not to exeed the ≈ 700
µA limit. Should this happen, it would result
in an inoperative speech funktion. This
circuit can not retrigger before the voltage
level at C1 drops below 2V or the line
voltage is below 1V. See fig. 17.
+Line
1
Tx
PBL 38 582
DC supply
R3
2
4
C1
-Line
Figure 17. Fast startup circuit.
PBL 385 73
Power supplies DCO and V+C.
Hook
switch
(See fig.18)
1
PBL 385 73 generates its own DC
supply V+C dependent of line current with
an internal shunt regulator. This regulator
senses the line voltage VL via R3 and line
current via R6 in order to set the correct
V+C so the circuit can generate the required
DC characteristic for a given line resistance
RLine and the line feeding data of the
exchange. A decoupling capacitor is
needed between pins +C and -L. The V+C
supply changes its voltage linearly with the
line current. It can be used to feed an
electret microphone. Caution must be taken though not to drain too much current
out of this output because it will affect the
internal quick start circuit by locking itself
into active state. (max. permissible current
drain 700µA)
Care has to be taken when deciding the resistance value of R3. See fig.6.
All resistances that are applied from +Line
to ground (-Line) will be in parallel, forming
the real impedance towards the line. This
will sometimes result in, that the ohmic
value of R3 is increased in order to comply
to the impedance specification towards the
line. The speech circuit sinks ≈ 1mA into
pin 4, which means that the working voltage
for the speech function V+ will decrease
with increasing R3, thus starving in the end
the circuit of its working voltage . This
dependency is often falsely taken as a sign
+Line
PBL 385 73
1.2V
+
1-10M
-
0-470Ω
7
VDC0
DC0
15k
+
6V
15k
11
-Line
Figure 18. DC - supply for external load.
of that the circuit does not work down to the
low line current specified, but in fact it is the
working voltage at pin 4 that has become
too low. It is obvious that this problem is
also connected into what kind of DCcharacteristic is set. See fig. 8.
The circuit has further a temperature
and line current compensated DC supply
DC0 . DC0 is a voltage supply for supplying
diallers, can be used for memory back up
because it does not leak any current back
into the circuit. Typical voltage 2.4V down
to line voltage of 4.1V, in case the line
voltage is lower than 4.1V calculate ; actual
line voltage minus 1.9V. In order to prevent noise entering the line, a series
resistor and a reservoir capaciotor is
recommended in for this output.The output
current given in the specification is 2 mA..
R18
4-8M
Hook
switch
1
VDD
R1
200Ω
CMOS
DIALLER
13
PBL 385 73
C8
1µF
9
1µF
10
MIC.
MUTE
R2
200Ω
AT
AM
AR
+
C9
1N4007
7
6
DTMF
5
8
3
R17
100Ω
3
4
5
6
7
8
9
*
0
#
C3
100nF
R4
18k
R7
910Ω
R11
6.2k
R8
560Ω
+ C7
47µF
R16
R15
R5
22k
R6
75Ω
R9
11k
+4
C6
47nF
R10
62k
R13
11k
C5
100nF
C4
100µF
11
12
2
GND
2
1N4007
14
DC-supply
1
4.7-47
µF
1N4007
1N4007
Telephone
line
R14
10Ω
R3
910Ω
15V
+ C1
47µF
C2
15nF
+
R12
1k
Figure 19. Typical standard DTMF dialling telephone application.
11
PBL 385 73
Few hints how to widen
some of the specification
limits. See figure 20.
V
16
How to increase the maximum allowable
line current ? The current capability can
be extended by using more sophisticated
cooling methods but can also be done as
follows:
The maximum line current capability is
mainly set by the power rating of the
package used for the speech circuit
chip.The power used in the speech circuit
loop is divided between the IC and the
resistor R6. When a resistor RA is added
into the loop it will drain power at high line
currents from the IC but observe that it will
also limit the receiver and transmitter signal swing on the line.These have to be
recalculated at introduction of a resistor RA.
Also observe that at high line currents the
R6 has to withstand close to 1 watt.
In case a higher DC - mask is desired, it
can be increased by connecting diodes in
series with the pin 4. Each diode will rise
the level by ≈ 0.7V. The diode in opposite
direction will preserve the function of the
fast start circuit.
+
V telephone line
V line
14
12
RA
V pin 1
WRA
1
10
PBL 38573
V pin 2
Wpackage
8
2
6
4
WR6
R6
2
I
20
40
60
80
11
100
L
mA
120
The rest like in figure 4.
Figure 20. Power distribution in the speech circuit using RA.
Zline
1
9
AT
AM
+
13
PBL 385 73
AR
10
50µA
14
DC-supply
7
6
5
3
8
2
12
11
+4
1k
V
How to determin the chip
temperature at various line
currents. See figures 21 a and b.
It is important to know the chip temperature when using the component at high line
currents but especially when using the SOpackage. Obviously the mounting of the
component on the PC-board has a great
influence on the heat transfer from the chip
to the ambient. A couple rules of thumb
are; leave as much copper foil under and
around the package as possible. Use, if
possible, temperature conducting glue at
assembly of the package. Do not mount
the R6 close to the circuit and sometimes
it´s good to split the R6 into several
components in order to avoid heat concentration.
A simple methode to ensure that the chip
temperature is within safe limits at the line
current in question is to measure the the
voltage at pin 10 to ground with pin 9
grounded. This measured voltage being
over 0.5 volts indicates a safe chip
temperature but going under this value
some additional calculations are to be
made to determine the real chip temperature.
12
10kΩ
75Ω
+
47µF
R6
R6 = 10k to calibrate the temperature characteristic of the diode at ≈ 0 line current.
Figure 21a. Chip temperature measuring circuit.
mV
690
SO - package
K=2mV/°C
500
25
120
°C
Figure 21b. Chip temperature expressed as a voltage across a diode on the chip
between pins 9 and 10.
PBL 385 73
Short about Radio Frequency
Interference RFI.
HF suppression at the microphone input.
The HF-signal at the microphone input can
be seen composed as of two components.
One component being the differential
(between pins 9 and 10) and the second
related to ground at pin 14. Of these two,
the first is the most serious, entering the
amplifier directly being amplified and
detected. The second component is less
serious because it affecting both inputs
alike and most of it will be balanced out of
the amplifier. There might be the case
where the HF-signal will have such an
amplitude that the amplifier can not balance
it out. Then components must be filtered
with capacitors and maybe resistors. It is
extremely important that everything that is
done at the input is in balance, otherways
the problem might get worse instead of
better. The extreme balance requirement
a)
goes all the way to the PCB-layout. Small
unbalance signals can be corrected with
capacitors marked with*) this requiring high
precision components. See fig 22a. The
solution shown is rather expensive but
with precision components it renders good
filtering at the input. If the main problem is
the signal between the inputs, try to
increase the 1nF capacitor but make the
others procentually smaller in order to
maintain the frequency response. A more
simple solution, that is sufficient in most of
the cases is also shown in fig. 22b.
b)
c)
+
10n
10n
8
8
*
9
100Ω
Mic.
100Ω
10
*
10n
9
M
1n
1n
+
11
M
Mic.
PBL385 73
<20n
10
11
9
1µ
Mic.
1n
M
+
+
PBL385 73
10n
10n
+
PBL385 73
1µ
200470Ω
10
11
Line
Line
Line
Dynamic microphone
8
200470Ω
+
10n
<20n
Dynamic microphone (simplified)
Electret microphone
Figure 22. RFI elimination at microphone amplifier input.
HF-suppression at the receiver output.
To find out if the problem originates in the
DTMF-generator disconnect the generator and disconnect the mute input at pin 5.
If the problem is small try to connect a
capacitor from mute input to -line pin 11.
DTMF circuits are sensitive to RFI because
of their high impedance at the input pins,
especially the keyboard inputs. These
inputs are not possible to filter with large
capacitors because of the keyboard scanning pulses (1µs) that will be loaded down.
used and effective counter measure to this
kind of RFI penetration is to shield the
telephone set, at least the bottom of it, that
is closest to the main PCB board by metal
foil or by spraying the plastic casing with
metallic matter. See figure 24. This
methode does not necessarily count out
the RFI components that
are
recommended earlier.
13
+
Other paths for the HF-signal to enter
the audible system.
To shield the keyboard will some times
help. The polarity guard bridge can also
act as a rectifier and demodulator, of the
HF-signals. Connect 1nF capacitors across
each diode in the bridge. There is a
capacitor across the line C10, this is for
RFI suppression but also to stabilise the
whole system.
The cappacitor C10 shoud be connected
like in figure 24. The frequencies at which
the RFI comes through are in the region of
10-1000MHz. The resistance of the C10
will be somewhere 0.01-10Ω hence even
the shortest lenght of connector on the
PCB board or wire wil be in the same
region of resistance and thus of greatest of
importance.These actions described above
should, when applied correctly, take care
of the RFI coming in from the telephone
line. The second way for the RFI to enter
the system is to penetrate the PCB board
capacitively. The test methode is to place
a metal sheet under the telephone set to
be tested and inject the sheet with RF
signal. The most
10-100Ω
<47n
+
The problem here is of the same kind as
at the microphone amplifier input but will
be easier to solve because of the much
lower impedance and level of gain. The
solution is shown in the fig. 23. No
capacitors should be connected directly
from pins 13 or 14 to ground because of
the low output impedance, series
resistance of at least 10Ω must be used if
there is a tendency to self oscillation.
Rx
14
-
10-100Ω
<47n
PBL385 73
12
11
- Line
Figure 23. RFI elimination at receiver
amplifier output
13
PBL 385 73
Radio interference originating
from mobile phones
The problem with direct radiated RFI has
accentuated nowadays because of the
growing numbers of mobile and especially
pocket telephones. Thus it is today rather
common that a RF transmitter with output
power of several watts in form of a mobile
telephone is placed quite close to an analog telephone. There is a simultaneous
even bigger problem coming from these
portable phones of digital time-multiplex
type like the GSM. The GSM signal consists
of 900 MHz carrier that is transmitted in
short signal bursts 1/8 of time and with a
repetition frequency of slightly higher than
200 Hz. This signal will be directly radiated
to all parts in a conventional telephone set.
All unlinear elements as most of the
semiconductors will envelope detect this
signal and thus feed the 200 Hz signal with
harmonics into all points of the telephone.
The methode to counteract this problem is
the same as before with a difference that it
has to be done with much more precision.
The principle is to attenuate the HF signal
to a level where the detected 200 Hz signal
is below a disturbing level especially at
high sensitive points like at the microphone
input.
Following aspects ought to be considered:
1).
2).
3).
4).
5).
6).
14
Do not make any points in the
circuitry more high impedive than
necessary.
Keep all cables, wires and tracks
on PC-board as short as possible.
Decouple all sensitive points to an
internal ground with capacitors
especially the microphone amplifier
input.
To include series elements like
resistors and inductors in all long
wires or cables that could act as
aerials. For ex. microphone cable,
earphone cable, cable to the
telephone network, mute wire and
cable to the keypad.
Comprehend that it is a question of
a HF- design,so that all used
decoupling components are well
suited to the frequencies at hand.
(up to several GHz).
HF- design includes also that
tracks on the PC-board act as
inductors and therefore it is the
more important that the decoupling
capacitors are placed directly
between the actual points and not
V and I protection
SIOV
5 - 10Ω
Line
in
The
electronic
circuitry
C10
Plastic
enclosure
Metallic shield, sprayed or foil
RF radiating measuring sheet.
RF-gen.
Figure 24. How to measure the RFI pickup.
via tracks on the board (See fig. 31).
Balanced points like a differential
microphone input may have to be
decoupled differentially between
the inputs and ”common mode” to
common ground each input
separately.
8). A virtual ground may have to be
created into which all outgoing
cables are decoupled in order to
bypass the RF- signal. See fig. 31.
9). Think that even overvoltage and
overcurrent protectors can be
acting as HF detectors.
10). Shields that are connected to the
internal ground can be of help.
11). Control that no already detected
signals from for ex. dialler enter the
speech circuit via the mute
function.
12). Try to reach a high packing density
on the PC-board.
7).
13). Connect components as close to
the IC as possible. Connect
especially decoupling capacitors
close to the ground pin of the IC.
The terminal circuits from Ericsson
Components are manufactured in IC
processes with large internal capacitors
on the chip to counteract RFI disturbanses
in every possible way.
The simplest method to test the
susceptibility of an apparatus to RFI is to
take a portable phone of an actual type
and move it transmitting across the phone,
cables and handset. Measure the signal
at earphone output as well as on the line.
Finally; to design an ordinary analog
telephone to fullfil todays requirements is
not a low frequency but a high frequency
task.
Not like
this
Like this
Figure 25. RFI elimination at PCB layout level.
Microphone
Earphone
Line
Virtual ground
Resistors
or
inductors
Figure 26. RFI elimination in the wiring.
Common ground of the telephone
PBL 385 73
75V
Transorber
a
HMPSA 92M
10Ω
HK
470K
470K
12V
0V
0V
10K
3.9M
4x1N4004
MPSA 42
220K
b
1N
4148
0V
680K
+ 0.1µF
1
2.2M
0V
620Ω
BC550
14
VDD
10 HKS
DP 11
9
MIC
1nF
180K
0.33µF+
0V
33pF
6 VSS
CMOS
DIALLER
91414
620Ω
47nF
100nF
SW1
B/M 5
8 XT
0V
C1
1
C3
3
6
5
8
3
12
2
11
+4
T
0V
C2
2
14
DC-supply
7
100Ω
0V
47nF
R15
DTMF 12
7 XT
REC
AR
10
MODE 13
3.58MHz
AT
AM
1nF
MUTE 9
P
33pF
13
PBL 385 73
47nF
18K
220nF
R1 R2 R3 R4
15 16 17 18
430Ω
68K
5.6K
430Ω
200R
3K6
680pF
1
2
3
4
5
6
7
8
9
750Ω
R16
+
3.3nF
47K
15nF
47Ω
10k
+ 100µ
330nF
100µ
6V
470nF
16V
0V
100µ
To increase the dc characteristic across the telephone set,
one or two diodes can be connected in pin 4. The third is
to maintain the quick start feature.
+ 16V
*
0
#
1k
Figure 27. Typical standard telephone application. ( Testboard TB 132 T )
Information given in this data sheet is believed to be
accurate and reliable. However no responsibility is
assumed for the consequences of its use nor for any
infringement of patents or other rights of third parties
which may result from its use. No license is granted by
implication or otherwise under any patent or patent
rights of Ericsson Components. These products are
sold only according to Ericsson Components' general
conditions of sale, unless otherwise confirmed in writing.
Specifications subject to change without
notice.
1522-PBL385 73
© Ericsson Components AB
June 1999
Ordering Information:
Ericsson Components AB
S-164 81 Kista-Stockholm, Sweden
Telephone: +46 8 757 50 00
Package
Temp. Range
Part No.
14-pin Plastic DIP
14-pin Plastic SO
-40 to +70°C
-40 to +70°C
PBL 385 73/1NS
PBL 385 73/1SOS
15