16-Bit, 4-Channel/8-Channel, 250 kSPS PulSAR ADC AD7682/AD7689 FEATURES APPLICATIONS Battery-powered equipment Medical instruments: ECG/EKG Mobile communications: GPS Personal digital assistants Power line monitoring Data acquisition Seismic data acquisition systems Instrumentation Process control FUNCTIONAL BLOCK DIAGRAM 0.5V TO 4.096V 0.1µF REFIN REF BAND GAP REF VDD 1.8V VIO TO VDD AD7682/ AD7689 TEMP SENSOR IN0 IN1 IN2 IN3 IN4 IN5 IN6 IN7 2.7V TO 5V 0.5V TO VDD 22µF CNV 16-BIT SAR ADC MUX ONE-POLE LPF SPI SERIAL INTERFACE SCK SDO DIN SEQUENCER COM 07353-001 16-bit resolution with no missing codes 4-channel (AD7682)/8-channel (AD7689) multiplexer with choice of inputs Unipolar single ended Differential (GND sense) Pseudobipolar Throughput: 250 kSPS INL: ±0.5 LSB typical, ±1.5 LSB maximum (±23 ppm or FSR) Dynamic range: 93.8 dB SINAD: 92.5 dB @ 20 kHz THD: −100 dB @ 20 kHz Analog input range: 0 V to VREF with VREF up to VDD Multiple reference types Internal selectable 2.5 V or 4.096 V External buffered (up to 4.096 V) External (up to VDD) Internal temperature sensor Channel sequencer, selectable 1-pole filter, busy indicator No pipeline delay, SAR architecture Single-supply 2.7 V to 5.5 V operation with 1.8 V to 5 V logic interface Serial interface compatible with SPI, MICROWIRE, QSPI, and DSP Power dissipation 3.5 mW @ 2.5 V/200 kSPS 12 mW @ 5 V/250 kSPS Standby current: 50 nA 20-lead 4 mm × 4 mm LFCSP package GND Figure 1. Table 1. Multichannel 14-/16-Bit PulSAR® ADC Type 14-Bit 16-Bit 16-Bit Channels 8 4 8 250 kSPS AD7949 AD7682 AD7689 500 kSPS AD7699 ADC Driver ADA4841-x ADA4841-x ADA4841-x GENERAL DESCRIPTION The AD7682/AD7689 are 4-channel/8-channel, 16-bit, charge redistribution successive approximation register (SAR) analogto-digital converters (ADCs) that operate from a single power supply, VDD. The AD7682/AD7689 contain all components for use in a multichannel, low power data acquisition system, including a true 16-bit SAR ADC with no missing codes; a 4-channel (AD7682) or 8-channel (AD7689), low crosstalk multiplexer useful for configuring the inputs as single ended (with or without ground sense), differential, or bipolar; an internal low drift reference (selectable 2.5 V or 4.096 V) and buffer; a temperature sensor; a selectable one-pole filter; and a sequencer that is useful when channels are continuously scanned in order. The AD7682/AD7689 use a simple SPI interface for writing to the configuration register and receiving conversion results. The SPI interface uses a separate supply, VIO, which is set to the host logic level. Power dissipation scales with throughput. Each AD7682/AD7689 is housed in a tiny 20-lead LFCSP with operation specified from −40°C to +85°C. Rev. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2008 Analog Devices, Inc. All rights reserved. AD7682/AD7689 TABLE OF CONTENTS Features .............................................................................................. 1 Typical Connection Diagrams.................................................. 17 Applications....................................................................................... 1 Analog Inputs ............................................................................. 18 Functional Block Diagram .............................................................. 1 Driver Amplifier Choice ........................................................... 19 General Description ......................................................................... 1 Voltage Reference Output/Input .............................................. 20 Revision History ............................................................................... 2 Power Supply............................................................................... 21 Specifications..................................................................................... 3 Supplying the ADC from the Reference.................................. 21 Timing Specifications....................................................................... 6 Digital Interface .............................................................................. 22 Absolute Maximum Ratings............................................................ 8 Configuration Register, CFG .................................................... 22 ESD Caution.................................................................................. 8 Pin Configurations and Function Descriptions ........................... 9 Read/Write Spanning Conversion Without a Busy Indicator............................................................................. 24 Typical Performance Characteristics ........................................... 11 Read/Write Spanning Conversion with a Busy Indicator..... 25 Terminology .................................................................................... 14 Application Hints ........................................................................... 26 Theory of Operation ...................................................................... 15 Layout .......................................................................................... 26 Overview...................................................................................... 15 Evaluating AD7682/AD7689 Performance ............................ 26 Converter Operation.................................................................. 15 Outline Dimensions ....................................................................... 27 Transfer Functions...................................................................... 16 Ordering Guide .......................................................................... 27 REVISION HISTORY 5/08—Revision 0: Initial Version Rev. 0 | Page 2 of 28 AD7682/AD7689 SPECIFICATIONS VDD = 2.5 V to 5.5 V, VIO = 2.3 V to VDD, VREF = VDD, all specifications TMIN to TMAX, unless otherwise noted. Table 2. Parameter RESOLUTION ANALOG INPUT Voltage Range Absolute Input Voltage Analog Input CMRR Leakage Current at 25°C Input Impedance 1 THROUGHPUT Conversion Rate Full Bandwidth 2 ¼ Bandwidth2 Transient Response ACCURACY No Missing Codes Integral Linearity Error Differential Linearity Error Transition Noise Gain Error 4 Gain Error Match Gain Error Temperature Drift Offset Error4 Offset Error Match Offset Error Temperature Drift Power Supply Sensitivity AD7682B/AD7689B Typ Max Conditions/Comments Min 16 Unipolar mode Bipolar mode Positive input, unipolar and bipolar modes Negative or COM input, unipolar mode Negative or COM input, bipolar mode fIN = 250 kHz Acquisition phase 0 −VREF/2 −0.1 −0.1 VREF/2 − 0.1 VDD = 4.5 V to 5.5 V VDD = 2.7 V to 4.5 V VDD = 2.3 V to 2.7 V VDD = 4.5 V to 5.5 V VDD = 2.7 V to 4.5 V VDD = 2.3 V to 2.7 V Full-scale step, full bandwidth Full-scale step, ¼ bandwidth 0 0 0 0 0 0 16 −1.5 −1 REF = VDD = 5 V −30 −2 −2 VDD = 5 V ± 5% Rev. 0 | Page 3 of 28 VREF/2 68 1 +VREF +VREF/2 VREF + 0.1 +0.1 VREF/2 + 0.1 V V dB nA 250 200 190 60 50 47 1.8 14.5 ±0.5 ±0.25 0.5 ±2 ±0.5 ±1 ±2 ±0.5 ±1 ±1.5 Unit Bits +1.5 +1.5 +30 +2 +2 kSPS kSPS kSPS kSPS kSPS kSPS μs μs Bits LSB 3 LSB LSB LSB LSB ppm/°C LSB LSB ppm/°C LSB AD7682/AD7689 Parameter AC ACCURACY 5 Dynamic Range Signal-to-Noise SINAD Total Harmonic Distortion Spurious-Free Dynamic Range Channel-to-Channel Crosstalk SAMPLING DYNAMICS −3 dB Input Bandwidth Aperture Delay INTERNAL REFERENCE REF Output Voltage REFIN Output Voltage 7 REF Output Current Temperature Drift Line Regulation Long-Term Drift Turn-On Settling Time EXTERNAL REFERENCE Voltage Range Current Drain TEMPERATURE SENSOR Output Voltage 8 Temperature Sensitivity DIGITAL INPUTS Logic Levels VIL VIH IIL IIH DIGITAL OUTPUTS Data Format 9 Pipeline Delay 10 VOL VOH Conditions/Comments Min AD7682B/AD7689B Typ Max Unit 93.8 93.5 92.3 88.8 92.5 33.5 dB 6 dB dB dB dB dB 90 87 91 88.4 −100 110 −125 dB dB dB dB dB Selectable VDD = 5 V 0.425 1.7 2.5 MHz ns 2.5 V, @ 25°C 4.096 V, @ 25°C 2.5 V, @ 25°C 4.096 V, @ 25°C 2.490 4.086 2.500 4.096 1.2 2.3 ±300 ±10 ±15 50 5 fIN = 20 kHz, VREF = 5 V fIN = 20 kHz, VREF = 4.096 V internal REF fIN = 20 kHz, VREF = 2.5 V internal REF fIN = 20 kHz, VREF = 5 V fIN = 20 kHz, VREF = 5 V −60 dB input fIN = 20 kHz, VREF = 4.096 V internal REF fIN = 20 kHz, VREF = 2.5 V internal REF fIN = 20 kHz fIN = 20 kHz fIN = 100 kHz on adjacent channel(s) 92.5 91 87.5 91 VDD = 5 V ± 5% 1000 hours CREF = 10 μF REF input REFIN input (buffered) 250 kSPS, REF = 5 V 2.510 4.106 V V V V μA ppm/°C ppm/V ppm ms VDD + 0.3 VDD − 0.2 50 V V μA 283 1 mV mV/°C 0.5 0.5 @ 25°C −0.3 0.7 × VIO −1 −1 ISINK = +500 μA ISOURCE = −500 μA VIO − 0.3 Rev. 0 | Page 4 of 28 +0.3 × VIO VIO + 0.3 +1 +1 V V μA μA 0.4 V V AD7682/AD7689 Parameter POWER SUPPLIES VDD VIO Standby Current 11, 12 Power Dissipation Energy per Conversion TEMPERATURE RANGE 13 Specified Performance Conditions/Comments Min Specified performance Specified performance Operating range VDD and VIO = 5 V, @ 25°C VDD = 2.5 V, 100 SPS throughput VDD = 2.5 V, 100 kSPS throughput VDD = 2.5 V, 200 kSPS throughput VDD = 5 V , 250 kSPS throughput VDD = 5 V, 250 kSPS throughput with internal reference 2.3 2.3 1.8 TMIN to TMAX −40 1 AD7682B/AD7689B Typ Max 5.5 VDD + 0.3 VDD + 0.3 50 1.7 1.75 3.5 12.5 15.5 50 2.1 4.1 15.9 19.2 +85 Unit V V V nA μW mW mW mW mW nJ °C See the Analog Inputs section. The bandwidth is set with the configuration register 3 LSB means least significant bit. With the 5 V input range, one LSB is 76.3 μV. 4 See the Terminology section. These specifications include full temperature range variation but not the error contribution from the external reference. 5 With VDD = 5 V, unless otherwise noted. 6 All specifications expressed in decibels are referred to a full-scale input FSR and tested with an input signal at 0.5 dB below full scale, unless otherwise specified. 7 This is the output from the internal band gap. 8 The output voltage is internal and present on a dedicated multiplexer input. 9 Unipolar mode: serial 16-bit straight binary. Bipolar mode: serial 16-bit twos complement. 10 Conversion results available immediately after completed conversion. 11 With all digital inputs forced to VIO or GND as required. 12 During acquisition phase. 13 Contact an Analog Devices, Inc., sales representative for the extended temperature range. 2 Rev. 0 | Page 5 of 28 AD7682/AD7689 TIMING SPECIFICATIONS VDD = 4.5 V to 5.5 V, VIO = 2.3 V to VDD, all specifications TMIN to TMAX, unless otherwise noted. Table 3. 1 Parameter Conversion Time: CNV Rising Edge to Data Available Acquisition Time Time Between Conversions CNV Pulse Width Data Write/Read During Conversion SCK Period SCK Low Time SCK High Time SCK Falling Edge to Data Remains Valid SCK Falling Edge to Data Valid Delay VIO Above 4.5 V VIO Above 3 V VIO Above 2.7 V VIO Above 2.3 V CNV Low to SDO D15 MSB Valid VIO Above 4.5 V VIO Above 3 V VIO Above 2.7 V VIO Above 2.3 V CNV High or Last SCK Falling Edge to SDO High Impedance CNV Low to SCK Rising Edge DIN Valid Setup Time from SCK Falling Edge DIN Valid Hold Time from SCK Falling Edge 1 Symbol tCONV tACQ tCYC tCNVH tDATA tSCK tSCKL tSCKH tHSDO tDSDO Min Typ Max 2.2 1.8 4 10 1.4 15 7 7 4 Unit μs μs μs ns μs ns ns ns ns 16 17 18 19 ns ns ns ns 15 17 18 22 25 ns ns ns ns ns ns ns ns tEN tDIS tCLSCK tSDIN tHDIN See Figure 2 and Figure 3 for load conditions. Rev. 0 | Page 6 of 28 10 4 4 AD7682/AD7689 VDD = 2.5 V to 4.5 V, VIO = 2.3 V to VDD, all specifications TMIN to TMAX, unless otherwise noted. Table 4. 1 Parameter Conversion Time: CNV Rising Edge to Data Available VDD = 2.7 V to 4.5 V VDD = 2.3 V to 2.7 V Acquisition Time Time Between Conversions VDD = 2.7 V to 4.5 V VDD = 2.3 V to 2.7 V CNV Pulse Width Data Write/Read During Conversion SCK Period SCK Low Time SCK High Time SCK Falling Edge to Data Remains Valid SCK Falling Edge to Data Valid Delay VIO Above 3 V VIO Above 2.7 V VIO Above 2.3 V CNV Low to SDO D15 MSB Valid VIO Above 3 V VIO Above 2.7 V VIO Above 2.3 V CNV High or Last SCK Falling Edge to SDO High Impedance CNV Low to SCK Rising Edge SDI Valid Setup Time from SCK Falling Edge SDI Valid Hold Time from SCK Falling Edge Min tCONV tCONV tACQ tCYC tCYC tCNVH tDATA tSCK tSCKL tSCKH tHSDO tDSDO Max Unit 3.2 3.4 μs μs μs μs μs μs ns μs ns ns ns ns 5 5.2 10 1.4 25 12 12 5 24 30 37 ns ns ns 21 27 35 50 ns ns ns ns ns ns ns tEN tDIS tCLSCK tSDIN tHDIN 10 5 5 See Figure 2 and Figure 3 for load conditions. 500µA IOL 1.4V TO SDO CL 50pF 500µA IOH Figure 2. Load Circuit for Digital Interface Timing 70% VIO 30% VIO tDELAY 2V OR VIO – 0.5V1 0.8V OR 0.5V2 2V OR VIO – 0.5V1 0.8V OR 0.5V2 2V IF VIO ABOVE 2.5V, VIO – 0.5V IF VIO BELOW 2.5V. 0.8V IF VIO ABOVE 2.5V, 0.5V IF VIO BELOW 2.5V. Figure 3. Voltage Levels for Timing Rev. 0 | Page 7 of 28 07353-003 tDELAY 1 2 Typ 1.8 07353-002 1 Symbol AD7682/AD7689 ABSOLUTE MAXIMUM RATINGS Table 5. Parameter Analog Inputs INx, 1 COM1 REF, REFIN Supply Voltages VDD, VIO to GND VDD to VIO DIN, CNV, SCK to GND 2 SDO to GND Storage Temperature Range Junction Temperature θJA Thermal Impedance (LFCSP) θJC Thermal Impedance (LFCSP) 1 2 Rating GND − 0.3 V to VDD + 0.3 V or VDD ± 130 mA GND − 0.3 V to VDD + 0.3 V −0.3 V to +7 V ±7 V −0.3 V to VIO + 0.3 V −0.3 V to VIO + 0.3 V −65°C to +150°C 150°C 47.6°C/W 4.4°C/W Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ESD CAUTION See the Analog Inputs section. CNV must be low at power up. See the Power Supply section. Rev. 0 | Page 8 of 28 AD7682/AD7689 20 19 18 17 16 20 19 18 17 16 VDD IN3 IN2 IN1 IN0 VDD NC IN1 NC IN0 PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS AD7682 TOP VIEW (Not to Scale) 15 14 13 12 11 VIO SDO SCK DIN CNV VDD REF REFIN GND GND 07353-004 NC = NO CONNECT 1 2 3 4 5 PIN 1 INDICATOR AD7689 TOP VIEW (Not to Scale) 15 14 13 12 11 VIO SDO SCK DIN CNV 07353-005 PIN 1 INDICATOR IN4 6 IN5 7 IN6 8 IN7 9 COM 10 1 2 3 4 5 NC 6 IN2 7 NC 8 IN3 9 COM 10 VDD REF REFIN GND GND Figure 5. AD7689 20-Lead LFCSP (QFN) Pin Configuration Figure 4. AD7682 20-Lead LFCSP (QFN) Pin Configuration Table 6. Pin Function Descriptions Pin No. 1, 20 AD7682 Mnemonic VDD AD7689 Mnemonic VDD Type 1 P 2 REF REF AI/O 3 REFIN REFIN AI/O 4, 5 6 GND NC GND IN4 P AI 7 IN2 IN5 AI 8 NC IN6 AI 9 IN3 IN7 AI 10 COM COM AI 11 CNV CNV DI 12 DIN DIN DI 13 SCK SCK DI Description Power Supply. Nominally 2.5 V to 5.5 V when using an external reference and decoupled with 10 μF and 100 nF capacitors. When using the internal reference for 2.5 V output, the minimum should be 2.7 V. When using the internal reference for 4.096 V output, the minimum should be 4.5 V. Reference Input/Output. See the Voltage Reference Output/Input section. When the internal reference is enabled, this pin produces a selectable system reference = 2.5 V or 4.096 V. When the internal reference is disabled and the buffer is enabled, REF produces a buffered version of the voltage present on the REFIN pin (VDD – 0.3 V maximum) useful when using low cost, low power references. For improved drift performance, connect a precision reference to REF (0.5 V to VDD). For any reference method, this pin needs decoupling with an external 10 μF capacitor connected as close to REF as possible. See the Reference Decoupling section. Internal Reference Output/Reference Buffer Input. See the Voltage Reference Output/Input section. When using the internal reference, the internal unbuffered reference voltage is present and needs decoupling with a 0.1μF capacitor. When using the internal reference buffer, apply a source between 0.5 V and 4.096 V that is buffered to the REF pin as described above. Power Supply Ground. AD7682: No connection. AD7689: Analog Input Channel 4. AD7682: Analog Input Channel 2. AD7689: Analog Input Channel 5. AD7682: No connection. AD7689: Analog Input Channel 6. AD7682: Analog Input Channel 3. AD7689: Analog Input Channel 7. Common Channel Input. All channels [3:0] or [7:0] can be referenced to a common mode point of 0 V or VREF/2 V. Convert Input. On the rising edge, CNV initiates the conversion. During conversion, if CNV is held high, the busy indictor is enabled. Data Input. This input is used for writing to the 14-bit configuration register. The configuration register can be written to during and after conversion. Serial Data Clock Input. This input is used to clock out the data on ADO and clock in data on DIN in an MSB first fashion. Rev. 0 | Page 9 of 28 AD7682/AD7689 Pin No. 14 AD7682 Mnemonic SDO AD7689 Mnemonic SDO Type 1 DO 15 VIO VIO P 16 17 IN0 NC IN0 IN1 AI AI 18 IN1 IN2 AI 19 NC IN3 AI Description Serial Data Output. The conversion result is output on this pin synchronized to SCK. In unipolar modes, conversion results are straight binary; in bipolar modes, conversion results are twos complement. Input/Output Interface Digital Power. Nominally at the same supply as the host interface (1.8 V, 2.5 V, 3 V, or 5 V). Analog Input Channel 0. AD7682: No connection. AD7689: Analog Input Channel 1. AD7682: Analog Input Channel 1. AD7689: Analog Input Channel 2. AD7682: No connection. AD7689: Analog Input Channel 3. 1 AI = analog input, AI/O = analog input/output, DI = digital input, DO = digital output, and P = power. Rev. 0 | Page 10 of 28 AD7682/AD7689 TYPICAL PERFORMANCE CHARACTERISTICS VDD = 2.5 V to 5.5 V, VREF = 2.5 V to 5 V, VIO = 2.3 V to VDD 1.5 1.5 1.0 1.0 0.5 DNL (LSB) INL (LSB) 0.5 0 0 –0.5 0 16,384 32,768 49,152 65,536 CODES –1.0 07353-009 –1.5 0 16,384 32,768 49,152 07353-006 –0.5 –1.0 65,536 CODES Figure 6. Integral Nonlinearity vs. Code, VREF = VDD = 5 V Figure 9. Differential Nonlinearity vs. Code, VREF = VDD = 5 V 200k 160k σ = 0.50 VREF = VDD = 5V 180k σ = 0.78 VREF = VDD = 2.5V 135,207 140k 160k 100k 120k COUNTS COUNTS 120k 135,326 124,689 140k 100k 80k 80k 63,257 60k 51,778 60k 40k 20k 20k 0 487 619 0 7FFA 7FFB 7FFC 7FFD 7FFE 7FFF 8000 0 0 8001 8002 CODE IN HEX 6649 1 0 07353-007 0 0 7FFB 7FFC 7FFD 7FFE 7FFF 8000 60 1 8002 8003 Figure 10. Histogram of a DC Input at Code Center 0 0 VREF = VDD = 5V fS= 250kSPS fIN = 19.9kHz SNR = 92.9dB SINAD = 92.4dB THD = –102dB SFDR = 103dB SECOND HARMONIC = –111dB THIRD HARMONIC = –104dB –40 –60 –80 VREF = VDD = 2.5V fs= 200kSPS fIN = 19.9kHz SNR = 88.0dB SINAD = 87.0dB THD = –89dB SFDR = 89dB SECOND HARMONIC = –105dB THIRD HARMONIC = –90dB –20 AMPLITUDE (dB of Full-Scale) –20 –100 –120 –140 –40 –60 –80 –100 –120 –140 –160 0 25 50 75 100 FREQUENCY (kHz) 125 Figure 8. 20 kHz FFT, VREF = VDD = 5 V –180 0 25 50 75 FREQUENCY (kHz) Figure 11. 20 kHz FFT, VREF = VDD = 2.5 V Rev. 0 | Page 11 of 28 100 07353-011 –160 07353-008 AMPLITUDE (dB of Full-Scale) 4090 8001 CODE IN HEX Figure 7. Histogram of a DC Input at Code Center –180 78 07353-010 40k 100 100 95 95 90 90 85 85 80 75 60 0 50 VDD = VREF VDD = VREF VDD = VREF VDD = VREF 70 65 100 150 200 FREQUENCY (kHz) 60 0 50 17.0 SNR @ 2kHz SINAD @ 2kHz SNR @ 20kHz SINAD @ 20kHz ENOB @ 2kHz ENOB @ 20kHz 16.5 200 130 –60 125 –65 120 16.0 88 15.0 86 14.5 –75 110 SFDR (dB) 15.5 –70 SFDR = 2kHz 115 ENOB (Bits) 90 –80 105 –85 100 95 SFDR = 20kHz –90 THD = 20kHz –95 90 84 14.0 82 –100 85 –105 THD = 2kHz 80 13.5 –110 75 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.5 5.0 13.0 REFERENCE VOLTAGE (V) –115 70 1.0 07353-013 80 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 –120 5.5 5.0 REFERENCE VOLTAGE (V) Figure 13. SNR, SINAD, and ENOB vs. Reference Voltage Figure 16. SFDR and THD vs. Reference Voltage 96 –90 fIN = 20kHz fIN = 20kHz VDD = VREF = 5V 94 –95 92 THD (dB) VDD = VREF = 5V 90 88 –100 VDD = VREF = 2.5V VDD = VREF = 2.5V –105 84 –55 –35 –15 5 25 45 65 TEMPERATURE (°C) 85 105 125 –110 –55 –35 –15 5 25 45 65 TEMPERATURE (°C) Figure 14. SNR vs. Temperature Figure 17. THD vs. Temperature Rev. 0 | Page 12 of 28 85 105 125 07353-017 86 07353-014 SNR (dB) SNR, SINAD (dB) 150 Figure 15. SINAD vs. Frequency 96 92 100 FREQUENCY (kHz) Figure 12. SNR vs. Frequency 94 = 5V, –0.5dB = 5V, –10dB = 2.5V, –0.5dB = 2.5V, –10dB THD (dB) 65 75 07353-016 70 = 5V, –0.5dB = 5V, –10dB = 2.5V, –0.5dB = 2.5V, –10dB 07353-041 VDD = VREF VDD = VREF VDD = VREF VDD = VREF 80 07353-012 SINAD (dB) SNR (dB) AD7682/AD7689 AD7682/AD7689 3000 100 2.5V INTERNAL REF 4.096V INTERNAL REF INTERNAL BUFFER, TEMP ON INTERNAL BUFFER, TEMP OFF EXTERNAL REF, TEMP ON EXTERNAL REF, TEMP OFF VIO VDD CURRENT (µA) 2500 THD (dB) –80 –90 –100 –120 0 50 = 5V, –0.5dB = 2.5V, –0.5dB = 2.5V, –10dB = 5V, –10dB 100 150 200 FREQUENCY (kHz) 94 1750 50 1500 40 1250 30 3.0 3.5 4.0 4.5 20 5.5 5.0 VDD SUPPLY (V) Figure 21. Operating Currents vs. Supply 3000 fIN = 20kHz 180 fS = 200kSPS 2750 160 VDD = VREF = 5V 93 VDD = 5V, INTERNAL 4.096V REF 2500 VDD CURRENT (µA) 92 SNR (dB) 70 60 Figure 18. THD vs. Frequency 95 80 2000 1000 2.5 07353-015 VDD = VREF VDD = VREF VDD = VREF VDD = VREF –110 2250 90 VIO CURRENT (µA) 2750 –70 fS = 200kSPS 07353-021 –60 91 90 89 VDD = VREF = 2.5V 88 140 2250 120 VDD = 5V, EXTERNAL REF 2000 100 1750 80 1500 60 VDD = 2.5, EXTERNAL REF 87 1250 86 40 –8 –6 –4 –2 1000 –55 07353-018 85 –10 0 INPUT LEVEL (dB) 25 45 65 85 105 20 125 25 VDD = 2.5V, 85°C 20 TDSDO DELAY (ns) 1 0 –1 UNIPOLAR ZERO UNIPOLAR GAIN BIPOLAR ZERO BIPOLAR GAIN –35 –15 5 15 VDD = 2.5V, 25°C 10 VDD = 5V, 85°C VDD = 5V, 25°C 5 VDD = 3.3V, 85°C 07353-023 2 VDD = 3.3V, 25°C 25 45 65 85 105 TEMPERATURE (°C) 125 0 07353-020 OFFSET ERROR AND GAIN ERROR (LSB) 5 Figure 22. Operating Currents vs. Temperature 3 –3 –55 –15 TEMPERATURE (°C) Figure 19. SNR vs. Input Level –2 –35 0 20 40 60 80 SDO CAPACITIVE LOAD (pF) 100 Figure 23. tDSDO Delay vs. SDO Capacitance Load and Supply Figure 20. Offset and Gain Errors vs. Temperature Rev. 0 | Page 13 of 28 120 07353-022 VIO AD7682/AD7689 TERMINOLOGY Least Significant Bit (LSB) The LSB is the smallest increment that can be represented by a converter. For an analog-to-digital converter with N bits of resolution, the LSB expressed in volts is LSB (V) = VREF 2N Integral Nonlinearity Error (INL) INL refers to the deviation of each individual code from a line drawn from negative full scale through positive full scale. The point used as negative full scale occurs ½ LSB before the first code transition. Positive full scale is defined as a level 1½ LSB beyond the last code transition. The deviation is measured from the middle of each code to the true straight line (see Figure 25). Differential Nonlinearity Error (DNL) In an ideal ADC, code transitions are 1 LSB apart. DNL is the maximum deviation from this ideal value. It is often specified in terms of resolution for which no missing codes are guaranteed. Offset Error The first transition should occur at a level ½ LSB above analog ground. The unipolar offset error is the deviation of the actual transition from that point. Signal-to-Noise Ratio (SNR) SNR is the ratio of the rms value of the actual input signal to the rms sum of all other spectral components below the Nyquist frequency, excluding harmonics and dc. The value for SNR is expressed in decibels. Signal-to-(Noise + Distortion) Ratio (SINAD) SINAD is the ratio of the rms value of the actual input signal to the rms sum of all other spectral components below the Nyquist frequency, including harmonics but excluding dc. The value for SINAD is expressed in decibels. Total Harmonic Distortion (THD) THD is the ratio of the rms sum of the first five harmonic components to the rms value of a full-scale input signal and is expressed in decibels. Spurious-Free Dynamic Range (SFDR) SFDR is the difference, in decibels, between the rms amplitude of the input signal and the peak spurious signal. Effective Number of Bits (ENOB) ENOB is a measurement of the resolution with a sine wave input. It is related to SINAD by the formula ENOB = (SINADdB − 1.76)/6.02 and is expressed in bits. Gain Error The last transition (from 111 … 10 to 111 … 11) should occur for an analog voltage 1½ LSB below the nominal full scale. The gain error is the deviation in LSB (or percentage of full-scale range) of the actual level of the last transition from the ideal level after the offset error is adjusted out. Closely related is the full-scale error (also in LSB or percentage of full-scale range), which includes the effects of the offset error. Aperture Delay Aperture delay is the measure of the acquisition performance. It is the time between the rising edge of the CNV input and the point at which the input signal is held for a conversion. Channel-to-Channel Crosstalk Channel-to-channel crosstalk is a measure of the level of crosstalk between any two adjacent channels. It is measured by applying a dc to the channel under test and applying a full-scale, 100 kHz sine wave signal to the adjacent channel(s). The crosstalk is the amount of signal that leaks into the test channel and is expressed in decibels. Reference Voltage Temperature Coefficient Reference voltage temperature coefficient is derived from the typical shift of output voltage at 25°C on a sample of parts at the maximum and minimum reference output voltage (VREF) measured at TMIN, T (25°C), and TMAX. It is expressed in ppm/°C as Transient Response Transient response is the time required for the ADC to accurately acquire its input after a full-scale step function is applied. Dynamic Range Dynamic range is the ratio of the rms value of the full scale to the total rms noise measured with the inputs shorted together. The value for dynamic range is expressed in decibels. TCVREF (ppm/ °C ) = VREF ( Max ) – VREF ( Min) VREF (25°C ) × (TMAX – TMIN ) × 10 6 where: VREF (Max) = maximum VREF at TMIN, T (25°C), or TMAX. VREF (Min) = minimum VREF at TMIN, T (25°C), or TMAX. VREF (25°C) = VREF at 25°C. TMAX = +85°C. TMIN = –40°C. Rev. 0 | Page 14 of 28 AD7682/AD7689 THEORY OF OPERATION INx+ SWITCHES CONTROL MSB 32,768C 16,384C LSB 4C 2C C SW+ C BUSY REF COMP GND 32,768C 16,384C 4C 2C C CONTROL LOGIC OUTPUT CODE C MSB LSB SW– 07353-026 CNV INx– OR COM Figure 24. ADC Simplified Schematic OVERVIEW CONVERTER OPERATION The AD7682/AD7689 are 4-channel/8-channel, 16-bit, charge redistribution successive approximation register (SAR) analog-todigital converters (ADCs). These devices are capable of converting 250,000 samples per second (250 kSPS) and power down between conversions. For example, when operating with an external reference at 1 kSPS, they consumes 17 μW typically, ideal for battery-powered applications. The AD7682/AD7689 are successive approximation ADCs based on a charge redistribution DAC. Figure 24 shows the simplified schematic of the ADC. The capacitive DAC consists of two identical arrays of 16 binary-weighted capacitors, which are connected to the two comparator inputs. The AD7682/AD7689 contain all of the components for use in a multichannel, low power data acquisition system, including • • • • • • 16-bit SAR ADC with no missing codes 4-channel/8-channel, low crosstalk multiplexer Internal low drift reference and buffer Temperature sensor Selectable one-pole filter Channel sequencer These components are configured through an SPI-compatible, 14-bit register. Conversion results, also SPI compatible, can be read after or during conversions with the option for reading back the current configuration. The AD7682/AD7689 provide the user with an on-chip trackand-hold and do not exhibit pipeline delay or latency. The AD7682/AD7689 are specified from 2.3 V to 5.5 V and can be interfaced to any 1.8 V to 5 V digital logic family. They are housed in a 20-lead, 4 mm × 4 mm LFCSP that combines space savings and allows flexible configurations. They are pin-for-pin compatible with the 16-bit AD7699 and 14-bit AD7949. During the acquisition phase, terminals of the array tied to the comparator input are connected to GND via SW+ and SW−. All independent switches are connected to the analog inputs. Thus, the capacitor arrays are used as sampling capacitors and acquire the analog signal on the INx+ and INx− (or COM) inputs. When the acquisition phase is complete and the CNV input goes high, a conversion phase is initiated. When the conversion phase begins, SW+ and SW− are opened first. The two capacitor arrays are then disconnected from the inputs and connected to the GND input. Therefore, the differential voltage between the INx+ and INx− (or COM) inputs captured at the end of the acquisition phase is applied to the comparator inputs, causing the comparator to become unbalanced. By switching each element of the capacitor array between GND and CAP, the comparator input varies by binary-weighted voltage steps (VREF/2, VREF/4, ... VREF/32,768). The control logic toggles these switches, starting with the MSB, to bring the comparator back into a balanced condition. After the completion of this process, the part returns to the acquisition phase, and the control logic generates the ADC output code and a busy signal indicator. Because the AD7682/AD7689 have an on-board conversion clock, the serial clock, SCK, is not required for the conversion process. Rev. 0 | Page 15 of 28 AD7682/AD7689 TRANSFER FUNCTIONS 011...111 111...111 011...110 011...101 111...110 111...101 The ideal transfer characteristic for the AD7682/AD7689 is shown in Figure 25 and for both unipolar and bipolar ranges with the internal 4.096 V reference. 100...010 000...010 100...001 000...001 100...000 000...000 –FSR –FSR + 1LSB –FSR + 0.5LSB +FSR – 1LSB +FSR – 1.5LSB ANALOG INPUT 07353-027 With the inputs configured for bipolar range (COM = VREF/2 or paired differentially with INx− = VREF/2), the data outputs are twos complement. TWOS STRAIGHT COMPLEMENT BINARY ADC CODE With the inputs configured for unipolar range (single ended, COM with ground sense, or paired differentially with INx− as ground sense), the data output is straight binary. Figure 25. ADC Ideal Transfer Function Table 7. Output Codes and Ideal Input Voltages Description FSR − 1 LSB Midscale + 1 LSB Midscale Midscale − 1 LSB −FSR + 1 LSB −FSR Unipolar Analog Input1 VREF = 4.096 V 4.095938 V 2.048063 V 2.048 V 2.047938 V 62.5 μV 0V Digital Output Code (Straight Binary Hex) 0xFFFF 0x8001 0x8000 0x7FFF 0x0001 0x0000 1 With COM or INx− = 0 V or all INx referenced to GND. With COM or INx− = VREF /2. 3 This is also the code for an overranged analog input ((INx+) − (INx−), or COM, above VREF − VGND). 4 This is also the code for an underranged analog input ((INx+) − (INx−), or COM, below VGND). 2 Rev. 0 | Page 16 of 28 Bipolar Analog Input2 VREF = 4.096 V 2.047938 V 62.5 μV 0V −62.5 μV −2.047938 V −2.048 V Digital Output Code (Twos Complement Hex) 0x7FFF 0x0001 0x00004 0xFFFF3 0x8001 0x8000 AD7682/AD7689 TYPICAL CONNECTION DIAGRAMS IN0 100nF REFIN VDD REF 0V TO VREF V– 100nF 100nF 10µF2 V+ ADA4841-x 3 1.8V TO VDD 5V VIO AD7682/AD7689 V+ INx 0V TO VREF ADA4841-x 3 DIN MOSI SCK SCK SDO MISO CNV SS V– 0V OR VREF /2 COM GND 07353-028 NOTES: 1. INTERNAL REFERENCE SHOWN. SEE VOLTAGE REFERENCE OUTPUT/INPUT SECTION FOR REFERENCE SELECTION. 2. CREF IS USUALLY A 10µF CERAMIC CAPACITOR (X5R). 3. SEE DRIVER AMPLIFIER CHOICE SECTION FOR ADDITIONAL RECOMMENDED AMPLIFIERS. 4. SEE THE DIGITAL INTERFACE SECTION FOR CONFIGURING AND READING CONVERSION DATA. Figure 26. Typical Application Diagram with Multiple Supplies 5V V+ 100nF 100nF 10µF2 REF REFIN VDD 1.8V TO VDD 100nF VIO ADA4841-x 3 IN0 AD7682/AD7689 V+ INx ADA4841-x 3 VREF /2 COM MOSI SCK SDO MISO CNV SS GND NOTES: 1. INTERNAL REFERENCE SHOWN. SEE VOLTAGE REFERENCE OUTPUT/INPUT SECTION FOR REFERENCE SELECTION. 2. CREF IS USUALLY A 10µF CERAMIC CAPACITOR (X5R). 3. SEE DRIVER AMPLIFIER CHOICE SECTION FOR ADDITIONAL RECOMMENDED AMPLIFIERS. 4. SEE THE DIGITAL INTERFACE SECTION FOR CONFIGURING AND READING CONVERSION DATA. 07353-029 VREF p-p DIN SCK Figure 27. Typical Application Diagram Using Bipolar Input Rev. 0 | Page 17 of 28 AD7682/AD7689 Unipolar or Bipolar 70 Figure 26 shows an example of the recommended connection diagram for the AD7682/AD7689 when multiple supplies are available. 65 60 55 CMRR (dB) Bipolar Single Supply Figure 27 shows an example of a system with a bipolar input using single supplies with the internal reference (optional different VIO supply). This circuit is also useful when the amplifier/signal conditioning circuit is remotely located with some common mode present. Note that for any input configuration, the inputs INx are unipolar and always referenced to GND (no negative voltages even in bipolar range). ANALOG INPUTS Input Structure Figure 28 shows an equivalent circuit of the input structure of the AD7682/AD7689. The two diodes, D1 and D2, provide ESD protection for the analog inputs, IN[7:0] and COM. Care must be taken to ensure that the analog input signal does not exceed the supply rails by more than 0.3 V because this causes the diodes to become forward biased and to start conducting current. These diodes can handle a forward-biased current of 130 mA maximum. For instance, these conditions may eventually occur when the input buffer supplies are different from VDD. In such a case, for example, an input buffer with a short circuit, the current limitation can be used to protect the part. VDD INx+ OR INx– OR COM D1 CPIN RIN 40 35 30 1 10 100 FREQUENCY (kHz) 1k 10k Figure 29. Analog Input CMRR vs. Frequency During the acquisition phase, the impedance of the analog inputs can be modeled as a parallel combination of the capacitor, CPIN, and the network formed by the series connection of RIN and CIN. CPIN is primarily the pin capacitance. RIN is typically 3.5 kΩ and is a lumped component made up of serial resistors and the on resistance of the switches. CIN is typically 27 pF and is mainly the ADC sampling capacitor. Selectable Low Pass Filter During the conversion phase, where the switches are opened, the input impedance is limited to CPIN. While the AD7682/ AD7689 are acquiring, RIN and CIN make a one-pole, low-pass filter that reduces undesirable aliasing effects and limits the noise from the driving circuitry. The low-pass filter can be programmed for the full bandwidth or ¼ of the bandwidth with CFG[6] as shown in Table 9. Note that the converters throughout must also be reduced by ¼ when using the filter. If the maximum throughput is used with the BW set to ¼, the converter acquisition time, tACQ, will be violated, resulting in increased THD. Input Configurations Figure 30 shows the different methods for configuring the analog inputs with the configuration register (CFG[12:10]). Refer to the Configuration Register, CFG, section for more details. CIN D2 07353-030 GND 45 07353-031 For this circuit, a rail-to-rail input/output amplifier can be used; however, the offset voltage vs. input common-mode range should be noted and taken into consideration (1 LSB = 62.5 μV with VREF = 4.096 V). Note that the conversion results are in twos complement format when using the bipolar input configuration. Refer to the AN-581 Application Note for additional details about using single-supply amplifiers. 50 Figure 28. Equivalent Analog Input Circuit This analog input structure allows the sampling of the true differential signal between INx+ and COM or INx+ and INx−. (COM or INx− = GND ± 0.1 V or VREF ± 0.1 V). By using these differential inputs, signals common to both inputs are rejected, as shown in Figure 29. Rev. 0 | Page 18 of 28 AD7682/AD7689 CH0+ IN0 CH0+ CH1+ IN1 CH1+ IN1 CH2+ IN2 CH2+ IN2 CH3+ IN3 CH3+ IN3 CH4+ IN4 CH4+ IN4 CH5+ IN5 CH5+ IN5 CH6+ IN6 CH6+ IN6 CH7+ IN7 CH7+ IN7 COM COM– COM GND A—8 CHANNELS, SINGLE ENDED channel pairs are always paired IN (even) = INx+ and IN (odd) = INx− regardless of CFG[7]. IN0 To enable the sequencer, CFG[2:1] are written to for initializing the sequencer. After CFG[13:0] are updated, DIN must be held low while reading data out (at least for Bit 13), or the CFG will begin updating again. While operating in a sequence, the CFG can be changed by writing 012 to CFG[2:1]. However, if changing CFG11 (paired or single channel) or CFG[9:7] (last channel in sequence), the sequence reinitializes and converts IN0 (or IN1) after CFG is updated. GND B—8 CHANNELS, COMMON REFERNCE Examples IN0 CH0+ (–) CH0– (+) IN1 CH0– (+) IN1 CH1+ (–) IN2 CH1+ (–) IN2 CH1– (+) IN3 CH1– (+) IN3 CH2+ (–) IN4 CH2+ IN4 CH2– (+) IN5 CH3+ IN5 CH3+ (–) IN6 CH4+ IN6 CH3– (+) IN7 CH5+ IN7 COM COM– As a first example, scan all IN[7:0] referenced to COM = GND with temperature sensor. 13 CFG - GND D—COMBINATION 13 CFG - Figure 30. Multiplexed Analog Input Configuraitons • • • 1 11 10 INCC 1 0 9 8 7 INx 1 1 1 6 BW - 5 - 4 3 REF - 2 1 SEQ 1 0 0 RB - 12 0 11 10 INCC 0 X 9 8 7 INx 1 0 X 6 BW - 5 - 4 3 REF - 2 1 SEQ 1 1 0 RB - Source Resistance The analog inputs can be configured as • 12 As a second example, scan three paired channels without temperature sensor and referenced to VREF/2. COM GND C—4 CHANNELS, DIFFERENTIAL Only the bits for input and sequencer are highlighted. IN0 07353-032 CH0+ (–) Figure 30A, single ended referenced to system ground; CFG[12:10] = 1112. Figure 30B, bipolar differential with a common reference point; COM = VREF/2; CFG[12:10] = 0102. Unipolar differential with COM connected to a ground sense; CFG[12:10] = 1102. Figure 30C, bipolar differential pairs with INx− referenced to VREF/2; CFG[12:10] = 00X2. Unipolar differential pairs with INx− referenced to a ground sense; CFG[12:10] = 10X2. In this configuration, the INx+ is identified by the channel in CFG[9:7]. Example: for IN0 = IN1+ and IN1 = IN1−, CFG[9:7] = 0002; for IN1 = IN1+ and IN0 = IN1−, CFG[9:7] = 0012. Figure 30D, inputs configured in any of the above combinations (showing that the AD7682/AD7689 can be configured dynamically). When the source impedance of the driving circuit is low, the AD7682/AD7689 can be driven directly. Large source impedances significantly affect the ac performance, especially total harmonic distortion (THD). The dc performances are less sensitive to the input impedance. The maximum source impedance depends on the amount of THD that can be tolerated. The THD degrades as a function of the source impedance and the maximum input frequency. DRIVER AMPLIFIER CHOICE Although the AD7682/AD7689 are easy to drive, the driver amplifier must meet the following requirements: • Sequencer The AD7682/AD7689 include a channel sequencer useful for scanning channels in a IN0 to INx fashion. Channels are scanned as singles or pairs, with or without the temperature sensor, after the last channel is sequenced. The sequencer starts with IN0 and finishes with INx set in CFG[9:7]. For paired channels, the channels are paired depending on the last channel set in CFG[9:7]. Note that the Rev. 0 | Page 19 of 28 The noise generated by the driver amplifier must be kept as low as possible to preserve the SNR and transition noise performance of the AD7682/AD7689. Note that the AD7682/ AD7689 have a noise much lower than most of the other 16-bit ADCs and, therefore, can be driven by a noisier amplifier to meet a given system noise specification. The noise from the amplifier is filtered by the AD7682/AD7689 analog input circuit low-pass filter made by RIN and CIN or by an external filter, if one is used. Because the typical noise of the AD7682/AD7689 is 35 μV rms (with VREF = 5 V), the SNR degradation due to the amplifier is AD7682/AD7689 SNRLOSS ⎛ ⎜ 35 = 20 log ⎜ ⎜ π ⎜ 35 2 + f −3dB ( Ne N ) 2 2 ⎝ ⎞ ⎟ ⎟ ⎟ ⎟ ⎠ AD7682/AD7689 and is thus useful for performing a system calibration. Note that, when using the temperature sensor, the output is straight binary referenced from the AD7682/AD7689 GND pin. where: f–3dB is the input bandwidth in megahertz of the AD7682/AD7689 (1.7 MHz in full BW or 425 kHz in ¼ BW) or is the cutoff frequency of an input filter, if one is used. N is the noise gain of the amplifier (for example, 1 in buffer configuration). eN is the equivalent input noise voltage of the op amp, in nV/√Hz. • • For ac applications, the driver should have a THD performance commensurate with the AD7682/AD7689. Figure 18 shows THD vs. frequency for the AD7682/AD7689. For multichannel, multiplexed applications on each input or input pair, the driver amplifier and the AD7682/ AD7689 analog input circuit must settle a full-scale step onto the capacitor array at a 16-bit level (0.0015%). In the amplifier data sheet, settling at 0.1% to 0.01% is more commonly specified. This may differ significantly from the settling time at a 16-bit level and should be verified prior to driver selection. Table 8. Recommended Driver Amplifiers Amplifier ADA4841-x AD8655 AD8021 AD8022 OP184 AD8605, AD8615 Typical Application Very low noise, small, and low power 5 V single supply, low noise Very low noise and high frequency Low noise and high frequency Low power, low noise, and low frequency 5 V single supply, low power The internal reference is temperature-compensated to within 15 mV. The reference is trimmed to provide a typical drift of 3 ppm/°C. External Reference and Internal Buffer For improved drift performance, an external reference can be used with the internal buffer. The external reference is connected to REFIN, and the output is produced on the REF pin. An external reference can be used with the internal buffer with or without the temperature sensor enabled. Refer to Table 9 for the register details. With the buffer enabled, the gain is unity and is limited to an input/output of 4.096 V. The internal reference buffer is useful in multiconverter applications because a buffer is typically required in these applications. In addition, a low power reference can be used because the internal buffer provides the necessary performance to drive the SAR architecture of the AD7682/AD7689. External Reference In any of the six voltage reference schemes, an external reference can be connected directly on the REF pin because the output impedance of REF is >5 kΩ. To reduce power consumption, the reference and buffer can be powered down independently or together for the lowest power consumption. However, for applications requiring the use of the temperature sensor, the reference must be active. Refer to Table 9 for register details. For improved drift performance, an external reference such as the ADR43x or ADR44x is recommended. Reference Decoupling VOLTAGE REFERENCE OUTPUT/INPUT The AD7682/AD7689 allow the choice of a very low temperature drift internal voltage reference, an external reference, or an external buffered reference. The internal reference of the AD7682/AD7689 provide excellent performance and can be used in almost all applications. There are six possible choices of voltage reference schemes briefly described in Table 9 with more details in each of the following sections. Internal Reference/Temperature Sensor The internal reference can be set for either 2.5 V or a 4.096 V output as detailed in Table 9. With the internal reference enabled, the band gap voltage is also present on the REFIN pin, which requires an external 0.1 μF capacitor. Because the current output of REFIN is limited, it can be used as a source if followed by a suitable buffer, such as the AD8605. Whether using an internal or external reference, the AD7682/ AD7689 voltage reference output/input, REF, has a dynamic input impedance and should therefore be driven by a low impedance source with efficient decoupling between the REF and GND pins. This decoupling depends on the choice of the voltage reference but usually consists of a low ESR capacitor connected to REF and GND with minimum parasitic inductance. A 10 μF (X5R, 1206 size) ceramic chip capacitor is appropriate when using the internal reference, the ADR43x/ADR44x external reference, or a low impedance buffer such as the AD8031 or the AD8605. The placement of the reference decoupling capacitor is also important to the performance of the AD7682/AD7689, as explained in the Layout section. The decoupling capacitor should be mounted on the same side as the ADC at the REF pin with a thick PCB trace. The GND should also connect to the reference decoupling capacitor with the shortest distance and to the analog ground plane with several vias. Enabling the reference also enables the internal temperature sensor, which measures the internal temperature of the Rev. 0 | Page 20 of 28 AD7682/AD7689 If desired, smaller reference decoupling capacitor values down to 2.2 μF can be used with a minimal impact on performance, especially on DNL. 10000 Regardless, there is no need for an additional lower value ceramic decoupling capacitor (for example, 100 nF) between the REF and GND pins. OPERATING CURRENT (µA) 1000 For applications that use multiple AD7682/AD7689s or other PulSAR devices, it is more effective to use the internal reference buffer to buffer the external reference voltage, thus reducing SAR conversion crosstalk. The voltage reference temperature coefficient (TC) directly impacts full scale; therefore, in applications where full-scale accuracy matters, care must be taken with the TC. For instance, a ±15 ppm/°C TC of the reference changes full scale by ±1 LSB/°C. VDD = 5V, INTERNAL REF 100 VDD = 5V, EXTERNAL REF 10 VDD = 2.5V, EXTERNAL REF 1 VIO 0.1 0.001 POWER SUPPLY 100 10 1k 10k SAMPLING RATE (sps) 100k 1M 07353-040 0.010 Figure 32. Operating Currents vs. Sampling Rate The AD7682/AD7689 use two power supply pins: an analog and digital core supply (VDD) and a digital input/output interface supply (VIO). VIO allows direct interface with any logic between 1.8 V and VDD. To reduce the supplies needed, the VIO and VDD pins can be tied together. The AD7682/AD7689 are independent of power supply sequencing between VIO and VDD. The only restriction is that CNV must be low when powering up the AD7682/AD7689. Additionally, it is very insensitive to power supply variations over a wide frequency range, as shown in Figure 31. 75 SUPPLYING THE ADC FROM THE REFERENCE For simplified applications, the AD7682/AD7689, with their low operating current, can be supplied directly using the reference circuit as shown in Figure 33. The reference line can be driven by • • The system power supply directly A reference voltage with enough current output capability, such as the ADR43x/ADR44x A reference buffer, such as the AD8605, which can also filter the system power supply, as shown in Figure 33 • 70 65 5V 5V 10Ω 5V 55 10kΩ 1µF 50 AD8605 1µF 10µF 0.1µF 1 45 REF 40 VDD VIO AD7689 35 1 10 100 FREQUENCY (kHz) 1k 10k 1OPTIONAL 07353-034 30 0.1µF REFERENCE BUFFER AND FILTER. Figure 33. Example of an Application Circuit Figure 31. PSRR vs. Frequency The AD7682/AD7689 power down automatically at the end of each conversion phase; therefore, the operating currents and power scale linearly with the sampling rate. This makes the part ideal for low sampling rates (even of a few hertz) and low battery-powered applications. Rev. 0 | Page 21 of 28 07353-035 PSSR (dB) 60 AD7682/AD7689 The SCK frequency required is calculated by DIGITAL INTERFACE The AD7682/AD7689 use a simple 4-wire interface and are compatible with SPI, MICROWIRE™, QSPI™, digital hosts, and DSPs, for example, Blackfin® ADSP-BF53x, SHARC®, ADSP219x, and ADSP-218x. The interface uses the CNV, DIN, SCK, and SDO signals and allows CNV, which initiates the conversion, to be independent of the readback timing. This is useful in low jitter sampling or simultaneous sampling applications. A 14-bit register, CFG[13:0], is used to configure the ADC for the channel to be converted, the reference selection, and other components, which are detailed in the Configuration Register, CFG, section. When CNV is low, reading/writing can occur during conversion, acquisition, and spanning conversion (acquisition plus conversion), as detailed in the following sections. The CFG word is updated on the first 14 SCK rising edges, and conversion results are read back on the first 15 (or 16 if busy mode is selected) SCK falling edges. If the CFG readback is enabled, an additional 14 SCK falling edges are required to read back the CFG word associated with the conversion results with the CFG MSB following the LSB of the conversion result. A discontinuous SCK is recommended because the part is selected with CNV low, and SCK activity begins to write a new configuration word and clock out data. Note that in the following sections, the timing diagrams indicate digital activity (SCK, CNV, DIN, SDO) during the conversion. However, due to the possibility of performance degradation, digital activity should occur only prior to the safe data reading/writing time, tDATA, because the AD7682/AD7689 provide error correction circuitry that can correct for an incorrect bit during this time. From tDATA to tCONV, there is no error correction and conversion results may be corrupted. The user should configure the AD7682/AD7689 and initiate the busy indicator (if desired) prior to tDATA. It is also possible to corrupt the sample by having SCK or DIN transitions near the sampling instant. Therefore, it is recommended to keep the digital pins quiet for approximately 30 ns before and 10 ns after the rising edge of CNV, using a discontinuous SCK whenever possible to avoid any potential performance degradation. Reading/Writing During Conversion, Fast Hosts When reading/writing during conversion (n), conversion results are for the previous (n − 1) conversion, and writing the CFG is for the next (n + 1) acquisition and conversion. f SCK ≥ Number _ SCK _ Edges t DATA The time between tDATA and tCONV is a safe time when digital activity should not occur, or sensitive bit decisions may be corrupt. Reading/Writing During Acquisition, Any Speed Hosts When reading/writing during acquisition (n), conversion results are for the previous (n − 1) conversion, and writing is for the (n + 1) acquisition. For the maximum throughput, the only time restriction is that the reading/writing take place during the tACQ (min) time. For slow throughputs, the time restriction is dictated by throughput required by the user, and the host is free to run at any speed. Thus for slow hosts, data access must take place during the acquisition phase. Reading/Writing Spanning Conversion, Any Speed Host When reading/writing spanning conversion, the data access starts at the current acquisition (n) and spans into the conversion (n). Conversion results are for the previous (n − 1) conversion, and writing the CFG is for the next (n + 1) acquisition and conversion. Similar to reading/writing during conversion, reading/writing should only occur up to tDATA. For the maximum throughput, the only time restriction is that reading/writing take place during the tACQ (min) + tDATA time. For slow throughputs, the time restriction is dictated by the user’s required throughput, and the host is free to run at any speed. Similar to the reading/writing during acquisition, for slow hosts, the data access must take place during the acquisition phase with additional time into the conversion. Note that data access spanning conversion requires the CNV to be driven high to initiate a new conversion, and data access is not allowed when CNV is high. Thus, the host must perform two bursts of data access when using this method. CONFIGURATION REGISTER, CFG The AD7682/AD7689 use a 14-bit configuration register (CFG[13:0]) as detailed in Table 9 for configuring the inputs, channel to be converted, one-pole filter bandwidth, reference, and channel sequencer. The CFG is latched (MSB first) on DIN with 14 SCK rising edges. The CFG update is edge dependent, allowing for asynchronous or synchronous hosts. After the CNV is brought high to initiate conversion, it must be brought low again to allow reading/writing during conversion. Reading/writing should only occur up to tDATA and, because this time is limited, the host must use a fast SCK. Rev. 0 | Page 22 of 28 AD7682/AD7689 • • • The register can be written to during conversion, during acquisition, or spanning acquisition/conversion and is updated at the end of conversion, tCONV (max). There is always a one deep delay when writing CFG. Note that, at power up, the CFG is undefined and two dummy conversions are required to update the register. To preload the CFG with a factory setting, hold DIN high for two conversions. Thus CFG[13:0] = 0x3FFF. This sets the AD7682/AD7689 for 13 CFG 12 INCC 11 INCC 10 INCC 9 INx 8 INx • IN[7:0] unipolar referenced to GND, sequenced in order Full bandwidth for one-pole filter Internal reference/temperature sensor disabled, buffer enabled No readback of CFG Table 9 summarizes the configuration register bit details. See the Theory of Operation section for more details. 7 INx 6 BW 5 REF 4 REF 3 REF 2 SEQ 1 SEQ 0 RB Table 9. Configuration Register Description Bit(s) [13] Name CFG [12:10] INCC [9:7] INx [6] BW [5:3] REF [2:1] SEQ 0 RB Description Configuration update. 0 = Keep current configuration settings. 1 = Overwrite contents of register. Input channel configuration. Selection of pseudobipolar, pseudodifferential, pairs, single-ended or temperature sensor. Refer to the Input Configurations section. Bit 12 Bit 11 Bit 10 Function 0 0 X Bipolar differential pairs; INx− referenced to VREF/2 ± 0.1 V. 0 1 0 Bipolar; INx referenced to COM = VREF/2 ± 0.1 V. 0 1 1 Temperature sensor. 1 0 X Unipolar differential pairs; INx− referenced to GND ± 0.1 mV. 1 1 0 Unipolar, IN0 to IN7 referenced to COM = GND ± 0.1 V (GND sense). 1 1 1 Unipolar, IN0 to IN7 referenced to GND. Input channel selection in binary fashion. AD7682 AD7689 Bit 9 Bit 8 Bit 7 Channel Bit 9 Bit 8 Bit 7 Channel 0 0 X IN0 0 0 0 IN0 0 1 X IN1 0 0 1 IN1 1 0 X IN2 … … … 1 1 X IN3 1 1 1 IN7 Select bandwidth for low-pass filter. Refer to the Selectable Low Pass Filter section. 0 = ¼ of BW, uses an additional series resistor to further bandwidth limit the noise. Maximum throughout must be reduced to ¼ also. 1 = Full BW. Reference/buffer selection. Selection of internal, external, external buffered, and enabling of the on-chip temperature sensor. Refer to the Voltage Reference Output/Input section. Bit 5 Bit 4 Bit 3 Function 0 0 0 Internal reference, REF = 2.5 V output. 0 0 1 Internal reference, REF = 4.096 V output. 0 1 0 External reference, temperature enabled. 0 1 1 External reference, internal buffer, temperature enabled. 1 1 0 External reference, temperature disabled. 1 1 1 External reference, internal buffer, temperature disabled. Channel sequencer. Allows for scanning channels in an IN0 to INx fashion. Refer to the Sequencer section. Bit 2 Bit 1 Function 0 0 Disable sequencer. 0 1 Update configuration during sequence. 1 0 Scan IN0 to INx (set in CFG[9:7]), then temperature. 1 1 Scan IN0 to INx (set in CFG[9:7]). Read back the CFG register. 0 = Read back current configuration at end of data. 1 = Do not read back contents of configuration. Rev. 0 | Page 23 of 28 AD7682/AD7689 to begin the CFG update. While CNV is low, both a CFG update and a data readback take place. The first 14 SCK rising edges are used to update the CFG, and the first 15 SCK falling edges clock out the conversion results starting with MSB − 1. The restriction for both configuring and reading is that they both occur before the tDATA time of the next conversion elapses. All 14 bits of CFG[13:0] must be written, or they are ignored. In addition, if the 16-bit conversion result is not read back before tDATA elapses, it is lost. READ/WRITE SPANNING CONVERSION WITHOUT A BUSY INDICATOR This mode is used when the AD7682/AD7689 are connected to any host using an SPI, serial port, or FPGA. The connection diagram is shown in Figure 34, and the corresponding timing is given in Figure 35. For SPI, the host should use CPHA = CPOL = 0. Reading/writing spanning conversion is shown, which covers all three modes detailed in the Digital Interface section. A rising edge on CNV initiates a conversion, forces SDO to high impedance, and ignores data present on DIN. After a conversion is initiated, it continues until completion irrespective of the state of CNV. CNV must be returned high before the safe data transfer time, tDATA, and then held high beyond the conversion time, tCONV, to avoid generation of the busy signal indicator. The SDO data is valid on both SCK edges. Although the rising edge can be used to capture the data, a digital host using the SCK falling edge allows a faster reading rate, provided it has an acceptable hold time. After the 16th (or 30th) SCK falling edge, or when CNV goes high (whichever occurs first), SDO returns to high impedance. If CFG readback is enabled, the CFG associated with the conversion result (n − 1) is read back MSB first following the LSB of the conversion result. A total of 30 SCK falling edges is required to return SDO to high impedance if this is enabled. After the conversion is complete, the AD7682/AD7689 enter the acquisition phase and power down. When the host brings CNV low after tCONV (max), the MSB is enabled on SDO. The host also must enable the MSB of CFG at this time (if necessary) CNV SS SDO MISO DIN MOSI SCK SCK FOR SPI USE CPHA = 0, CPOL = 0. 07353-036 DIGITAL HOST AD7682/ AD7689 Figure 34. Connection Diagram for the AD7682/AD7689 Without a Busy Indicator tCYC > tCONV tCONV tCONV tDATA tDATA tCNVH RETURN CNV HIGH FOR NO BUSY RETURN CNV HIGH FOR NO BUSY CNV tACQ (QUIET TIME) CONVERSION (n – 1) UPDATE (n) CFG/SDO SCK 14 CFG LSB DIN 16/ 30 15 1 tEN END CFG (n) SDO LSB + 1 tDIS END DATA (n – 2) tSDIN CFG MSB tDSDO MSB LSB tDIS 15 16/ 30 CFG LSB X X tHDIN CFG MSB – 1 BEGIN CFG (n + 1) tHSDO tEN 14 2 ACQUISITION (n + 1) UPDATE (n + 1) CFG/SDO SEE NOTE tCLSCK X X (QUIET TIME) CONVERSION (n) ACQUISITION (n) tEN END CFG (n + 1) SEE NOTE MSB – 1 BEGIN DATA (n – 1) LSB + 1 tDIS END DATA (n – 1) NOTES: 1. THE LSB IS FOR CONVERSION RESULTS OR THE CONFIGURATION REGISTER CFG (n – 1) IF 15 SCK FALLING EDGES = LSB OF CONVERSION RESULTS. 29 SCK FALLING EDGES = LSB OF CONFIGURATION REGISTER. ON THE 16TH OR 30TH SCK FALLING EDGE, SDO IS DRIVEN TO HIGH IMPENDANCE. Figure 35. Serial Interface Timing for the AD7682/AD7689 Without a Busy Indicator Rev. 0 | Page 24 of 28 LSB tDIS 07353-037 ACQUISITION (n - 1) AD7682/AD7689 update. While CNV is low, both a CFG update and a data readback take place. The first 14 SCK rising edges are used to update the CFG, and the first 16 SCK falling edges clock out the conversion results starting with the MSB. The restriction for both configuring and reading is that they both occur before the tDATA time elapses for the next conversion. All 14 bits of CFG[13:0] must be written or they are ignored. Also, if the 16-bit conversion result is not read back before tDATA elapses, it is lost. READ/WRITE SPANNING CONVERSION WITH A BUSY INDICATOR This mode is used when the AD7682/AD7689 are connected to any host using an SPI, serial port, or FPGA with an interrupt input. The connection diagram is shown in Figure 36, and the corresponding timing is given in Figure 37. For SPI, the host should use CPHA = CPOL = 1. Reading/writing spanning conversion is shown, which covers all three modes detailed in the Digital Interface section. The SDO data is valid on both SCK edges. Although the rising edge can be used to capture the data, a digital host using the SCK falling edge allows a faster reading rate, provided it has an acceptable hold time. After the optional 17th SCK falling edge, SDO returns to high impedance. Note that, if the optional SCK falling edge is not used, the busy feature cannot be detected if the LSB for the conversion is low. A rising edge on CNV initiates a conversion, forces SDO to high impedance, and ignores data present on DIN. After a conversion is initiated, it continues until completion irrespective of the state of CNV. CNV must be returned low before the safe data transfer time, tDATA, and then held low beyond the conversion time, tCONV, to generate the busy signal indicator. When the conversion is complete, SDO transitions from high impedance to low with a pull-up to VIO, which can be used to interrupt the host to begin data transfer. If CFG readback is enabled, the CFG associated with the conversion result (n − 1) is read back MSB first following the LSB of the conversion result. A total of 31 SCK falling edges is required to return SDO to high impedance if this is enabled. After the conversion is complete, the AD7682/AD7689 enter the acquisition phase and power down. The host must enable the MSB of CFG at this time (if necessary) to begin the CFG VIO AD7682/ AD7689 DIGITAL HOST SDO MISO IRQ DIN MOSI SCK SCK FOR SPI USE CPHA = 1, CPOL = 1. 07353-038 SS CNV Figure 36. Connection Diagram for the AD7682/AD7689 with a Busy Indicator tCYC tCONV tACQ tDATA tCNVH tDATA CNV CONVERSION (n – 1) (QUIET TIME) CONVERSION (n – 1) UPDATE (n) CFG/SDO SCK 15 16 17/ 31 (QUIET TIME) CONVERSION (n) ACQUISITION (n) UPDATE (n + 1) CFG/SDO SEE NOTE 1 2 15 16 17/ 31 X X X tHDIN ACQUISITION (n + 1) tSDIN X X CFG CFG MSB MSB –1 X tDIS END CFG (n) SDO LSB +1 END DATA (n – 2) BEIGN CFG (n + 1) MSB LSB tEN tHSDO tDSDO tEN MSB –1 BEGIN DATA (n – 1) LSB +1 tDIS LSB END DATA (n – 1) SEE NOTE NOTES: 1. THE LSB IS FOR CONVERSION RESULTS OR THE CONFIGURATION REGISTER CFG (n – 1) IF 16 SCK FALLING EDGES = LSB OF CONVERSION RESULTS. 30 SCK FALLING EDGES = LSB OF CONFIGURATION REGISTER. ON THE 17TH OR 31st SCK FALLING EDGE, SDO IS DRIVEN TO HIGH IMPENDANCE. OTHERWISE, THE LSB REMAINS ACTIVE UNTIL THE BUSY INDICATOR IS DRIVEN LOW. Figure 37. Serial Interface Timing for the AD7682/AD7689 with a Busy Indicator Rev. 0 | Page 25 of 28 tDIS END CFG (n + 1) tEN 07353-039 DIN AD7682/AD7689 APPLICATION HINTS LAYOUT The printed circuit board that houses the AD7682/AD7689 should be designed so that the analog and digital sections are separated and confined to certain areas of the board. The pinout of the AD7682/AD7689, with all its analog signals on the left side and all its digital signals on the right side, eases this task. Avoid running digital lines under the device because these couple noise onto the die unless a ground plane under the AD7682/AD7689 is used as a shield. Fast switching signals, such as CNV or clocks, should not run near analog signal paths. Crossover of digital and analog signals should be avoided. At least one ground plane should be used. It can be common or split between the digital and analog sections. In the latter case, the planes should be joined underneath the AD7682/AD7689. The AD7682/AD7689 voltage reference input REF has a dynamic input impedance and should be decoupled with minimal parasitic inductances. This is done by placing the reference decoupling ceramic capacitor close to, ideally right up against, the REF and GND pins and connecting them with wide, low impedance traces. Finally, the power supplies VDD and VIO of the AD7682/ AD7689 should be decoupled with ceramic capacitors, typically 100 nF, placed close to the AD7682/AD7689 and connected using short, wide traces to provide low impedance paths and reduce the effect of glitches on the power supply lines. EVALUATING AD7682/AD7689 PERFORMANCE Other recommended layouts for the AD7682/AD7689 are outlined in the documentation of the evaluation board for the AD7682/AD7689 (EVAL-AD7682CBZ/EVAL-AD7689CBZ). The evaluation board package includes a fully assembled and tested evaluation board, documentation, and software for controlling the board from a PC via the evaluation controller board, EVAL-CONTROL BRD3. Rev. 0 | Page 26 of 28 AD7682/AD7689 OUTLINE DIMENSIONS 0.60 MAX 4.00 BSC SQ 0.60 MAX 15 PIN 1 INDICATOR 20 16 1 PIN 1 INDICATOR 3.75 BSC SQ 0.50 BSC 2.65 2.50 SQ 2.35 EXPOSED PAD (BOTTOM VIEW) 5 TOP VIEW 12° MAX SEATING PLANE 10 6 0.25 MIN 0.80 MAX 0.65 TYP 0.30 0.23 0.18 0.05 MAX 0.02 NOM COPLANARITY 0.08 0.20 REF COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-1 012508-B 1.00 0.85 0.80 0.50 0.40 0.30 11 Figure 38. 20-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 4 mm × 4 mm Body, Very Thin Quad (CP-20-4) Dimensions shown in millimeters ORDERING GUIDE Model AD7682BCPZ1 AD7682BCPZRL71 AD7689ACPZ 1 AD7689ACPZRL71 AD7689BCPZ1 AD7689BCPZRL71 EVAL-AD7682CBZ1 EVAL-AD7689CBZ1 EVAL-CONTROL BRD3 2 1 2 Integral Nonlinearity ±2 LSB max ±2 LSB max ±6 LSB max ±6 LSB max ±2 LSB max ±2 LSB max No Missing Code 16 bits 16 bits 15 bits 15 bits 16 bits 16 bits Temperature Range –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C Package Description 20-Lead QFN (LFCSP_VQ) 20-Lead QFN (LFCSP_VQ) 20-Lead QFN (LFCSP_VQ) 20-Lead QFN (LFCSP_VQ) 20-Lead QFN (LFCSP_VQ) 20-Lead QFN (LFCSP_VQ) Evaluation Board Evaluation Board Controller Board Package Option CP-20-4 CP-20-4 CP-20-4 CP-20-4 CP-20-4 CP-20-4 Z = RoHS Compliant Part. This controller board allows a PC to control and communicate with all Analog Devices evaluation boards whose model numbers end in CB. Rev. 0 | Page 27 of 28 Ordering Quantity Tray, 490 Reel, 1,500 Tray, 490 Reel, 1,500 Tray, 490 Reel, 1,500 AD7682/AD7689 NOTES ©2008 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D07353-0-5/08(0) Rev. 0 | Page 28 of 28