APPLICATION NOTE U-137

U-137
APPLICATION NOTE
PRACTICAL CONSIDERATIONS IN HIGH PERFORMANCE
MOSFET, IGBT and MCT GATE DRIVE CIRCUITS
BILL ANDREYCAK
INTRODUCTION
The switchmode power supply industry’s trend towards higher conversion frequencies is justified by the
dramatic improvement in obtaining higher power densities. And as these frequencies are pushed towards
and beyond one megahertz, the Mosfet transition periods can become a significant portion of the total
switching period. Losses associated with the overlap of switch voltage and current not only degrade the
overall power supply efficiency, but warrant consideration from both a thermal and packaging standpoint.
A/though brief, each of the Mosfet switching transitions can be further reduced if driven from from a high
speed, high current totem-pole driver - one designed exclusively for this application. This paper will highlight
three such devices; the UC1708 and UC1710 high current Mosfet driver ICs, and the UC1711 high speed
driver. Other Mosfet driver ICs and typical application circuits are featured in UNITRODE Application Note
U-118.
EFFECTIVE GATE CAPACITANCE
The Mosfet input capacitance (Ciss) is frequently
misused as the load represented by a power mosfet
to the gate driver IC. In reality, the effective input
capacitance of a Mosfet (Ceff) is much higher, and
must be derived from the manufacturers’ published
total gate charge (Qg) information. Even the specified maximum values of the gate charge parameter
do not accurately reflect the driver’s instantaneous
loads during a given switching transition. Fortunately,
FET manufacturers provide a curve for the gate-tosource voltage (Vgs) versus total gate charge in
their datasheets. This will be segmented into four
time intervals of interest per switching transition.
Each of these will be analyzed to determine the
effective gate capacitance and driver requirements
for optimal performance.
adjusting the gate charge numbers accordingly.
Both turn-on and turn-off trasnsitions are shown with
the respective drain currents and drain-to-source
voltages.
TURN-ON WAVEFORMS
Gate voltage vs time
Inadequate gate drive is generally the result
TOTAL GATE CHARGE (Qg)
First, a typical high power Mosfet “Gate Charge
versus Gate-to-Source Voltage” curve will be examined. An IRFP460 device has been selected and
this curve is applicable to most other Fet devices by
3-314
Figure 1.
APPLICATION NOTE
U-137
INTERVAL t0-t1
INTERVAL t2-t3
The time required to bring the gate voltage from zero
to its threshold Vgs(th) can be expressed as a delay
time. Both the voltage across the switching device
and current through it are uneffected during this
interval.
Beginning at time t2 the drain-to-source voltage
starts to fall which introduces the “Miller” capacitance effects (Cgd) from the drain to the Mosfet gate.
The result is the noticeable plateau in the gate
voltage waveform from time t2 until t3 while a charge
equal to Qgd is admitted. It is here that most drive
circuits are taxed to their limits. The interval concludes at time t3 when the drain voltage approaches
its minimum.
INTERVAL t1-t2
This period starts at time t1 when the gate voltage
has reached Vgs(th) and drain current begins to
flow. Current continues to rise until essentially
reaching its final value at time t2. While this occurred,
the gate to source voltage had also been increasing.
The drain-to-source voltage remains unchanged at
Vds(off). Power in the Mosfet is wasted by the
simultaneous overlap of voltage and current.
INTERVAL t3-t4
During this final interval of interest the gate voltage
rises from the plateau of the prior region up to its final
drive voltage. This increasing gate voltage decreases
Rds(on), the Mosfet drain-to-source resistance.
Bringing the gate voltage above 10 to 12 volts,
however, has little effect on further reducing Rds(on).
SUMMARY OF INTERVAL WAVEFORMS AND DRIVER LIMITATIONS
INTERVAL
to-t1
t1-t2
t2-t3
t3-t4
Vgs(t)
0-threshold
thrs-plateau
V(plateau)
rising
ID(t)
Vds(t)
DRIVER LIMITATIONS
0
rising
lon(dc)
lon(dc)
Vds(off)
Vds(off)
falling
lon*Rds(t)
Slew rate (dv/dt)
Slew rate (dv/dt)
Peak current l(max)
Peak I & dv/dt
INTERVAL t4-t3
The beginning of the turn-off cycle can be described
as a delay from the final drive voltage (Vgs(on)) the
the plateau region. Both the drain voltage and current waveforms remain unchanged while the devices effective resistance (Rds(on)) increases as
the gate voltage decreases.
TURN-OFF WAVEFORMS
Gate voltage vs. time
INTERVAL t3-t2
Once the plateau is reached at time t3, the gate
voltage remains constant until time t2. Gate charge
due to the Miller effect is being removed, an amount
equal to Qgd. The drain voltage rises to its off state
amplitude, Vds(off), while the drain current continues to flow and equals l(on). This lossy transition
ends at time t2.
INTERVAL t2-t1
Once the Miller charge is completely removed, the
gate voltage is reduced from the plateau to the
threshold voltage causing the drain current to fall
from l(on) to zero. Transition power loss ends at
time t1 when the gate threshold is crossed.
Figure 2
The intervals during turn-off are basically the same
as those described for turn-on, however the sequence and corresponding waveforms are reversed.
INTERVAL t1 -t0
This brief period is of little interest in the turn-off
sequence since the device is off at time t1.
3-315
APPLICATION NOTE
SUMMARY OF INTERVAL WAVEFORMS AND DRIVER LIMITATIONS
INTERVAL
t4-t3
t3-t2
t2-t1
t1-t0
Vgs(t)
falling
V(plateau)
Vplat-thrsh
thrsh-0
ID(t)
Vds(t)
DRIVER LIMITATIONS
Ion(dc)
lon(dc)
falling
0
Ion” Rds(t)
falling
Vds(off)
Vds(off)
Peak I and dv/dt
Peak Current I (max)
Slew rate (dv/dt)
Slew rate (dv/dt)
Ploss = Vds(off)*l(on)*t(trans)/t(period)
FET Transition Power Loss
During each of the FET turn-on and turn-off sequences power is lost due to the switching device’s
simultaneous overlap of drain - source voltage and
drain current. Since both the FET voltage and current are externally controlled by the application, the
driver IC can only reduce the power losses by
making the transition times as brief as possible.
Minimization of these losses simply requires a
competent driver IC, one able to provide high peak
currents with high voltage slew rates.
A review of the prior transition waveforms indicates
that power is lost between the times of t1 and t3.
While t2 serves as the pivot point for which waveform
is rising or falling, as the equations show its irrelavent
in the power loss equation. For the purpose of
brevity, the waveform of interest can be approximated as a triangle while the other waveform is
constant. The duration between times t1 and t3 can
now be defined as the net transition time, t(tran), with
a conversion period of t(period)
This relationship displays the need for fast transitions at any switching frequency, and is of significant
concern at one megaHertz. Minimization of the FET
transition power loss can be achieved with high
current drivers.
GATE CHARGE
Each division of the transition interval has an associated gate charge which can be derived from the
FET manufacturers datasheets. Since there are
three basic shapes to the Vgs curve, the interval
from t0 to t1 can be lumped together with that of the
t1 to t2 period. For most large FET geometries, the
amount of charge in the t0 to - t1 span is negligible
anyway. This simplification allows an easy calculation of the effective gate capacitance for each interval along with quantifying the peak current required
to traverse in a given amount of time.
During the two intervals from t1 to t3:
0.5 * l(on) * Vds(off) * t(2-1)
Ploss =
t(period)
Charge can be represented as the product of capacitance multiplied by voltage, or current multiplied
by time. The effective gate capacitance is determined
by dividing the required gate charge (Qg) by the
gate voltage during a given interval. Likewise, the
current necessary to force a transition within a
specified time is obtained by dividing the gate charge
by the desired time.
0.5 * Vds(off) * l(on) *t(3-2)
Cgs (effective) = delta Qg / delta Vgs
Ploss =
t(period)
Ig(required) = delta Qg / t(transition)
Combining the two equations with t(tran)
= t3-1 results in a net loss of:
UC1710
The “MILLER KILLER”
0.5 * Vds(off) * l(on) * t(trans)
Ploss =
t(period)
Since these loses are incurred twice percycle, first at
turn-on and then again at turn-off, the net result is a
doubling of the power loss.
3-316
High peak gate drive currents are desirable in paralleled FET applications, typical of a high power
switching section or power factor correction stage.
Dubbed as “the Miller Killer”, the UC1710 boasts a
guaranteed 6 amp peak output current. This hefty
driver current minimizes the FET parasitic “Miller”
effects which would otherwise result in poor transi-
U -137
APPLICATION NOTE
tion performance. Higher currents are possible with
this driver, however the limiting factor soon becomes the parasitic series inductance of the FET
package (15 nH) and the layout interconnection of
20 nH/inch. An RF type arrangement of the PC
board layout is an absolute MUST to realize this
device’s full potential.
UC 1708 BLOCK DIAGRAM
UC 1710 BLOCK DIAGRAM
I
The UC1708 is a unique blend of the high speed
attributes of the UC1711 along with the higher peak
current capability of the UC1710. This dual
noninverting driver accepts positive TTL/CMOS logic
from control circuits and provides 3 amp peak outputs from each totem pole.
The UC1710 has “no-load” rise and fall times of 20
nanoseconds (or less) which do not change significantly with any loads under 3 nanoFarads. It’s also
specified into a load capacitance of 30 nanoFarads,
roughly equivalent to what is represented by three
paralleled “size 6” FET devices. Propagation delays
are brief with typical values specified at 35 nanoseconds from either input to a ten percent change in
output voltage.
Propagation delays are under 25 nanoseconds while
rise and fall times typically run 35 nanoseconds into
2.2 nanoFarads. The output stage design is a “no
float” version which incorporates a self biasing
technique to hold the outputs low during undervoltage
lockout, even with Vin removed.
In the 16 pin DIL package, the device features a
remote ENABLE and SHUTDOWN function in addition to seperate signal and power grounds. The
ENABLE function places the device in a low current
standby mode and the SHUTDOWN circuitry is high
speed logic directly to the outputs.
UC1708 / 1710 / 1711 PERFORMANCE COMPARISON
TABLE 1.
PARAMETER
LOAD
UC1708
UC1710
UC1711
Propagation Delay
t(plh)
input to 10% output
0
1.0 nF
2.2 nF
30 nF
25
25
25
30
10
15
20
Raise time
t(tlh)
10% to 90% rise
0
1.0 nF
2.2 nF
30 nF
25
30
40
20
Propagation Delay
t(phl)
input to 90% output
0
1.0 nF
2.2 nF
30 nF
25
25
25
30
Fall Time
t(thl)
90% to 10% fall
0
1.0 nF
2.2 nF
30 nF
25
30
40
3-317
30
30
25
85
12
25
40
3
5
30
30
15
20
85
7
25
40
APPLICATION NOTE
U-137
TRANSITION PERFORMANCE
Using the table above, the driver output slew rates
and average current delivered can be calculated.
The figures can be compared to lower power opamps or comparators to gain a perspective on the
relative speed of these high performance drivers.
The UC1708 delivers output slew rates (dv/dt) in the
order of 300 to 480 volts per microsecond, at average load currents of under one amp, depending on
the load. The high speed UC1711 exhibits similar
characteristics under loaded conditions, but can
achieve a no load slew rate of over 1700 volts per
microsecond - nearly 2 volts per nanosecond.
For higher power applications, the UC1710 “Miller
Killer” will produce an average current of 4.5 amps
AT slew rates of 150 volts per microsecond. With
lighter loads it will deliver an average current of 1.5
amps at a slew rate of approximately 500 volts per
microsecond. In most applications, the UC1710 will
easily outperform “homebrew” discrete mosfet
transistor totem pole drive techniques.
Each device in this new generation of MOSFET
drivers is significantly more responsive than the
earlier counterparts for a given application - whether
it’s higher speed (UC1711), higher peak current
(UC1710) or a combination of both (UC1708).
Optimization of a driver for this type of application
can be difficult. In general, the MOSFET driver IC
output stage is designed to switch as fast as the
manufacturer’s process will allow.
CROSS CONDUCTION
There are numerous tradeoffs involved in the design
of these drivers beyond the obvious choices of
number of outputs and peak current capability. Crossconduction is defined as the conduction of current
through both of the totem pole transistors simultaneously from Vin to ground. It is an unproductive loss
in the output stage which results in unnecessary
heating of the driver and wasted power. Cross conduction is the result of turning one transistor ON
before the opposing one is fully off, a compromise
often necessary to minimize the input to output
propagation delays.
An interesting observation is that cross-conduction
is less of a concern with large capacitive loads
(FETs) than with unloaded or lightly loaded driver
outputs. Any capacitive load will reduce the slew of
the output stage, slowing down its dv/dt. This causes
a portion of the cross conduction current to flow from
the load, rather than from the input supply through
the driver’s opposite output transistor. The power
loss associated with a drivers inherent cross-conduction is unchanged with large capacitive loads,
however it is not caused by a “shoot-through” of
supply current.
DRIVER CONSIDERATIONS
DRIVER PERFORMANCE
As previously demonstrated, the ideal MOSFET
gate drive IC is a unique blend of both high speed
switching and high peak current capability. Initially,
the high speed is required to bring the gate voltage
from zero to the plateau, but the current is low. Once
the plateau is intersected, the driver voltage is fairly
constant, and the IC must switch modes. Instantly,
the driver current snaps to its maximum as charge is
injected to overcome the FET’s Miller effects. Finally,
a combination of both high slew rate and high current
is needed to complete the gate drive cycle.
There are a variety of applications for MOSFET
drivers - each with its own unique set of speed and
peak current requirements. Most general purpose
drivers feature 1.5 amp peak totem-pole outputs
which deliver rise and fall times of approximately 40
nanoseconds into 1 nanoFarad. Propagation delays
are in the neighborhood of 40 to 50 nanoseconds,
making these devices quite adaptable to numerous
power supply and motor control applications. These
specifications can be used for a comparison to those
of a new series of higher speed and higher current
devices, specifically, the UC1708, UC1710 and the
UC1711 power MOSFET drivers. Each member in
this group of “third” generation driver ICs features
significant performance improvements over their
predecessors with one parameter optimized for a
specific set of applications.
At turn-off this sequence is reversed, first demanding both high slew rate and high current simultaneously. This is followed by the plateau region which
is limited only by the maximum driver current. Finally, there is high speed discharge of the gate to
zero volts.
3-318
U-137
APPLICATION NOTE
MOSFET DRIVER IC FEATURE AND PERFORMANCE OVERVIEW
TABLE 2.
UC1708
UC1770
UC1711
Number of outputs
Peak output current (per output)
2
3A
1
6A
2
I.5 A
Noninverting input-output logic
Inverting input-output logic
YES
YES
YES
YES
Maximum supply voltage Vcc
Typical supply current ICC (1.)
35v
16ma
20V
30ma
40V
17ma
Remote Enable
Shutdown Input
YES
YES
YES (2)
Feature
Seperate grounds, signal and power
Seperate Vin and Vc pins
8 pin DIL package
16 pin DIL package
5 pin TO-220 package
YES (3)
YES (3)
YES (3)
YES
YES
YES
YES
YES
Note 1. Typical Vc plus Vcc current measured at 200KHZ, 50% duty cycle and no load
Note 2. Using the device’s other input
Note 3. Package dependent
UC1711 BLOCK DIAGRAM
PROPAGATION DELAYS
The power supply industry’s trend towards higher
power densities has thrust switching frequencies
well beyond one megaHertz in many low to medium
power systems. With a one microsecond total conversion period, or less, the FET switching transitions
should be in the order of low tens of nanoseconds
to yield high efficiency. Additionally, the propagation
delays from the driver input to output should be
around ten nanoseconds for quick response.
UC1711
The UC1711 device features typical propagation
delays of three and ten nanoseconds at no load,
depending on the transition. Coupled with dual 1.5
amp peak totem-pole outputs, this device is optimized for high frequency FET drive applications. Its
all NPN Schottky transistor construction is not only
fast, but radiation tolerant as well.
3-319
YES
YES
APPLICATION NOTE
U-137
GATE DRIVE POWER CONSIDERATIONS
The gate power utilized in charging and discharging
a capacitor at frequency “F” is:
Perhaps the most popular misconception in the
power supply industry is that a FET gates require NO
power from the auxiliary supply - that both turn-on
and turn-off are miraculously power free. Another
fallacy is that the driver consumes all the measured
supply current, ICc, and none of it is used to transition
the gates. Obviously, both of these statements are
false.
Substituting the gate charge for capacitance multiplied by voltage (Q=C*V) in this equation results in:
In reality, the power required by the gate itself can be
quite substantial in high frequency applications.
Calculation of this begins by listing the specified total
gate charge for the FET device, Qg.
The gate power required verses FET size and
switching frequencies is tabulated for some common
applications in Table 3. Table 4. transforms this
power into driver input current at a nominal 12 volt
bias.
P(cap) = C * VA2 * F
P(gate) = Qg * V * F
GATE POWER (mW) VS. SWITCHING FREQUENCY AND FET SIZE
SWITCHING FREQUENCY (kHz)
FET
SIZE
50
100
150
200
250
500
750
1MEG
SIZE 1
10
18
28
36
46
90
136
180
SIZE 2
16
30
46
60
76
153
226
300
SIZE 3
28
54
82
108
136
275
406
504
SIZE 4
48
96
144
192
240
480
720
960
SIZE 5
100
200
300
400
550
1W
1.5W
2W
SIZE 6
144
288
432
576
720
1.4W
2W
>2w
Table 3.
DC SUPPLY CURRENT (mA) VS. SWITCHING FREQUENCY AND FET SIZE
SWITCHING FREQUENCY (kHz)
50
100
150
200
250
500
750
1 MEG
SIZE 1
1
1
2
4
5
6
10
12
SIZE 2
1
2
4
5
6
10
16
20
SIZE 3
2
4
6
8
10
16
26
36
SIZE 4
4
8
10
12
16
32
48
64
SIZE 5
8
14
20
26
32
66
100
130
SIZE 6
10
20
28
38
48
96
144
190
Table 4.
3-320
APPLICATION NOTE
U-137
The driver output stage can be modelled as a
resistance to the respective auxiliary supply rail
driving an ideal FET capacitor. All of the power used
to charge and discharge the MOSFET gate capacitor is completely transferred into heat by the driver.
This gate power loss adds to the driver’s own power
loss - resulting in a net driver power dissipation
equal to it’s input voltage, Vcc, multiplied by the sum
of the gate and driver currents, lg + ICC. This can
be calculated or determined empirically by measuring
the driver DC input voltage and current.
This term is then added to the ambient temperature
to yield the resulting junction temperature, Tj.
If the driver is thermally attached to a heatsink or
“cold plate”, then the thermal impedance from the
device junction to it’s package case, theta JC (Ojc),
is used to determine the thermal rise. Likewise, this
thermal rise is added to the heatsink temperature to
determine the junction temperature. In either case,
the maximum junction temperature (tj(max)) should
be determined and checked against the device’s
absolute maximum specification.
THERMAL CONSIDERATIONS
Proper IC package selection and/or device
heatsinking is the only method available to insure a
safe operating junction temperature, tj. All IC’s are
specified and graded for various junction temperature
ranges, and priced accordingly. As a precaution, it
should be noted that using a device outside its
tested temperature range can result in poor performance, parameters which run outside their specifications, and quite possibly - no operation at all.
JUNCTION TEMPERATURE
The junction temperature of the driver IC is obtained
by first calculating the device’s thermal rise above
the ambient temperature. This is obtained by multiplying the average input power (Vin*lin) by the
device’s thermal impedance to air, theta JA (Oja).
Average supply currents for each of the three drivers
of interest varies primarily with the switching frelisting each driver
quency. Rather than
independently, an rough approximation of 25
milliamps will be used as the driver current, regardless of the specific device utilized and switching
frequency. In addition, a typical supply voltage of 12
volts results in a power dissipation by the driver itself
of 300 milliwatts.
The calculated gate power of Table 5. has been
added to the estimated 300mW of device power to
formulate Table 6. - the driver total power dissipation. This is of particular interest in selecting a driver
package (8 pin, TO-220, etc.) and heat sink determination for a specific maximum junction temperature, or
rise. Typical junction temperature rises vs. frequency
and FET size for a IC package, and recommendations are shown in table 7.
AVERAGE POWER DISSIPATION (mW) VS. FREQUENCY AND FET SIZE
SWITCHING FREQUENCY (kHz)
FET SIZE
50
100
150
200
250
500
750
1MEG
SIZE 1
310
318
328
336
346
390
436
480
SIZE 2
316
330
346
360
376
452
526
600
SIZE 3
328
354
382
408
436
570
706
840
SIZE 4
348
396
444
492
540
780
1.0W
1.3w
SIZE 5
400
500
600
700
800
900
1.7W
2.4W
SIZE 6
444
588
732
876
1.OW
1.7W 2.5W
3.1w
Table 5.
3-321
APPLICATION NOTE
U-137
PACKAGE RECOMMENDATIONS
SWITCHING FREQUENCY (kHz)
For P(diss) = or < 500mW
A: 8 pin DIL, <40 C rise
B: 8 pin DIL, <45 C rise
C: 8 pin DIL, <50 C rise
For P(diss) = or > 500mW
(using heatsink)
D: 8 pin DIL, <40 C rise
E: 8 pin DIL, <50 C rise
For P(diss) > 500mW
F: TO-220 recommended
500 750
1 MEG
200
250
B
B
B
C
D
D
B
B
C
D
D
D
B
C
D
D
D
E
F
B
C
D
D
D
F
F
F
SIZE 5
C
D
D
E
F
F
F
F
SIZE 6
D
D
E
F
F
F
F
F
50
100
SIZE 1
A
A
SIZE 2
A
SIZE 3
B
SIZE 4
150
B
Table 6.
HIGH POWER APPLICATIONS
Most high power applications require the use of
“monster” MOSFETs or several large FETs in parallel for each switch. Generally, these are low to
medium frequency applications (less than 200kHz)
where obtaining a low Rds(on) is of primary concern
to minimize the DC switch loss. It is not uncommon
to find two, three and even four large devices used
in parallel, although some of these combinations are
unlikely from a cost versus performance standpoint.
Table seven displays the individual FET device
characteristics and several popular parallel arrangements. Listed in descending order is Rds(on)
at room temperature and the total gate charge
required. This will ultimately be used to determine
the gate drive current in Table 8., total power dissipation in Table 9., and driver IC recommendation in
Table 10 for various applications.
PARALLELED MOSFET CHARACTERISTICS - TABLE 7.
MOSFET
ARRANGEMENT
Rds(on)
effective
Qg(nC)
total
1 X SIZE 4
1 X SIZE 5
1 X SIZE 6
2 X SIZE 4 (1)
3XSlZE4(1)
4 X SIZE 4 (1)
0.85
0.40
0.27
0.425
0.283
0.213
63
130
190
126
189
252
MOSFET
ARRANGEMENT
2X
2X
3X
4X
3X
4X
SIZE
SIZE
SIZE
SIZE
SIZE
SIZE
Rds(on)
effective
Qg(nC)
total
0.200
0.135
0.133
0.100
0.090
0.068
260
380
390
520
570
760
5
6
5 (1)
5 (1)
6 (2)
6 (2)
1. Consider another selection 2. Consider a “Monster” FET
AVERAGE SUPPLY CURRENT (mA) VS. FREQUENCY AND FET SELECTION
SWITCHING FREQUENCY (kHz)
FET
ARRANGEMENT
Rds
mohm
25
50
75
100
150
200
2 X SIZE 5
200
31
39
45
51
65
77
2 X SIZE 6
135
35
45
53
63
83
101
3 X SIZE 6
90
39
53
69
73
91
139
4 X SIZE 6
68
45
63
82
101
139
177
*Includes 25mA of driver supply current
Table 8
3-322
APPLICATION NOTE
U-137
POWER DISSIPATION (mW) VS. FREQUENCY AND APPLICATION
SWITCHING FREQUENCY (kHz)
FET
ARRANGEMENT
Rds
mohm
25
50
75
100
150
200
2 X SIZE 5
200
372
468
540
612
780
924
2 X SIZE 6
135
420
540
636
756
1.0W
1.2W
3 X SIZE 6
90
468
636
828
876
1.1W
1.7W
4 X SIZE 6
68
540
756
984
1.2W
1.7W
2.1W
Table 9.
* Includes 300mW of driver dissipation
DRIVER IC AND PACKAGE SELECTION GUIDE
Selection Guide
for < 50 C rise
A: 8 pin DIL or
20 pin PLCC
B: 8 pin DIL
with heatsink.
or TO-220.
C: TO-220 with
heatsink
SWITCHING FREQUENCY (kHz)
FET
ARRANGEMENT
Rds
mohm
25
50
75
100
150
200
2 X SIZE 5
200
A
B
C
C
C
C
2 X SIZE 6
135
B
C
C
C
C
C
3 X SIZE 6
90
B
C
C
C
C
C
4 X SIZE 6
68
C
C
C
C
C
C
Table 10.
UC1710 DRIVER PERFORMANCE
Table 11. shows the typical response of the UC1710
“Miller Killer” driving a single APT5025BN (size 6)
device and paralleled MOSFET combinations for
reference.
Although capacitive in nature, the FET “Miller” effects and demands on the driver differ significantly
than a true capacitor load as previously described.
UC1710 RISE, FALL AND DELAY TIMES VS. LOADS
TEST
CONDITIONS
Tp
LH
Tt
LH
Tp
HL
Tt
HL
Tp
+Tt
LH
Tt
+TP
HL
NO LOAD
VDS
28
12
36
12
40
50
ONE
0
28
26
38
30
54
68
APT5025
350
28
35
40
30
63
70
TWO
0
28
38
40
36
66
76
APT5025
350
28
48
42
38
76
80
THREE
0
28
48
42
48
76
90
APT5025
350
28
60
44
58
88
92
Table 11.
3-323
U-137
APPLICATION NOTE
PERFORMANCE COMPARISONS
“HOMEBREW” TOTEM-POLES VS.
INTEGRATED CIRCUIT DRIVERS
The prior lack of “off-the-shelf” high current or high
speed drivers had prompted many to design their
own gate drive circuits. Traditionally, an NPN-PNP
emitter follower arrangement had been used in
lower frequency applications as shown in Figure 7.
input drive waveform is above Vgs(th) of the N
device and below that of the P device. One technique
to minimize the cross conduction peak current is to
add some resistance between the FETs. While this
does minimize the “shoot-through” current, it also
limits the peak current available to the load. This
somewhat defeats the purpose of using the
MOSFETs in the first place to deliver high currents.
The resistor serves an additional purpose of damping
the gate drive oscillations during the transitions. In a
practical application, two resistors can be used in
the place of one with the center-tap connecting to
the FET gate, or load as shown in figure 9.
Figure 7
Figure 8
For higher speed applications, a P and N channel
FET pair can be used as shown in figure 8. The
circuit is configured with the P channel MOS as the
upper side switch to simplify the auxiliary bias.
Otherwise, a gate drive potential of ten volts above
the auxiliary bias would required.
The performance of the circuit in figure 9 was
evaluated and compared to that of the UC1710
driver into a 30 nanoFarad load. A size three P type
FET and a size two N channel device were connected
in series with two one-half ohm resistors to limit the
shoot-through current. These FETs were driven
from the UC1711 dual driver which can deliver 3
Amp peak gate drive currents for rapid transitions.
The results of this test are shown in figure 10.
Driver Performance into 30nF load
Unfortunately, this configuration has a few drawbacks First, it leads to an inverting logic flow from
the driver input to its output, complicating matters
especially during power-up and power-down sequences. Without a clever undervoltage lockout
circuit the main power switch will tend to be ON as
the auxiliary supply voltage is raised or lowered
while the PWM is OFF.
Cross conduction of both FETs is unavoidable with
this configuration due to the difference between the
gate threshold voltages of each device. Both P and
N channel devices are cross conducting while their
3-324
Lines: solid=UC3710, dashed=discrete
Figure 10. - VERT: 5V/DIV: HORIZ: 50 nS/DIV
APPLICATION NOTE
U-137
The test results indicate very similar performance
into this load from either technique. Obviously, the
“homebrew” approach utilizes a total of three devices in comparison to a single UC1710 driver to
obtain essentially the same high speed performance.
Additionally, the cost of the P channel FET alone
may exceed the price of the UC1710 device, not to
mention the difference in PC board real estate.
As a final note, the discrete FET approach
required over 10 milliamps more supply current
than the single UC1710 driver or a increase in
supply current of twenty percent. Results of
this test shown in figures 11 and 12.
RISE AND FALL TRANSITION PERFORMANCE INTO 30 nF
FALL TIMES (Fig 12.)
RISE TIMES (Fig 11.)
PHOTO SCALES (BOTH): VERT=2V/DIV, HORIZ=10 nS/DIV
LINES: SOLID = UC3710; DASHED = DISCRETE CIRCUIT OF FIGURE 9.
POWER DEVICES
IGBTs and MCTs: While existing generations
of power MOSFETs continue to be enchanced
for lower RDS(on) and faster recovery internal
diodes, alternative new devices have also been
introduced. Among the most popular, and
viable for high voltage high power applications
are IGBTs (Insulated Gate Bipolar transistors)
and MCTs (MOS Controlled Thyristors). Although
frequently drawn as an NPN structure, the IGBT
actually resembles a PNP bipolar transistor with
an internal MOS device to control the base
drive. Indicative by its description, the MCT is
essentially an SCR structure also utilitzing a
MOS drive stage. Both devices offer significant
cost advantages over MOSFETs for a given
power capability.
MOSFET, IGBT and MCT Gate Drives: There are
numerous reasons for driving the MOSFET gate
to a negative potential during the device’s off
state. Degradation of the gate turn-on threshold
over time and especially following high levels of
irradiation are amongst the most common.
However, with IGBTs, the important concern is
the ability to keep the device off following turnoff with a high drain current flowing. On larger
IGBT’s with ratings up to 300 Amps, inductive
effects caused by the device’s package alone
can “kick” the effective gate-to-emitter voltage
positive by several Volts at the die - even with
the gate shorted to the emitter at the package
terminals Actually, this is the result of the high
current flowing in the emitter lead (package)
inductance which can less than 1nH. The
corresponding voltage drop changes polarity at
turn off, thus pulling the emitter below the
gate, or ground. If high enough, a fast turn off
will be followed by a parasitic turn-on of the
3-325
APPLICATION NOTE
U-137
switch, and potential destruction of the
semiconductor. Applying the correct amplitude
of negative gate voltage can insure proper
operation under these high current turn-off
conditions. Also, the negative bias protects
against turn-on from high dv/dt related changes
that could couple into the gate through the
“Miller” capacitance.
Figure 13 - IGBT and MCT Diagrams
Unlike power MOSFET switches, IGBT
transconductance continues to increase with
gate voltage. While most MOSFET devices peak
with about 10 to 12 Volts at the gate, IGBT
performance steadily improves up to the
suggested 16 Volt maximum gate voltage.
Typically, most IGBT manufacturers recommend
a negative drive voltage between -5 and -15V.
Generally, it is most convenient to derive a
negative voltage equal in amplitude to the
positive supply rail, and ±15V is common.
The gate charge required by an IGBT (for a given
voltage and current rating) is noticeably less
than that of a MOSFET. Part of this is due to the
better utilization of silicon which allows the
IGBT die to be considerably smaller than its FET
counterpart. Additionally, the IGBT (being a
bipolar transistor) does not suffer from the
severe “Miller” effects of the MOS devices,
easing the drive requirements in a given
application.
However, because of their
advantages, most available IGBTs have fairly
high gate charge demands - simply because of
their greater power handling capability.
In contrast, MCTs (MOS Controlled Thyristors)
exhibit the highest silicon utilization level among
power switching devices. While relatively new
to the market, these devices are quickly gaining
acceptance in very high power (above several
kilowatts) applications because of their high
voltage (1000V) and high current (to 1000A)
capability. Recently introduced parts boast
maximum ratings to one megawatt, ideal for
large industrial motor drives and high power
distribution-even at the substation level. These
devices are essentially MOS controlled SCRs and
are intended for low frequency switchmode
conversion. They will most likely replace high
power discrete transistors, Darlingtons and SCRs
because of their higher efficiency and lower
cost.
Gate Charge and Effective Capacitance
with Negative Bias: While several MOSFET
and IGBT manufacturers recommend negative
gate voltages in the device’s off state, few
publish any curves or information about gate
charge characteristics when the gate is below
zero Volts. This complicates the gate drive
circuit design as each IGBT, MOSFET or MCT
switch must be evaluated by the user over the
ranges of operation conditions. A test fixture
as shown in Figure 14 can be used to provide
empirical generalizations for devices of interest.
A switched constant current source/sink has
been configured using a simple dual op-amp to
drive a “constant” 1mA at the device under test
(DUT). Gate voltage versus time can be
monitored which provides the exact gate charge
requirements for a given device. Any application
specific requirements can also be accommodated
by modifying the test circuit with external
circuitry.
Negative Gate Charge - Empirical Data:
Several MOSFET, IGBT and MCT gate charge
measurements were taken to establish the
general characteristics with negative gate charge
and effective capacitance during this third
quadrant operation was calculated and compared
to of the first quadrant specifications from the
manufacturers data sheets.
Figure 15
demonstrates the general relationships of gate
charges for comparison.
Both the IGBT and MCT have similar negative
bias gate charge requirements as with an applied
positive bias. The MOSFET, however, exhibits a
slightly reduced gate charge in its negative bias
region, somewhere between 70 and 75 percent
of its positive bias charge. The MOSFET’s more
significant “Miller” effect in the first quadrant is
responsible for this since the higher effective
capacitance during the plateau region does not
occur with negative bias.
3-326
U-137
APPLICATION NOTE
Figure 14 - Gate Charge Test Circuit
VERT 5V/DIV
HORIZ 5OuS/DIV
Figure 15 - Gate Charge Comparison Low to
High Transition
VERT 5V/DIV
Total Gate Power - Negative Drive Voltage
Applications: All of the previously presented
gate power equations still apply, however they
must be modified to include the additional
charge requirements of the negative supply
For the sake of simplicity, a
voltage.
multiplication factor can be used for recalculation
of the exact figures. When identical amplitudes
of positive and negative supply voltages are
used, for example ±15V, then the gate power
utilized can be simply multiplied by a factor of
two. This completes the process for the IGBTs
and MCTs. The total MOSFET gate charge, on
the other hand, should only be multiplied by a
factor of 1.7 to 1.75 to accommodate the
reduced negative bias demands. Additionally, if
a negative supply voltage different than the
positive rail voltage is used, for example +15
and -5, then the scaling factor must be adjusted
accordingly. In this case, the new total gate
power would be 1+ (-5/-15) or 1.33 times the
initial 0-15V gate power for IGBTs and MCTs.
The negative drive voltage scaling factor (-5/15) would be multiplied by the 70 to 75% index
if a MOSFET were used instead of an IGBT or
MCT. This would result in a 1.23 to 1.25 times
net increase over the initial (0-15V) gate power
demand.
HORIZ 50uS/DIV
Figure 16 - Gate Drive Comparison High to
Low Transition
3-327
APPLICATION NOTE
U-137
SUMMARY
The need for higher speed and higher current FET
driver ICs has become increasingly apparent as
power conversion switching frequencies are pushed
towards and beyond one megaHertz. Likewise, the
quest for higher overall efficiencies has resulted in
creation of large, even “monster” size MOSFET
geometries. These industry trends have stimulated
the development of innovative MOSFET driver ICs
- ones which would significantly outperform any of
their predecessors, including discrete versions.
A new generation of high speed and high current
MOSFET drivers has been presented. Each optimized for a unique blend of these attributes, the
UC1708, UC1710 and the UC1711 devices suc-
cessfully conquer the challenges of obtaining
rapid transitions in MOSFET gate drive circuits.
REFERENCES
UNITRODE Application Note U-118, ”New Driver
IC’s Optimize High Speed Power MOSFET Switching
Characteristics”, UNITRODE LINEAR IC DATABOOK, IC 600
INTERNATIONAL RECTIFIER Application Notes
AN-937, AN-947 and Datasheets, I.R. HEXFET
Power MOSFET Designers Manual HDB-4
ADVANCED POWER TECHNOLOGY Databook
1989
HIGH CURRENT FET DRIVER CIRCUITS
TYPE
DESCRIPTION
KEY FEATURES
UC1705/3705
High Speed Power Driver
(Single ended)
1.5A
High
Low
Low
UC1706/3706
Dual High Current
MOSFET Compatible
Output Driver
16 Pin
DIL
“Batwing”
UC1707/3707
Dual Uncommitted High
Current MOSFET
Compatible Output Driver
Dual, 1.5A Totem Pole Outputs
Parallel or Push-Pull Conversion
(1706Series)
Internal Overlap Protection
Analog, Latched Shutdown
High-Speed, Power MOSFET Compatible
Thermal Shutdown Protection
5 to 40V Operation
Low Quiescent Current
UC1708/3708
Dual Non-Inverting
Power Driver
3.0 Peak Current Totem Pole Output
5 to 35V Operation
25n Set Rise and Fall Times
25n Set Propagation Delays
Thermal Shutdown and Under-Voltage
Protection
High-Speed, Power MOSFET Compatible
Efficient High Frequency Operation
Low-Cross-Conduction Current Spike
Enable and Shutdown Functions
Wide Input Voltage Range
ESD Protection to 2kV
8-Pin
DIL
16-Pin
DIL
UC1709/3709
Dual High
Speed FET Driver
1.5A Source/Sink Drive
Pin Compatible with 0026
40ns Rise and Fall into 1000pF
Low Quiescent Current
UC1710/3710
High CurrentSpeed
FET Driver
10A Peak Current Capability
40ns Rise and Fall Times
40ns Delay Times (1Nf)
Low Saturation Voltage
UC1711/3711
Dual Ultra High
Speed FET Driver
25nS Rise and Fall into 1000pF
15nS Propagation Delay
1.5Amp Source or Sink Output Drive
Operation with 5V to 35V Supply
High-Speed Schottky NPN Process
8-PIN Mini-DIP Package
Radiation Hard
8-Pin
DIL
UC3724
UC3725
(PAIR)
Isolated High Side Drive
for N-Channel
Power MOSFET Gates
Fully Isolated Drive for High Voltage
0% to 100% Duty Cycle
600kHz Carrier Capability
Local Current Limiting Feature
8-Pin
DIL
(Pair)
CORPORATION
7 CONTINENTAL BLVD.. MERRIMACK, NH 03054
TEL. (603) 424-2410 l FAX (603) 424-3460
3-328
TotemPole Output
Speed MOSFET Compatible
Quiescent Current
Cost Package
PACKAGE
8 Pin
DIL
5 Pin
TO-220
8-Pin
DIL
8- Pin
DIL
5- Pin
TO-220
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