200322A.pdf

APPLICATION NOTE
APN1012: VCO Designs for Wireless Handset and
CATV Set-Top Applications
Introduction
Voltage Controlled Oscillators (VCOs) have come to the forefront
of RF designs together with the first PLL circuits. In the era
before the PLL, oscillators were mostly free running, and only in
rare cases were varactors used for modulation or temperature
compensation. Nowadays, we rarely see free running oscillators,
instead they have become varactor-controlled oscillators. This is
because most RF applications require band coverage, which can
be realized through the PLL circuit requiring two sources of RF
power. The reference source frequency is often a VCXO or TCXO,
while the other frequency is controlled by the PLL phase detector.
Usually, both VCXO/TCXO and the RF VCO are voltage controlled
oscillators. The difference between a reference oscillator and a
tuned VCO is that the former usually has a very high Q resonator,
which allows for very stable oscillation, while the latter has a
lower Q resonator, allowing a relatively high tuning range. In reference oscillators, varactors are used for fine tuning or
temperature compensation (TCXO). In tunable oscillators, varactors are used to change (tune) the frequency. In some VCOs,
varactors may be used also for modulation, for example in a
DECT system where modulation is used to generate a constant
envelope GMSK signal.
Although it is a small part of the RF design, the VCO is often a
major headache for designers. The goal, in this application note,
is to show how Skyworks products and services may help you to
overcome VCO concerns and help to make your design among
the best products on the market.
Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com
200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005
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APPLICATION NOTE • APN1012
VCOs in Digital Wireless Phones
verted or demodulated into a digital I/Q signal using a lower frequency IF VCO. The transmitter path is either directly modulated
at 900 MHz or uses a dual conversion scheme requiring at least
two VCOs.
Consider the hypothetical wireless handset phone. Today, the
handset is a dual-band (cellular/PCS) and multimode system
employing many VCO functions. There are many ways to realize
these functions, making it virtually impossible to specify the frequency and tuning range for all designs. However, there are
certain common features that are outlined in Figure 1.
When dual-band requirements are needed, up to eight or more
VCOs may be required to satisfy specific frequency plans. This is
often a technically and economically restrictive solution. Many
designers try to solve this over-VCOed problem using both smart
frequency planning and multiband VCOs, as shown in Figure 1.
In a typical receiver, dual conversion superheterodyne solutions
are usually employed. They convert either 900 MHz (cellular) or
1.8 GHz (PCS) down to the SAW frequency range, which may be
between 90–400 MHz. Further, this signal is either downcon-
PLL
Control
PLL
Control
PLL
RF VCO
PLL
I
LNA
BPF
BPF
π/2
BPF
Q
RF VCO Ranging:
400–1900 MHz
T/R
Switch
Coupler
I
PA
BPF
IF VCO
Ranging:
100–400 MHz
BPF
BPF
Σ
π/2
Q
RF VCO
PLL
RF
Detector
PLL
PLL
Control
IF VCO
PLL
Control
Figure 1. VCOs in a Digital Wireless Phone
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July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A
APPLICATION NOTE • APN1012
Fundamental Low Noise Colpitts VCO
Without delving deeply into phase noise theory, we note that
phase noise is inversely proportional to the power bypassed
through the feedback loop, and the loaded Q of the tank circuit.
Thus, the more power lost on the way to the transistor base, the
higher the noise. It is clear that varactor loss plays a crucial role
in the phase noise property of the VCO. If phase noise is an issue,
the varactor series resistance should be carefully considered.
The characteristic feature of the Colpitts VCO is that it uses a
capacitive divider for the feedback consisting of C1 and C2, and
an inductive branch including a parallel resonator and series
capacitor C3. The parallel resonator includes inductive element
M1 (that may be a discrete inductor for lower frequencies or a
length of microstrip line for RF) and a capacitive branch, consisting of a varactor and a series capacitor(s). The entire
inductive branch should have inductive impedance at the frequency of oscillation, otherwise there will be no oscillation. This
means that the resonant frequency should be higher than the
oscillation frequency.
There is an additional concern because phase noise is not only a
function of varactor loss. The varactor capacitance voltage characteristic has a crucial impact on phase noise as well. With a
higher capacitance ratio, the varactor’s coupling to the resonator
is reduced, resulting in lower resonator current. Therefore, a
hyperabrupt varactor having higher series resistance is often a
better choice than a lower capacitance ratio abrupt varactor
having lower series resistance.
Note that the resonator current circulates through the varactor,
series capacitor C11 and inductor M1 and is the largest current in
the tank circuit. Because of this, losses introduced in this current
path are the crucial ones with respect to phase noise.
M6
5 x 0.2 mm
C11
1.5 p (0603)
C7
0.5 p (0402)
R2
2.7 k
M3
4 x 0.35 mm
C3
0.75 p
D1
SMV1493
M2
6.5 x 0.2 mm
M1
C10
15 p
R4
3.6 k
Q2
2SC5008 (44)
Q1
2SC5007 (34)
C9
C12
R1
1p
150
VCC
R3
200
C4
M4
2 x 0.2 mm
C1
2p
C2
1.5 p
C8
C5 0.75 p
0.75 p
RF Out
C6
0.75 p
M5
2 x 0.2 mm
4 x 04 mm (Trim)
VTUNE
Low RS, Low Voltage
Hyperabrupt Varactor
Figure 2. Low Noise, High-Performance Colpitts VCO
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200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005
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APPLICATION NOTE • APN1012
Differential VCO for Integration with an RF IC
and a differential VCO. In the Colpitts case, the resonant frequency is always higher than the oscillation frequency; in the
differential VCO the resonant and oscillation frequencies may
coincide. Thus the loaded Q of the circuit becomes significantly
higher, and the feedback loop losses are increased due to the
higher resonant currents. When this happens, the differential VCO
is more vulnerable to resonator loss than the Colpitts VCO and
usually shows 5–10 dB higher noise if compared to an equivalent
Colpitts case.
Designs based on RF IC solutions, with built-in VCOs, often
employ a differential VCO configuration. One possible differential
VCO configuration is shown in Figure 3. In this case, the tank circuit is formed by C3, C4 and a resonator C8, C9, D1, M1. Here
again, the resonator current plays a decisive role in phase noise
definition. Thus, phase noise is strongly dependent on resonator
loss. Capacitors C3 and C4 help establish the correct phase shift
value in the feedback loop moving oscillations closer to the resonant frequency. This is the principal difference between a Colpitts
VCC 3 V
RF IC
R4
3k
R2
3 k R7
3k
L1
33 n
R1
47
C1
100 p
C4
1.5 p
C5
1.5 p
RF Out
V2
V1
C2
2 p R9
3.9 k
R3
9.1 k
C7
100 p
C3
4p
R6
3k
R5
3k
R10
51
C9
6p
V3
C8
2p
R8
20
M1
4 x 0.5 mm
M2
6.5 x 0.2 mm
D1
SMV1493-079
VVAR
C6
100 p
Low RS, Low Voltage
Hyperabrupt Varactor
Figure 3. Differential VCO for the Integration with the RF IC
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APPLICATION NOTE • APN1012
Dual-Band Switchable VCO Schematic
time. Thus, a trick is used — connecting a capacitor C11 in parallel with C6 when a lower band resonator is selected. This
provides significant improvement in phase noise since C6 may
then be optimized for the best performance at high band, and C11
at the lower band.
One way to improve design economics in the multi-VCO requirement is to employ band switching in the VCO. If the frequency
switching required isn’t very large (say within 20%), it may usually be realized within the same tank circuit, by switching “on” or
“off” an additional capacitor or inductor. However, if the required
switching is more than 30%, it becomes very difficult to satisfy
both wideband and low noise requirements in a single design.
One possible solution is to use two separate tank resonator circuits switched with two PIN diodes. In this case, the feedback
needs to be optimized to fit both band requirements at the same
Another important feature of this switching scheme is that the
PIN diodes are not in the resonator current path. Because of this,
phase noise is not sensitive to the PIN diode resistance. This is
fortunate, since it means less forward current is needed. In addition, any noise on the PIN diode bias current (common for the
noisy digital environment of today’s phones) would not cause significant modulation noise.
J3
1
C4
100 pF
R4
1.5 k
VSW_Low
J4
D2
SMV1139-011
VTUNE
R1
300
C3
100 pF
C8
10 pF
L1
12 nH
C1
8 pF
P2
SMP1320-011
R6
3k
SMP1320-011
C2
470 pF
C5
100 pF
Q1
NE68519
C9
100 pF
P1
R3
1.5 k
D1
SMV1408-011
Low RS, Low Voltage
Hyperabrupt Varactor
1
VCC 3 V
CCC
100 pF
L2
56 nH
J1
1
Low Current Switching
PIN Diodes
R7
6.8 k
C6
20 pF
C7
15 pF
C10
20 pF
R5
100
J5
1
RF
C11
30 pF
R2
1.5 k
J2
1
VSW_High
Figure 4. Dual-Band Switchable VCO Schematic
Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com
200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005
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APPLICATION NOTE • APN1012
Dual-Band Switchable RF VCO
As mentioned before, relatively small (less than 20%) frequency
switching may be achieved inside the same tank circuit by connecting or disconnecting capacitors (and sometimes inductors).
The PIN diode D2 performs a tricky task — it adds more capacitance in parallel with the existing parallel capacitance of the
resonator, and also adds more capacitance in parallel with the
VVAR
existing series capacitor. This technique is used to overcome the
problem of increased resonator Q, when connecting additional
parallel capacitance, by decreasing it with higher series capacitance. It allows D2 to keep phase noise near its optimum at both
bands. Another PIN diode in the output matching circuit tunes the
buffer to a frequency doubler mode when working in PCS band.
VCC (3 V)
VCTL2
D1: SMV1493-079
D2, D3: SMP1322-079
RF Out
D3
D2
D1
VCTL1
PINs are Used in Both Resonator Tank
and Output Matching Circuits
Figure 5. Dual-Band Switchable RF VCO
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APPLICATION NOTE • APN1012
VCOs in a Set-Top Cable Downconverter
The typical set-top downconverter is a dual-conversion receiver
employing upconversion and downconversion techniques to
overcome image problems in a wideband environment of
50–1000 MHz. In the dual up/downconversion scheme, the
problem of image channel and input filtering virtually does not
exist because there is no signal at the image channel. The
image channel is always higher than the highest frequency of
the cable signal.
Two RF VCOs are required for dual downconversion. The first is a
wideband VCO tuned from 1100–2000 MHz with a control voltage
from 1–20 V. The other is a narrow-band VCO, which may use a
CDR, coaxial dielectric resonator, at 1144 MHz. In a digital
system, the second IF signal may be further demodulated,
requiring an additional 44 MHz VCO.
The specific action of the wideband VCO is its wideband tuning
requirement. Let us consider some possible solutions for the
wideband VCO.
AGC
54–860 MHz
40 dB
75 Ω
HPF
LPF
2nd Mixer
1100 MHz
45.75/44 MHz
BPF
dB
Upstream
filter
LPF
1st Mixer
PLL
BPF
PLL
1154–1960 MHz
PLL
Control
Upstream
Low-Distortion PIN Diode
Attenuator
1144/1145 MHz
PLL
Control
Wideband VCO Tuning in
1100–2000 MHz Range
Narrow-Band VCO
@ 1145 MHz
Figure 6. VCOs in a Set-Top Cable Down-Converter
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200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005
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APPLICATION NOTE • APN1012
Wideband Colpitts VCO Schematic
band. Back-to-back varactors are often used to minimize parasitic mounting capacitance (between mounting pads and
adjoining components). This circuit is usually designed to minimize any parasitic parallel capacitance that may be caused by
component pads or transmission lines close to the inductive path.
The unique action of the wideband Colpitts VCO is in its tank circuit design, which uses an inductor with a varactor connected in
series and no parallel capacitor, in contrast to the low noise
Colpitts VCO described in Figure 7. The feedback capacitors are
optimized to the best power flatness over the entire frequency
VCC = 5 V
ICC = 9 mA
3.3 k
NE68519
VTUNE
560 p
320 x 30 mils
5.6 nH
1p
SMV1265-011
2p
3k
3k
300 p
9.1 k
SMV1265-011
RF Output
1.62 p
200
High C (V) Ratio
Hyperabrupt Varactors
Figure 7. Wideband Colpitts VCO Schematic
part of the design. Skyworks new SMV1265-011 varactor is
specifically designed to fit this wideband application.
7
2.2
2.1
2.0
1.9
1.8
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.0
0.9
6
5
4
3
POUT (dBm)
Useful Tuning
Range:
980–2120 MHz
Frequency (GHz)
A carefully designed layout with minimum parasitic capacitances
may show large frequency coverage, for example 980–2120 MHz
as the performance indicates. The varactor selection is a crucial
2
1
0
0
5
10
15
20
25
30
Varactor Voltage (V)
Fexp
Fmodel
POUT_exp
POUT_model
Figure 8. Wideband Colpitts VCO Performance
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APPLICATION NOTE • APN1012
Wideband Balanced VCO Schematic
An even wider tuning range may be achieved with a balanced
VCO configuration. The reason for its wider tuning performance is
that the phase response of this VCO’s active element is flatter
C6
VCC1
1000
5–8 V
over the range of tuning compared to a Colpitts VCO. This allows
the tank circuit more control over the oscillation frequency.
The best results are achieved with back-to-back connected
SMV1265 varactors, where there is 820–2120 MHz coverage.
R10
2.4 k
R3
120
R6
R5
V2
C5
100
T1
NE68119
R12
5
0
R7
C4
51
100
820
R8
1k
820
R4
120
V1
16 x 0.4 mm
NE68119
L1
R1
R2
L2
33 nH
33
33
33 nH
A
C2
C1
10
10
T2
15 x 0.7 mm
D1
D2
High C (V) Ratio
Hyperabrupt Varactors
T3
SMV1265
3 X 0.7 mm
R9
1k
SMV1265
R11
1000
C3
100
VVAR1
Figure 9. Wideband Balanced VCO Schematic
8
2.2
Measured
Measured @ 7 V
6
2.0
Simulations
1.6
Useful Tuning
Range:
820–2120 MHz
1.4
1.2
Power (dBm)
Frequency (GHz)
4
1.8
2
Simulated @ 7 V
0
-2
-4
Measured @ 5 V
1.0
-6
0.8
-8
0
5
10
15
20
25
30
0
5
10
15
20
Varactor Voltage (V)
Varactor Voltage (V)
Frequency Tuning
Power Response
25
30
Figure 10. Wideband Balanced VCO Performance
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APPLICATION NOTE • APN1012
Varactor Fundamentals
Let us consider some fundamental properties of varactors. A varactor is a specially designed P-N junction diode, whose
capacitance changes significantly in reverse bias mode. There
are three important parameters characterizing varactors. The first
is the capacitance ratio at two reverse voltages; this value characterizes the tuning ability of the varactor capacitance and is one
of the most important parameters. The second is the value of
capacitance at a given voltage. The third is the series resistance
of the varactor.
The structure of the basic varactor, called an abrupt junction varactor, is shown in Figure 11. Generally, it is built as a P++ - N N+ structure, using epitaxial N-growth on the N+ substrate with
a constant doping level in the N-region. The lower doped Nregion is the active area where the electron concentration
changes, depending on the reverse voltage applied between the
anode and cathode of the varactor. There are certain limitations
on the level of doping in the N-region, which is usually defined by
the required capacitance ratio of the varactor. Because of this,
the conductance of the N-area is a major contributor to the
varactor’s series resistance. Note that as the reverse voltage is
increased, the series resistance (due to the N-area) will decrease
along with the capacitance.
The hyperabrupt junction varactor has a more complicated
doping profile. Because of much higher doping on the P++
border, the electron concentration changes much more abruptly
compared to an abrupt junction. As a result, the capacitance of
the hyperabrupt diode at zero bias is much higher than for the
abrupt diode. Therefore, the capacitance change vs. reverse bias
becomes significantly higher for hyperabrupt diodes. The tradeoff for this better capacitance ratio is increased series resistance.
The reason is that the doping level of the N-area has been
reduced to keep average doping level over the N-region the same
as the abrupt diode level. There are many ways to bring the
series resistance in the hyperabrupt diode to as low a level as
possible. Modern state-of-the-art hyperabrupt diodes for low
noise VCOs have series resistance almost as low as discrete
ceramic capacitors.
Abrupt Junction
Doping
Level
Hyperabrupt Junction
P++
N+
Doping
Level
P++
N+
N
N
Depth from Anode
V0
V1
V2
VSAT
Electron Concentration
Electron Concentration
Depth from Anode
V0
V1
V2
VSAT
Figure 11. Varactor Fundamentals
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APPLICATION NOTE • APN1012
Varactor Packaging
Most high-volume discrete applications require varactors in lowcost, small surface mount plastic packages. Skyworks provides a
large variety of both plastic and ceramic packages. The recent,
most advanced, miniature plastic package, SC-79, shown in
Figure 12, is as small as 0402 discrete components.
24 mil (0.62 mm)
62 mil (1.58 mm)
SC-79
Figure 12. Varactor Packaging
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APPLICATION NOTE • APN1012
Relative Capacitance Change vs. Temperature
residual variation bar is shown for a typical ±100 ppm device.
This has a possible total capacitance change of over 2%. When
comparing the overall effect of temperature on varactors and
ceramic capacitors, the coupling of the devices to the resonator
circuit should be considered.
Figure 13 shows typical relative capacitance variations vs. temperature for different reverse voltages. It indicates a total
capacitance change of 5–6% in the range of -40°C to +80°C. In
comparison, a temperature compensated, ceramic capacitor
Percentage of Variation (%)
5
VVAR = -1 V
4
VVAR = -2.5 V
3
2
VVAR = -4 V
1
0
-1
Deviation Range for a Typical
Temperature Compensated
Discrete Ceramic
Multilayer Capacitor
-2
-3
-4
Consider
Varactor
Coupling!!
-40
-20
0
20
40
60
80
Temperature (°C)
Figure 13. Relative Capacitance Change vs. Temperature for Hyperabrupt Varactors
The coupling coefficient may be derived from the known (or typical) values of the tuning frequency and varactor capacitance
variation. Note that the total temperature drift in this case is
about 0.5%, as compared to 1% maximum drift caused by tem-
perature compensated discrete ceramic capacitors. Even those
numbers are extremely small when compared to the temperature
drifts caused by a VCO transistor.
CSER2
CRES
CDIV1
LRES
CBE
CCB
CVAR
CDIV2
CCE
For the Typical Wireless Case:
f = 1.6 ± -0.04 GHz
Using SMV1235-011 Varactor:
The Total Temperature Drifts
Due to Varactor in the
-40 to +85°C Becomes:
K=2
C
∆f
x
=2
f
∆C VAR
0.08 GHz
1.6 GHz
x
8 pF
≈ 0.24
3.4 pF
∆fT
≈ 0.54 %
f
Compare to the Discrete Capacitor!
Figure 14. Varactor Temperature Effect on the Oscillation Frequency
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APPLICATION NOTE • APN1012
Varactor SPICE Model
package capacitance. For ideal abrupt junction varactors, the
parameters are constant and may be defined from physical
theory. However, for actual abrupt or hyperabrupt varactors, these
values are not constant. In these cases, we use the same equation, to fit its parameters, for the best compliance with measured
capacitance vs. voltage response.
To model a varactor in most commercial simulators, we recommend the available PN-junction diode SPICE model. We specify
the barrier junction capacitance parameters CGO, VJ, and M,
instead of the default parameters. In addition, we add a value of
CP, in parallel with the junction capacitor, which is not the
CV =
SPICE Model for SMV1142-011
CGO
V
⎞M
⎛
1 + VAR ⎟
⎜⎝
VJ ⎠
+ CP
Figure 15. Typical Varactor SPICE Model
Because of formalization, parameters describing the junction
capacitance of hyperabrupt varactors may be significantly different from the default values used in the SPICE simulators for
the ideal silicon PN-junction.
For example, typical hyperabrupt varactor SMV1235 was fitted
with M = 4 as opposed to 0.5, which follows silicon PN diode
theory. Note that some SPICE simulators offer fixed default values
of M = 0.5 which can’t be changed. In this case, a diode model
may not be used, however, direct nonlinear capacitance may be
used as defined in the given formula.
20
Capacitance (pF)
SMV1235
16.13/(1-Vv/8)^4 + 2
15
CV =
10
16.13
+2
⎛
VVAR ⎞ 4
⎜1+
⎟
8 ⎠
⎝
SMV1235
Approximation
5
0
0
2
4
6
8
10
12
Varactor Voltage
Figure 16. C (V) Curve Fitting for Typical Hyperabrupt Varactor
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APPLICATION NOTE • APN1012
Super-hyperabrupt Varactor Modeling
formula to provide good approximation, not only within a given
subrange, but also for certain extensions beyond it. The extension
margin is defined from previously estimated RF signal amplitude.
Such interleaving ensures that the formula would work well, not
only in terms of DC bias, but for large signal RF analysis as well.
To overcome limitations of the “standard” PN-junction SPICE
model for hyperabrupt and super-hyperabrupt devices, such as
the SMV1265, an interleaving technique is used. In this technique, the entire capacitance reverse voltage range is broken into
several subranges. These subranges are small enough for the
100
1.0
Capacitance (pF)
SMV1265
0.8
10
Approximation
Measured
0.6
0.4
1.0
0.2
0.1
0
5
10
15
20
Interleaving of Splines
0
25
30
Varactor Voltage
M = pwl (VVAR 0 2 2.5 2 2.500009 25 6.5 25 6.50009 7.3 11 7.3 11.0009 1.8 40 1.8)
VJ = pwl (VVAR 0 4 2.5 4 2.500009 68 6.5 68 6.50009 14 11 14 11.0009 1.85 40 1.85)
CP = pwl (VVAR 0 0 2.5 0 2.500009 0 6.5 0 6.50009 0.9 11 0.9 11.0009 0.56 40 0.56)
CJO = pwl (VVAR 0 22.5 2.5 22.5 2.500009 21 6.5 21 6.50009 20 11 20 11.0009 20 40 20)*10
-12
RS = pwl (VVAR 0 2.4 3 2.4 4 2.3 5 2.2 6 2 7 1.85 8 1.76 9 1.7 10 1.65 11 1.61 12 1.5 40 1.5)
Figure 17. Piece-Wise Curve Fitting for High C (V) Ratio Varactors
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APPLICATION NOTE • APN1012
VCO Modeling Concept
For the purpose of modeling and analysis, a VCO design may be
simulated as an amplifier with parallel feedback. This analysis
involves measuring loop gain using a specific idealized direc-
tional coupler called OSCTEST in Libra IV. (For Harmonica users
there is an application note showing how to implement OSCTEST
function using S-parameters file. Refer to your Harmonica vendor
for more information).
Feedback Model of Colpitts VCO
Loop Gain
Observation
Plane
Simplified Colpitts VCO
Amplifier
L
C1
C
I1
C2
I2
V2
RL
V1
Tank Circuit
Figure 18. VCO Modeling Concept
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APPLICATION NOTE • APN1012
The major goal of the large signal, open loop VCO analysis is to
observe the magnitude (defined in dB) and the phase of the open
loop voltage gain Ku, to identify particular features of the
designed VCO.
First, we need to establish the optimum conditions for the oscillations in a given tuning range. Second, we need to find out
whether there are possibilities for parasitic oscillations both in
the lower and higher frequency ranges. If there are parasitic
oscillations, some preventive measures should be taken. Third,
we need to find ways to make both QL and the loop power (PIN)
as high as possible to facilitate phase noise performance. Finally,
other features of the VCO need to be addressed, among them
load pulling and VCC pushing.
Oscillation Happens at the 0 dB Gain
and 0 Loop Phase Shift
4
200
100
3
Mag (Ku) dB
Arg (Ku)
100
2
1
0
Mag (Ku)
0
0
Oscillation
Point
-1
VVAR = 30 V
50
VVAR = 0 V
-50
-100
-2
-100
-3
Resonator at Different
Varactor Voltages
-150
-200
-4
-200
1.0
1.5
2.0
2.5
Frequency (GHz)
3.0
3.5
Transitor X at PIN = 10 dBm
-250
0.5
Parallel Resonance:
When it nears the oscillation
point, tank circuit losses
increase; noise increase and
power decrease follow
1.0
1.5
2.0
Frequency
Figure 19. Typical Loop Gain Results for the Colpitts VCO
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2.5
APPLICATION NOTE • APN1012
Wideband Colpitts VCO Model
The OSCTEST component interrupts the oscillator feedback,
allowing the designer to analyze the VCO as an ordinary two-port
circuit (amplifier). To observe the loop response, we define the
open loop voltage gain Ku. For more details, please refer to the
VCO application notes listed in the References section. The varactor model is defined as a PN-junction diode SPICE model for
large signal, harmonic balance analysis. The transistor is
described by the Gumel Poon SPICE model with parameters provided by the vendor.
Transistor
Subcircuit
Seamless
VCO Loop
Opener
Varactor
Model
Figure 20. Wideband Colpitts VCO Model
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APPLICATION NOTE • APN1012
Differential VCO Fundamentals
The differential VCO utilizes paired transistors in common emitter
and common base configurations. The phase balance condition
for sustaining oscillations requires significantly lower phase shift
in comparison to a Colpitts design (ideally 0 degrees vs. 180
degrees). This makes it possible to use a resonator tuned to the
exact resonant frequency. However, the feedback losses may be
higher because the higher resonating currents will cause
increased ohmic losses.
Two Transistors in the
Loop Give Advantage
of Higher Loop Gain
Common-collector Common-base Phase
Shift Ideally Is 0
Resonator Works at its
Parallel Resonance, Giving
Best Phase Slope Performance
Figure 21. Concept of Differential VCO
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APPLICATION NOTE • APN1012
Balanced VCO Fundamentals
bases to drive them with a 180° phase shift. The emitter current IFB forms the feedback loop, carrying an amplified energy
surplus that is needed to sustain resonant current IRES and coupling current ICPL through the emitter base path. Unlike a
Colpitts VCO, this circuit does not require frequency dependent
feedback to match the internal transistor, high frequency phase
shifts. When properly compensated for wideband performance
with interbase inductor, LB, this circuit will be more broadband
than a Colpitts VCO.
Fundamental properties of the balanced VCO are more clearly
understood using the simplified circuit diagram shown in Figure
22. The transistors are in common collector configuration. This
is characterized by high input impedance, looking from the
transmission line and referenced as LB. Capacitor CBP simulates
the transmission lines and the grounding effect of the mounting
pads. Coupling current ICPL circulates between the transistor
Collector Currents
are Shifted 180°
In Phase. That’s
Why We Call It
“Balanced”
ICPL
IRES
LPAR
Low pass Matching
Serves to Improve
High-frequency
Performance
Re2
CBP
CBP
Le2
Rcol
LB
LSER
Rcol
DVAR
CBP
Le1
CBP
Re1
IFB
Figure 22. Balanced VCO Fundamentals
References
“Varactor SPICE Models for RF VCO Applications.” Applications
Note APN1004, Skyworks Solutions Inc., 1998.
“A Colpitts VCO for Wideband (0.95 GHz–2.15 GHz) Set-Top TV
Tuner Applications.” Applications Note APN1006, Skyworks
Solutions Inc., 1998.
“A Balanced Wideband VCO for Set-Top TV Tuner Applications.”
Applications Note APN1005, Skyworks Solutions Inc., 1998.
“Wideband VCO for Set-Top Applications.” Microwave Journal,
April 1999.
“Circuit Models for Plastic Packaged Microwave Diodes.”
Applications Note APN1001, Skyworks Solutions, Inc.
“Design with PIN Diodes.” Applications Note APN1002, Skyworks
Solutions, Inc.
For the availability of the above materials, visit the Skyworks Web
site at: www.skyworksinc.com.
“Switchable Dual-Band 170/420 MHz VCO For Hand-Set Cellular
Applications.” Applications Note APN1007, Skyworks Solutions
Inc., 1998.
“A Wideband General Purpose PIN Attenuator.” Applications Note
APN1003, Skyworks Solutions Inc., 1999.
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19
APPLICATION NOTE • APN1012
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