APPLICATION NOTE APN1012: VCO Designs for Wireless Handset and CATV Set-Top Applications Introduction Voltage Controlled Oscillators (VCOs) have come to the forefront of RF designs together with the first PLL circuits. In the era before the PLL, oscillators were mostly free running, and only in rare cases were varactors used for modulation or temperature compensation. Nowadays, we rarely see free running oscillators, instead they have become varactor-controlled oscillators. This is because most RF applications require band coverage, which can be realized through the PLL circuit requiring two sources of RF power. The reference source frequency is often a VCXO or TCXO, while the other frequency is controlled by the PLL phase detector. Usually, both VCXO/TCXO and the RF VCO are voltage controlled oscillators. The difference between a reference oscillator and a tuned VCO is that the former usually has a very high Q resonator, which allows for very stable oscillation, while the latter has a lower Q resonator, allowing a relatively high tuning range. In reference oscillators, varactors are used for fine tuning or temperature compensation (TCXO). In tunable oscillators, varactors are used to change (tune) the frequency. In some VCOs, varactors may be used also for modulation, for example in a DECT system where modulation is used to generate a constant envelope GMSK signal. Although it is a small part of the RF design, the VCO is often a major headache for designers. The goal, in this application note, is to show how Skyworks products and services may help you to overcome VCO concerns and help to make your design among the best products on the market. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 1 APPLICATION NOTE • APN1012 VCOs in Digital Wireless Phones verted or demodulated into a digital I/Q signal using a lower frequency IF VCO. The transmitter path is either directly modulated at 900 MHz or uses a dual conversion scheme requiring at least two VCOs. Consider the hypothetical wireless handset phone. Today, the handset is a dual-band (cellular/PCS) and multimode system employing many VCO functions. There are many ways to realize these functions, making it virtually impossible to specify the frequency and tuning range for all designs. However, there are certain common features that are outlined in Figure 1. When dual-band requirements are needed, up to eight or more VCOs may be required to satisfy specific frequency plans. This is often a technically and economically restrictive solution. Many designers try to solve this over-VCOed problem using both smart frequency planning and multiband VCOs, as shown in Figure 1. In a typical receiver, dual conversion superheterodyne solutions are usually employed. They convert either 900 MHz (cellular) or 1.8 GHz (PCS) down to the SAW frequency range, which may be between 90–400 MHz. Further, this signal is either downcon- PLL Control PLL Control PLL RF VCO PLL I LNA BPF BPF π/2 BPF Q RF VCO Ranging: 400–1900 MHz T/R Switch Coupler I PA BPF IF VCO Ranging: 100–400 MHz BPF BPF Σ π/2 Q RF VCO PLL RF Detector PLL PLL Control IF VCO PLL Control Figure 1. VCOs in a Digital Wireless Phone Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 2 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Fundamental Low Noise Colpitts VCO Without delving deeply into phase noise theory, we note that phase noise is inversely proportional to the power bypassed through the feedback loop, and the loaded Q of the tank circuit. Thus, the more power lost on the way to the transistor base, the higher the noise. It is clear that varactor loss plays a crucial role in the phase noise property of the VCO. If phase noise is an issue, the varactor series resistance should be carefully considered. The characteristic feature of the Colpitts VCO is that it uses a capacitive divider for the feedback consisting of C1 and C2, and an inductive branch including a parallel resonator and series capacitor C3. The parallel resonator includes inductive element M1 (that may be a discrete inductor for lower frequencies or a length of microstrip line for RF) and a capacitive branch, consisting of a varactor and a series capacitor(s). The entire inductive branch should have inductive impedance at the frequency of oscillation, otherwise there will be no oscillation. This means that the resonant frequency should be higher than the oscillation frequency. There is an additional concern because phase noise is not only a function of varactor loss. The varactor capacitance voltage characteristic has a crucial impact on phase noise as well. With a higher capacitance ratio, the varactor’s coupling to the resonator is reduced, resulting in lower resonator current. Therefore, a hyperabrupt varactor having higher series resistance is often a better choice than a lower capacitance ratio abrupt varactor having lower series resistance. Note that the resonator current circulates through the varactor, series capacitor C11 and inductor M1 and is the largest current in the tank circuit. Because of this, losses introduced in this current path are the crucial ones with respect to phase noise. M6 5 x 0.2 mm C11 1.5 p (0603) C7 0.5 p (0402) R2 2.7 k M3 4 x 0.35 mm C3 0.75 p D1 SMV1493 M2 6.5 x 0.2 mm M1 C10 15 p R4 3.6 k Q2 2SC5008 (44) Q1 2SC5007 (34) C9 C12 R1 1p 150 VCC R3 200 C4 M4 2 x 0.2 mm C1 2p C2 1.5 p C8 C5 0.75 p 0.75 p RF Out C6 0.75 p M5 2 x 0.2 mm 4 x 04 mm (Trim) VTUNE Low RS, Low Voltage Hyperabrupt Varactor Figure 2. Low Noise, High-Performance Colpitts VCO Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 3 APPLICATION NOTE • APN1012 Differential VCO for Integration with an RF IC and a differential VCO. In the Colpitts case, the resonant frequency is always higher than the oscillation frequency; in the differential VCO the resonant and oscillation frequencies may coincide. Thus the loaded Q of the circuit becomes significantly higher, and the feedback loop losses are increased due to the higher resonant currents. When this happens, the differential VCO is more vulnerable to resonator loss than the Colpitts VCO and usually shows 5–10 dB higher noise if compared to an equivalent Colpitts case. Designs based on RF IC solutions, with built-in VCOs, often employ a differential VCO configuration. One possible differential VCO configuration is shown in Figure 3. In this case, the tank circuit is formed by C3, C4 and a resonator C8, C9, D1, M1. Here again, the resonator current plays a decisive role in phase noise definition. Thus, phase noise is strongly dependent on resonator loss. Capacitors C3 and C4 help establish the correct phase shift value in the feedback loop moving oscillations closer to the resonant frequency. This is the principal difference between a Colpitts VCC 3 V RF IC R4 3k R2 3 k R7 3k L1 33 n R1 47 C1 100 p C4 1.5 p C5 1.5 p RF Out V2 V1 C2 2 p R9 3.9 k R3 9.1 k C7 100 p C3 4p R6 3k R5 3k R10 51 C9 6p V3 C8 2p R8 20 M1 4 x 0.5 mm M2 6.5 x 0.2 mm D1 SMV1493-079 VVAR C6 100 p Low RS, Low Voltage Hyperabrupt Varactor Figure 3. Differential VCO for the Integration with the RF IC Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 4 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Dual-Band Switchable VCO Schematic time. Thus, a trick is used — connecting a capacitor C11 in parallel with C6 when a lower band resonator is selected. This provides significant improvement in phase noise since C6 may then be optimized for the best performance at high band, and C11 at the lower band. One way to improve design economics in the multi-VCO requirement is to employ band switching in the VCO. If the frequency switching required isn’t very large (say within 20%), it may usually be realized within the same tank circuit, by switching “on” or “off” an additional capacitor or inductor. However, if the required switching is more than 30%, it becomes very difficult to satisfy both wideband and low noise requirements in a single design. One possible solution is to use two separate tank resonator circuits switched with two PIN diodes. In this case, the feedback needs to be optimized to fit both band requirements at the same Another important feature of this switching scheme is that the PIN diodes are not in the resonator current path. Because of this, phase noise is not sensitive to the PIN diode resistance. This is fortunate, since it means less forward current is needed. In addition, any noise on the PIN diode bias current (common for the noisy digital environment of today’s phones) would not cause significant modulation noise. J3 1 C4 100 pF R4 1.5 k VSW_Low J4 D2 SMV1139-011 VTUNE R1 300 C3 100 pF C8 10 pF L1 12 nH C1 8 pF P2 SMP1320-011 R6 3k SMP1320-011 C2 470 pF C5 100 pF Q1 NE68519 C9 100 pF P1 R3 1.5 k D1 SMV1408-011 Low RS, Low Voltage Hyperabrupt Varactor 1 VCC 3 V CCC 100 pF L2 56 nH J1 1 Low Current Switching PIN Diodes R7 6.8 k C6 20 pF C7 15 pF C10 20 pF R5 100 J5 1 RF C11 30 pF R2 1.5 k J2 1 VSW_High Figure 4. Dual-Band Switchable VCO Schematic Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 5 APPLICATION NOTE • APN1012 Dual-Band Switchable RF VCO As mentioned before, relatively small (less than 20%) frequency switching may be achieved inside the same tank circuit by connecting or disconnecting capacitors (and sometimes inductors). The PIN diode D2 performs a tricky task — it adds more capacitance in parallel with the existing parallel capacitance of the resonator, and also adds more capacitance in parallel with the VVAR existing series capacitor. This technique is used to overcome the problem of increased resonator Q, when connecting additional parallel capacitance, by decreasing it with higher series capacitance. It allows D2 to keep phase noise near its optimum at both bands. Another PIN diode in the output matching circuit tunes the buffer to a frequency doubler mode when working in PCS band. VCC (3 V) VCTL2 D1: SMV1493-079 D2, D3: SMP1322-079 RF Out D3 D2 D1 VCTL1 PINs are Used in Both Resonator Tank and Output Matching Circuits Figure 5. Dual-Band Switchable RF VCO Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 6 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 VCOs in a Set-Top Cable Downconverter The typical set-top downconverter is a dual-conversion receiver employing upconversion and downconversion techniques to overcome image problems in a wideband environment of 50–1000 MHz. In the dual up/downconversion scheme, the problem of image channel and input filtering virtually does not exist because there is no signal at the image channel. The image channel is always higher than the highest frequency of the cable signal. Two RF VCOs are required for dual downconversion. The first is a wideband VCO tuned from 1100–2000 MHz with a control voltage from 1–20 V. The other is a narrow-band VCO, which may use a CDR, coaxial dielectric resonator, at 1144 MHz. In a digital system, the second IF signal may be further demodulated, requiring an additional 44 MHz VCO. The specific action of the wideband VCO is its wideband tuning requirement. Let us consider some possible solutions for the wideband VCO. AGC 54–860 MHz 40 dB 75 Ω HPF LPF 2nd Mixer 1100 MHz 45.75/44 MHz BPF dB Upstream filter LPF 1st Mixer PLL BPF PLL 1154–1960 MHz PLL Control Upstream Low-Distortion PIN Diode Attenuator 1144/1145 MHz PLL Control Wideband VCO Tuning in 1100–2000 MHz Range Narrow-Band VCO @ 1145 MHz Figure 6. VCOs in a Set-Top Cable Down-Converter Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 7 APPLICATION NOTE • APN1012 Wideband Colpitts VCO Schematic band. Back-to-back varactors are often used to minimize parasitic mounting capacitance (between mounting pads and adjoining components). This circuit is usually designed to minimize any parasitic parallel capacitance that may be caused by component pads or transmission lines close to the inductive path. The unique action of the wideband Colpitts VCO is in its tank circuit design, which uses an inductor with a varactor connected in series and no parallel capacitor, in contrast to the low noise Colpitts VCO described in Figure 7. The feedback capacitors are optimized to the best power flatness over the entire frequency VCC = 5 V ICC = 9 mA 3.3 k NE68519 VTUNE 560 p 320 x 30 mils 5.6 nH 1p SMV1265-011 2p 3k 3k 300 p 9.1 k SMV1265-011 RF Output 1.62 p 200 High C (V) Ratio Hyperabrupt Varactors Figure 7. Wideband Colpitts VCO Schematic part of the design. Skyworks new SMV1265-011 varactor is specifically designed to fit this wideband application. 7 2.2 2.1 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 0.9 6 5 4 3 POUT (dBm) Useful Tuning Range: 980–2120 MHz Frequency (GHz) A carefully designed layout with minimum parasitic capacitances may show large frequency coverage, for example 980–2120 MHz as the performance indicates. The varactor selection is a crucial 2 1 0 0 5 10 15 20 25 30 Varactor Voltage (V) Fexp Fmodel POUT_exp POUT_model Figure 8. Wideband Colpitts VCO Performance Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 8 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Wideband Balanced VCO Schematic An even wider tuning range may be achieved with a balanced VCO configuration. The reason for its wider tuning performance is that the phase response of this VCO’s active element is flatter C6 VCC1 1000 5–8 V over the range of tuning compared to a Colpitts VCO. This allows the tank circuit more control over the oscillation frequency. The best results are achieved with back-to-back connected SMV1265 varactors, where there is 820–2120 MHz coverage. R10 2.4 k R3 120 R6 R5 V2 C5 100 T1 NE68119 R12 5 0 R7 C4 51 100 820 R8 1k 820 R4 120 V1 16 x 0.4 mm NE68119 L1 R1 R2 L2 33 nH 33 33 33 nH A C2 C1 10 10 T2 15 x 0.7 mm D1 D2 High C (V) Ratio Hyperabrupt Varactors T3 SMV1265 3 X 0.7 mm R9 1k SMV1265 R11 1000 C3 100 VVAR1 Figure 9. Wideband Balanced VCO Schematic 8 2.2 Measured Measured @ 7 V 6 2.0 Simulations 1.6 Useful Tuning Range: 820–2120 MHz 1.4 1.2 Power (dBm) Frequency (GHz) 4 1.8 2 Simulated @ 7 V 0 -2 -4 Measured @ 5 V 1.0 -6 0.8 -8 0 5 10 15 20 25 30 0 5 10 15 20 Varactor Voltage (V) Varactor Voltage (V) Frequency Tuning Power Response 25 30 Figure 10. Wideband Balanced VCO Performance Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 9 APPLICATION NOTE • APN1012 Varactor Fundamentals Let us consider some fundamental properties of varactors. A varactor is a specially designed P-N junction diode, whose capacitance changes significantly in reverse bias mode. There are three important parameters characterizing varactors. The first is the capacitance ratio at two reverse voltages; this value characterizes the tuning ability of the varactor capacitance and is one of the most important parameters. The second is the value of capacitance at a given voltage. The third is the series resistance of the varactor. The structure of the basic varactor, called an abrupt junction varactor, is shown in Figure 11. Generally, it is built as a P++ - N N+ structure, using epitaxial N-growth on the N+ substrate with a constant doping level in the N-region. The lower doped Nregion is the active area where the electron concentration changes, depending on the reverse voltage applied between the anode and cathode of the varactor. There are certain limitations on the level of doping in the N-region, which is usually defined by the required capacitance ratio of the varactor. Because of this, the conductance of the N-area is a major contributor to the varactor’s series resistance. Note that as the reverse voltage is increased, the series resistance (due to the N-area) will decrease along with the capacitance. The hyperabrupt junction varactor has a more complicated doping profile. Because of much higher doping on the P++ border, the electron concentration changes much more abruptly compared to an abrupt junction. As a result, the capacitance of the hyperabrupt diode at zero bias is much higher than for the abrupt diode. Therefore, the capacitance change vs. reverse bias becomes significantly higher for hyperabrupt diodes. The tradeoff for this better capacitance ratio is increased series resistance. The reason is that the doping level of the N-area has been reduced to keep average doping level over the N-region the same as the abrupt diode level. There are many ways to bring the series resistance in the hyperabrupt diode to as low a level as possible. Modern state-of-the-art hyperabrupt diodes for low noise VCOs have series resistance almost as low as discrete ceramic capacitors. Abrupt Junction Doping Level Hyperabrupt Junction P++ N+ Doping Level P++ N+ N N Depth from Anode V0 V1 V2 VSAT Electron Concentration Electron Concentration Depth from Anode V0 V1 V2 VSAT Figure 11. Varactor Fundamentals Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 10 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Varactor Packaging Most high-volume discrete applications require varactors in lowcost, small surface mount plastic packages. Skyworks provides a large variety of both plastic and ceramic packages. The recent, most advanced, miniature plastic package, SC-79, shown in Figure 12, is as small as 0402 discrete components. 24 mil (0.62 mm) 62 mil (1.58 mm) SC-79 Figure 12. Varactor Packaging Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 11 APPLICATION NOTE • APN1012 Relative Capacitance Change vs. Temperature residual variation bar is shown for a typical ±100 ppm device. This has a possible total capacitance change of over 2%. When comparing the overall effect of temperature on varactors and ceramic capacitors, the coupling of the devices to the resonator circuit should be considered. Figure 13 shows typical relative capacitance variations vs. temperature for different reverse voltages. It indicates a total capacitance change of 5–6% in the range of -40°C to +80°C. In comparison, a temperature compensated, ceramic capacitor Percentage of Variation (%) 5 VVAR = -1 V 4 VVAR = -2.5 V 3 2 VVAR = -4 V 1 0 -1 Deviation Range for a Typical Temperature Compensated Discrete Ceramic Multilayer Capacitor -2 -3 -4 Consider Varactor Coupling!! -40 -20 0 20 40 60 80 Temperature (°C) Figure 13. Relative Capacitance Change vs. Temperature for Hyperabrupt Varactors The coupling coefficient may be derived from the known (or typical) values of the tuning frequency and varactor capacitance variation. Note that the total temperature drift in this case is about 0.5%, as compared to 1% maximum drift caused by tem- perature compensated discrete ceramic capacitors. Even those numbers are extremely small when compared to the temperature drifts caused by a VCO transistor. CSER2 CRES CDIV1 LRES CBE CCB CVAR CDIV2 CCE For the Typical Wireless Case: f = 1.6 ± -0.04 GHz Using SMV1235-011 Varactor: The Total Temperature Drifts Due to Varactor in the -40 to +85°C Becomes: K=2 C ∆f x =2 f ∆C VAR 0.08 GHz 1.6 GHz x 8 pF ≈ 0.24 3.4 pF ∆fT ≈ 0.54 % f Compare to the Discrete Capacitor! Figure 14. Varactor Temperature Effect on the Oscillation Frequency Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 12 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Varactor SPICE Model package capacitance. For ideal abrupt junction varactors, the parameters are constant and may be defined from physical theory. However, for actual abrupt or hyperabrupt varactors, these values are not constant. In these cases, we use the same equation, to fit its parameters, for the best compliance with measured capacitance vs. voltage response. To model a varactor in most commercial simulators, we recommend the available PN-junction diode SPICE model. We specify the barrier junction capacitance parameters CGO, VJ, and M, instead of the default parameters. In addition, we add a value of CP, in parallel with the junction capacitor, which is not the CV = SPICE Model for SMV1142-011 CGO V ⎞M ⎛ 1 + VAR ⎟ ⎜⎝ VJ ⎠ + CP Figure 15. Typical Varactor SPICE Model Because of formalization, parameters describing the junction capacitance of hyperabrupt varactors may be significantly different from the default values used in the SPICE simulators for the ideal silicon PN-junction. For example, typical hyperabrupt varactor SMV1235 was fitted with M = 4 as opposed to 0.5, which follows silicon PN diode theory. Note that some SPICE simulators offer fixed default values of M = 0.5 which can’t be changed. In this case, a diode model may not be used, however, direct nonlinear capacitance may be used as defined in the given formula. 20 Capacitance (pF) SMV1235 16.13/(1-Vv/8)^4 + 2 15 CV = 10 16.13 +2 ⎛ VVAR ⎞ 4 ⎜1+ ⎟ 8 ⎠ ⎝ SMV1235 Approximation 5 0 0 2 4 6 8 10 12 Varactor Voltage Figure 16. C (V) Curve Fitting for Typical Hyperabrupt Varactor Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 13 APPLICATION NOTE • APN1012 Super-hyperabrupt Varactor Modeling formula to provide good approximation, not only within a given subrange, but also for certain extensions beyond it. The extension margin is defined from previously estimated RF signal amplitude. Such interleaving ensures that the formula would work well, not only in terms of DC bias, but for large signal RF analysis as well. To overcome limitations of the “standard” PN-junction SPICE model for hyperabrupt and super-hyperabrupt devices, such as the SMV1265, an interleaving technique is used. In this technique, the entire capacitance reverse voltage range is broken into several subranges. These subranges are small enough for the 100 1.0 Capacitance (pF) SMV1265 0.8 10 Approximation Measured 0.6 0.4 1.0 0.2 0.1 0 5 10 15 20 Interleaving of Splines 0 25 30 Varactor Voltage M = pwl (VVAR 0 2 2.5 2 2.500009 25 6.5 25 6.50009 7.3 11 7.3 11.0009 1.8 40 1.8) VJ = pwl (VVAR 0 4 2.5 4 2.500009 68 6.5 68 6.50009 14 11 14 11.0009 1.85 40 1.85) CP = pwl (VVAR 0 0 2.5 0 2.500009 0 6.5 0 6.50009 0.9 11 0.9 11.0009 0.56 40 0.56) CJO = pwl (VVAR 0 22.5 2.5 22.5 2.500009 21 6.5 21 6.50009 20 11 20 11.0009 20 40 20)*10 -12 RS = pwl (VVAR 0 2.4 3 2.4 4 2.3 5 2.2 6 2 7 1.85 8 1.76 9 1.7 10 1.65 11 1.61 12 1.5 40 1.5) Figure 17. Piece-Wise Curve Fitting for High C (V) Ratio Varactors Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 14 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 VCO Modeling Concept For the purpose of modeling and analysis, a VCO design may be simulated as an amplifier with parallel feedback. This analysis involves measuring loop gain using a specific idealized direc- tional coupler called OSCTEST in Libra IV. (For Harmonica users there is an application note showing how to implement OSCTEST function using S-parameters file. Refer to your Harmonica vendor for more information). Feedback Model of Colpitts VCO Loop Gain Observation Plane Simplified Colpitts VCO Amplifier L C1 C I1 C2 I2 V2 RL V1 Tank Circuit Figure 18. VCO Modeling Concept Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 15 APPLICATION NOTE • APN1012 The major goal of the large signal, open loop VCO analysis is to observe the magnitude (defined in dB) and the phase of the open loop voltage gain Ku, to identify particular features of the designed VCO. First, we need to establish the optimum conditions for the oscillations in a given tuning range. Second, we need to find out whether there are possibilities for parasitic oscillations both in the lower and higher frequency ranges. If there are parasitic oscillations, some preventive measures should be taken. Third, we need to find ways to make both QL and the loop power (PIN) as high as possible to facilitate phase noise performance. Finally, other features of the VCO need to be addressed, among them load pulling and VCC pushing. Oscillation Happens at the 0 dB Gain and 0 Loop Phase Shift 4 200 100 3 Mag (Ku) dB Arg (Ku) 100 2 1 0 Mag (Ku) 0 0 Oscillation Point -1 VVAR = 30 V 50 VVAR = 0 V -50 -100 -2 -100 -3 Resonator at Different Varactor Voltages -150 -200 -4 -200 1.0 1.5 2.0 2.5 Frequency (GHz) 3.0 3.5 Transitor X at PIN = 10 dBm -250 0.5 Parallel Resonance: When it nears the oscillation point, tank circuit losses increase; noise increase and power decrease follow 1.0 1.5 2.0 Frequency Figure 19. Typical Loop Gain Results for the Colpitts VCO Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 16 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A 2.5 APPLICATION NOTE • APN1012 Wideband Colpitts VCO Model The OSCTEST component interrupts the oscillator feedback, allowing the designer to analyze the VCO as an ordinary two-port circuit (amplifier). To observe the loop response, we define the open loop voltage gain Ku. For more details, please refer to the VCO application notes listed in the References section. The varactor model is defined as a PN-junction diode SPICE model for large signal, harmonic balance analysis. The transistor is described by the Gumel Poon SPICE model with parameters provided by the vendor. Transistor Subcircuit Seamless VCO Loop Opener Varactor Model Figure 20. Wideband Colpitts VCO Model Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 17 APPLICATION NOTE • APN1012 Differential VCO Fundamentals The differential VCO utilizes paired transistors in common emitter and common base configurations. The phase balance condition for sustaining oscillations requires significantly lower phase shift in comparison to a Colpitts design (ideally 0 degrees vs. 180 degrees). This makes it possible to use a resonator tuned to the exact resonant frequency. However, the feedback losses may be higher because the higher resonating currents will cause increased ohmic losses. Two Transistors in the Loop Give Advantage of Higher Loop Gain Common-collector Common-base Phase Shift Ideally Is 0 Resonator Works at its Parallel Resonance, Giving Best Phase Slope Performance Figure 21. Concept of Differential VCO Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 18 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A APPLICATION NOTE • APN1012 Balanced VCO Fundamentals bases to drive them with a 180° phase shift. The emitter current IFB forms the feedback loop, carrying an amplified energy surplus that is needed to sustain resonant current IRES and coupling current ICPL through the emitter base path. Unlike a Colpitts VCO, this circuit does not require frequency dependent feedback to match the internal transistor, high frequency phase shifts. When properly compensated for wideband performance with interbase inductor, LB, this circuit will be more broadband than a Colpitts VCO. Fundamental properties of the balanced VCO are more clearly understood using the simplified circuit diagram shown in Figure 22. The transistors are in common collector configuration. This is characterized by high input impedance, looking from the transmission line and referenced as LB. Capacitor CBP simulates the transmission lines and the grounding effect of the mounting pads. Coupling current ICPL circulates between the transistor Collector Currents are Shifted 180° In Phase. That’s Why We Call It “Balanced” ICPL IRES LPAR Low pass Matching Serves to Improve High-frequency Performance Re2 CBP CBP Le2 Rcol LB LSER Rcol DVAR CBP Le1 CBP Re1 IFB Figure 22. Balanced VCO Fundamentals References “Varactor SPICE Models for RF VCO Applications.” Applications Note APN1004, Skyworks Solutions Inc., 1998. “A Colpitts VCO for Wideband (0.95 GHz–2.15 GHz) Set-Top TV Tuner Applications.” Applications Note APN1006, Skyworks Solutions Inc., 1998. “A Balanced Wideband VCO for Set-Top TV Tuner Applications.” Applications Note APN1005, Skyworks Solutions Inc., 1998. “Wideband VCO for Set-Top Applications.” Microwave Journal, April 1999. “Circuit Models for Plastic Packaged Microwave Diodes.” Applications Note APN1001, Skyworks Solutions, Inc. “Design with PIN Diodes.” Applications Note APN1002, Skyworks Solutions, Inc. For the availability of the above materials, visit the Skyworks Web site at: www.skyworksinc.com. “Switchable Dual-Band 170/420 MHz VCO For Hand-Set Cellular Applications.” Applications Note APN1007, Skyworks Solutions Inc., 1998. “A Wideband General Purpose PIN Attenuator.” Applications Note APN1003, Skyworks Solutions Inc., 1999. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 19 APPLICATION NOTE • APN1012 Copyright © 2002, 2003, 2004, 2005, Skyworks Solutions, Inc. All Rights Reserved. Information in this document is provided in connection with Skyworks Solutions, Inc. (“Skyworks”) products or services. These materials, including the information contained herein, are provided by Skyworks as a service to its customers and may be used for informational purposes only by the customer. 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