A8652, A8653 Datasheet

A8652, A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
FEATURES AND BENEFITS
DESCRIPTION
•
•
•
•
•
•
•
•
•
•
•
The A8652/53 is a high output current synchronous buck
regulator that provides tight load regulation over a wiring
harness without the need for remote sense lines. This remote load
regulation is achieved with an integrated open-loop correction
scheme that, given a known wiring harness resistance, adjusts
the output voltage based on the measured load current and a
user-programmable gain, achieving ±2% accuracy at 500 mV
of correction. The Remote Load Regulation control includes a
115% regulated voltage clamp in conjunction with a dynamic
overvoltage protection, with OVP threshold changing with the
correction voltage. The A8652/53 includes a user-configurable
load-side current limit to fold back the output voltage during an
output overcurrent condition. The A8652/53 regulates nominal
input voltages from 4 to 36 V and remains operational when
VIN drops as low as 2.6 V. When the input voltage approaches
the output voltage, the duty cycle is maximized to maintain
the output voltage.
•
•
•
•
Automotive AEC-Q100 qualified
Cable and wiring drop compensation
Dynamic voltage correction with controller
Integrated high-side and low-side switching MOSFETs
Programmable load-side current limit
Maximized duty cycle for low dropout operation
Operating input voltage range: 4 V to 36 V
UVLO STOP threshold is at 2.6 VTYP
Withstands surge voltages up to 40 V
Continuous loading: 2.6 A for A8653; 1 A for A8652
Adjustable switching frequency (fSW): 100 kHz to
2.2 MHz
Synchronization to external clock: 100 kHz to 2.2 MHz
Frequency dithering for lower EMI signature
External adjustable compensation network
Stable with ceramic output capacitors
Continued on next page...
The A8652/53 features include externally set soft-start time,
external compensation network, an EN input to enable VOUT,
a SYNC/FSET input to synchronize or set the PWM switching
PACKAGES:
16-Pin eTSSOP (suffix LP) with exposed thermal pad
Continued on next page...
APPLICATIONS
•
•
•
•
Not to scale
VIN
CIN
2 × 4.7 µF
Automotive USB Power Ports
Rear Seat Entertainment
Navigation Systems
Motorcycle Clusters
BOOT
VIN
CBOOT
100 nF
GND
LO
RSEN
VOUT
RWIRE/2
SW
CO
RGADJ
EN
RWIRE/2
GADJ
SYNC/FSET
A8652/3
ISEN+
CF
(optional)
RFSET
SS
ISENRIADJ
CSS
22 nF
IADJ
COMP
CP
RPU
10 kΩ
RZ
CZ
RF
(optional)
FB
POK
Typical Application Diagram 1
A8652/53-DS, Rev.2
RFB1
24.9 kΩ
RFB2
4.75 kΩ
VLOAD
CLOAD
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
FEATURES AND BENEFITS (continued)
• Pre-bias startup compatible
• Power OK (POK) output
• Dynamic overvoltage protection, pulse-by-pulse current limit,
hiccup mode short-circuit, and thermal protections
• Open-circuit and adjacent pin short-circuit tolerant
• Short-to-ground tolerant at every pin
• USB3 charging capability: 2.6 A (A8653)
• USB2 capability: 1 A (A8652)
DESCRIPTION (continued)
frequency, and a Power OK output to indicate when VOUT is within
regulation and there is no load-side current limit condition. Protection
features include VIN undervoltage lockout, pulse-by-pulse current
limit, hiccup mode short-circuit protection, dynamic overvoltage
protection, and thermal shutdown. The A8652/53 provides opencircuited, adjacent pin short-circuit, and short-to-ground protection
at every pin to satisfy the most demanding automotive and nonautomotive applications.
The A8652/53 device is available in a 16-pin eTSSOP package with
exposed pads for enhanced thermal dissipation. It is lead (Pb) free,
with 100% matte-tin lead frame plating. The maximum junction
temperature (TJ(max)) is 150ºC.
VIN
CIN
2 × 4.7 µF
BOOT
VIN
CBOOT
100 nF
GND
RSEN
20 mΩ
LO 10 µH
VOUT
RWIRE/2
62.5 mΩ
SW
CO
2 × 22 µF
RIADJ
20 kΩ
EN
RWIRE/2
62.5 mΩ
IADJ
SYNC/FSET
A8653
RFSET
52.3 kΩ
ISEN+
CF
(optional)
RF
(optional)
RGE
CSS
22 nF
VG
GADJ
RGADJ
20 kΩ
COMP
5V
RFB1
24.9 kΩ
RPU
10 kΩ
RZ
POK
FB
CZ
CLOAD
100 µF
ISEN-
SS
CP
VLOAD
RFB2
4.75 kΩ
Typical Application Diagram 2 with Dynamic Voltage Correction Control at Pin GADJ
Selection Guide
Part Number
A8652KLPTR-T
A8653KLPTR-T
Packing
Package
4000 pieces per 13-inch reel
4.4 mm × 5 mm, 1.2 mm nominal height
16-pin eTSSOP with exposed thermal pad
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
SPECIFICATIONS
Absolute Maximum Ratings1
Characteristic
Symbol
Notes
Rating
VIN, EN, SS
Unit
–0.3 to 40
V
–0.3 to VIN + 0.3
V
VIN ≤ 36 V, t < 50 ns
–1 to VIN + 2
V
Continuous
VSW – 0.3 to
VSW + 5.5
V
< 1 ms
VSW – 0.3 to
VSW + 7
V
–0.3 to 6.5
V
ISEN+ to ISEN– Differential Voltage
–0.3 to 0.3
V
All other pins
–0.3 to 5.5
V
TJ(max)
150
ºC
Tstg
–55 to 150
ºC
SW to GND 2
BOOT Pin Above SW Pin
VSW
VBOOT
ISEN+ and ISEN– Pins
Maximum Junction Temperature
Storage Temperature Range
Continuous
ISEN+ and ISEN– Pins
1 Stresses
beyond those listed in this table may cause permanent damage to the device. The absolute maximum ratings are stress ratings only, and functional operation of
the device at these or any other conditions beyond those indicated in the Electrical Characteristics table is not implied. Exposure to absolute-maximum-rated conditions for
extended periods may affect device reliability
2 SW has internal clamp diodes to GND and VIN. Applications that forward bias these diodes should take care not to exceed the IC package power dissipation limits.
Thermal Characteristics
Characteristic
Package Thermal Resistance
3 Additional
Symbol
RθJA
Test Conditions3
LP Package, 4-layer PCB based on JEDEC standard
Value
Unit
34
ºC/W
thermal information available on the Allegro website.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
PINOUT DIAGRAM AND TERMINAL LIST TABLE
EN
1
16 BOOT
VIN
2
15 SW
SS
3
14 PGND
GADJ
4
FB
5
PAD
13 PGND
12 ISEN+
IADJ
6
11 ISEN–
SGND
7
10 POK
COMP
8
9
SYNC/FSET
Package LP, 16-Pin eTSSOP Pinout Diagram
Terminal List Table
Symbol
Number
Function
EN
1
Enable input. This pin is used to turn the converter on or off: set this pin high to turn the converter on or set this pin low to
turn the converter off. May be connected to VIN.
VIN
2
Power input for the control circuits and the drain of the internal high-side N-channel MOSFET. A high quality ceramic
capacitor should be placed very close to this pin.
SS
3
Soft-Start pin. Connect a capacitor, CSS, from this pin to GND to set the soft-start time. This capacitor also determines
the hiccup period during overcurrent.
GADJ
4
This pin is used to set the gain of the differential current sense amplifier with ISEN+/ISEN– pins. A resistor from this pin
to GND set the amplifier gain. Together with load sense resistor, it sets the desired voltage correction at the specified
load condition. Grounding GADJ disables Remote Load Regulation function.
FB
5
Feedback (negative) input to the error amplifier. Connect a resistor divider from the converter output node (VOUT) to this
pin to program the output voltage.
IADJ
6
Active current limit adjust pin. A resistor from this pin to GND sets the current limit. When the load current exceeds this
limit, the output voltage will decrease at the predefined slope.
SGND
7
Signal (quiet) GND.
COMP
8
Output of the error amplifier and compensation node for the control loop. Connect a series RC network from this pin to
GND for loop compensation.
SYNC/FSET
9
Frequency setting and synchronization pin. A resistor, RFSET, from this pin to GND sets the PWM switching frequency.
POK
10
Power OK output signal. This pin is an open-drain output that transitions from low impedance to high impedance when
the output is within the final regulation voltage and no load side current limit exists.
ISEN–
11
Negative current-sensing pin to the internal current sense amplifier, connected to the load side of the external current
sensing resistor.
ISEN+
12
Positive current-sensing pin to the internal current sense amplifier, connected to the inductor side of the external current
sensing resistor.
PGND
13, 14
SW
115
The source of the high-side N-channel MOSFET. The output inductor (LO) should be connected to this pin. LO should be
placed as close as possible to this pin and connected with relatively wide traces.
BOOT
16
High-side gate drive boost input. Connect a 100 nF ceramic capacitor from BOOT to SW.
PAD
–
Exposed pad of the package providing enhanced thermal dissipation. This pad must be connected to the ground
plane(s) of the PCB with at least 6 vias, directly in the pad.
Power GND.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
VIN
BOOT REG
VIN
Regulator
VREF
VREG
UVLO
EN
BOOT
Current Sense
Amp
TSD
Isen
OCP
500 nA
EN
SYNC/FSET
Protection & Fault
OVP
OSC
SW
Adj
5 µs
CLK
80 mV
80 mΩ
FB
PWM
Control
Logic
PWM
COMP
Error Amp
VREG
55 mΩ
Σ
COMP
Slope
Comp
PGND
Ramp
Offset
20 µA
5 µA
HICCUP
LOGIC
400 mV
2 kΩ
FAULT
SS
REF_ADJ
POK
UV
920 mV
OV
VREF
800 mV
Sense
Amp
ISEN+
ISEN–
Adj
GADJ
SGND
IADJ
Functional Block Diagram
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
ELECTRICAL CHARACTERISTICS: Valid at 4 V ≤ VIN ≤ 36 V; TA = 25ºC; • indicates specifications guaranteed
–40°C ≤ TA = TJ ≤ 150°C (unless noted otherwise).
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
INPUT VOLTAGE SPECIFICATIONS
Operating Input Voltage Range 2
4
–
36
V
UVLO Start Threshold
VUVLO(START)
VIN
VIN rising
●
–
3.4
3.7
V
UVLO Stop Threshold
VUVLO(STOP)
VIN falling
–
2.6
2.9
V
UVLO Hysteresis
VUVLO(HYS)
–
800
–
mV
–
3
6.5
mA
VIN = 12 V, VEN ≤ 0.4 V, –40°C < TA = TJ <
85°C
–
1
240
µA
VIN = 12 V, VEN ≤ 0.4 V, TA = TJ = 125°C
–
40
900
µA
792
800
808
mV
INPUT CURRENTS
Input Quiescent Current 1
Input Sleep Supply Current 1
IQ
IQSLEEP
VEN = 5 V, VFB = 1 V, no PWM switching
●
VOLTAGE REGULATION
Feedback Voltage Accuracy 3
Feedback Voltage Accuracy with Cable
Compensation 3
Error Amp Clamp Voltage 3
Output Voltage Setting Range 3
VFB
VFB = VCOMP, VGADJ = 0 V, –40°C < TA = TJ <
125°C
VFB = VCOMP, VGADJ = 0 V
●
788
800
812
mV
VFB(ACC)
VFB = VCOMP, VISEN+ – VISEN– = 25 mV,
VOUT = 5 V, RGADJ = 20 kΩ, RIADJ = 20 kΩ
●
808
825
842
mV
VFB(CLAMP)
VFB = VCOMP, VISEN+ – VISEN– = 55 mV,
VOUT =5 V, RGADJ = 7.5 kΩ, RIADJ = 20 kΩ
900
920
940
mV
3.3
–
5.75
V
VOUT
VIN = 5.7 V, IO = 2.6 A, fSW = 500 kHz
Output Dropout Voltage 3
VO(PWM)
VIN = 7.3 V, IO = 2.6 A, fSW = 2 MHz
VIN = 5.5 V, IO = 1 A, fSW = 500 kHz
VIN = 6.8 V, IO = 1 A, fSW = 2 MHz
A8653
A8652
●
4.9
–
–
V
●
4.9
–
–
V
●
4.9
–
–
V
●
4.9
–
–
V
ERROR AMPLIFIER
Feedback Input Bias Current 1
IFB
Open-Loop Voltage Gain
AVOL
Transconductance
gmEA
Output Current
IEA
–100
–
–8
nA
VCOMP = 1.2 V
–
65
–
dB
400 mV < VFB
550
750
950
0 V < VFB < 400 mV
275
375
475
–
±75
–
VCOMP = 1.2 V
µA/V
µA
INTERNAL MOSFET PARAMETERS
High-Side MOSFET On-Resistance 3
SW Node Rising Slew Rate
RDSON(HS)
dV/dt
TA = 25°C, IDS = 100 mA
–
80
–
mΩ
12 V < VIN < 16 V
–
0.75
–
V/ns
–10
0
10
µA
–
55
–
mΩ
SW Leakage 1
ISW(LEAK)
VEN ≤ 0.4 V, VSW = 5 V, VIN = 12 V, TJ = 25°C
Low-Side MOSFET On-Resistance 3
RDSON(LS)
TA = 25°C, IDS = 100 mA
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
ELECTRICAL CHARACTERISTICS (continued): Valid at 4 V ≤ VIN ≤ 36 V; TA = 25ºC; • indicates specifications guaranteed –40°C ≤ TA = TJ ≤ 150°C (unless noted otherwise).
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
RFSET = 261 kΩ
–
RFSET = 61.9 kΩ
375
100
–
kHz
415
457
kHz
OSCILLATOR
PWM Switching Frequency
fSW
RFSET = 10.5 kΩ
–
2000
–
kHz
PWM Frequency Dithering
fDITHER
No dithering with FSET synchronization
–
±13
–
%
Minimum Controllable On-Time
tON(MIN)
VIN = 12 V, IOUT = 1 A
–
95
135
ns
Minimum Switch Off-Time
tOFF(MIN)
VIN = 12 V, IOUT = 1 A
–
100
135
ns
FSET SYNCHRONIZATION TIMING
Synchronization Frequency Range
fSW_MULT
100
–
2200
kHz
Synchronization Input Off-Time
tSYNC_OFF
0.2
–
1.3
µs
tr(SYNC)
–
10
15
ns
tf(SYNC)
–
10
15
ns
Synchronization Input Rise
Time3
Synchronization Input Fall Time3
Synchronization Rising Threshold
VSYNC(HI)
VSYNC rising
Synchronization Falling Threshold
VSYNC(LO)
VSYNC falling
–
–
2
V
0.5
–
0.7
V
3.3
4
4.62
A
CURRENT LOOP
Peak Inductor (Pulse-by-Pulse) Current
Limit
Load-Side Current Limit
COMP to SW Current Gain
IPK_LIM(MINON) tON = tON(MIN)
IPK_LIM(MINOFF) tON = 1/fSW – tOFF(MIN), No Sync
IOUT_LIM
RIADJ = 20 kΩ, RSEN = 20 mΩ,
VOUT = 5 V
gmPOWER
A8653
A8652
●
1.5
1.8
2.1
A
A8653
●
2.4
3.2
4
A
A8652
●
0.9
1
1.5
A
A8653
2.5
3
3.3
A
A8652
1
1.2
1.4
A
A8653
–
6.3
–
A/V
A8652
–
3.2
–
A/V
–
0.056
–
A/µs
A8653
0.09
0.24
0.43
A/µs
RFSET = 261 kΩ, 100 kHz
RFSET = 61.9 kΩ, 415 kHz
Slope Compensation
SE
●
RFSET = 10.5 kΩ, 2 MHz
–
1.3
–
A/µs
RFSET = 261 kΩ, 100 kHz
–
0.035
–
A/µs
0.07
0.15
0.23
A/µs
–
0.8
–
A/µs
RFSET = 61.9 kΩ, 415 kHz
RFSET = 10.5 kΩ, 2 MHz
A8652
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
ELECTRICAL CHARACTERISTICS (continued): Valid at 4 V ≤ VIN ≤ 36 V; TA = 25ºC; • indicates specifications guaranteed –40°C ≤ TA = TJ ≤ 150°C (unless noted otherwise).
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
–
200
275
mV
–
3.3
–
V
–30
–20
–10
µA
1
2.2
5
µA
SOFT-START
SS FAULT/HICCUP Reset Voltage
VSS(RST)
SS Maximum Charge Voltage
VSS(MAX)
SS Startup (Source)
Current 1
VSS falling due to RSS(FLT)
ISS(SU)
HICCUP = FAULT = 0
SS Hiccup (Sink) Current 1
ISS(HIC)
HICCUP = 1
SS Pull-Down Resistance
RSS(FLT)
FAULT = 1 or EN = 0
–
2
–
kΩ
0 V < VFB < 200 mV
–
fSW/4
–
–
200 mV < VFB < 400 mV
–
fSW/2
–
–
400 mV < VFB
–
fSW
–
–
SS Switching Frequency
fSS
HICCUP MODE
Hiccup OCP Enable Threshold
VHIC(EN)
VSS rising
–
2.3
–
V
Hiccup, OCP Count
OCPLIM
VSS > 2.3 V, OCP pulses
–
240
–
counts
BOOTUV
–
64
–
counts
BOOTOPEN
–
7
–
counts
–
–
0.4
V
Hiccup, BOOT Shorted Count
Hiccup, BOOT Open Count
POWER OK (POK) OUTPUT
POK Output Voltage
POK Leakage
VPOK
IPOK = 4 mA
IPOK(LEAK)
VPOK = 5 V
●
–
–
5
µA
POK UV Threshold
VPOK(UV)
VFB falling
●
715
740
760
mV
POK UV Hysteresis
VPOK(UV,HYS)
1
POK OV Threshold
POK OV Hysteresis
ISEN+ OV Threshold
ISEN+ OV Hysteresis
POK Delay
VPOK(OV)
–
10
–
mV
VFB rising, VGADJ = 0 V
840
880
920
mV
VFB rising, VISEN+ – VISEN– = 25 mV,
VOUT = 5 V, RGADJ = 20 kΩ, RIADJ = 20 kΩ
865
905
950
mV
VFB rising, VISEN+ – VISEN– = 55 mV,
VOUT = 5 V, RGADJ = 7.5 kΩ, RIADJ = 20 kΩ
–
1
–
V
–
10
–
mV
ISEN+ rising, VGADJ = 0 V
5.4
5.65
6
V
ISEN+ rising, VISEN+ – VISEN– = 25 mV,
VOUT = 5 V, RGADJ = 20 kΩ, RIADJ = 20 kΩ
5.6
5.8
6.1
V
ISEN+ rising, VISEN+ – VISEN– = 55 mV,
VOUT = 5 V, RGADJ = 7.5 kΩ, RIADJ = 20 kΩ
–
6.45
–
V
–
60
–
mV
VPOK(OV,HYS)
VISEN(OV)
VISEN(OV,HYS)
td(POK)
VFB rising only
–
7
–
PWM
cycles
155
170
185
ºC
–
20
–
ºC
THERMAL PROTECTION
TSD Rising Threshold
TSD Hysteresis 3
TSD
TSDHYS
PWM stops immediately and COMP is pulled
low and SS is reset
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
8
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
ELECTRICAL CHARACTERISTICS (continued): Valid at 4 V ≤ VIN ≤ 36 V; TA = 25ºC; • indicates specifications guaranteed –40°C ≤ TA = TJ ≤ 150°C (unless noted otherwise).
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
EN INPUT THRESHOLDS
EN High Threshold
VEN(H)
EN rising
–
1.41
2
V
EN Low Threshold
VEN(L)
EN falling
0.7
1.36
–
V
EN Delay
td(EN)
EN transitioning low, VOUT < 25%
–
60
–
PWM
cycles
EN = 5 V
–
500
–
nA
EN Input Bias Current 1
IEN_BIAS
1 For
input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin or node.
2 Thermally limited depending on input voltage, output voltage, duty cycle, regulator load currents, PCB layout, and airflow.
3 Ensured by design and characterization, not production tested.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
9
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
815
4.05
810
4.00
3.95
805
IPK_LIM(MINON) (A)
Reference Voltage, VREF (mV)
TYPICAL PERFORMANCE CHARACTERISTICS
800
3.90
3.85
795
3.80
790
3.75
3.70
-50
785
-50
-25
0
25
50
75
100
125
150
175
-25
0
25
50
75
100
125
150
175
Temperature (ºC)
Temperature (ºC)
Pulse-by-Pulse Current Limit at tON(MIN) (IPK_LIM(MINON))
versus Temperature
Reference Voltage versus Temperature
3.50
1.6
EN Rising Threshold
EN Falling Threshold
3.25
1.4
START, UVLOSTART
EN Thresholds (V)
VIN UVLO Thresholds (V)
1.5
STOP, UVLOSTART
3.00
2.75
1.3
1.2
1.1
1.0
2.50
0.9
-50
-25
0
25
50
75
100
125
150
175
-50
-25
0
25
Temperature (ºC)
VIN UVLO START and STOP Thresholds versus
Temperature
75
100
125
150
175
EN Rising and Falling Thresholds versus Temperature
8.3
925
8.2
900
8.1
875
8.0
850
825
POK Overvoltage
800
POK Undervoltage
POK Delay (µs)
POK OV & UV Thresholds at FB (mV)
50
Temperature (ºC)
7.9
7.8
7.7
7.6
775
7.5
750
7.4
725
7.3
7.2
700
-50
-25
0
25
50
75
Temperature (ºC)
100
125
150
175
POK Overvoltage and Undervoltage Thresholds at FB
versus Temperature (POK OV test with VGADJ = 0 V)
-50
-25
0
25
50
75
100
125
150
175
Temperature (ºC)
POK Delay Time versus Temperature
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10
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
6.00
2.5
5.95
5.5
Input Quiescent Current (mA)
ISEN+ OV Thresholds (V)
5.90
5.85
5.80
5.75
5.70
5.65
5.60
5.0
4.5
4.0
3.5
5.55
5.50
3.0
-50
0
-25
25
75
50
125
100
150
175
-50
0
-25
25
100
125
94
4.00
92
3.50
90
Efficiency (%)
96
4.50
3.00
2.50
2.00
88
86
84
VIN = 8 V
82
1.50
VIN = 12 V
80
1.00
VOUT
0.50
78
VLOAD
VIN = 16 V
0
0.5
1.5
1.0
Load (A)
0.00
3.0
3.5
4.5
4.0
175
Quiescent Current IQ versus Temperature
5.00
2.5
150
Temperature (ºC)
ISEN+ Overvoltage Thresholds versus Temperature
(test with VGADJ = 0 V)
Voltage (V)
75
50
Temperature (ºC)
5.0
5.5
6.0
2.0
2.5
3.0
Efficiency versus Output Current
(Typical Design A in Table 3)
VIN (V)
Low VIN Dropout Operation at 5 Ω Load
5.2 V
5.2 V
VOUT
VOUT
5.1 V
5.1 V
VLOAD
5.0 V
4.9 V
4.9 V
25 mA/µs
1 A/div
VLOAD
5.0 V
25 mA/µs
1 A/div
IOUT
IOUT
C4
C2
100 µs/div
100 µs/div
Transient Response 0 to 1 A Load Step
Transient Response 1 to 2 A Load Step
(Typical Design A in Table 3)
(Typical Design A in Table 3)
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11
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
FUNCTIONAL DESCRIPTION
Overview
The A8652/53 is a synchronous PWM buck regulator that integrates low RDS(on) high-side and low-side N-channel MOSFETs.
It is designed to remain operational when input voltage falls as
low as 2.6 V. The A8652/53 employs peak current mode control
to provide superior line and load regulation, pulse-by-pulse current limit, fast transient response and simple compensation. The
A8652/53 incorporates a Cable Drop Compensation (Remote
Load Regulation) function in its current mode control architecture to adjust the output voltage according to the load current,
offsetting the voltage drop introduced by the wiring harness. The
reference voltage in the feedback loop is adjusted relative to the
voltage across the sensing resistor at the load side. When the
load current increases, it causes the reference voltage at the error
amplifier to increase and the output voltage to follow. The gain of
the voltage correction is configurable using the GADJ and IADJ
pins. Such features provide flexibility in setting the amount of
output voltage correction and the load current limit.
The features of the A8652/53 include Remote Load Regulation, an internal precision reference, an adjustable switching
frequency, a transconductance error amplifier, an enable input,
integrated top and bottom switching MOSFETs, adjustable softstart time, pre-bias startup, and a Power OK output. Protection
features of A8652/53 include VIN undervoltage lockout, pulse-bypulse overcurrent protection, BOOT overvoltage and undervoltage protection, hiccup mode short-circuit protection, dynamic
overvoltage protection, and thermal shutdown. In addition, the
A8652/53 provides open-circuit, adjacent pin short-circuit, and
pin-to-ground short-circuit protection.
Reference Voltage
The A8652/53 incorporates an internal precision reference that
allows output voltages as low as 0.8 V. The accuracy of the
internal reference is ±1% from –40°C to 125°C and ±1.5% across
from –40°C to 150°C when the Remote Load Regulation is disabled. The output voltage of the regulator is programmed with a
resistor divider between VOUT and the FB pin of the A8652/53.
Oscillator/Switching Frequency and
Synchronization
The PWM switching frequency of the A8652/53 is adjustable
from 100 kHz to 2.2 MHz and has an accuracy of about ±10%
over the operating temperature range. Connecting a resistor
from the FSET/SYNC pin to GND, as shown in the Applications
Schematic, sets the switching frequency. An FSET resistor with
±1% tolerance is recommended. A graph of switching frequency
versus FSET resistor value is shown in the Component Selection
section of this datasheet. The A8652/53 will suspend operation if
the FSET pin is shorted to GND or left open.
FSET/SYNC pin also can be used as a synchronization input that
accepts an external clock to switch the A8652/53 from 100 kHz
to 2.2 MHz and scales the slope compensation according to the
synchronization frequency. When being used as a synchronization input, the applied clock pulses must satisfy the pulse width,
duty cycle, and rise/fall time requirements shown in the Electrical
Characteristics shown in this datasheet.
Remote Load Regulation Control and
Transconductance Error Amplifier
The Remote Load Regulation control in the A8652/53 provides
improved load regulation at the remote load by increasing the
voltage reference of the error amplifier to correct for the voltage
drop introduced by wiring harness to the load. The amount of
voltage correction is user-programmable with external configuration resistors, allowing the A8652/53 to be applied to wiring
harnesses that have up to 750 mV IR drops at full load. The
Remote Load Regulation controller has a variety of protection
features, including a load-side current limit, a maximum regulation voltage, and protection in the event of open pin or shorted
pin conditions.
The Remote Load Regulation voltage correction and protection
features interface with the error amplifier, which is a four-terminal input device with three positive inputs and one negative input,
as shown in Figure 1. The negative input is simply connected to
the FB pin and is used to sense the feedback voltage for regulation. The error amplifier performs an “analog OR” selection
between its positive inputs, operating according to the positive
input with the lowest potential. The three positive inputs are used
for soft-start, steady-state regulation, and the 15% maximum
regulation voltage. The error amplifier regulates to the soft-start
pin voltage minus 400 mV during startup, the sum of A8652/53
internal reference (VREF) and the Remote Load Regulation
correction (REF_ADJ) during normal operation, or the 920 mV
maximum.
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12
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
The gain of the current sense to the REF_ADJ (GADJ) signal is set
by the ratio of resistance to GND on the GADJ pin and the IADJ
pin in conjunction with Rsen:
400 mV
SS
GADJ =
Error Amp
COMP
920 mV
REF_ADJ
VREF
800 mV
REF_ADJ
Rsen × RIADJ
=
RGADJ
IOUT
(1)
This allows the user to calibrate the voltage correction to the IR
drop of the wiring harness, as shown in Figure 3. This calibration results in improved load regulation at the end of the wiring
harness.
Larger
Rsen × RIADJ / RGADJ
VOUT
FB
5.5 V
Figure 1: A8652/53 Error Amplifier
5.0 V
Current
Sense
Amp
ISEN+
VREF
800 mV
Smaller
Rsen × RIADJ / RGADJ
920 mV
Error
Amp
ISEN-
REF_ADJ
Σ
Σ
IOUT
Gain Adj
GADJ
0A
1.0 A
Figure 3: Voltage Correction Gain Adjustment
RGADJ
IADJ
RIADJ
Figure 2: Remote Load Regulation Control
The amount of the voltage correction for the wiring harness is
generated by the Remote Load Regulation control circuit and
fed to the error amplifier via the REF_ADJ signal, as shown in
Figures 1 and 2. The Remote Load Regulation controller generates REF_ADJ according to the load current sensed by ISEN+
and ISEN– and the gain set by the configuration resistors. The
current sense resistor (Rsen) is connected on the load side between
the regulator output capacitor and the load terminal and can have
a value between 20 and 50 mΩ.
To configure the voltage correction gain, the load-side current
limit (IOUT_LIM) must first be set by the following equation (also
referring to Table 1):
RIADJ =
1200
IOUT_LIM × Rsen
(2)
The voltage correction gain is based on RWIRE, which is the sum
of the wiring harness supply and return resistive paths as detailed
in the typical application diagram. Given the gain of the FB pin
voltage divider (AFB = VOUT/VFB), RWIRE and RIADJ the desired
voltage correction gain is set by the following equation (referring
to Figure 4):
RGADJ =
Rsen
× AFB × RIADJ
RWIRE
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(3)
13
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
For example, for a 5 V application with a 20 mΩ current sense
resistor and a 3 A load side current limit the IADJ configuration
resistor would be 20 kΩ. To correct for a 125 mV wire harness
drop at 1 A (RWIRE =125 mΩ) given RIADJ = 20 kΩ, the GADJ
configuration resistor should be 20 kΩ.
Rsen = 50 mΩ, RIADJ = 20 kΩ (1.2 A)
RWIRE (mΩ)
Table 1: RIADJ Resistor Selection vs. IOUT_LIM
800
700
Rsen = 50 mΩ, RIADJ = 23.7 kΩ (1.0 A)
600
Rsen = 50 mΩ, RIADJ = 30.1 kΩ (0.8 A)
500
400
RIADJ (kΩ)
Rsen = 20 mΩ
Rsen = 50 mΩ
15.8
3.80
1.52
16.9
3.55
1.42
200
100
3.45
1.38
3.37
1.35
18.2
3.30
1.32
18.7
3.21
1.28
19.1
3.14
1.26
19.6
3.06
1.22
20.0
3.00
1.20
20.5
2.93
1.17
21.0
2.86
1.14
21.5
2.79
1.12
22.1
2.71
1.09
22.6
2.65
1.06
23.2
2.59
1.03
23.7
2.53
1.01
24.3
2.47
0.99
26.7
2.25
0.90
30.1
1.99
0.80
34.8
1.72
0.69
40.2
1.49
0.60
As will be discussed in further detail below, altering the GADJ
resistance with an external voltage proportionally adjusts the
voltage correction gain (Method 1, refer to Typical Application
Diagram 2). On the other hand, altering the IADJ resistance with
an external voltage proportionally adjusts the load-side current
limit, and inversely proportionally adjusts the voltage correction
gain (Method 2).
0
0
10
20
30
RGADJ (kΩ)
40
50
60
300
Rsen = 20 mΩ, RIADJ = 20 kΩ (3.0 A)
Rsen = 20 mΩ, RIADJ = 23.7 kΩ (2.5 A)
250
Rsen = 20 mΩ, RIADJ = 30.1 kΩ (2.0 A)
200
RWIRE (mΩ)
17.4
17.8
300
150
100
50
0
0
10
20
30
RGADJ (kΩ)
40
50
60
Figure 4: RGADJ Selection vs RWIRE for Given Rsen, RIADJ
This can be very useful, for instance when one “universal”
design is created for multiple platforms, where the expected wiring resistance can vary widely. The design can use this method
in conjunction with the system controller such that the degree of
voltage correction can be set via software.
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14
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
The minimum and maximum voltage correction can be adjusted
with the ratio of RGADJ and RG_E for a given controlling voltage
(VG) range.
1.0 V
A8652/53
Similarly, as shown in Figure 7, Method 2 can inversely proportionally adjust the voltage correction gain for a fixed GADJ
resistance, at the same time proportionally adjusts the load side
current limit (IOUT_LIM) by applying the control voltage (VI) to
the IADJ pin. The equivalent resistance RIADJ at pin IADJ is:
ig
1
GADJ
VG
RI_eqv =
RG_E
(R1
IADJ
RGADJ
Figure 5: Dynamic Voltage Correction Adjustment
at Pin GADJ
Figure 5 above illustrates how to adjust the amount of voltage
correction by controlling VG. The equivalent resistance RG_eqv at
pin GADJ with respect to GND now becomes:
–
)
VI – 1
RI_E
(6)
Thus the voltage correction gain can be kept the same by applying a separate control voltage to the GADJ pin to keep the GADJ
and IADJ resistor ratio the same if only load-side current limit
needs to be adjusted.
1.0 V
A8652/53
1
RG_eqv =
(R
1
GADJ
–
)
VG – 1
RG_E
(4)
ia
The voltage correction gain (GADJ) then varies linearly with the
applied VG for given RGADJ and RG_E. Normalizing this gain to
the case with only RGADJ connecting to pin GADJ results in:
Gnorm = 1 +
RGADJ
R
– GADJ VG
RG_E
RG_E
IADJ
RI_E
(5)
VI
RIADJ
Gnorm
Figure 7: Simultaneous Voltage Correction and LoadSide Current Limit Adjustment at Pin IADJ
1 + RGADJ / RG_E
If a fixed load-side current limit is desired it is simpler to use
GADJ pin to dynamically control the amount of voltage correction because of linear control and only altering the voltage
correction gain.
1
The GADJ and IADJ pins are designed for a resistance range
of 10 to 34 kΩ, and therefore the controlling voltage VG and VI
must be limited as follows:
0
0
1
VG(V)
10 kΩ < RG_eqv < 34 kΩ
(7)
10 kΩ < RI_eqv < 34 kΩ
(8)
and
Figure 6: Normalized Gain Gnorm vs. VG
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15
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
In addition to the voltage correction circuitry, Figure 2 also
details the load-side current limit (IOUT_LIM), which is configured independently by the resistance to GND on the IADJ pin.
As shown by Figure 8 and 9, when the load current exceeds
IOUT_LIM, POK is pulled low to flag the condition, and the output
voltage is decreased at the same rate as the voltage correction (set
by Rsen, RGADJ and RIADJ) to protect against unstable behavior.
Figure 8 and 9 also details the operation of peak inductor current
limit (IPK_LIM), which monitors the inductor current and will
enter into hiccup mode after 240 counts of exceeding IPK_LIM for
robust protection and against the output voltage shorted to GND.
2.6 V – UVLOfall
5.75 V
VOUT
5.0 V
5.5 V
IPK_LIM
4.0 A
IOUT_LIM
2.5 A
2.0 A
IOUT
0
1A
1.5 A
SS
12 V
VIN
12 V
VIN
2.6 V – UVLOfall
VOUT
5.0 V
5.75 V
POK
IPK_LIM
4.0 A
Figure 9: Excessive Voltage Correction Gain
IOUT_LIM
The Remote Load Regulation controller is also robust against
pin faults, such as adjacent pins shorting, shorting pins to GND,
or pin open faults. When a pin fault is detected, either the IADJ
or GADJ pin, the A8652/53 will default to the 800 mV reference, not applying any voltage correction to the output voltage.
Given this, the GADJ pin can be connected to GND to disable the
Remote Load Regulation function.
3.0 A
IOUT
0
1A
SS
POK
t
5.25 V
t
Figure 8: POK and Load-Side Current Limit Timing
In addition to current protection, the A8652/53 also includes a
115% (5.75 V) regulation voltage limit on the error amplifier.
This protection feature prevents the excessive output voltages
during fault conditions, and is therefore set above the operating
range of the Remote Load Regulation. However, if the voltage
correction gain is too high, the 115% voltage limit will impede
the Remote Load Regulation operation, as shown in Figure 9.
The VOUT waveform shows the operation point of the Remote
Load Regulation controller set by REF_ADJ, but as illustrated,
the 115% voltage regulation limit at the error amplifier clips the
output voltage to 5.75 V.
Compensation Components
To stabilize the regulator, a series RC compensation network
(RZ and CZ) must be connected from the error amplifier output
(COMP pin) to GND as shown in the applications schematic. In
most instances, an additional low-value capacitor (CP) should be
connected in parallel with the RZ-CZ compensation network to
reduce the loop gain at very high frequencies. However, if the CP
capacitor is too large, the phase margin of the converter may be
reduced. How to calculate RZ, CZ and CP is covered in the Component Selection section of this datasheet. When selecting the
compensation components, the load decoupling capacitance and
the wiring resistance must be taken into consideration.
If a fault occurs or the regulator is disabled, the COMP pin is
pulled to GND via the approximately 1 kΩ internal resistor and
PWM switching is inhibited.
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A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Slope Compensation
Power MOSFETs
The A8652/53 incorporates internal slope compensation to allow
PWM duty cycles above 50% for a wide range of input/output
voltages, switching frequencies, and inductor values. As shown
in the functional block diagram, the slope compensation signal
is added to the sum of the current sense and PWM Ramp Offset.
The amount of slope compensation is scaled with the switching
frequency when programming the frequency with a resistor or
with an external clock.
The A8652/53 includes an 80 mΩ, high-side N-channel MOSFET capable of delivering up to 4 A typical. The A8652/53 also
includes a 55 mΩ, low-side N-channel MOSFET to provide
synchronous rectification.
The value of the output inductor should be chosen such that slope
compensation rate SE is between 0.5× and 1× the falling slope of
the inductor current (SF).
Pulse-Width Modulation (PWM) Mode
Current Sense Amplifier
The A8652/53 incorporates a high-bandwidth current sense
amplifier to monitor the current through the top MOSFET. This
current signal is used to regulate the peak current when the top
MOSFET is turned on. The current signal is also used by the
protection circuitry for the pulse-by-pulse current limit (IPK_LIM)
and hiccup mode short-circuit protection.
Low Dropout Operation and Undervoltage
Lockout
The Undervoltage Lockout behavior is described in the following
Protection Features section.
The A8652/53 is designed to allow operation when input voltage drops as low as 2.6 V, which is the UVLO STOP threshold.
When the input voltage falls towards the nominal output voltage,
the high-side switch can remain on for maximum on-time to keep
regulating the output. This is accomplished by decreasing the fSW
switching frequency. In this way, the dropout from the input to
output voltage is minimized.
Sleep Mode with Enable input
The A8652/53 provides a shutdown function via the EN pin.
When this pin is low, the A8652/53 is shut down and the
A8652/53 will enter a “sleep mode” where the internal control
circuits will be shut off and draw less current from VIN. If EN
goes high, the A8652/53 will turn on and provided there are no
fault conditions, soft-start will be initiated and VOUT will ramp to
its final voltage in a time set by the soft-start capacitor (CSS). To
automatically enable the A8652/53, the EN pin may be connected
directly to VIN.
When the A8652/53 is disabled via the EN input being low or a
Fault condition, the A8652/53 output stage is tri-stated by turning
off both the upper and lower MOSFETs.
The A8652/53 employs fixed-frequency, peak current mode
control to provide excellent load and line regulation, fast transient
response, and simple compensation.
A high-speed comparator and control logic is included in
A8652/53. The inverting input of the PWM comparator is connected to the output of the error amplifier. The non-inverting
input is connected to the sum of the current sense signal, the
slope compensation signal, and a DC PWM Ramp offset voltage
(VPWM(OFFSET)).
At the beginning of each PWM cycle, the CLK signal sets the
PWM flip-flop, the bottom MOSFET is turned off, the top MOSFET is turned on, and the inductor current increases. When the
voltage at the non-inverting of PWM comparator rises above the
error amplifier output COMP, the PWM flip-flop is reset and the
top MOSFET is turned off, the bottom MOSFET is turned on and
the inductor current decreases.
The PWM flip-flop is reset-dominant, so the error amplifier may
override the CLK signal in certain situations.
BOOT Regulator
The A8652/53 includes a regulator to charge its boot capacitor.
The voltage across the boot capacitor is typically 5 V. If the boot
capacitor is missing, the A8652/53 will detect a boot overvoltage. Similarly, if the boot capacitor is shorted, the A8652/53 will
detect a boot undervoltage. Also, the boot regulator has a current
limit to protect itself during a short-circuit condition.
Soft-Start (Startup) and Inrush Current Control
The soft-start function controls the inrush current at startup. The
soft-start pin, SS, is connected to GND via a capacitor. When the
A8652/53 is enabled and all faults are cleared, the soft-start pin
will source the charging current ISS(SU), and the voltage on the
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17
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
soft-start capacitor, CSS, will ramp upward from 0 V. When the
voltage at the soft-start pin exceeds the Soft-Start COMP Release
Threshold (VSS(RELEASE), typically 400 mV) the error amplifier
will ramp up its output voltage above the PWM Ramp Offset. At
that instant, the top and bottom MOSFETs will begin switching.
There is a small delay (td(SS)) between the moments of EN pin
transitioning high and the soft-start voltage reaching 400mV to
initiate PWM switching.
Once the A8652/53 begins PWM switching, the error amplifier
will regulate the voltage at the FB pin to the soft-start pin voltage
minus approximately 400 mV. During the active portion of softstart, the voltage at the SS pin will rise from 400 mV to 1.2 V
(a difference of 800 mV), the voltage at the FB pin will rise from
0 V to 800 mV, and the regulator output voltage will rise from
0 V to the setpoint determined by the feedback resistor divider.
During startup, the PWM switching frequency is reduced to 25%
of fSW while VFB is below 200 mV. If VFB is above 200 mV but
below 400 mV, the switching frequency is 50% of fSW. At the
same time, the transconductance of the error amplifier, gmEA,
is reduced to 1/2 of nominal value when VFB is below 400 mV.
When VFB is above 400 mV, the switching frequency will be
fSW and the error amplifier gain will be the nominal value. The
reduced switching frequencies and error amplifier gain are necessary to help improve output regulation and stability when VOUT
is at very low voltage. When VOUT is very low, the PWM control
loop requires on-time near the minimum controllable on-time and
extra low duty cycles that are not possible at the nominal switching frequency.
When the voltage at the soft-start pin reaches approximately
1.2 V, the error amplifier will “switch over” and begin regulating
the voltage at the FB pin to the A8652/53 adjusted reference voltage. The voltage at the soft-start pin will continue to rise to the
internal LDO regulator output voltage.
If the A8652/53 is disabled or a fault occurs, the internal fault
latch is set and the capacitor at the SS pin is discharged to ground
very quickly through a 2 kΩ pull-down resistor. The A8652/53
will clear the internal fault latch when the voltage at the SS pin
decays to approximately 200 mV. However, if the A8652/53
enters hiccup mode, the capacitor at the SS pin is slowly discharged through a current sink, ISS(HIC). Therefore, the soft-start
capacitor CSS not only controls the startup time but also the time
between soft-start attempts in hiccup mode.
Pre-Biased Startup
If the output of the buck regulator is pre-biased at certain output
voltage level, the A8652/53 will modify the normal startup
routine to prevent discharging the output capacitors. As described
in the Soft-Start (Startup) and Inrush Current Control section,
the error amplifier usually becomes active when the voltage at
the soft-start pin exceeds 400 mV. If the output is pre-biased, the
voltage at the FB pin will be non-zero. The A8652/53 will not
start switching until the voltage at SS pin rises to approximately
VFB + 400 mV. From then on, the error amplifier becomes active,
the voltage at the COMP pin rises, PWM switching starts, and
VOUT will ramp upward from the pre-bias level.
Power OK (POK) Output
The Power OK (POK) output is an open-drain output, so an
external pull-up resistor must be connected. POK remains high
when the voltage at the FB pin is within regulation and the loadside current limit Iout_LIM is not triggered. The POK output is
pulled low under the conditions below:
1. VFB(RISING) < 92.5% of the reference voltage VREF
2. VFB(RISING) is larger than the sum of the adjusted reference
voltage (i.e. VREF + VREF_ADJ or 920 mV, whichever is
lower) and 80 mV
3. Load current exceeds the load-side current limit Iout_LIM
4. EN is low for more than 32 PWM cycles
5. VIN UVLO event occurs
6. Thermal shutdown event occurs
Once the load-side current limit is triggered, POK will go low
even if the output voltage has not yet dropped due to the current
limit. If the A8652/53 is running and EN is kept low for more
than 32 PWM cycles, POK will fall low and remain low only as
long as the internal circuitry is able to enhance the open-drain
output device. Once VIN fully collapses, POK will return to the
high-impedance state. Hysteresis is included in POK comparators
to prevent chattering due to the ripple effects on comparators’
input terminals.
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A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
PROTECTION FEATURES
The A8652/53 was designed to satisfy the most demanding
automotive and nonautomotive applications. In this section, a
description of each protection feature is described and Table 2
summarizes the protections and their operation.
Undervoltage Lockout Protection (UVLO)
An Undervoltage Lockout (UVLO) comparator in the A8652/53
monitors the voltage at the VIN pin and keeps the regulator disabled if the voltage is below the START threshold (VUVLO(START),
VIN rising) or the STOP threshold (VUVLO(STOP), VIN falling). The
UVLO comparator incorporates some hysteresis, VUVLO(HYS), to
help reduce ON/OFF cycling of the regulator due to the resistive
or inductive drops in the VIN path during heavy loading or during
startup.
Pulse-by-Pulse Overcurrent Protection (OCP)
The A8652/53 monitors the current in the upper MOSFET, and
if this current exceeds the pulse-by-pulse overcurrent threshold
(IPK_LIM), then the upper MOSFET is turned off. Normal PWM
operation resumes on the next clock pulse from the oscillator. The
A8652/53 includes leading-edge blanking to prevent falsely triggering the pulse-by-pulse current limit when the upper MOSFET
is turned on.
4.4
4.2
4.0
3.8
ILIM (A)
Figure 10 shows the typical and worst case pulse-by-pulse current
limits versus duty cycles at 2 MHz, 550 kHz, and 100 kHz.
The exact current the buck regulators can support is heavily
dependent on duty cycle (VIN, VOUT), ambient temperature,
thermal resistance of the PCB, airflow, component selection, and
nearby heat sources.
Overcurrent Protection (OCP) and Hiccup
Mode
An OCP counter and hiccup mode circuit protect the buck regulator when the output of the regulator is shorted to ground or when
the load current is too high. When the voltage at the SS pin is
below the Hiccup OCP Threshold, the hiccup mode counter is
disabled. Two conditions must be met for the OCP counter to be
enabled and begin counting:
1. VSS > VHIC(EN) (2.3 V) and
2. VCOMP clamped at its maximum voltage (OCL = 1)
As long as these two conditions are met, the OCP counter
remains enabled and will count pulses from the overcurrent
comparator. If the COMP voltage decreases (OCL = 0) the OCP
counter is cleared. If the OCP counter reaches OCPLIM counts
(240), a hiccup latch is set and the COMP pin is quickly pulled
down by a relatively low resistance (1 kΩ).
4.6
3.6
3.4
3.2
3.0
MAX_550 kHz
TYP_550 kHz
MIN_550 kHz
MAX_100 kHz
TYP_100 kHz
MIN_100 kHz
MAX_2 MHz
TYP_2 MHz
MIN_2 MHZ
2.8
2.6
2.4
2.2
Because of the addition of the slope compensation ramp to the
inductor current, the A8652/53 can deliver more current at lower
duty cycles than at higher duty cycles to activate pulse-by-pulse
overcurrent protection. Also the slope compensation is not a perfectly linear function of switching frequency, so the current limit
at lower switching frequency is larger compared with the limit at
higher switching frequency for a given duty cycle.
0
5
10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100
Duty Cycle (%)
Figure 10: Pulse-by-Pulse Current Limit vs. Duty Cycle
at 100 kHz (long dashed lines), 550 kHz (solid lines) and 2 MHz (short
dashed lines)
The hiccup latch also enables a small current sink connected to
the SS pin (ISS(HIC)). This causes the voltage at the soft start pin
to slowly ramp downward. When the voltage at the soft-start pin
decays to a low-enough level (VSS(RST), 200 mVTYP), the hiccup
latch is cleared and the small current sink turned off. At that
instant, the SS pin will begin to source current (ISS(SU)) and the
voltage at the SS pin will ramp upward. This marks the beginning of a new, normal soft-start cycle as described earlier. When
the voltage at the soft-start pin exceeds the error amp voltage
by approximately 400 mV, the error amp will force the voltage
at the COMP pin to quickly slew upward and PWM switching
will resume. If the short circuit at the regulator’s output remains,
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115 Northeast Cutoff
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19
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
another hiccup cycle will occur. Hiccups will repeat until the
short circuit is removed or the converter is disabled. If the short
circuit is removed, the A8652/53 will soft-start normally and the
output voltage will automatically recover to the desired level.
Thus Hiccup mode is a very effective protection for the overload
condition. It can avoid false trigger for a short-term overload. On
the other hand, for the extended overload, the average power dissipation during Hiccup operation is very low to keep the controller cool and enhance the reliability.
Note that OCP is the only fault that results in Hiccup mode being
ignored while VSS < 2.3 V.
BOOT Capacitor Protection
The A8652/53 monitors the voltage across the BOOT capacitor to detect if the capacitor is missing or short-circuited. If the
BOOT capacitor is missing, the regulator will enter Hiccup mode
after 7 PWM cycles. If the BOOT capacitor is short-circuited, the
regulator will enter Hiccup mode after 64 PWM cycles.
For a BOOT fault, Hiccup mode will operate virtually the same
as described previously for an output short-circuit fault (OCP),
with SS ramping up and down as a timer to initiate repeated
soft-start attempts. BOOT faults are nonlatched condition, so the
A8652/53 will automatically recover when the fault is corrected.
Dynamic Overvoltage Protection (OVP)
In addition to the error amp regulation voltage clamp
(VEA(CLAMP)) at 115%, the A8652/53 includes a dynamic
overvoltage protection feature where the overvoltage threshold
changes with the correction voltage. As shown in Figure 11
below, the A8652/53 also includes an overvoltage comparator
that monitors the FB pin and the sum of the adjusted reference
voltage (i.e. VREF + VREF_ADJ or 920 mV, whichever is lower)
and 80 mV. In this way, the overvoltage threshold will dynamically change with the amount of the correction voltage and the
threshold is always 0.5 V above the output voltage. For example,
OVP threshold is 5.5 V when VOUT = 5 V at IOUT = 0 A; the OVP
threshold will be 5.75 V when VOUT = 5.25 V at IOUT = 1 A; if
VOUT = 5.75 V at certain load, the OVP threshold will become
6.25 V. When the Remote Load Regulation function is disabled
due to some reason (e.g., IADJ or GADJ pin fault or general non-
FB
OV
5 µs
80 mV
800 mV
VREF
920 mV
REF_ADJ
Figure 11: Dynamic Overvoltage Protection
USB buck application), the OVP threshold will be 5.5 V; when
the output voltage reaches the 115% regulation limit, the OVP
threshold will reach the maximum value of 6.25 V.
When the voltage at the FB pin exceeds the overvoltage threshold
(VPOK(OV)), A8652/53 will stop PWM switching, i.e. both high
and side switches will be turned off, and POK will be pulled low.
In most cases, the error amplifier will be able to maintain regulation since the synchronous output stage has excellent sink and
source capability. However the error amplifier and its regulation
voltage clamp are not effective when the FB pin is disconnected
or when the output is shorted to the input supply. When the FB
pin is disconnected from the feedback resistor divider, a tiny
internal current source will force the voltage at the FB pin to rise
above VPOK(OV) and disable the regulator, preventing the load
from being significantly overvoltage. If a higher external voltage is accidently shorted to the A8652/53’s output, VFB will rise
above the overvoltage threshold, triggering an OVP event and
thus protecting the low-side switch. In either case, if the conditions causing the overvoltage are corrected, the regulator will
automatically recover.
To provide additional protection when the battery is shorted to
the load terminal, a 40 V Schottky diode (D) can be inserted after
sensing resistor Rsen to block the high voltage entering into IC,
and a Zener diode is placed at FB pin, as shown in Figure 12. The
test result was shown in Figure 12b when Battery = 36 V was
shorted to VOUT.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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20
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Rsen
20 mΩ
LO 10 µH
CO
2 × 22 µF
VOUT
D
RWIRE/2
62.5 mΩ
RWIRE/2
62.5 mΩ
VLOAD
CLOAD
100 µF
ISEN+
ISENRFB1
FB
5.1 V
RFB2
Figure 12a: Protection Circuitry for
Load Short-to-Battery
FB
VOUT
Thermal Shutdown (TSD)
The A8652/53 monitors its junction temperature and will stop
PWM switching and pull POK low if it becomes too hot. Also, to
prepare for a restart, the SS and COMP pins will be pulled low
until VSS < VSS(RST). TSD is a non-latched fault so the A8652/53
will automatically recover if the junction temperature decreases
by approximately 20°C.
Pin-to-Ground and Pin-to-Pin Short Protections
The A8652/53 was designed to satisfy the most demanding
automotive and nonautomotive applications. For example, the
A8652/53 was carefully designed “up front” to withstand a short
circuit to ground at each pin without suffering damage.
In addition, care was taken when defining the A8652/53’s pinouts
to optimize protection against pin-to-pin adjacent short-circuits.
For example, logic pins and high-voltage pins were separated as
much as possible. Inevitably, some low-voltage pins were located
adjacent to high-voltage pins. In these instances, the low-voltage
pins were designed to withstand increased voltages, with clamps
and/or series input resistance, to prevent damage to the A8652/53.
POK
Figure 12b: Test Results when Battery = 36 V is
Shorted to VOUT
Ch1: VOUT (10 V/div); Ch2: VFB (2 V/div);
Ch4: VPOK (5 V/div); 50 ms/div
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115 Northeast Cutoff
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21
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Table 2: Summary of A8652/53 Fault Modes and Operation
During Fault Count, before Hiccup
Dropout
Foldback
POK
BOOT
Charging
LATCH
RESET
CCM
according to
COMP
fSW/4 or fSW/2
based on VFB
Depends on
VOUT and
ISEN
Not
affected
No
Auto,
remove
short
CCM
according to
COMP
CCM
according to
COMP
Not affected
Pulled Low
immediately
Not
affected
No
Auto
Pulled low only by
hiccup
Forced off
Immediately
Forced off
Immediately
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Not
affected
No
Auto,
remove
short
Hiccup at the
end of blankOn
Pulled low only by
hiccup
Forced off
Immediately
One Shot
Diode
Emulation
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Not
affected
No
(option
avail
able)
Auto,
remove
short
Die is too hot
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Forced off
Immediately
Forced off
Immediately
Dropout
Foldback
Reset
Pulled Low
Immediately
Off
No
Auto,
Cool
Down
Boot
Capacitor
Greater than
7V
BOOT capacitor
Open
Hiccup, after 7
latched faults
Pulled low by
hiccup
CCM
according to
COMP
CCM
according to
COMP
Dropout
Foldback
Disabled by
Hiccup
Depends on
VOUT and
ISEN
Off for rest
of period
–
–
Boot
Capacitor On
Fault
BOOT Capacitor
Open
Hiccup, after 7
latched faults
Pulled low by
hiccup
CCM
according to
COMP
CCM
according to
COMP
Dropout
Foldback
Disabled by
Hiccup
Depends on
VOUT and
ISEN
Off only
during
hiccup
No
Auto,
replace
capacitor
Boot
Capacitor
Overcurrent
BOOT to GND
Short
Not affected
Not affected
Not affected
Pulsed at
minOff
Not affected
Depends on
VOUT and
ISEN
Off until
fault clears
–
–
Boot
Capacitor
Low Voltage
Normal Low VIN
Operation
Not affected
Not affected
Not affected
Active during
minOff period
Not affected
Depends on
VOUT and
ISEN
On
–
–
Boot
Capacitor
Undervoltage
BOOT Capacitor
Short
Not affected
Not affected
Forced Off
Immediately
Active during
minOff period
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
On
–
–
Low-Side
Switch
Undervoltage
Low VIN
Not affected
Not affected
Forced Off
Immediately
Forced Off
Immediately
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Not
affected
–
–
VREG
Undervoltage
Low VIN
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Forced Off
Immediately
Forced Off
Immediately
Dropout
Foldback
Reset
Pulled Low
Immediately
Off
No
Auto
VIN
Undervoltage
Low VIN
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Forced Off
Immediately
One Shot
Diode
Emulation
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Off
No
Auto
Fault Mode
Fault Cases
VSS
Positive
Overcurrent
Protection
1. Excessive IOUT
2. VOUT Shorted
to GND
3. SW Soft Short
To GND
Load-Side
Current Limit
Excessive IOUT
Negative
Overcurrent
Protection
SW Hard
Short to GND
VCOMP
High-Side
Switch
Low-Side
Switch
Hiccup, after 240
faults of OCL
Clamped to
achieve ILIM, and
pulled low only by
hiccup
CCM
according to
COMP
Not affected
Not affected
1. Excessive
Negative IOUT
2. Inductor Short
Hiccup, after 1
fault of LSOC
SW to GND hard
Short
Thermal
Shutdown
Continued on next page...
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
22
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Table 2: Summary of A8652/53 Fault Modes and Operation (continued)
During Fault Count, before Hiccup
Fault Mode
Hiccup Delay
(after fault
count is
reached)
Hiccup
Restart or
Startup (after
VSS returns to
VSS(RST))
Fault Cases
Hiccup
Startup Hiccup
FB
Overvoltage
1. VOUT to VIN
Short
2. FB pin Open
ISENP
Overvoltage
1. VOUT to VIN
short
2. FB to GND
short
VSS
VCOMP
High-Side
Switch
Low-Side
Switch
Dropout
Foldback
POK
BOOT
Charging
LATCH
RESET
–
–
Discharged with
ISS(HIC) until VSS
< VSS(RST)
Pulled Low until
VSS < VSS(RST)
Forced Off at
Start of Period
One Shot
Diode
Emulation
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Not
affected,
(off only
for boot
capacitor
faults)
(sleep opt
available)
Charged with
ISS(SU)
Released
from 0 V, then
responds to VSS↑
CCM after
VCOMP > 400
mV
CCM after
VCOMP >
400 mV
(pulsed at
minOff)
Dropout
Foldback
Reset
Depends on
VOUT and
ISEN
Not
affected
–
–
Not affected
Not affected
Forced Off
Immediately
One Shot
Diode
Emulation
Dropout
Foldback
Reset
Pulled Low
Immediately
Off
No
Auto,
VFB to
normal
range
Not affected
Not affected
Forced Off
Immediately
One Shot
Diode
Emulation
Dropout
Foldback
Reset
Pulled Low
Immediately
Off
No
Auto,
VFB to
normal
range
FB
Undervoltage
Startup
Not affected
Not affected
CCM
according to
COMP
CCM
according to
COMP
Not affected
Pulled Low
Immediately
Not
affected
No
Auto,
VFB to
normal
range
Feedback
Less Than
400 mV
Startup
Not affected
Not affected
fSW/2
CCM
according to
COMP
Not affected
Pulled Low
Not
affected
–
–
Feedback
Less Than
200 mV
Startup VOUT to
GND Short
Not affected
Not affected
fSW/4
CCM
according to
COMP
fSW already
at 1/4
Pulled Low
Not
affected
–
–
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Pulled Low
Immediately &
latched until VSS
< VSS(RST)
Forced Off
Immediately
One Shot
Diode
Emulation
Not affected
Depends on
VOUT and
ISEN
Off
–
–
FSET
Resistor Fault
1.FSET to GND
short
2.FSET pulled
high
3.Low R
4.High R
GADJ or
IADJ Resistor
Fault
ADJ to GND short
ADJ pulled high
Low R High R
Not affected
Not affected
CCM
according to
COMP
CCM
according to
COMP
Not affected
Depends on
VOUT and
ISEN
Not
affected
–
–
SS shorted
to VIN
SS to VIN short
Clamped to
Zener voltage
internally
Not affected
CCM
according to
COMP
CCM
according to
COMP
Not affected
Depends on
VOUT and
ISEN
Not
affected
–
–
SS shorted to
GND
SS to GND short
At GND
Loop response
only
CCM
according to
COMP
CCM
according to
COMP
Not affected
Depends on
VOUT and
ISEN
Not
affected
–
–
COMP
shorted to
GND
COMP to GND
short
Not affected
At GND
CCM
according to
COMP
CCM
according to
COMP
Not affected
Depends on
VOUT and
ISEN
Not
affected
–
–
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23
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
DESIGN AND COMPONENT SELECTION
Setting the Output Voltage
The output voltage of the regulator is determined by connecting
a resistor divider from the output node (VOUT) to the FB pin as
shown in Figure 13. There are tradeoffs when choosing the value
of the feedback resistors. If the series combination (RFB1 + RFB2)
is too low, then the light load efficiency of the regulator will be
reduced. To maximize the efficiency, it is best to choose higher
values of resistors. On the other hand, if the parallel combination
(RFB1//RFB2) is too high, then the regulator may be susceptible to
noise coupling onto the FB pin. 1% resistors are recommended to
maintain the output voltage accuracy.
1.8 × 10
3
1.6 × 10
3
1.4 × 10
3
1.2 × 103
1 × 10
3
800
600
400
CFB
FB Pin
Frequency (kHz)
2.0 × 103
200
VOUT
0
10
RFB1
20
30
40
50
60
70
80
90 100 110 120 130 140 150 160
RFSET (kΩ)
RFB2
Figure 14: PWM Switching Frequency versus RFSET
the FSET resistor, RFSET (x-axis).
For a desired switching frequency (fSW), the FSET resistor can be
calculated using equation 11, where fSW is in kHz and RFSET is in
kΩ.
Figure 13: Connecting a Feedback Divider to
Set the Output Voltage
The feedback resistors must satisfy the ratio shown in equation
below to produce a desired output voltage, VOUT.
VOUT
RFB1
–1
RFB2 = 0.8 V
RFSET =
(9)
A phase lead capacitor (CFB) can be connected in parallel with
RFB1 to increase the phase and gain margins. It adds a zero and
pole to the compensation network and boosts the loop phase
at the crossover frequency. In general, CFB should be less than
25 pF. If CFB is too large, it will have no effect.
If CFB is used, CFB can be calculated from equation 10:
CFB =
1
2πRFB1fc
(10)
where fc is crossover frequency.
PWM Switching Frequency (fSW, RFSET)
The PWM switching frequency is set by connecting a resistor
from the FSET pin to ground. Figure 14 is a graph showing the
relationship between the typical switching frequency (y-axis) and
26000
– 2.2
fSW
(11)
When the PWM switching frequency is chosen, the designer
should be aware of the minimum controllable on-time, tON(MIN),
of the A8652/53. If the system’s required on-time is less than
the minimum controllable on-time, pulse skipping will occur
and the output voltage will have increased ripple or oscillations. The PWM switching frequency should be calculated using
equation 12, where VOUT is the output voltage, tON(MIN) is the
minimum controllable on-time of the A8652/53 (See EC table),
and VIN(MAX) is the maximum required operational input voltage
(not the peak surge voltage).
fSW <
VOUT
tON(MIN) × VIN(MAX)
(12)
If the A8652/53 synchronization function is employed, the base
switching frequency should be chosen such that pulse skipping
will not occur at the maximum synchronized switching frequency
according to equation 12 (i.e. 1.5 × fSW is less than the result
from equation 12).
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115 Northeast Cutoff
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24
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Output Inductor (LO)
For a peak current mode regulator, it is common knowledge that
without adequate slope compensation, the system will become
unstable when the duty cycle is near or above 50%. However,
the slope compensation in the A8652/53 is a fixed value (SE).
Therefore, it is important to calculate an inductor value so the
falling slope of the inductor current (SF) will work well with the
A8652/53 slope compensation. Equations 13 and 14 can be used
to calculate a range of values for the output inductor based on
the well-known approach of providing slope compensation that
matches 50% to 100% of the down slope of the inductor current.
In equation 13, use the slope compensation (SE), which is a function of switching frequency according to equation 14.
VOUT
VOUT
≤
L
O ≤
2 × SE
SE
2
SE = 0.0445 × fSW + 0.5612 × fSW
2
SE = 0.0237 × fSW + 0.3529 × fSW
(13)
(14a for
A8653)
(14b for
A8652)
SE is in A/µs, fSW is in MHz, and LO will be in µH.
If equations 13 or 14 yield an inductor value that is not a standard
value, then the next highest available value should be used. The
final inductor value should allow for 10%-20% of initial tolerance and 20%-30% of inductor saturation.
The saturation current of the inductor should be higher than the
peak current capability of the A8652/53. Ideally, for output shortcircuit conditions, the inductor should not saturate at the highest
pulse-by-pulse current limit at minimum duty cycle. This may be
too costly. At the very least, the inductor should not saturate at
the peak operating current according to equation 15. In equation 15, VIN(MAX) is the maximum continuous input voltage.
IPEAK = 4.62 –
SE × VOUT
1.15 × fSW × VIN(MAX)
(15a) for
A8653
IPEAK = 2.1 –
SE × VOUT
1.15 × fSW × VIN(MAX)
(15b) for
A8652
Subtracting half of the inductor ripple current from equation 15
gives an interesting equation to predict the typical DC load capability of the regulator at a given duty cycle (D),
IOUT(DC)≤ 4.62 –
V × (1 – D)
SE × D
– OUT
fSW
2 × fSW × LO
(16a) for
A8653
IOUT(DC)≤ 2.1 –
SE × D
V × (1 – D)
– OUT
fSW
2 × fSW × LO
(16b) for
A8652
After an inductor is chosen, it should be tested during output
short-circuit conditions. The inductor current should be monitored using a current probe. A good design should ensure neither
the inductor nor the regulator are damaged when the output is
shorted to ground at maximum input voltage and the highest
expected ambient temperature.
Output Capacitors
The output capacitors filter the output voltage to provide an
acceptable level of ripple voltage and they store energy to help
maintain voltage regulation during a load transient. The voltage
rating of the output capacitors must support the output voltage
with sufficient design margin. The output voltage ripple (ΔVOUT)
is a function of the output capacitors parameters: CO, ESRCO,
ESLCO.
VOUT = IL × ESRCO +
IL
VIN – VOUT
× ESLCO +
LO
8fSWCO
(17)
The type of output capacitors will determine which terms of
equation 17 are dominant.
For ceramic output capacitors, the ESRCO and ESLCO are virtually zero, so the output voltage ripple will be dominated by the
third term of equation 17.
VOUT =
IL
8fSWCO
(18)
To reduce the voltage ripple of a design using ceramic output
capacitors, simply increase the total capacitance, reduce the
inductor current ripple (i.e. increase the inductor value), or
increase the switching frequency.
For electrolytic output capacitors, the value of capacitance will be
relatively high, so the third term in equation 17 will be very small
and the output voltage ripple will be determined primarily by the
first two terms of equation 17.
VOUT = IL × ESRCO +
VIN – VOUT
× ESLCO
LO
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Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
(19)
25
A8652,
A8653
To reduce the voltage ripple of a design using electrolytic output
capacitors, simply decrease the equivalent ESRCO and ESLCO
by using a high(er) quality capacitor, or add more capacitors in
parallel, or reduce the inductor current ripple (i.e. increase the
inductor value).
The ESR of some electrolytic capacitors can be quite high so
Allegro recommends choosing a quality capacitor for which the
ESR or the total impedance is clearly documented in the datasheet. Also, the ESR of electrolytic capacitors usually increases
significantly at cold ambients, as much as 10×, which increases
the output voltage ripple and in most cases reduces the stability
of the system.
The transient response of the regulator depends on the quantity
and type of output capacitors. In general, minimizing the ESR of
the output capacitance will result in a better transient response.
The ESR can be minimized by simply adding more capacitors in
parallel or by using higher quality capacitors. At the instant of a
fast load transient (di/dt), the output voltage will change by the
amount:
VOUT = ILOAD × ESRCO +
di
ESLCO
dt
(20)
After the load transient occurs, the output voltage will deviate
from its nominal value for a short time. This time will depend
on the system bandwidth, the output inductor value, and output
capacitance. Eventually, the error amplifier will bring the output
voltage back to its nominal value.
The speed at which the error amplifier will bring the output
voltage back to its setpoint mainly depends on the closed-loop
bandwidth of the system. A higher bandwidth usually results in
a shorter time to return to the nominal voltage. However, with
a higher bandwidth system, it may be more difficult to obtain
acceptable gain and phase margins. Selection of the compensation components (RZ, CZ, CP) are discussed in more detail in the
Compensation Components section of this datasheet.
Input Capacitors
Three factors should be considered when choosing the input
capacitors. First, they must be chosen to support the maximum
expected input surge voltage with adequate design margin.
Second, the capacitor RMS current rating must be higher than
the expected RMS input current to the regulator. Third, they must
have enough capacitance and a low enough ESR to limit the input
voltage dV/dt to something much less than the hysteresis of the
IRMS / IOUT
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
0.55
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0.00
0
10
20
30
60
40
50
Duty Cycle (%)
70
80
90
100
Figure 15: Input Capacitor Ripple vs. Duty Cycle
VIN pin UVLO circuitry (VUVLO(HYS), nominally 800 mV for the
A8652/53) at maximum loading and minimum input voltage.
The input capacitors must deliver the RMS current according to:
IRMS = IO D × (1 – D)
(21)
where the duty cycle D is D ≈ VOUT / VIN. Figure 15 shows the
normalized input capacitor RMS current versus duty cycle. To
use this graph, simply find the operational duty cycle (D) on the
x-axis and determine the input/output current multiplier on the
y-axis. For example, at a 20% duty cycle, the input/output current
multiplier is 0.40. Therefore, if the regulator is delivering 2.6 A of
steady-state load current, the input capacitor(s) must support 0.40
× 2.6 A or 1.04 ARMS.
The input capacitor(s) must limit the voltage deviations at the
VIN pin to something significantly less than the A8652/53 UVLO
hysteresis during maximum load and minimum input voltage.
The minimum input capacitance can be calculated as follows:
CIN ≥
IOUT × D × (1 – D)
0.85 × fSW × ΔVIN(MIN)
(22)
where ΔVIN(MIN) is chosen to be much less than the hysteresis
of the VIN UVLO comparator (ΔVIN(MIN) ≤ 150 mV is recommended), and fSW is the nominal PWM frequency.
The D × (1 – D) term in equation 20 has an absolute maximum
value of 0.25 at 50% duty cycle. So, for example, a very conservative design based on IOUT = 2.6 A, fSW = 85% of 425 kHz,
D × (1 – D) = 0.25, and ΔVIN = 150 mV,
CIN ≥
2.6 A × 0.25
= 12 µF
361 kHz × 150 mV
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26
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
A good design should consider the DC bias effect on a ceramic
capacitor: as the applied voltage approaches the rated value, the
capacitance value decreases. This effect is very pronounced with
the Y5V and Z5U temperature characteristic devices (as much as
90% reduction) so these types should be avoided. The X5R and
X7R type capacitors should be the primary choices due to their
stability versus both DC bias and temperature.
To avoid prematurely triggering Hiccup mode, the soft-start
capacitor (CSS) should be calculated according to equation below,
}
ILIM
ILOAD
Output
Capacitor
Current (ICO)
For all ceramic capacitors, the DC bias effect is even more
pronounced on smaller case sizes, so a good design will use the
largest affordable case size (i.e. 1206 or 1210). Also, it is advisable to select input capacitors with plenty of design margin in
the voltage rating to accommodate the worst-case transient input
voltage (such as a load dump as high as 40 V for automotive
applications).
tSS
Bootstrap Capacitor
A bootstrap capacitor must be connected between the BOOT and
SW pins to provide the floating gate drive to the high-side MOSFET. Usually, 100 nF is an adequate value. This capacitor should
be a high-quality ceramic capacitor, such as an X5R or X7R, with
a voltage rating of at least 16 V.
Soft-Start and Hiccup Mode Timing (CSS)
The soft-start time of the A8652/53 is determined by the value of
the capacitance at the soft-start pin (CSS).
When the A8652/53 is enabled, the voltage at the soft-start pin
will start from 0 V and will be charged by the soft-start current
(ISS(SU)). However, PWM switching will not begin instantly
because the voltage at the soft-start pin must rise above 400 mV.
The soft-start delay (td(SS)) can be calculated using equation
below,
td(SS) = C SS ×
(400I mV)
SS(SU)
(23)
If the A8652/53 is starting with a very heavy load, a very fast
soft-start time may cause the regulator to exceed the pulse-bypulse overcurrent threshold. This occurs because the sum of the
full load current, the inductor ripple current, and the additional
current required to charge the output capacitors,
ICO = CO × VOUT / tSS
is higher than the pulse-by-pulse current threshold, as shown in
Figure 16. This phenomena is more pronounced when using highvalue electrolytic type output capacitors.
Figure 16: Output Current (ICO) During Startup
CSS ≥
ISS(SU) × VOUT × CO
0.8 V × ICO
(24)
where VOUT is the output voltage, CO is the output capacitance,
ICO is the amount of current allowed to charge the output capacitance during soft-start (recommend 0.1 A < ICO < 0.3 A). Higher
values of ICO result in faster soft-start times. Howewer, lower
values of ICO ensure that Hiccup mode is not falsely triggered.
Allegro recommends starting the design with an ICO of 0.1 A and
increasing it only if the soft-start time is too slow. If a nonstandard capacitor value for CSS is calculated, the next larger value
should be used.
The output voltage ramp time (tSS) can be calculated by using
either of the following methods:
COUT
CSS
tSS = VOUT × I or 0.8 V × I
SS(SU)
CO
(25)
When the A8652/53 is in hiccup mode, the soft-start capacitor is used as a timing capacitor and sets the hiccup period. The
soft-start pin charges the soft-start capacitor with ISS(SU) during
a startup attempt and discharges the same capacitor with ISS(HIC)
between startup attempts. Because the ratio of ISS(SU)/ISS(HIC) is
approximately 4:1, the time between hiccups will be about four
times as long as the startup time. Therefore, the effective duty
cycle will be very low and the junction temperature will be kept
low.
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27
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Remote Load Regulation Control Components Compensation Components (RZ, CZ, CP)
To compensate the voltage drop across the wiring harness, the
wire resistance RWIRE must be know as a priori. The current sense
resistor Rsen,which is connected in series with the load after the
output capacitor CO, is recommended to have a value between
20 and 50 mΩ, considering the tradeoff between measurement
accuracy and power dissipation.
When the load side current limit IOUT_LIM is set, the resistor
RIADJ at pin IADJ can be calculated from the equation below:
1,200
RIADJ =
IOUT_LIM × Rsen
The needed amount of voltage correction should be equal to the
voltage drop across the wiring harness:
IOUT × RWIRE = IOUT × Rsen ×
RIADJ
× AFB
RGADJ
Thus for given Rsen, RIADJ, RWIRE, the resistance RGADJ at pin
GADJ can be determined from the equation above:
Rsen × RIADJ
RGADJ =
× AFB
RWIRE
Where AFB = VOUT / VFB is the gain of the FB pin voltage
divider.
If the dynamic voltage correction adjustment at pin GADJ is
desired (refer to Figure 5), then RGADJ in the equation above
should be replaced with the equivalent resistance RG_eqv at pin
GADJ:
RG_eqv =
(R
1
GADJ
1
V –1
– G
RG_E
)
Similarly the equivalent resistance RIADJ at pin IADJ can be
calculated below if an external voltage VI is applied at pin IADJ
as shown in Figure 7:
RI_eqv =
(
1
1
V –1
– I
RIADJ
RI_E
)
The GADJ and IADJ pins are designed for a resistance range
of 10 to 34 kΩ, and therefore the controlling voltage VG and VI
must be limited as follows:
10 kΩ < RG_eqv < 34 kΩ
and,
To compensate the system, it is important to understand where
the buck power stage, load resistance, and output capacitance
form their poles and zeroes in frequency. Also, it is important
to understand that the (Type II) compensated error amplifier
introduces a zero and two more poles and where these should be
placed to maximize system stability, provide a high bandwidth,
and optimize the transient response.
First, consider the power stage of the A8652/53, the output
capacitors, and the load resistance. This circuitry is commonly
referred as the “control to output” transfer function. The low frequency gain of this section depends on the COMP to SW current
gain (gmPOWER), and the value of the load resistor (RL). The DC
gain (GCO(0 HZ)) of the control-to-output is
GCO(0 Hz) = gmPOWER × RL
(26)
The control to output transfer function has a pole (fP1) formed by
the output capacitance (COUT) and load resistance (RL) at
fP1 =
1
2π × RL × COUT
(27)
The control to output transfer function also has a zero (fZ1)
formed by the output capacitance (COUT) and its associated ESR
fZ1 =
1
2π × ESR × COUT
(28)
For a design with very low ESR type output capacitors (i.e.
ceramic or OSCON output capacitors), the ESR zero, fZ1, is
usually at a very high frequency, so it can be ignored. On the
other hand, if the ESR zero falls below or near the 0 dB crossover
frequency of the system (as is the case with electrolytic output
capacitors), then it should be cancelled by the pole formed by the
CP capacitor and the RZ resistor (discussed and identified later as
fP3).
Next, consider the feedback resistor divider, (RFB1 and RFB2), the
error amplifier (gmEA), and its compensation network RZ/CZ/CP.
It greatly simplifies the transfer function derivation if RO >> RZ,
and CZ >> CP (where RO is the error amplifier output impedance). In most cases, RO > 2 MΩ, 1 kΩ < RZ < 100 kΩ, 220 pF <
CZ < 47 nF, and CP < 50 pF, so the following equations are very
accurate.
The low frequency gain of the control section (GC(0 Hz)) is formed
by the feedback resistor divider and the error amplifier. It can be
calculated using equation 29:
10 kΩ < RI_eqv < 34 kΩ
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A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
RFB2
× gmEA × RO
RFB1 + RFB2
VFB
=
× gmEA × RO
VOUT
VFB
× AVOL
=
VOUT
A Generalized Tuning Procedure
GC(0 Hz) =
(29)
where
VOUT is the output voltage,
VFB is the reference voltage (0.8 V),
gmEA is the error amplifier transconductance (750 µA/V),
and
RO is the error amplifier output impedance (AVOL/gmEA)
The transfer function of the Type-II compensated error amplifier
has a (very) low frequency pole (fP2) dominated by the output
error amplifier’s output impedance RO and the CZ compensation
capacitor,
1
fP2 =
2π × RO × CZ
(30)
The transfer function of the Type-II error amplifier also has a
low frequency zero (fZ2) dominated by the RZ resistor and the CZ
capacitor.
fZ2 =
1
2π × RZ × CZ
(31)
Lastly, the transfer function of the Type-II compensated error
amplifier has a (very) high frequency pole (fP3) dominated by the
RZ resistor and the CP capacitor
fP3 =
1
2π × RZ × CP
(32)
Placing fZ2 just above fP1 will result in excellent phase margin,
but relatively slow transient recovery time.
The magnitude and phase of the entire system are simply the sum
of the error amplifier response and the control-to-output response.
1. Choose the system bandwidth, fC, the frequency at which the
magnitude of the gain will cross 0 dB. Recommended values
for fC based on the PWM switching frequency are fSW/20
< fC < fSW/7.5. A higher value of fC will generally provide
a better transient response while a lower value of fC will be
easier to obtain higher gain and phase margins.
2. Calculate the RZ resistor value to set the desired system
bandwidth (fC),
VOUT
2 × π × COUT
RZ = fC ×
×
(33)
VFB
gmPOWER × gmEA
3. Determine the frequency of the pole (fP1) formed by COUT
and RL by using equation 27 (repeated here).
1
fP1 =
2π × RL × COUT
4. Calculate a range of values for the CZ capacitor,
4
1
< CZ <
2 × π × RZ × fc
2 × π × RZ × 1.5 × fP1
(34)
To maximize system stability (i.e. have the most gain margin), use a higher value of CZ. To optimize transient recovery
time at the expense of some phase margin, use a lower value
of CZ.
5. Calculate the frequency of the ESR zero (fZ1) formed by the
output capacitor(s) by using equation 28 (repeated here).
1
fZ1 =
2π × ESR × COUT
A.If fZ1 is at least 1 decade higher than the target crossover
frequency (fC), then fZ1 can be ignored. This is usually
the case for a design using ceramic output capacitors. Use
equation 32 to calculate the value of CP by setting fP3 to
either 5 × fC or fSW/2, whichever is higher.
B. On the other hand, if fZ1 is near or below the target
crossover frequency (fC) then use equation 32 to calculate
the value of CP by setting fP3 equal to fZ1. This is usually
the case for a design using high ESR electrolytic output
capacitors.
Referring to Typical Application Diagram 1 on page 1, several
typical designs are provided in Table 3 with VLOAD = 5 V.
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29
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Table 3: Recommended Components Values for Several Typical Designs
Design
fSW
RFSET
LO
CO
RZ + CZ // CP
Rsen
RGADJ
RIADJ
A8653 (set IOUT_LIM = 2.75 A, RWIRE = 125 mΩ)
A
500 kHz
52.3 kΩ
10 µH (74437368100)
44 µF
14 kΩ + 2.7 nF//33 pF
20 mΩ
20.0 kΩ
20.0 kΩ
B
1 MHz
23.7 kΩ
6.8 µH (74437368068)
44 µF
14 kΩ + 2.7 nF//33 pF
20 mΩ
20.0 kΩ
20.0 kΩ
C
2 MHz
10.5 kΩ
6.8 µH (74437368068)
32 µF
37.4 kΩ + 1.8 nF//4.7 pF
20 mΩ
20.0 kΩ
20.0 kΩ
A8652 (set IOUT_LIM = 1.2 A, RWIRE = 200 mΩ)
D
500 kHz
52.3 kΩ
33 µH (74437368330)
20 µF
14 kΩ + 2.7 nF//33pF
50 mΩ
31.6 kΩ
20.0 kΩ
E
2 MHz
10.5 kΩ
10 µH (74437368100)
20 µF
24.3 kΩ + 2.2 nF//4.7 pF
50 mΩ
31.6 kΩ
20.0 kΩ
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30
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
POWER DISSIPATION AND THERMAL CALCULATIONS
The power dissipated in the A8652/53 is the sum of the power
dissipated from the VIN supply current (PIN), the power dissipated
due to the switching of the high-side power MOSFET (PSW1), the
power dissipated due to the RMS current being conducted by the
high-side MOSFET (PCOND1) and low-side MOSFET (PCOND2),
and the power dissipated by both gate drivers (PDRIVER).
The power dissipated from the VIN supply current can be calculated using equation 35,
PIN = VIN × IQ + (VIN – VGS ) × (QG1 + QG2 ) × fSW (35)
where
Similarly, the conduction losses dissipated by the low-side MOSFET while it is conducting can be calculated by the following
equation:
PCOND2 = I
2
RMS,FET
(
× R DS(ON)LS = 1 –
VOUT
I
2
× I OUT + L × R DS(ON)LS
VIN
12
(38)
)(
2
)
where
IOUT is the regulator output current,
ΔIL is the peak-to-peak inductor ripple current,
RDS(ON)HS is the on-resistance of the high-side MOSFET,
VIN is the input voltage,
IQ is the input quiesent current drawn by the A8652/53 (see
EC table),
VGS is the MOSFET gate drive voltage (typically 5 V),
QG1 and QG2 is the internal high-side and low-side MOSFET
gate charges (approximately 5.8 nC and 10.4 nC, respectively), and
fSW is the PWM switching frequency.
The power dissipated by the high-side MOSFET during PWM
switching can be calculated using equation 36,
VIN × IOUT × (tr + tf ) × fSW
PSW1 =
2
(36)
where
RDS(ON)LS is the on-resistance of the low-side MOSFET
The RDS(ON) of the both MOSFETs have some initial tolerance
plus an increase from self-heating and elevated ambient temperatures. A conservative design should accomodate an RDS(ON) with
at least a 15% initial tolerance plus 0.39%/°C increase due to
temperature.
The power dissipated from the low-side MOSFET body diode
during the non-overlap time can be calculated as follows:
PNO = VSD × IOUT × 2 × tNO × fSW (39)
where
VSD is the source-to-drain voltage of the low-side MOSFET
(typically 0.60 V), and
tNO is the non-overlap time (15 ns(typ))
VIN is the input voltage,
IOUT is the regulator output current,
The sum of the power dissipated by the internal gate driver can be
calculated using equation 40,
fSW is the PWM switching frequency, and
tr and tf are the rise and fall times measured at the SW node.
where
The exact rise and fall times at the SW node will depend on the
external components and PCB layout, so each design should be
measured at full load. Approximate values for both tr and tf range
from 10 to 20 ns.
The power dissipated by the high-side MOSFET while it is conducting can be calculated using equation 37,
PCOND1 = I
2
RMS,FET
× R DS(ON)HS =
VOUT
I
2
× I OUT + L × R DS(ON)HS
VIN
12
(37)
( )(
2
)
PDRIVER = (QG1 + QG2) × VGS × fSW (40)
VGS is the gate drive voltage (typically 5 V),
QG1 and QG2 is the gate charges to drive high-side and
low-side MOSFETs to VGS = 5 V (about 5.8 nC and 10.4 nC
respectively), and
fSW is the PWM switching frequency.
Finally, the total power dissipated by the A8652/53 (PTOTAL) is
the sum of the previous equations,
PTOTAL = PIN + PSW1 + PCOND1 + PCOND2 + PNO + PDRIVER (41)
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A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
The average junction temperature can be calculated with the
equation below,
TJ = PTOTAL × RθJA + TA
(42)
where
PTOTAL is the total power dissipated from equation 40,
RθJA is the junction-to-ambient thermal resistance (34°C/W
on a 4-layer PCB), and
TA is the ambient temperature.
The maximum junction temperature will be dependent on how
efficiently heat can be transferred from the PCB to the ambient air. It is critical that the thermal pad on the bottom of the IC
should be connected to at least one ground plane using multiple
vias.
As with any regulator, there are limits to the amount of heat that
can be dissipated before risking thermal shutdown. There are
tradeoffs between ambient operating temperature, input voltage,
output voltage, output current, switching frequency, PCB thermal
resistance, airflow, and other nearby heat sources. Even a small
amount of airflow will reduce the junction temperature considerably.
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A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
PCB COMPONENT PLACEMENT AND ROUTING
A good PCB layout is critical for the A8652/53 to provide clean,
stable output voltages. Follow these guidelines to ensure a good
PCB layout. Figure 17 shows a typical buck converter schematic
with the critical power paths/loops. Figure 18 shows an example
PCB component placement and routing with the same critical
power paths/loops from the schematic.
1. The current sensing traces from ISEN+ and ISEN– are most
sensitive to noise. A Kelvin connection is strongly recommended for the low ohmic current sensing resistor. The loop
formed by ISEN+ and ISEN– traces should be minimal. The
loop should be away from the noisy switching areas and can
be placed on the bottom layer of PCB where it is away from
the high di/dt and dv/dt areas. This critical loop is shown as
trace 1 in Figure 17 and 18.
2. Place the ceramic input capacitors as close as possible to the
VIN pin and GND pins to make the loop area minimal, and
the traces of the input capacitors to VIN pin should be short
and wide to minimize the inductance. This critical loop is
shown as trace 2 in Figure 17 and 18. The larger input capacitor can be located further away from VIN pin. The input
capacitors and A8652/53 IC should be on the same side of the
board with traces on the same layer.
3. The loop from the input supply and capacitors, through the
high-side MOSFET, into the load via the output inductor, and
back to ground should be minimized with relatively wide
traces.
4. When the high-side MOSFET is off, free-wheeling current flows from ground, through the synchronous low-side
MOSFET, into the load via the output inductor, and back to
ground. This loop should be minimized and have relatively
wide traces, shown as trace 4 in Figure 17 and 18.
5. Place the output capacitors relatively close to the output
inductor (LO) and the A8652/53. Ideally, the output capacitors, output inductor and the controller IC A8652/53 should
be on the same layer. Connect the output inductor and the
output capacitors with a fairly wide trace. The output capacitors must use a ground plane to make a very low-inductance
connection to the GND.
6. Place the output inductor (LO) as close as possible to the SW
pin with short and wide traces. This critical trace is shown as
trace 3 in Figure 17 and 18. The SW node voltage transitions
from 0 V to VIN and with a high dv/dt rate. This node is the
root cause of many noise issues. It is suggested to minimize
the SW copper area to minimize the coupling capacitance
between SW node and other noise-sensitive nodes. However,
the SW node area cannot be too small in order to conduct
high current. A ground copper area can be placed beneath
the SW node to provide additional shielding. Also, keep low
level analog signals (like FB, COMP) away from the SW
polygon.
7. Place the feedback resistor divider (RFB1 and RFB2) very
close to the FB pin. Make the ground side of RFB2 as close as
possible to the A8652/53.
8. Place the compensation components (RZ, CZ, and CP) as
close as possible to the COMP pin. Also make the ground
side of CZ and CP as close as possible to the A8652/53.
9. Place the FSET resistor as close as possible to the SYNC/
FSET pin. Place the soft-start capacitor CSS as close as possible to the SS pin.
10.The output voltage sense trace (from VOUT to RFB1) should
be connected as close as possible to the load to obtain the
best load regulation.
11. Place the bootstrap capacitor (CBOOT) near the BOOT pin and
keep the routing from this capacitor to the SW polygon as
short as possible.
12.When connecting the input and output ceramic capacitors,
use multiple vias to GND and place the vias as close as possible to the pads of the components. Do not use thermal reliefs
around the pads for the input and output ceramic capacitors.
13.To minimize PCB losses and improve system efficiency, the
input and output traces should be as wide as possible and be
duplicated on multiple layers, if possible.
14.The thermal pad under the A8652/53 IC should be connected
to the GND plane (preferably on the top and bottom layer)
with as many vias as possible. Allegro recommends vias
with an approximately 0.25 to 0.30 mm hole and a 0.13 and
0.18 mm ring.
15.EMI/EMC issues are always a concern. Allegro recommends
having locations for an RC snubber from SW to ground. The
resistor should be 0805 or 1206 size.
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33
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
2
VIN
1
ISENRFB1
1
ISEN+
CIN
FB
LO
SW
SS
Rsen
3
VOUT
Load
VREG
SYNC/FSET
CO
COMP
CSS
RFB2
4
CP
RFSET
CZ RZ
SGND
PGND
2
Figure 17: Typical Synchronous Buck Regulator with Current Sensing Resistor for Remote Load Regulation
A single-point ground is recommended, which could be the exposed thermal pad under the IC.
Figure 18a: Example PCB Component Placement and Routing, Top Layer
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34
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Figure 18b: Example PCB Component Placement and Routing, Bottom Layer
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35
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
PACKAGE OUTLINE DRAWING
For Reference Only – Not for Tooling Use
(Reference MO-153 ABT)
Dimensions in millimeters. NOT TO SCALE
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
0.65
0.45
8º
0º
5.00 ±0.10
16
16
0.20
0.09
1.70
B
3 NOM
4.40 ±0.10
6.40 ±0.20
A
1
3.00
6.10
0.60 ±0.15
1.00 REF
2
3 NOM
1 2
0.25 BSC
Branded Face
C
16X
0.10 C
0.30
0.19
3.00
SEATING PLANE
GAUGE PLANE
C
PCB Layout Reference View
SEATING
PLANE
1.20 MAX
0.65 BSC
NNNNNNN
0.15
0.00
YYWW
LLLL
A Terminal #1 mark area
B Exposed thermal pad (bottom surface); dimensions may vary with device
C Reference land pattern layout (reference IPC7351 SOP65P640X110-17M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
D Branding scale and appearance at supplier discretion
1
D
Standard Branding Reference View
N = Device part number
= Supplier emblem
Y = Last two digits of year of manufacture
W= Week of manufacture
L = Characters 5-8 of lot number
Figure 19: Package LP, 16-Pin eTSSOP with Exposed Thermal Pad
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
36
A8652,
A8653
Wide Input Voltage, Synchronous USB Buck Regulator
with Remote Load Regulation
Revision Table
Number
Date
–
February 6, 2015
1
March 2, 2016
2
April 7, 2016
Description
Initial Release
Corrected Output Voltage Accuracy symbol (page 6), Synchronization Input Rise and Fall Time units and
Threshold symbols (page 7), SS Maximum Charge Voltage (page 8), and miscellaneous editorial changes.
Test specifications changed.
Copyright ©2016, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of
Allegro’s product can reasonably be expected to cause bodily harm.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
37
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