STMICROELECTRONICS VIPER100B

VIPer100B
VIPer100BSP

SMPS PRIMARY I.C.
PRELIMINARY DATA
T YPE
VIPer100B/BSP
V DSS
In
R DS(on)
400V
6A
1.1 Ω
10
1
PENTAWATT HV
FEATURE
■
ADJUSTABLE SWITCHING FREQUENCY UP
TO 200KHZ
■
CURRENT MODE CONTROL
■
SOFT START AND SHUT DOWN CONTROL
■
AUTOMATIC BURST MODE OPERATION IN
STAND-BY CONDITION ABLE TO MEET
”BLUE ANGEL” NORM (<1W TOTAL POWER
CONSUMPTION)
■
INTERNALLY TRIMMED ZENER
REFERENCE
■
UNDERVOLTAGE LOCK-OUT WITH
HYSTERESIS
■
INTEGRATED START-UP SUPPLY
■
AVALANCHE RUGGED
■
OVERTEMPERATURE PROTECTION
■
LOW STAND-BY CURRENT
■
ADJUSTABLE CURRENT LIMITATION
PowerSO-10
PENTAWATT HV
(022Y)
DESCRIPTION
VIPer100B/100BSP, made using VIPower M0
Technology, combines on the same silicon chip a
state-of-the-art PWM circuit together with an
optimized high voltage avalanche rugged Vertical
Power MOSFET (400 V / 6 A). Typical
applications cover off line power supplies with
a secondary power capability of 100 W in a US
mains lines configuration. It is compatible from
both primary or secondary regulation loop despite
using around 50% less components when
compared with a discrete solution. Burst mode
operation is an additional feature of this device,
offering the possibility to operate in stand-by
mode without extra components.
BLOCK DIAGRAM
DRAIN
OSC
ON/OFF
OSCILLATOR
SECURITY
LATCH
UVLO
LOGIC
V DD
R/S
FF
Q
S
PWM
LATCH
S
R1 FF Q
R2 R3
OVERTEMP.
DETECTOR
0.5 V
+
_
1.7 µ s
DELAY
25 0 ns
BLANKING
FC00231
+
4.5 V
COMP
January 2000
0. 5V/A
CURRENT
AMPLIFIER
ERROR
_ AMPLIFIER
13 V
0.5V
_
+ +
_
SOURCE
1/20
VIPER100B/BSP
ABSOLUTE MAXIMUM RATING
Symb ol
V DS
ID
VDD
V OSC
V COMP
I COMP
V esd
I D(AR)
P tot
Tj
T s tg
Parameter
o
Continuous Drain-Source Voltage (T j = 25 to 125 C)
Maximum Current
Supply Voltage
Voltage Range Input
Voltage Range Input
Maximum Continuous Current
Electrostatic discharge (R = 1.5 KΩ C = 100pF)
Avalanche Drain-Source Current, Repetitive or Not-Repetitive
(T C = 100 o C, Pulse W idth Limited by T J max, δ <1%)
o
Power Dissipation at Tc = 25 C
Junction Operating Temperature
Storage Temperature
Value
-0.3 to 400
Internally Limited
0 to 15
0 to V DD
0 to 5
±2
4000
3
Unit
V
A
V
V
V
mA
V
A
82
Internally Limited
-65 to 150
W
C
o
C
o
THERMAL DATA
R t hj-ca se
R th j-a mb.
Thermal Resistance Junction-case
PENT AW ATT-HV
PowerSO-10(*)
1.4
1.4
o
C/W
50
o
C/W
Max
Thermal Resistance Ambient-case
Max
60
(*) When mounted using the minimum recommended pad size on FR-4 board.
CONNECTION DIAGRAMS (Top View)
PENTAWATT HV
PENTAWATT HV (022Y)
PowerSO-10
CURRENT AND VOLTAGE CONVENTIONS
IDD
ID
VDD
IOSC
DRAIN
OSC
13V
+
COMP SOURCE
VDD
VDS
ICOMP
VOSC
VCOMP
FC00020
2/20
VIPER100B/BSP
ORDERING NUMBERS
PENT AW ATT HV
PENTAWATT HV (022Y)
Pow erSO -10
VIPer100B
VIPer100B (022Y)
VIPer100BSP
PINS FUNCTIONAL DESCRIPTION
DRAIN PIN:
Integrated power MOSFET drain pin. It provides
internal bias current during start-up via an
integrated high voltage current source which is
switched off during normal operation. The device
is able to handle an unclamped current during its
normal operation, assuring self protection against
voltage surges, PCB stray inductance, and
allowing a snubberless operation for low output
power.
SOURCE PIN:
Power MOSFET source pin. Primary side circuit
common ground connection.
VDD PIN :
This pin provides two functions :
- It corresponds to the low voltage supply of the
control part of the circuit. If VDD goes below 8V,
the start-up current source is activated and the
output power MOSFET is switched off until the
VDD voltage reaches 11V. During this phase,
the internal current consumption is reduced,
the VDD pin is sourcing a current of about 2mA
and the COMP pin is shorted to ground. After
that, the current source is shut down, and the
device tries to start up by switching again.
- This pin is also connected to the error amplifier,
in order to allow primary as well as secondary
regulation configurations. In case of primary
regulation, an internal 13V trimmed reference
voltage is used to maintain VDD at 13V. For
secondary regulation, a voltage between 8.5V
and 12.5V will be put on VDD pin by
transformer design, in order to stuck the output
of the transconductance amplifier to the high
state. The COMP pin behaves as a constant
current source, and can easily be connected to
the output of an optocoupler. Note that any
overvoltage due to regulation loop failure is still
detected by the error amplifier through the VDD
voltage, which cannot overpass 13V. The
output voltage will be somewhat higher than
the nominal one, but still under control.
COMP PIN :
This pin provides two functions :
- It is the output of the error transconductance
amplifier, and allows for the connection of a
compensation network to provide the desired
transfer function of the regulation loop. Its
bandwidth can be easily adjusted to the
needed value with usual components value. As
stated
above,
secondary
regulation
configurations are also implemented through
the COMP pin.
- When the COMP voltage is going below 0.5V,
the shut-down of the circuit occurs, with a zero
duty cycle for the power MOSFET. This feature
can be used to switch off the converter, and is
automatically activated by the regulation loop
(whatever is the configuration) to provide a
burst mode operation in case of negligible
output power or open load condition.
OSC PIN :
An RT-CT network must be connected on that pin
to define the switching frequency. Note that
despite the connection of RT to VDD, no
significant frequency change occurs for VDD
varying from 8V to 15V. It provides also a
synchronisation capability, when connected to an
external frequency source.
3/20
VIPER100B/BSP
AVALANCHE CHARACTERISTICS
Symb ol
Parameter
Max Valu e
Unit
I D(a r)
Avalanche Current, Repetitive or Not-Repetitive
(pulse width limited by Tj max, δ < 1%)
3
A
E (ar)
Single Pulse Avalanche Energy
(starting Tj = 25 o C, I D = ID(ar ))
60
mJ
(see fig.12)
ELECTRICAL CHARACTERISTICS (TJ = 25 oC, VDD = 13 V, unless otherwise specified)
POWER SECTION
Symb ol
BV DSS
I DSS
R DS( on)
Parameter
Test Con ditio ns
Drain-Source Voltage
I D = 1 mA
Off-State Drain Current
V COMP = 0 V
V DS = 400 V
Min.
VCOMP = 0 V
Typ .
Max.
400
Un it
V
o
T J = 125 C
0.9
1
mA
1.1
2
Ω
Ω
Static Drain Source on
Resistance
ID = 4 A
ID = 4 A
tf
Fall Time
I D = 0.2 A V i n = 300 V (1)
(see fig.3)
100
ns
tr
Rise Time
ID = 4 A
(see fig. 3)
50
ns
Output Capacitance
V DS = 25 V
180
pF
C OSS
o
T J = 100 C
Vi n = 300 V (1)
(1) On Inductive Load, Clamped.
SUPPLY SECTION
Symb ol
Parameter
Test Con ditio ns
Min.
Typ .
Max.
Un it
I DDch
Start-up Charging
Current
I DD0
Operating Supply Current V DD = 12 V,
(see fig. 2)
F SW = 0 KHz
I DD1
Operating Supply Current V DD = 12 V,
F SW = 100 KHz
15.5
mA
I DD2
Operating Supply Current V DD = 12 V,
F SW = 200 KHz
19
mA
V
V DD = 5 V
V DS = 70 V
(see fig. 2 and fig. 15)
-2
12
V DDo ff
Undervoltage Shutdown
(see fig. 2)
8
V DDo n
Undervoltage Reset
(see fig. 2)
11
VDDhyst
Hysteresis Start-up
(see fig. 2)
4/20
2.4
3
mA
16
12
mA
V
V
VIPER100B/BSP
ELECTRICAL CHARACTERISTICS (continued)
OSCILLATOR SECTION
Symb ol
F SW
Parameter
Oscillator Frequency
Total Variation
Test Con ditio ns
C T =2.4 nF
R T = 8.2 KΩ
V DD = 9 to15 V
C T ± 5%
with R T ± 1%
(see fig. 6 and fig. 9)
Min.
Typ .
Max.
Un it
90
100
110
KHz
V OSCih
Oscillator Peak Voltage
7.1
V
V OSCi l
Oscillator Valley Voltage
3.7
V
ERROR AMPLIFIER SECTION
Symb ol
V DDr e
∆V DDreg
Parameter
Vg Regulation Point
Test Cond ition s
I COMP = 0 mA
(see fig.1)
Min.
Typ .
Max.
Un it
12.6
13
13.4
V
o
Total Variation
T J = 0 to 100 C
GBW
Unity G ain Bandwidth
From Input = VDD to Output = V COMP
CO MP pin is open (see fig. 10)
A VOL
Open Loop Voltage
Gain
CO MP pin is open (see fig. 10)
DC T ransconductance
V COMP = 2.5 V
V COMPL O
Output Low Level
I COMP = -400 µA
V DD = 14 V
V COMPHI
Output High Level
I COMP = 400 µA
V DD = 12 V
4.5
V
I COMPLO
Output Low Current
Capability
V COMP = 2.5 V
V DD = 14 V
-600
µA
I COMPHI
Output High Current
Capability
V COMP = 2.5 V
V DD = 12 V
600
µA
Gm
(see fig. 1)
2
%
150
KHz
45
52
dB
1.1
1.5
1.9
0.2
mA/V
V
PWM COMPARATOR SECTION
Symb ol
H ID
V COMPof f
I Dpeak
Parameter
Test Cond ition s
∆V COMP /∆IDpea k
V COMP = 1 to 3 V
V COMP offset
I Dp eak = 10 mA
Peak Current Limitation V DD = 12 V
COMP pin open
Min.
Typ .
Max.
Un it
0.35
0.5
0.65
V/A
11
A
0.5
6
8
td
Current Sense Delay
to turn-off
tb
Blanking Time
250
Minimum on T ime
350
t on( mi n)
ID = 1 A
V
250
ns
360
ns
ns
SHUTDOWN AND OVERTEMPERATURE SECTION
Symb ol
Parameter
V COMPth
Restart threshold
(see fig. 4)
0.5
Disable Set Up T ime
(see fig. 4)
1.7
T t sd
Thermal Shutdown
Temperature
(see fig. 8)
T hyst
Thermal Shutdown
Hysteresis
(see fig. 8)
t DI Ssu
Test Cond ition s
Min.
140
Typ .
Max.
Un it
5
µs
V
170
o
C
40
o
C
5/20
VIPER100B/BSP
Figure 1: VDD Regulation Point
ICOMP
Figure 2: Undervoltage Lockout
IDD
Slope =
Gm in mA/V
ICOMPHI
IDD0
VDD
VDS = 70 V
Fsw = 0
VDDhyst
0
VDDoff
ICOMPLO
VDD
VDDon
IDDch
VDDreg
FC00150
Figure 3: Transition Time
FC00170
Figure 4: Shut Down Action
VOS C
ID
t
VCOMP
10% Ipeak
VDS
tDIS s u
t
VCOMPth
t
90% VD
ID
10% VD
t
tf
t
tr
ENABLE
FC00160
ENABLE
DISABLE
FC 00 0 6 0
Figure 5: Breakdown Voltage vs Temperature
Figure 6: Typical Frequency Variation
FC0 0180
FC0 019 0
1.15
(%)
BVDS S
(Nor malize d)
1
0
1.1
-1
-2
1.05
-3
1
-4
0.95
6/20
-5
0
20
40 60 80 100 120
Temp erature ( C)
0
20
40 60 80 100 120 140
Temperature ( C)
VIPER100B/BSP
Figure 7: Start-up Waveforms
Figure 8: Overtemperature Protection
Tj
Ttsd
Tts d-Thys t
t
Vdd
Vddo n
Vddoff
t
Id
t
Vco mp
t
SC1 019 1
7/20
VIPER100B/BSP
Figure 9: Oscillator
VDD
Rt
For RT > 1.2 KΩ:
2.3
FSW =
D
RT CT MAX
OSC
CLK
~360Ω
Ct
DMAX = 1 −
550
RT − 150
Recommended DMAX values:
100KHz: > 80%
200KHz: > 70%
FC00050
Maximum duty cycle vs Rt
FC00040
1
0.9
Dmax
0.8
0.7
0.6
0.5
1
2
3
5
10
20
30
50
Rt (kΩ)
Oscillator frequency vs Rt and Ct
FC00030
1,000
Ct = 1.5 nF
500
Frequency (kHz)
Ct = 2.7 nF
300
Ct = 4.7 nF
200
Ct = 10 nF
100
50
30
1
2
3
5
Rt (kΩ)
8/20
10
20
30
50
VIPER100B/BSP
Figure 10: Error Amplifier Frequency Response
FC00200
60
RCOMP = +∞
Voltage Gain (dB)
RCOMP = 270k
40
RCOMP = 82k
RCOMP = 27k
20
RCOMP = 12k
0
(20)
0.001
0.01
0.1
1
10
Frequency (kHz)
100
1,000
Figure 11: Error Amplifier Phase Response
FC00210
200
RCOMP = +∞
150
RCOMP = 270k
Phase (°)
RCOMP = 82k
RCOMP = 27k
100
RCOMP = 12k
50
0
(50)
0.001
0.01
0.1
1
10
Frequency (kHz)
100
1,000
9/20
VIPER100B/BSP
Figure 12: Avalanche Test Circuit
L1
1mH
2
VDD
1
3
DRAIN
OSC
13V
BT1
0 to 20V
+
COMP
BT2
12V
C1
47uF
16V
Q1
2 x STHV102FI in parallel
R1
SOURCE
5
4
47
GENERATOR INPUT
500us PULSE
U1
VIPer100B
R2
1k
R3
100
FC00195B
10/20
VIPER100B/BSP
Figure 13: Off Line Power Supply With Auxiliary Supply Feedback
F1
C1
BR1
TR2
TR1
D2
AC IN
L2
+Vcc
D1
R9
C2
C7
C9
R1
C3
GND
D3
C10
R7
C4
R2
VDD
DRAIN
OSC
13V
VIPer100B
+
COMP SOURCE
C5
C6
C11
R3
FC00081B
Figure 14: Off Line Power Supply With Optocoupler Feedback
F1
BR1
TR2
C1
TR1
D2
AC IN
L2
+Vcc
D1
R9
C2
C7
C9
R1
C3
GND
D3
C10
R7
C4
R2
VDD
DRAIN
OSC
13V
VIPer100B
+
COMP SOURCE
C5
C11
C6
R3
R6
ISO1
R4
C8
U2
R5
FC00091B
11/20
VIPER100B/BSP
OPERATION DESCRIPTION :
CURRENT MODE TOPOLOGY:
The current mode control method, like the one
integrated in the VIPer100B/BSPuses two control
loops - an inner current control loop and an outer
loop for voltage control. When the Power
MOSFET output transistor is on, the inductor
current (primary side of the transformer) is
monitored with a SenseFET technique and
converted into a voltage VS proportional to this
current. When VS reaches VCOMP (the amplified
output voltage error) the power switch is switched
off. Thus, the outer voltage control loop defines
the level at which the inner loop regulates peak
current through the power switch and the primary
winding of the transformer.
Excellent open loop D.C. and dynamic line
regulation is ensured due to the inherent input
voltage feedforward characteristic of the current
mode control. This results in an improved line
regulation, instantaneous correction to line
changes and better stability for the voltage
regulation loop.
Current mode topology also ensures good
limitation in the case of short circuit. During a first
phase the output current increases slowly
following the dynamic of the regulation loop. Then
it reaches the maximum limitation current
internally set and finally stops because the power
supply on VDD is no longer correct. For specific
applications the maximum peak current internally
set can be overridden by externally limiting the
voltage excursion on the COMP pin. An
integrated blanking filter inhibits the PWM
comparator output for a short time after the
integrated Power MOSFET is switched on. This
function prevents anomalous or premature
termination of the switching pulse in the case of
current spikes caused by primary side
capacitance or secondary side rectifier reverse
recovery time.
STAND-BY MODE
Stand-by operation in nearly open load condition
automatically leads to a burst mode operation
allowing voltage regulation on the secondary
side. The transition from normal operation to
burst mode operation happens for a power PSTBY
given by :
1
2
PSTBY = LP ISTBY FSW
2
12/20
Where:
LP is the primary inductance of the transformer.
FSW is the normal switching frequency.
ISTBY is the minimum controllable current,
corresponding to the minimum on time that the
device is able to provide in normal operation. This
current can be computed as :
(tb + td) VIN
ISTBY =
LP
tb + td is the sum of the blanking time and of the
propagation time of the internal current sense
and comparator, and represents roughly the
minimum on time of the device. Note that PSTBY
may be affected by the efficiency of the converter
at low load, and must include the power drawn on
the primary auxiliary voltage.
As soon as the power goes below this limit, the
auxiliary secondary voltage starts to increase
above the 13V regulation level forcing the output
voltage of the transconductance amplifier to low
state (VCOMP < VCOMPth). This situation leads to
the shutdown mode where the power switch is
maintained in the off state, resulting in missing
cycles and zero duty cycle. As soon as VDD gets
back to the regulation level and the VCOMPth
threshold is reached, the device operates again.
The above cycle repeats indefinitely, providing a
burst mode of which the effective duty cycle is
much lower than the minimum one when in
normal operation. The equivalent switching
frequency is also lower than the normal one,
leading to a reduced consumption on the input
mains lines. This mode of operation allows the
VIPer100B/BSP to meet the new German ”Blue
Angel” Norm with less than 1W total power
consumption for the system when working in
stand-by. The output voltage remains regulated
around the normal level, with a low frequency
ripple corresponding to the burst mode. The
amplitude of this ripple is low, because of the
output capacitors and of the low output current
drawn in such conditions.The normal operation
resumes automatically when the power get back
to higher levels than PSTBY.
HIGH
VOLTAGE
START-UP
CURRENT
SOURCE
An integrated high voltage current source
provides a bias current from the DRAIN pin
during the start-up phase. This current is partially
absorbed by internal control circuits which are
VIPER100B/BSP
in short circuit.
The external capacitor CVDD on the VDD pin must
be sized according to the time needed by the
converter to start up, when the device starts
switching. This time tSS depends on many
parameters, among which transformer design,
output capacitors, soft start feature and
compensation network implemented on the
COMP pin. The following formula can be used for
defining the minimum capacitor needed:
IDD tSS
CVDD >
placed into a standby mode with reduced
consumption and also provided to the external
capacitor connected to the VDD pin. As soon as
the voltage on this pin reaches the high voltage
threshold VDDon of the UVLO logic, the device
turns into active mode and starts switching. The
start up current generator is switched off, and the
converter should normally provide the needed
current on the VDD pin through the auxiliary
winding of the transformer, as shown on figure
15.
In case of abnormal condition where the auxiliary
winding is unable to provide the low voltage
supply current to the VDD pin (i.e. short circuit on
the output of the converter), the external
capacitor discharges itself down to the low
threshold voltage VDDoff of the UVLO logic, and
the device get back to the inactive state where
the internal circuits are in standby mode and the
start up current source is activated. The converter
enters a endless start up cycle, with a start-up
duty cycle defined by the ratio of charging current
towards discharging when the VIPer100B/BSP
tries to start. This ratio is fixed by design to 2 to
15, which gives a 12% start up duty cycle while
the power dissipation at start up is approximately
0.6 W, for a 230 Vrms input voltage. This low
value of start-up duty cycle prevents the stress of
the output rectifiers and of the transformer when
VDDhyst
where:
IDD is the consumption current on the VDD pin
when switching. Refer to specified IDD1 and IDD2
values.
tSS is the start up time of the converter when the
device begins to switch. Worst case is generally
at full load.
VDDhyst is the voltage hysteresis of the UVLO
logic. Refer to the minimum specified value.
Soft start feature can be implemented on the
COMP pin through a simple capacitor which will
be also used as the compensation network. In
this case, the regulation loop bandwidth is rather
low, because of the large value of this capacitor.
In case a large regulation loop bandwidth is
mandatory, the schematics of figure 16 can be
Figure 15: Behaviour of the high voltage current source at start-up
VDD
2 mA
VDDon
VDDoff
15 mA
3 mA
VDD
1 mA
DRAIN
15 mA
C VDD
Ref.
t
Auxiliary primary
winding
UNDERVOLTAGE
LOCK OUT LOGIC
VIP er100B
Sta rt up duty cycle ~ 10%
SOURCE
FC00100B
13/20
VIPER100B/BSP
used. It mixes a high performance compensation
network together with a separate high value soft
start capacitor. Both soft start time and regulation
loop bandwidth can be adjusted separately.
If the device is intentionally shut down by putting
the COMP pin to ground, the device is also
performing start-up cycles, and the VDD voltage is
oscillating between VDDon and VDDoff. This voltage
can be used for supplying external functions,
provided that their consumption doesn’t exceed
0.5mA. Figure 17 shows a typical application of
this function, with a latched shut down. Once the
”Shutdown” signal has been activated, the device
remains in the off state until the input voltage is
removed.
TRANSCONDUCTANCE ERROR AMPLIFIER
The VIPer100B/BSP includes a transconductance
error amplifier. Transconductance Gm is the
change in output current (ICOMP) versus change
in input voltage (VDD). Thus:
∂ ICOMP
Gm =
∂ VDD
The output impedance ZCOMP at the output of this
amplifier (COMP pin) can be defined as:
∂ VCOMP
∂ VCOMP
1
ZCOMP =
=
x
∂ ICOMP Gm
∂ VDD
This last equation shows that the open loop gain
AVOL can be related to Gm and Z COMP:
AVOL = Gm x ZCOMP
where Gm value for VIPer100B/BSP is 1.5 mA/V
typically.
Figure 16: Mixed Soft Start and Compensation
Gm is well defined by specification, but ZCOMP
and therefore AVOL are subject to large
tolerances. An impedance Z can be connected
between the COMP pin and ground in order to
define more accurately the transfer function F of
the error amplifier, according to the following
equation, very similar to the one above:
F(S) = Gm x Z(S)
The error amplifier frequency response is
reported in figure 10 for different values of a
simple resistance connected on the COMP pin.
The unloaded transconductance error amplifier
shows an internal ZCOMP of about 330 KΩ. More
complex impedance can be connected on the
COMP pin to achieve different compensation
laws. A capacitor will provide an integrator
function, thus eliminating the DC static error, and
a resistance in series leads to a flat gain at higher
frequency, insuring a correct phase margin. This
configuration is illustrated on figure 18.
As shown in figure 18 an additional noise filtering
capacitor of 2.2 nF is generally needed to avoid
any high frequency interference.
It can be also interesting to implement a slope
compensation when working in continuous mode
with duty cycle higher than 50%. Figure 19 shows
such a configuration. Note that R1 and C2 build
the classical compensation network, and Q1 is
injecting the slope compensation with the correct
polarity from the oscillator sawtooth.
EXTERNAL CLOCK SYNCHRONIZATION:
The OSC pin provides a synchronisation
capability, when connected to an external
Figure 17: Latched Shut Down
D2
D3
VIPer100B
VDD
R1
VDD
OSC
13V
Q2
R3
+
13V
D1
+
COMP
AUXILIARY
WINDING
R3
SOURCE
R2
R1
C4
R2
C1
R4
Shutdown
+ C2
FC0 0131B
14/20
DRAIN
-
OSC
COMP SOURCE
+ C3
VIPer100B
DRAIN
Q1
D1
FC00110B
VIPER100B/BSP
frequency source. Figure 20 shows one possible
schematic to be adapted depending the specific
needs. If the proposed schematic is used, the
pulse duration must be kept at a low value (500ns
is sufficient) for minimizing consumption. The
optocoupler must be able to provide 20mA
through the optotransistor.
PRIMARY PEAK CURRENT LIMITATION
The primary IDPEAK current and, as resulting
effect, the output power can be limited using the
simple circuit shown in figure 21. The circuit
based on Q1, R1 and R2 clamps the voltage on
the COMP pin in order to limit the primary peak
current of the device to a value:
Figure 18: Typical Compensation Network
IDPEAK =
VCOMP − 0.5
HID
where:
VCOMP = 0.6 x
R1 + R2
R2
The suggested value for R1+R2 is in the range of
220KΩ.
OVER-TEMPERATURE PROTECTION:
Over-temperature protection is based on chip
temperature sensing. The minimum junction
temperature at which over-temperature cut-out
occurs is 140oC while the typical value is 170oC.
The device is automatically restarted when the
junction temperature decreases to the restart
temperature threshold that is typically 40oC below
Figure 19: Slope Compensation
VIPer100B
VDD
DRAIN
R2
-
OSC
13V
R1
VIPer100B
+
VDD
COMP SOURCE
DRAIN
-
OSC
13V
+
COMP SOURCE
C2
R1
C2
C1
Q1
C1
C3
R3
F C00121B
FC0 0141B
Figure 20:External Clock Synchronization
Figure 21:Current Limitation Circuit Example
VIPer10 0B
VDD
DRAIN
-
OSC
13V
+
VIPer100B
VDD
COMP SOURCE
DRAIN
-
OSC
13V
+
COMP SOURCE
R1
10 kΩ
Q1
R2
FC00220B
FC0 0240B
15/20
VIPER100B/BSP
Figure 22: Recommended layout
T1
D1
C7
D2
R1
2
3
VDD
C1
To secondary
filteringandload
DRAIN
-
1 OSC
13V
Frominput
diod
es bridge
C5
+
COMP SOURCE
5
U1
VIPer100B
R2
4
C6
C2
C3
ISO1
C4
FC00500B
LAYOUT CONSIDERATIONS
Some simple rules insure a correct running of
switching power supplies. They may be classified
into two categories:
- To minimise power loops: the way the switched
power current must be carefully analysed and
the corresponding paths must present the
smallest inner loop area as possible. This
avoids radiated EMC noises, conducted EMC
noises by magnetic coupling, and provides a
better efficiency by eliminating parasitic
inductances, especially on secondary side.
- To use different tracks for low level signals and
16/20
power ones. The interferences due to a mixing
of signal and power may result in instabilities
and/or anomalous behaviour of the device in
case of violent power surge (Input
overvoltages, output short circuits...).
In case of VIPer, these rules apply as shown on
figure 22. The loops C1-T1-U1, C5-D2-T1,
C7-D1-T1 must be minimised. C6 must be as
close as possible from T1. The signal
components C2, ISO1, C3 and C4 are using a
dedicated track to be connected directly to the
source of the device.
VIPER100B/BSP
PENTAWATT HV (VERTICAL) MECHANICAL DATA
DIM.
mm
TYP.
MIN.
4.30
1.17
2.40
0.35
0.60
4.90
7.42
9.30
A
C
D
E
F
G1
G2
H1
H2
H3
L
L1
L2
L3
L5
L6
L7
M
M1
R
V4
Diam.
MAX.
4.80
1.37
2.80
0.55
0.80
5.28
7.82
9.70
10.40
10.40
17.30
15.22
21.85
22.82
3.00
15.80
6.60
3.10
8.16
10.05
16.60
14.60
21.20
22.20
2.60
15.10
6.00
2.50
7.56
inch
TYP.
MIN.
0.169
0.046
0.094
0.014
0.024
0.193
0.292
0.366
0.396
0.653
0.575
0.835
0.874
0.102
0.594
0.236
0.098
0.298
0.50
90o
MAX.
0.189
0.054
0.110
0.022
0.031
0.208
0.308
0.382
0.409
0.409
0.681
0.599
0.860
0.898
0.118
0.622
0.260
0.122
0.321
0.020
90
3.70
3.90
0.146
0.154
L
E
L1
M1
A
M
R
C
D
Resin
between
leads
L6
L7
V4
H2
H3
H1
G1
G2
F
L2
L5
Diam
L3
P023H3
17/20
VIPER100B/BSP
PENTAWATT HV 022Y(VERTICAL HIGH PITCH) MECHANICAL DATA
DIM.
mm
TYP.
MIN.
4.30
1.17
2.40
0.35
0.60
4.90
7.42
9.30
A
C
D
E
F
G1
G2
H1
H2
H3
L
L1
L3
L5
L6
L7
M
M1
R
V4
Diam.
MAX.
4.80
1.37
2.80
0.55
0.80
5.28
7.82
9.70
10.40
10.40
17.42
15.22
21.52
3.00
15.80
6.60
3.10
5.70
10.05
16.42
14.60
20.52
2.60
15.10
6.00
2.50
5.00
inch
TYP.
MIN.
0.169
0.046
0.094
0.014
0.024
0.193
0.292
0.366
0.396
0.646
0.575
0.808
0.102
0.594
0.236
0.098
0.197
0.50
o
90
MAX.
0.189
0.054
0.110
0.022
0.031
0.208
0.308
0.382
0.409
0.409
0.686
0.599
0.847
0.118
0.622
0.260
0.122
0.224
0.020
90o
3.70
3.90
0.146
0.154
L
L1
E
M1
A
M
R
D
C
Resin
between
leads
L6
L7
V4
H2
H3
H1
G1
G2
F
L5
Diam
L3
P023H2
18/20
VIPER100B/BSP
PowerSO-10 MECHANICAL DATA
mm
DIM.
MIN.
inch
TYP.
MAX.
MIN.
TYP.
MAX.
A
3.35
3.65
0.132
0.144
A1
0.00
0.10
0.000
0.004
B
0.40
0.60
0.016
0.024
C
0.35
0.55
0.013
0.022
D
9.40
9.60
0.370
0.378
D1
7.40
7.60
0.291
e
1.27
0.300
0.050
E
9.30
9.50
0.366
0.374
E1
7.20
7.40
0.283
0.291
E2
7.20
7.60
0.283
0.300
E3
6.10
6.35
0.240
0.250
E4
5.90
6.10
0.232
0.240
F
1.25
1.35
0.049
0.053
14.40
0.543
1.80
0.047
h
0.50
H
13.80
L
1.20
q
0.002
1.70
α
0.567
0.071
0.067
0o
8o
B
0.10 A B
10
=
E4
=
=
=
E1
=
E3
=
E2
=
E
=
=
=
H
6
=
=
1
5
e
0.25
B
SEATING
PLANE
DETAIL ”A”
A
C
M
Q
h
D
= D1 =
=
=
SEATING
PLANE
A
F
A1
A1
L
DETAIL ”A”
α
0068039-C
19/20
VIPER100B/BSP
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20/20