19-2091; Rev 0; 8/01 High-Efficiency, Current-Mode, Inverting PWM Controller Features ♦ 90% Efficiency The MAX1846 is available in an ultra-compact 10-pin µMAX package. Operation at high frequency, compatibility with ceramic capacitors, and inverting topology without transformers allow for a compact design. Compatibility with electrolytic capacitors and flexibility to operate down to 100kHz allow users to minimize the cost of external components. The high-current output drivers are designed to drive a P-channel MOSFET and allow the converter to deliver up to 30W. The MAX1847 features clock synchronization and shutdown functions. The MAX1847 can also be configured to operate as an inverting flyback controller with an Nchannel MOSFET and a transformer to deliver up to 70W. The MAX1847 is available in a 16-pin QSOP package. Current-mode control simplifies compensation and provides good transient response. Accurate current-mode control and over current protection are achieved through low-side current sensing. ♦ Internal Soft-Start ♦ +3.0V to +16.5V Input Range ♦ -2V to -200V Output ♦ Drives High-Side P-Channel MOSFET ♦ 100kHz to 500kHz Switching Frequency ♦ Current-Mode, PWM Control ♦ Electrolytic or Ceramic Output Capacitor ♦ The MAX1847 also offers: Synchronization to External Clock Shutdown N-Channel Inverting Flyback Option Ordering Information PART TEMP. RANGE PIN-PACKAGE MAX1846EUB -40°C to +85°C 10 µMAX MAX1847EEE -40°C to +85°C 16 QSOP Typical Operating Circuit POSITIVE VIN Applications Cellular Base Stations P Networking Equipment Optical Networking Equipment VL NEGATIVE VOUT IN EXT SLIC Supplies MAX1846 MAX1847 CO DSL Line Driver Supplies Industrial Power Supplies COMP CS Automotive Electronic Power Supplies Servers FREQ PGND REF GND FB Pin Configurations appear at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX1846/MAX1847 General Description MAX1846/MAX1847 high-efficiency PWM inverting controllers allow designers to implement compact, lownoise, negative-output DC-DC converters for telecom and networking applications. Both devices operate from +3V to +16.5V input and generate -2V to -200V output. To minimize switching noise, both devices feature a current-mode, constant-frequency PWM control scheme. The operating frequency can be set from 100kHz to 500kHz through a resistor. MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller ABSOLUTE MAXIMUM RATINGS IN, SHDN to GND ...................................................-0.3V to +20V PGND to GND .......................................................-0.3V to +0.3V VL to PGND for VIN ≤ 5.7V...........................-0.3V to (VIN + 0.3V) VL to PGND for VIN > 5.7V .......................................-0.3V to +6V EXT to PGND ...............................................-0.3V to (VIN + 0.3V) REF, COMP to GND......................................-0.3V to (VL + 0.3V) CS, FB, FREQ, POL, SYNC to GND .........................-0.3V to +6V Continuous Power Dissipation (TA = +70°C) 10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW 16-Pin QSOP (derate 8.3mW/°C above +70°C)...........696mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 16.5 V PWM CONTROLLER Operating Input Voltage Range UVLO Threshold 3.0 VIN rising 2.8 2.95 2.6 FB Threshold No load -12 0 12 mV FB Input Current VFB = -0.1V -50 -6 50 nA Load Regulation CCOMP = 0.068µF, VOUT = -48V, IOUT = 20mA to 200mA (Note 1) -1 0 % Line Regulation CCOMP = 0.068µF, VOUT = -48V, VIN = +8V to +16.5V, IOUT = 100mA UVLO Hysteresis 60 Current-Limit Threshold CS = GND Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V SHDN = GND, VIN = +3.0V to +16.5V mV 0.04 85 CS Input Current Shutdown Supply Current 2.74 V VIN falling 100 % 115 mV 10 20 µA 0.75 1.2 mA 10 25 µA 1.25 1.264 V -2 -15 mV 4.25 4.65 V -20 -60 mV REFERENCE AND VL REGULATOR REF Output Voltage IREF = 50µA REF Load Regulation IREF = 0 to 500µA VL Output Voltage IVL = 100µA VL Load Regulation IVL = 0.1mA to 2.0mA 2 1.236 3.85 _______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller (V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted.) OSCILLATOR Oscillator Frequency RFREQ = 500kΩ ±1% 90 100 110 RFREQ = 147kΩ ±1% 255 300 345 RFREQ = 76.8kΩ ±1% Maximum Duty Cycle 500 RFREQ = 500kΩ ±1% 93 96 97 RFREQ = 147kΩ ±1% 85 88 90 % 93 % 200 ns 200 ns 550 kHz RFREQ = 76.8kz ±1% SYNC Input Signal Duty-Cycle Range 80 7 Minimum SYNC Input Logic Low Pulse Width SYNC Input Rise/Fall Time kHz 50 (Note 2) SYNC Input Frequency Range 100 DIGITAL INPUTS POL, SYNC, SHDN Input High Voltage 2.0 V POL, SYNC, SHDN Input Low Voltage POL, SYNC Input Current POL, SYNC = GND or VL VSHDN = +5V or GND SHDN Input Current VSHDN = +16.5V -12 0.45 V 20 40 µA -4 0 1.5 6 µA SOFT-START Soft-Start Clock Cycles Cycles 1024 Soft-Start Levels 64 EXT OUTPUT EXT Sink/Source Current EXT On-Resistance VIN = +5V, VEXT forced to +2.5V 1 EXT high or low, tested with 100mA load, VIN = +5V 2 5 A EXT high or low, tested with 100mA load, VIN = +3V 5 10 Ω Note 1: Production test correlates to operating conditions. Note 2: Guaranteed by design and characterization. _______________________________________________________________________________________ 3 MAX1846/MAX1847 ELECTRICAL CHARACTERISTICS (continued) MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller ELECTRICAL CHARACTERISTICS (V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN MAX UNITS 3.0 16.5 V PWM CONTROLLER Operating Input Voltage Range UVLO Threshold VIN rising 2.95 V VIN falling 2.6 FB Threshold No load -20 20 mV FB Input Current VFB = -0.1V -50 50 nA Load Regulation CCOMP = 0.068µF, VOUT = -48V, IOUT= 20mA to 200mA (Note 1) -2 0 % 85 115 mV Current Limit Threshold CS Input Current CS = GND 20 µA Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V 1.2 mA Shutdown Supply Current SHDN = GND, VIN = +3.0V to +16.5V 25 µA REFERENCE AND VL REGULATOR REF Output Voltage IREF = 50µA REF Load Regulation IREF = 0 to 500µA VL Output Voltage IVL = 100µA VL Load Regulation IVL = 0.1mA to 2.0mA 1.225 3.85 1.275 V -15 mV 4.65 V -60 mV OSCILLATOR Oscillator Frequency Maximum Duty Cycle RFREQ = 500kΩ ±1% 84 116 RFREQ = 147kΩ ±1% 255 345 RFREQ = 500kΩ ±1% 93 98 RFREQ = 147kΩ ±1% 84 93 7 93 % 200 ns 200 ns 550 kHz SYNC Input Signal Duty-Cycle Range Minimum SYNC Input Logic Low Pulse Width SYNC Input Rise/Fall Time SYNC Input Frequency Range (Note 2) 100 kHz % DIGITAL INPUTS POL, SYNC, SHDN Input High Voltage 2.0 POL, SYNC, SHDN Input Low Voltage 4 _______________________________________________________________________________________ V 0.45 V High-Efficiency, Current-Mode, Inverting PWM Controller (V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS POL, SYNC Input Current MIN MAX UNITS 40 µA POL, SYNC = GND or VL V SHDN = +5V or GND SHDN Input Current -12 0 V SHDN = +16.5V µA 6 EXT OUTPUT EXT On-Resistance EXT high or low, IEXT = 100mA, VIN = +5V 7.5 EXT high or low, IEXT = 100mA, VIN = +3V 12 Ω Note 3: Parameters to -40°C are guaranteed by design and characterization. Typical Operating Characteristics (Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise noted.) VIN = 16.5V 50 40 70 VIN = 3.3V 60 VIN = 3V 50 40 50 40 30 30 20 20 10 10 APPLICATION CIRCUIT A 1 10 100 VOUT = -5V 1000 VOUT = -12V APPLICATION CIRCUIT B 0 10,000 VIN = 16.5V 60 20 0 VIN = 12V 70 30 10 1 10 LOAD CURRENT (mA) 100 1000 1 10,000 10 -11.94 1.6 1.4 1.2 -11.96 -12.02 1.258 VREF (V) IIN (mA) -12.00 1.262 1.254 1.0 -11.98 1000 REFERENCE VOLTAGE vs. TEMPERATURE MAX1846/7 toc05 -11.92 100 LOAD CURRENT (mA) SUPPLY CURRENT vs. SUPPLY VOLTAGE MAX1846/7 toc04 -11.90 VOUT = -48V APPLICATION CIRCUIT C 0 LOAD CURRENT (mA) OUTPUT VOLTAGE LOAD REGULATION OUTPUT VOLTAGE (V) 80 MAX1846/7 toc06 60 80 90 EFFICIENCY (%) 70 VIN = 5V 90 EFFICIENCY (%) EFFICIENCY (%) VIN = 5V 100 MAX1846/7 toc03 MAX1846/7 toc01 90 80 EFFICIENCY vs. LOAD CURRENT EFFICIENCY vs. LOAD CURRENT 100 MAX1846/7 toc02 EFFICIENCY vs. LOAD CURRENT 100 0.8 0.6 1.250 1.246 -12.04 0.4 -12.06 1.242 0.2 -12.08 APPLICATION CIRCUIT B -12.10 0 100 200 300 VFB = -0.1V VIN = 5V 400 LOAD CURRENT (mA) 1.238 0 500 600 0 2 4 6 8 VIN (V) 10 12 14 16 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) _______________________________________________________________________________________ 5 MAX1846/MAX1847 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (continued) (Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise noted.) VL VOLTAGE vs. TEMPERATURE VL LOAD REGULATION 4.340 4.300 1.255 4.27 4.26 VL (V) VL (V) VREF (V) 4.260 1.250 4.220 MAX1846/7 toc09 MAX1846/7 toc07 1.260 MAX1846/7 toc08 REFERENCE LOAD REGULATION 4.25 4.24 4.180 1.245 4.23 4.140 IVL = 0 4.100 1.240 100 200 300 400 500 0 20 40 60 80 0 100 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 TEMPERATURE (°C) IVL (mA) SHUTDOWN SUPPLY CURRENT vs. TEMPERATURE OPERATING CURRENT vs. TEMPERATURE SWITCHING FREQUENCY vs. RFREQ 10 8 VIN = 3V 6 4 500 12 MAX1846/7 toc12 A A: VIN = 3V, VOUT = -12V 400 10 fOSC (kHz) VIN = 10V VIN = 16.5V 14 MAX1846/7 toc11 14 MAX1846/7 toc10 8 APPLICATION CIRCUIT A B: VIN = 5V, VOUT = -5V C: VIN = 16.5V, VOUT = -5V 6 B 300 200 4 100 2 2 0 C 0 -40 -20 0 20 60 40 TEMPERATURE (°C) 80 100 0 -40 -20 0 20 40 60 80 100 0 TEMPERATURE (°C) SWITCHING FREQUENCY vs. TEMPERATURE EXT RISE/FALL TIME vs. CAPACITANCE 301 100 TIME (ns) 120 299 298 300 400 FALL TIME 60 296 40 295 20 600 5V/div SHDN 0 VOUT 80 297 500 MAX1846/7 toc15 140 300 200 EXITING SHUTDOWN 160 MAX1846/7 toc13 302 100 RFREQ (kΩ) MAX1846/7 toc14 SHUTDOWN SUPPLY CURRENT (µA) -20 IREF (µA) 16 12 4.22 -40 OPERATING CURRENT (mA) 0 FREQUENCY (kHz) MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller 5V/div RISE TIME RFREQ = 147kΩ ±1% 294 0 -40 6 VIN = 12V -20 0 20 60 40 TEMPERATURE (°C) 80 100 0 2000 4000 6000 CAPACITANCE (pF) 8000 1A/div IL 10,000 APPLICATION CIRCUIT B 1ms/div _______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller HEAVY-LOAD SWITCHING WAVEFORM ENTERING SHUTDOWN MAX1846/7 toc16 MAX1846/7 toc17 SHDN VOUT 5V/div 0 100mV/div 5V/div 1A/div IL VOUT LX 1A/div IL 10V/div APPLICATION CIRCUIT B 1ms/div APPLICATION CIRCUIT B 1µs/div ILOAD = 600mA LIGHT-LOAD SWITCHING WAVEFORM MAX1846/7 toc18 VOUT 100mV/div 1A/div IL LX 10V/div APPLICATION CIRCUIT B 1µs/div ILOAD = 50mA LOAD-TRANSIENT RESPONSE LOAD-TRANSIENT RESPONSE MAX1846/7 toc19 MAX1846/7 toc20 ILOAD ILOAD VOUT VOUT 200mV/div IL 500mA/div 500mV/div IL 1A/div APPLICATION CIRCUIT B 2ms/div ILOAD = 10mA to 400mA APPLICATION CIRCUIT C 400µs/div ILOAD = 4mA to 100mA _______________________________________________________________________________________ 7 MAX1846/MAX1847 Typical Operating Characteristics (continued) (Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise noted.) High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847 Pin Description PIN MAX1847 — 1 POL 1 2 VL FUNCTION Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS lowside FET in transformer-based applications. VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND. 2 3 FREQ Oscillator Frequency Set Input. Connect a resistor (RFREQ) from FREQ to GND to set the internal oscillator frequency from 100kHz (RFREQ = 500kΩ) to 500kHz (RFREQ = 76.8kΩ). RFREQ is still required if an external clock is used at SYNC. See Setting the Operating Frequency section. 3 4 COMP Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network from COMP to GND for loop compensation. See Design Procedure. 4 5 REF 5 6 FB — 7,9 N.C. — 8 SHDN Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect to IN for normal operation. 6 10,11 GND Analog Ground. Connect to PGND. 7 12 PGND 8 13 CS Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND. Positive Current-Sense Input. Connect a current-sense resistor (RCS) between CS and 9 14 EXT External MOSFET Gate-Driver Output. EXT swings from IN to PGND. 10 15 IN — 8 NAME MAX1846 16 SYNC 1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic capacitor from REF to GND. Feedback Input. Connect FB to the center of a resistor-divider connected between the output and REF. The FB threshold is 0. No Connection Power-Supply Input Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set the internal oscillator frequency with RFREQ. Drive SYNC with a logic-level clock input signal to externally set the converter’s operating frequency. DC-DC conversion cycles initiate on the rising edge of the input clock signal. Note that when driving SYNC with an external signal, RFREQ must still be connected to FREQ. _______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller 3 x 22µF 10V VIN +3V to +5.5V 22kΩ FDS6375 CMSH5-40 2 0.47µF IN VL 8 16 SHDN EXT SYNC CS MAX1847 4 N.C. COMP PGND 10kΩ 3 0.22µF 5 150kΩ VOUT -12V AT 400mA 15 47µF 16V 10µH DO5022P-103 14 47µF 16V 13 Sanyo 16TPB47M 7, 9 0.02Ω 1W 12 R1 95.3kΩ 1% FREQ REF FB POL 1 GND 10, 11 6 R2 10.0kΩ 1% 1200pF 0.1µF _______________________________________________________________________________________ 9 MAX1846/MAX1847 Typical Application Circuit High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847 Functional Diagram IN EXT SHDN MAX1847 ONLY STARTUP CIRCUITRY PGND EXT DRIVER VL VL REGULATOR UNDERVOLTAGE LOCK OUT POL SYNC MAX1847 ONLY MAX1846 MAX1847 CONTROL CIRCUITRY OSCILLATOR FREQ ERROR COMPARATOR COMP CS FB GM CURRENTSENSE AMPLIFIER ERROR AMPLIFIER SOFT-START PGND SLOPE COMP REF REFERENCE GND 10 ______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller PWM Controller The architecture of the MAX1846/MAX1847 currentmode PWM controller is a Bi-CMOS multi-input system that simultaneously processes the output-error signal, the current-sense signal, and a slope-compensation ramp (Functional Diagram). Slope compensation prevents subharmonic oscillation, a potential result in current-mode regulators operating at greater than 50% duty cycle. The controller uses fixed-frequency, current-mode operation where the duty ratio is set by the input-to-output voltage ratio. The current-mode feedback loop regulates peak inductor current as a function of the output error signal. Internal Regulator The MAX1846/MAX1847 incorporate an internal lowdropout regulator (LDO). This LDO has a 4.25V output and powers all MAX1846/MAX1847 functions (excluding EXT) for the primary purpose of stabilizing the performance of the IC over a wide input voltage range (+3V to +16.5V). The input to this regulator is connected to IN, and the dropout voltage is typically 100mV, so that when VIN is less than 4.35V, VL is typically VIN minus 100mV. When the LDO is in dropout, the MAX1846/MAX1847 still operate with VIN as low as 3V. For best performance, it is recommended to connect VL to IN when the input supply is less than 4.5V. Undervoltage Lockout The MAX1846/MAX1847 have an undervoltage lockout circuit that monitors the voltage at VL. If VL falls below the UVLO threshold (2.8V typ), the control logic turns the P-channel FET off (EXT high impedance). The rest of the IC circuitry is still powered and operating. When VL increases to 60mV above the UVLO threshold, the IC resumes operation from a start up condition (soft-start). Soft-Start The MAX1846/MAX1847 feature a “digital” soft-start that is preset and requires no external capacitor. Upon startup, the FB threshold decrements from the reference voltage to 0 in 64 steps over 1024 cycles of fOSC or fSYNC. See the Typical Operating Characteristics for a scope picture of the soft-start operation. Soft-start is implemented: 1) when power is first applied to the IC, 2) when exiting shutdown with power already applied, and 3) when exiting undervoltage lockout. Shutdown (MAX1847 only) The MAX1847 shuts down to reduce the supply current to 10µA when SHDN is low. In this mode, the internal reference, error amplifier, comparators, and biasing circuitry turn off. The EXT output becomes high impedance and the external pullup resistor connected to EXT pulls VEXT to VIN, turning off the P-channel MOSFET. When in shutdown mode, the converter’s output goes to 0. Frequency Synchronization (MAX1847 only) The MAX1847 is capable of synchronizing its switching frequency with an external clock source. Drive SYNC with a logic-level clock input signal to synchronize the MAX1847. A switching cycle starts on the rising edge of the signal applied to SYNC. Note that the frequency of the signal applied to SYNC must be higher than the default frequency set by RFREQ. This is required so that the internal clock does not start a switching cycle prematurely. If SYNC is inactive for an entire clock cycle of the internal oscillator, the internal oscillator takes over the switching operation. Choose RFREQ such that fOSC = 0.9 ✕ fSYNC. EXT Polarity (MAX1847 only) The MAX1847 features an option to utilize an N-channel MOSFET configuration, rather than the typical P-channel MOSFET configuration (Figure 1). In order to drive the different polarities of these MOSFETs, the MAX1847 is capable of reversing the phase of EXT by 180 degrees. When driving a P-channel MOSFET, connect POL to GND. When driving an N-Channel MOSFET, connect POL to VL. These POL connections ensure the proper polarity for EXT. For design guidance in regard to this application, refer to the MAX1856 data sheet. Design Procedure Initial Specifications In order to start the design procedure, a few parameters must be identified: the minimum input voltage expected (V IN(MIN) ), the maximum input voltage expected (VIN(MAX)), the desired output voltage (VOUT), and the expected maximum load current (ILOAD). Calculate the Equivalent Load Resistance This is a simple calculation used to shorten the verification equations: RLOAD = VOUT / ILOAD ______________________________________________________________________________________ 11 MAX1846/MAX1847 Detailed Description The MAX1846/MAX1847 current-mode PWM controller use an inverting topology that is ideal for generating output voltages from -2V to -200V. Features include shutdown, adjustable internal operating frequency or synchronization to an external clock, soft-start, adjustable current limit, and a wide (+3V to +16.5V) input range. MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller VIN +12V 12µF VP1-0190 25V 12.2µH 1:4 CMR1U-02 0.47µF 8 1 2 VL POL 15 IN SHDN 16 SYNC MAX1847 0.033µF 270kΩ 150kΩ 3 5 COMP 14 470Ω 13 100pF 100V 12µF 100V EXT CS 4 VOUT -48V AT 100mA IRLL2705 N.C. PGND 7, 9 0.05Ω 0.5W 383kΩ 1% 12 FREQ REF FB 6 GND 10, 11 1800pF 10.0kΩ 1% 0.1µF Figure 1. Using an N-Channel MOSFET (MAX1847 only) Calculate the Duty Cycle The duty cycle is the ratio of the on-time of the MOSFET switch to the oscillator period. This is determined by the ratio of the input voltage to the output voltage. Since the input voltage typically has a range of operation, a minimum (DMIN) and maximum (DMAX) duty cycle is calculated by: DMIN = DMAX = − VOUT + VD VIN(MAX) − VSW − VLIM − VOUT + VD − VOUT + VD VIN(MIN) − VSW − VLIM − VOUT + VD where VD is the forward drop across the output diode, VSW is the drop across the external FET when on, and V LIM is the current-limit threshold. To begin with, assume VD = 0.5V for a Schottky diode, VSW = 100mV, and VLIM = 100mV. Remember that VOUT is negative when using this formula. Setting the Output Voltage The output voltage is set using two external resistors to form a resistive-divider to FB between the output and REF (refer to R1 and R2 in Figure 1). VREF is nominally 12 1.25V and the regulation voltage for FB is nominally 0. The load presented to the reference by the feedback resistors must be less than 500µA. This is to guarantee that VREF is in regulation (see Electrical Characteristics Table). Conversely, the current through the feedback resistors must be large enough so that the leakage current of the FB input (50nA) is insignificant. Therefore, select R2 so that IR2 is between 50µA and 250µA. IR2 = VREF / R2 where VREF = 1.25V. A typical value for R2 is 10kΩ. Once R2 is selected, calculate R1 with the following equation: R1 = R2 x (-VOUT / VREF) Setting the Operating Frequency The MAX1846/MAX1847 are capable of operating at switching frequencies from 100kHz to 500kHz. Choice of operating frequency depends on a number of factors: 1) Noise considerations may dictate setting (or synchronizing) f OSC above or below a certain frequency or band of frequencies, particularly in RF applications. ______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller 3) 4) 5) Higher frequencies allow the use of smaller value (hence smaller size) inductors and capacitors. Higher frequencies consume more operating power both to operate the IC and to charge and discharge the gate of the external FET. This tends to reduce the efficiency at light loads. Higher frequencies may exhibit lower overall efficiency due to more transition losses in the FET; however, this shortcoming can often be nullified by trading some of the inductor and capacitor size benefits for lower-resistance components. High-duty-cycle applications may require lower frequencies to accommodate the controller minimum off-time of 0.4µs. Calculate the maximum oscillator frequency with the following formula: fOSC(MAX) = VIN(MIN) − VSW − VLIM VIN(MIN) − VSW − VLIM − VOUT + VD 1 × t OFF(MIN) tions, most circuits are more efficient and economical operating in continuous mode, which refers to continuous current in the inductor. In continuous mode there is a trade-off between efficiency and transient response. Higher inductance means lower inductor ripple current, lower peak current, lower switching losses, and, therefore, higher efficiency. Lower inductance means higher inductor ripple current and faster transient response. A reasonable compromise is to choose the ratio of inductor ripple current to average continuous current at minimum duty cycle to be 0.4. Calculate the inductor ripple with the following formula: IRIPPLE = ( 0.4 × ILOAD(MAX) × VIN(MAX) − VSW − VLIM − VOUT + VD (VIN(MAX) − VSW − VLIM ) ) Then calculate an inductance value: L = (VIN(MAX) / IRIPPLE) x (DMIN / fOSC) Choose the closest standard value. Once again, remember that VOUT is negative when using this formula. Remember that VOUT is negative when using this formula. Determining Peak Inductor Current The oscillator frequency is set by a resistor, RFREQ, connected from FREQ to GND. The relationship between fOSC (in Hz) and RFREQ (in Ω) is slightly nonlinear, as illustrated in the Typical Operating Characteristics. Choose the resistor value from the graph and check the oscillator frequency using the following formula: The peak inductor current required for a particular output is: fOSC = ( ) ( ) 1 ILPEAK = ILDC + (ILPP / 2) where ILDC is the average DC input current and ILPP is the inductor peak-to-peak ripple current. The ILDC and ILPP terms are determined as follows: ILDC = ( ) 2 −19 5.21 × 10 −7 + 1.92 × 10 −11 × R × (RFREQ ) FREQ − 4.86 × 10 ILPP = External Synchronization (MAX1847 only) The SYNC input provides external-clock synchronization (if desired). When SYNC is driven with an external clock, the frequency of the clock directly sets the MAX1847’s switching frequency. A rising clock edge on SYNC is interpreted as a synchronization input. If the sync signal is lost, the internal oscillator takes over at the end of the last cycle, and the frequency is returned to the rate set by RFREQ. Choose RFREQ such that fOSC = 0.9 x fSYNC. Choosing Inductance Value The inductance value determines the operation of the current-mode regulator. Except for low-current applica- ILOAD × ( − VOUT + VD ) VIN(MIN) − VSW − VLIM (VIN(MIN) − VSW − VLIM ) × (− VOUT + VD ) L × fOSC × ( − VOUT + VD ) where L is the selected inductance value. The saturation rating of the selected inductor should meet or exceed the calculated value for ILPEAK, although most coil types can be operated up to 20% over their saturation rating without difficulty. In addition to the saturation criteria, the inductor should have as low a series resistance as possible. For continuous inductor current, the power loss in the inductor resistance (PLR) is approximated by: PLR ~ (ILOAD x VOUT / VIN)2 x RL where RL is the inductor series resistance. ______________________________________________________________________________________ 13 MAX1846/MAX1847 2) MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller Once the peak inductor current is calculated, the current sense resistor, RCS, is determined by: RCS = 85mV / ILPEAK For high peak inductor currents (>1A), Kelvin-sensing connections should be used to connect CS and PGND to R CS . Connect PGND and GND together at the ground side of RCS. A lowpass filter between RCS and CS may be required to prevent switching noise from tripping the current-sense comparator at heavy loads. Connect a 100Ω resistor between CS and the high side of RCS, and connect a 1000pF capacitor between CS and GND. Checking Slope-Compensation Stability In a current-mode regulator, the cycle-by-cycle stability is dependent on slope compensation to prevent subharmonic oscillation at duty cycles greater than 50%. For the MAX1846/MAX1847, the internal slope compensation is optimized for a minimum inductor value (LMIN) with respect to duty cycle. For duty cycles greater then 50%, check stability by calculating LMIN using the following equation: [( ) LMIN = VIN(MIN) × RCS / MS [ ] ] × (2 × DMAX − 1) / (1 − DMAX ) where VIN(MIN) is the minimum expected input voltage, Ms is the Slope Compensation Ramp (41 mV/µs) and DMAX is the maximum expected duty cycle. If LMIN is larger than L, increase the value of L to the next standard value that is larger than L MIN to ensure slope compensation stability. Power MOSFET Selection The MAX1846/MAX1847 drive a wide variety of P-channel power MOSFETs (PFETs). The best performance, especially with input voltages below 5V, is achieved with low-threshold PFETs that specify on-resistance with a gate-to-source voltage (VGS) of 2.7V or less. When selecting a PFET, key parameters include: 1) Total gate charge (QG) 2) Reverse transfer capacitance (CRSS) 3) On-resistance (RDS(ON)) 4) Maximum drain-to-source voltage (VDS(MAX)) 5) Minimum threshold voltage (VTH(MIN)) At high switching rates, dynamic characteristics (parameters 1 and 2 above) that predict switching losses may have more impact on efficiency than R DS(ON), which predicts DC losses. QG includes all capacitance 14 associated with charging the gate. In addition, this parameter helps predict the current needed to drive the gate at the selected operating frequency. The power MOSFET in an inverting converter must have a high enough voltage rating to handle the input voltage plus the magnitude of the output voltage and any spikes induced by leakage inductance. Choose RDS(ON)(MAX) specified at VGS < VIN(MIN) to be one to two times RCS. Verify that VIN(MAX) < VGS(MAX) and VDS(MAX) > VIN(MAX) - VOUT + VD. Choose the riseand fall-times (tR, tF) to be less than 50ns. Output Capacitor Selection The output capacitor (COUT) does all the filtering in an inverting converter. The output ripple is created by the variations in the charge stored in the output capacitor with each pulse and the voltage drop across the capacitor’s equivalent series resistance (ESR) caused by the current into and out of the capacitor. There are two properties of the output capacitor that affect ripple voltage: the capacitance value, and the capacitor’s ESR. The output ripple due to the output capacitor’s value is given by: VRIPPLE-C = (ILOAD ✕ DMAX ✕ TOSC ) / COUT The output ripple due to the output capacitor’s ESR is given by: VRIPPLE-R = ILPP ✕ RESR These two ripple voltages are additive and the total output ripple is: VRIPPLE-T = VRIPPLE-C + VRIPPLE-R The ESR-induced ripple usually dominates this last equation, so typically output capacitor selection is based mostly upon the capacitor’s ESR, voltage rating, and ripple current rating. Use the following formula to determine the maximum ESR for a desired output ripple voltage (VRIPPLE-D): RESR = VRIPPLE-D / ILPP Select a capacitor with ESR rating less than RESR. The value of this capacitor is highly dependent on dielectric type, package size, and voltage rating. In general, when choosing a capacitor, it is recommended to use low-ESR capacitor types such as ceramic, organic, or tantalum capacitors. Ensure that the selected capacitor has sufficient margin to safely handle the maximum ripple current (ILPP) and the maximum output voltage. Choosing Compensation Components The MAX1846/MAX1847 are externally loop-compensated devices. This provides flexibility in designs to accommodate a variety of applications. Proper com- ______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller Calculating Poles and Zeros The MAX1846/MAX1847 current-mode architecture takes the double pole caused by the inductor and output capacitor and shifts one of these poles to a much higher frequency. This makes loop compensation easier. To compensate these devices, we must know the center frequencies of the right-half plane zero (zRHP) and the higher frequency pole (pOUT2). Calculate the zRHP frequency with the following formula: 2 − (1 − D MAX ) × VIN(MIN) − VOUT × RLOAD ZRHP = (2π × VOUT × L) ( ) The calculations for pOUT2 are very complex. For most applications where VOUT does not exceed -48V (in a negative sense), the pOUT2 will not be lower than 1/8th of the oscillator frequency and is generally at a higher frequency than zRHP. Therefore: pOUT2 ≥ 0.125 ✕ fOSC A pole is created by the output capacitor and the load resistance. This pole must also be compensated and its center frequency is given by the formula: pOUT1 = 1 / (2π ✕ RLOAD ✕ COUT) Finally, there is a zero introduced by the ESR of the output capacitor. This zero is determined from the following equation: zESR = 1 / (2π ✕ COUT ✕ RESR) Calculating the Required Pole Frequency To ensure stability of the MAX1846/MAX1847, the introduced pole (PDOM) by the compensation network must roll-off the error amplifier gain to 1 before z RHP or POUT2 occurs. First calculate the DC open-loop gain to determine the frequency of the pole to introduce. ( ) 2 G × R × 1 − D O ( MAX ) × VIN(MIN) − VOUT M × R LOAD ADC = R × VIN(MIN) + TOSC (1 − DMAX ) CS B× VIN(MIN) × RLOAD × RCS + MS1 2L ( ) where: B is the feedback divider attenuation factor = (-VOUT / VREF), G M is the error amplifier transconductance = 400 µA/V, RO is the error amplifier output resistance = 3 MΩ, M S1 is the slope compensation factor = [(1.636A / µs) ✕ RCS], RCS is the selected current sense resistor, L is the selected inductance value If zRHP is at a lower frequency than pOUT2, the required dominant pole frequency is given by: pDOM = zRHP / ADC Otherwise the required dominant pole frequency is: pDOM = pOUT2 / ADC Determining the Compensation Component Values Using p DOM, calculate the compensation capacitor required: CCOMP = 1 / (2π ✕ RO ✕ pDOM) Select the next largest standard value of capacitor and then calculate the compensation resistor required to cancel out the output-capacitor-induced pole (pOUT1) determined previously. A zero is needed to cancel the output-induced pole and the frequency of this zero must equal pOUT1. Therefore: zCOMP = pOUT1 RCOMP = RLOAD ✕ COUT / CCOMP Choose the nearest lower standard value of the resistor. Now check the final values selected for the compensation components: pCOMP = 1 / [2π ✕ CCOMP x (RO + RCOMP)] In order for pCOMP to compensate the loop, the openloop gain must reach unity at a lower frequency than the right-half-plane zero or the second output pole, whichever is lower in frequency. If the second output pole and the right-half-plane zero are close together in frequency, the higher resulting phase shift at unity gain ______________________________________________________________________________________ 15 MAX1846/MAX1847 pensation of the control loop is important to prevent excessive output ripple and poor efficiency caused by instability. The goal of compensation is to cancel unwanted poles and zeros in the DC-DC converter’s transfer function created by the power-switching and filter elements. More precisely, the objective of compensation is to ensure stability by ensuring that the DCDC converter’s phase shift is less than 180° by a safe margin, at the frequency where the loop gain falls below unity. One method for ensuring adequate phase margin is to introduce corresponding zeros and poles in the feedback network to approximate a single-pole response with a -20dB/decade slope all the way to unity-gain crossover. MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller may require a larger compensation capacitor than calculated. It might take more than a couple of iterations to obtain a suitable combination. Finally, the zero introduced by the output capacitor’s ESR must be compensated. This is accomplished by placing a capacitor between REF and FB creating a pole directly in the feedback loop. Calculate the value of this capacitor using the frequency of zESR and the selected feedback resistor values with the formula: R + R2 CFB = RESR × COUT × 1 R1 × R2 Applications Information Maximum Output Power The maximum output power that the MAX1846/MAX1847 can provide depends on the maximum input power available and the circuit’s efficiency: POUT(MAX) = Efficiency ✕ PIN(MAX) Furthermore, the efficiency and input power are both functions of component selection. Efficiency losses can be divided into three categories: 1) resistive losses across the inductor, MOSFET on-resistance, currentsense resistor, and the ESR of the input and output capacitors; 2) switching losses due to the MOSFET’s transition region, and charging the MOSFET’s gate capacitance; and 3) inductor core losses. Typically 80% efficiency can be assumed for initial calculations. The required input power depends on the inductor current limit, input voltage, output voltage, output current, inductor value, and the switching frequency. The maximum output power is approximated by the following formula: PMAX = [VIN - (VLIM + ILIM x RDS(ON))] x ILIM x [1 - (LIR / 2)] x [(-VOUT + VD) / (VIN - VSW - VLIM - VOUT + VD)] where I LIM is the peak current limit and LIR is the inductor current-ripple ratio and is calculated by: LIR = ILPP / ILDC Again, remember that V OUT for the MAX1846/ MAX1847 is negative. Diode Selection The MAX1846/MAX1847’s high-switching frequency demands a high-speed rectifier. Schottky diodes are recommended for most applications because of their fast recovery time and low forward voltage. Ensure that the diode’s average current rating exceeds the peak inductor current by using the diode manufacturer’s data. Additionally, the diode’s reverse breakdown voltage must 16 exceed the potential difference between VOUT and the input voltage plus the leakage inductance spikes. For high output voltages (-50V or more), Schottky diodes may not be practical because of this voltage requirement. In these cases, use an ultrafast recovery diode with adequate reverse-breakdown voltage. Input Filter Capacitor The input capacitor (CIN) in inverting converter designs reduces the current peaks drawn from the input supply and reduces noise injection. The source impedance of the input supply largely determines the value of CIN. High source impedance requires high input capacitance, particularly as the input voltage falls. Since inverting converters act as “constant-power” loads to their input supply, input current rises as the input voltage falls. Consequently, in low-input-voltage designs, increasing C IN and/or lowering its ESR can add as much as 5% to the conversion efficiency. Bypass Capacitor In addition to CIN and COUT, other ceramic bypass capacitors are required with the MAX1846/MAX1847. Bypass REF to GND with a 0.1µF or larger capacitor. Bypass VL to GND with a 0.47µF or larger capacitor. All bypass capacitors should be located as close to their respective pins as possible. PC Board Layout Guidelines Good PC board layout and routing are required in highfrequency-switching power supplies to achieve good regulation, high efficiency, and stability. It is strongly recommended that the evaluation kit PC board layouts be followed as closely as possible. Place power components as close together as possible, keeping their traces short, direct, and wide. Avoid interconnecting the ground pins of the power components using vias through an internal ground plane. Instead, keep the power components close together and route them in a “star” ground configuration using component-side copper, then connect the star ground to internal ground using multiple vias. Main Application Circuits The MAX1846/MAX1847 are extremely versatile devices. Figure 2 shows a generic schematic of the MAX1846. Table 1 lists component values for several typical applications. These component values also apply to the MAX1847. The first two applications are featured in the MAX1846/MAX1847 EV Kit. ______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847 VIN APPLICATION B ONLY CIN 22k P D1 1 0.47µF VOUT 10 IN VL EXT CS 9 COUT L1 8 MAX1846 RCS 3 CCOMP RCOMP RFREQ 2 4 COMP PGND R1 7 FREQ REF FB 5 R2 GND CFB 6 0.1µF NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS. Figure 2. MAX1846 Main Application Circuit Table 1. Component List for Main Application Circuits A B C D Input (V) CIRCUIT ID 12 3 to 5.5 12 12 Output (V) -5 -12 -48 -72 Output (A) 2 0.4 0.1 0.1 0.047 0.22 0.068 0.1 CCOMP (µF) CIN (µF) 3 x 10 3 x 22 10 10 COUT (µF) 2 x 100 2 x 47 47 33 CFB (pF) 390 1200 1800 1800 R1 (kΩ) (1%) 40.2 95.3 383 576 R2 (kΩ) (1%) 10 10 10 10 RCOMP (kΩ) 8.2 10 150 1800 RCS (Ω) 0.02 0.02 0.05 0.05 RFREQ (kΩ) D1 L1 (µH) P1 150 150 150 150 CMSH5-40 CMSH5-40 CMR1U-02 CMR1U-02 10 10 47 82 FDS6685 FDS6375 IRFR5410 IRFR5410 ______________________________________________________________________________________ 17 MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller Component Suppliers SUPPLIER COMPONENT PHONE WEBSITE Capacitors 803-946-0690 www.avxcorp.com Diodes 516-435-1110 www.centralsemi.com Coilcraft Inductors 847-639-6400 www.coilcraft.com Dale Resistors 402-564-3131 www.vishay.com/brands/dale/main.html Fairchild MOSFETs 408-721-2181 www.fairchildsemi.com International Rectifier MOSFETs 310-322-3331 www.irf.com IRC Resistors 512-992-7900 www.irctt.com AVX Central Semiconductor Kemet Capacitors 864-963-6300 www.kemet.com MOSFETs, Diodes 602-303-5454 www.onsemi.com Capacitors, Resistors 201-348-7522 www.panasonic.com On Semiconductor Panasonic Sanyo Capacitors 619-661-6835 www.secc.co.jp Siliconix MOSFETs 408-988-8000 www.siliconix.com Sprague Capacitors 603-224-1961 www.vishay.com/brands/sprague/main.html Sumida Inductors 847-956-0666 www.remtechcorp.com Vitramon Resistors 203-268-6261 www.vishay.com/brands/vitramon/main.html Note: Please indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers. Pin Configurations TOP VIEW VL 1 FREQ 2 COMP 3 REF 4 FB Chip Information TRANSISTOR COUNT: 2441 PROCESS TECHNOLOGY: BiCMOS 10 IN MAX1846 5 10-PIN µMAX POL 1 16 SYNC VL 2 15 IN 9 EXT 8 CS 7 PGND COMP 4 6 REF 5 12 PGND FB 6 11 GND N.C. 7 10 GND GND 14 EXT FREQ 3 MAX1847 SHDN 8 13 CS 9 N.C. 16-PIN QSOP 18 ______________________________________________________________________________________ High-Efficiency, Current-Mode, Inverting PWM Controller 10LUMAX.EPS ______________________________________________________________________________________ 19 MAX1846/MAX1847 Package Information High-Efficiency, Current-Mode, Inverting PWM Controller QSOP.EPS MAX1846/MAX1847 Package Information (continued) Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2001 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.