iW1692 Low-Power Off-line Digital PWM Controller 1.0 Features 2.0 Description • Primary-side feedback eliminates opto-isolators and simplifies design • Multi-mode operation for highest overall efficiency • Built-in cable drop compensation • Very tight output voltage regulation • No external loop compensation components required • Complies with CEC/EPA/IEC no load power consumption and average efficiency regulations • Built-in output constant-current control with primary-side feedback • Low start-up current (10 µA typical) • Built-in soft start • Built-in short circuit protection • AC line under/overvoltage and output overvoltage protection • 40 kHz PWM switching frequency The iW1692 is a high performance AC/DC power supply controller which uses digital control technology to build peak current mode PWM flyback power supplies. The device provides high efficiency along with a number of key built-in protection features while minimizing the external component count and bill of material cost. The iW1692 removes the need for secondary feedback circuitry while achieving excellent line and load regulation. It also eliminates the need for loop compensation components while maintaining stability over all operating conditions. Pulse-by-pulse waveform analysis allows for a loop response that is much faster than traditional solutions, resulting in improved dynamic load response. The built-in power limit function enables optimized transformer design in universal off-line applications and allows for a wide input voltage range. The low start-up power and PFM operation at light load ensure that the iW1692 is ideal for applications targeting the newest regulatory standards for standby power. 3.0 Applications • Low power AC/DC adapter/chargers for cell phones, PDAs, digital still cameras • PFM operation at light load • Built-in ISENSE pin short protection • Standby supplies for televisions, DVDs, set-top boxes and other consumer electronics • Space-saving SOT-23 package L N + + + VOUT RTN + 4 VCC 5 ISENSE 6 VIN OUTPUT 3 GND 2 VSENSE 1 U1 iW1692 Figure 2.0.1 iW1692 Typical Application Circuit MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 4.0 Pinout Description iW1692 1 VSENSE 2 GND 3 OUTPUT VIN 6 ISENSE 5 VCC 4 Pin # Name Type Pin Description 1 VSENSE Input Voltage sense input from the auxiliary winding. 2 GND Ground Ground connection. 3 OUTPUT Output Gate drive output for the external power MOSFET switch. 4 VCC Input Supply voltage. 5 ISENSE Input Primary current sense. Used for cycle-by-cycle peak current control and limit. 6 VIN Input Senses average rectified input voltage. 5.0 Absolute Maximum Ratings Absolute maximum ratings are the parametric values or ranges which can cause permanent damage if exceeded. For maximum safe operating conditions, refer to Electrical Characteristics in Section 6.0. Parameter Symbol Value Units DC supply voltage range (pin 4, ICC = 20mA max) VCC -0.3 to 18 V DC supply current at VCC pin ICC 20 mA Output (pin 3) -0.3 to 18 V VSENSE input (pin 1) -0.3 to 4.0 V ISENSE input (pin 5) -0.3 to 4.0 V VIN input (pin 6) -0.3 to 18 V Power dissipation at TA ≤ 25°C PD 400 mW Maximum junction temperature TJ (MAX) 125 °C Storage temperature TSTG –65 to 150 °C Lead temperature during IR reflow for ≤ 15 seconds TLEAD 260 °C θJA 240 °C/W ESD rating per JEDEC JESD22-A114 (HBM) 2,000 V Latch-Up test per JEDEC 78 ±100 mA Thermal resistance junction-to-ambient MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 6.0 Electrical Characteristics VCC = 12 V, -40°C ≤ TA ≤ 85°C, unless otherwise specified (Note 1) Parameter Symbol Test Conditions Min Typ Max Unit 366 407 448 mV 10 15 µA VIN SECTION (Pin 6) Start-up voltage threshold VINST TA= 25°C, positive edge Start-up current IINST VIN = 10 V, CVCC = 10 µF ROUTPUT = 10 kW to GND Shutdown low voltage threshold VUVDC TA= 25°C 216 240 264 mV Shutdown high voltage threshold VOVDC TA= 25°C 1.834 1.988 2.123 V Input impedance ZIN After start-up IBVS VSENSE = 2 V 20 kW VSENSE SECTION (Pin 1) Input leakage current 1 μA Nominal voltage threshold VSENSE(NOM) TA=25°C, negative edge 1.523 1.538 1.553 V Output OVP threshold VSENSE(MAX) TA=25°C, negative edge 1.683 1.700 1.717 V OUTPUT SECTION (Pin 3) Output low level ON-resistance RDS(ON)LO ISINK = 5 mA 45 100 W Output high level ON-resistance RDS(ON)HI ISOURCE = 5 mA 65 100 W Rise time (Note 2) tR TA = 25°C, CL = 330 pF 10% to 90% 40 75 ns Fall time (Note 2) tF TA = 25°C, CL = 330 pF 90% to 10% 40 75 ns Output switching frequency fS ILOAD > 15% of maximum 40 44 kHz 16 V 36 VCC SECTION (Pin 4) Maximum operating voltage VCC(MAX) Start-up threshold VCC(ST) VCC rising 11.0 12.0 13.2 V Undervoltage lockout threshold VCC(UVL) VCC falling 5.5 6.0 6.6 V 2.5 3.5 mA Operating current ICCQ CL = 330 pF, VSENSE = 1.5 V ISENSE SECTION (Pin 5) Peak limit threshold VPEAK 1000 mV CC limit threshold VCC-TH 900 mV Notes: Note 1. Adjust VCC above the start-up threshold before setting at 12 V. Note 2. These parameters are not 100% tested, guaranteed by design and characterization. MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 7.0 Typical Performance Characteristics 12.4 VCC Start-up Threshold (V) VCC Supply Current (mA) 2.8 2.6 12.3 2.4 2.2 12.2 2.0 12.1 1.8 1.6 VCC = 12 V TA = 25° 0 200 400 600 800 Load Capacitance (pF) 1000 12.0 Figure 7.0.1 Supply Current vs. Load Capacitance Internal Reference Voltage (V) Switching Frequency (kHz) 0 25 50 Ambient Temperature (°C) 75 100 2.015 42 40 38 VCC = 12 V -50 -25 0 25 50 Ambient Temperature (°C) 75 100 2.010 2.005 2.000 1.995 Figure 7.0.2 Switching Frequency vs. Temperature -25 Figure 7.0.3 Start-Up Threshold vs. Temperature 44 36 -50 VCC = 12 V -50 -25 0 25 50 Ambient Temperature (°C) 75 100 Figure 7.0.4 Internal Reference vs. Temperature MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 8.0 Functional Block Diagram VIN 4 6 VIN_A 0.2 V ~ 2.0 V VSENSE 1 Start-up VINSW VINSW Signal Conditioning VCC ADC Gate Driver Digital Logic Control VVMS OUTPUT 3 VFB VOCP GND – + 2 1.0 V ISENSE 5 DAC IPEAK VIPK 0.2 V ~ 0.9 V + – – Figure 8.0.1 iW1692 Functional Block Diagram 9.0 Theory of Operation The iW1692 is a digital controller which uses a new, proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs. This results in a low-cost solution for low power AC/DC adapters. The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency. Furthermore, iWatt’s digital control technology enables fast dynamic response, tight output regulation, and full featured circuit protection with primary-side control. Furthermore, accurate secondary constant-current operation is achieved without the need for any secondary-side sense and control circuits. Referring to the block diagram in Figure 8.0.1, the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal. The system loop is internally compensated inside the digital logic control, and no external analog components are required for loop compensation. The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability. iWatt’s digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design. This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost, smallest size and high performance output control. The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation. Additional built-in protection features include overvoltage protection (OVP), output short circuit protection (SCP), AC low line brown out, over current protection, single pin fault protection and ISENSE fault detection. MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 9.1 Pin Detail Start-up Sequencing Pin 1 – VSENSE Sense signal input from auxiliary winding. This provides the secondary voltage feedback used for output regulation. VIN Pin 2 – GND Analog, digital and power ground. VCC Pin 3 – OUTPUT Gate drive signal for the external power MOSFET switch. VINSW OFF ON Pin 4 – VCC Power supply for the controller during normal operation. The controller starts up when VCC reaches 12 V (typical) and shuts-down when the VCC voltage is below 6 V (typical). A 100 nF decoupling capacitor should be connected between the VCC pin and GND. Pin 5 – ISENSE Primary current sense. Pin 6 – Vin Sense signal input representing the instantaneous rectified line voltage. VIN is used for line regulation. The internal impedanace is 20 kW and the scale factor is 0.0043. It also provides input undervoltage and overvoltage protection. This pin also provides the supply current to the IC during start-up. 9.2 Start-up Prior to start-up the VIN pin charges up the VCC capacitor, through the diode between VIN and VCC. When VCC is fully charged to a voltage higher than VCC(ST) threshold, then the VIN_SW turns on and the analog-to-digital converter begins to sense the input voltage. The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST. The iW1692 incorporates an internal soft-start function. The soft-start time is set at 3.0 ms. Once the VIN pin voltage has reached its turn-on threshold, the iW1692 starts switching, but limits the on-time to a percentage of the maximum ontime. During the first 1 ms, the on-time is limited to 25%. During the next 1 ms, the on-time is limited to 50% and during the last 1 ms, the on-time is limited to 75%. If at any time the VCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset. At this time the VIN_SW switches off so that the VCC capacitor can be charged up again. VIN Impedance = 20kΩ VCC(ST) ENABLE 200µs Figure 9.2.1 Start-up Sequencing Diagram 9.3 Understanding Primary Feedback Figure 9.3.1 illustrates a simplified flyback converter. When the switch Q1 conducts during tON, the current ig is directly drawn from rectified sinusoid vg. The energy Eg is stored in the primary winding. The rectifying diode D1 is reverse biased and the load current IO is supplied by the secondary capacitor CO. When Q1 turns off, D1 conducts and the stored energy Eg(t) is delivered to the output. iin(t) + ig(t) id(t) D1 NS NP vg(t) vin(t) VO + CO IO NAUX – VAUX TS(t) Q1 Figure 9.3.1 Simplified Flyback Converter In order to regulate the output voltage within a tight specification, the information about the output voltage and load current needs to be accurately sensed. In the DCM flyback converter, this information can be read via the auxiliary winding or the primary magnetizing inductance (LM). During the Q1 on-time, the load current is supplied from the output filter capacitor CO. The voltage across the primary winding is vg(t), assuming the voltage dropped across Q1 is zero. The current in Q1 ramps up linearly at a rate of: dig (t ) dt MK-4008-B 11/20/07 = vg (t ) LM (9.1) Page iW1692 Low-Power Off-line Digital PWM Controller At the end of on-time, the current has ramped up to: ig (t ) = vg (t ) × tON (t ) LM (9.2) 9.4 Understanding CC and CV mode This current represents a stored energy of: Eg = LM × ig (t ) 2 2 (9.3) When Q1 turns off, ig(t) in LM forces a reversal of polarities on all windings. Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off, the primary current transfers to the secondary at an amplitude of: id (t ) = NP × ig (t ) NS (9.4) The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB. The VFB signal precisely represents the output voltage and is used to regulate the output voltage. Assuming the secondary winding is master, the auxiliary winding is slave. See equation 9.5 The constant current mode (CC mode) is useful in battery charging applications. During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop, while avoiding continuous conduction mode. To achieve this regulation the iW1692 senses the load current indirectly through the primary current. The primary current is detected by the ISENSE pin through a resistor from the MOSFET source to ground (RSS). This resistor value is given by: RSS = N × KC 2 × I OUTMAX (9.6) N is the ratio of primary turns to secondary turns of the transformer and KC is given as 0.264 V. 9.5 Constant Voltage Operation VAUX After soft-start is completed, the digital control block measures the output conditions. If the ISENSE signal is not consistently over 0.9 V, then the device will operate in constant voltage mode. 0V VAUX = -VG x NAUX NP If no voltage is detected on VSENSE after 20 pulses, it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down. Figure 9.3.2 Auxiliary Voltage Waveforms As long as calculated TON for CV is less than the TON in CC the IC operates in constant voltage mode. The auxiliary voltage is given by: VAUX = N AUX (VO + ∆V ) NS (9.5) and reflects the output voltage as shown in Figure 9.3.2. The voltage at the load differs from the secondary voltage by a diode drop and IR losses. The diode drop is a function of current, as are IR losses. Thus, if the secondary voltage is always read at a constant secondary current, the difference between the output voltage and the secondary voltage is a fixed ΔV. Furthermore, if the voltage can be read when the secondary current is small, ΔV is small. 9.6 Constant Current Operation The iW1692 has been designed to work in constant-current mode for battery charging applications. If the output voltage drops, but does not go below 20% of the nominal designed value, the device operates in this mode. MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 9.10 Voltage Protection Functions CV mode The iW1692 includes functions that protect against input and output overvoltage. CC mode Output Voltage VNOM Output Current IOUT(CC) The input voltage is monitored by the VIN pin and the output voltage is monitored by the VSENSE pin. If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles, the iW1692 shuts-down immediately. However, the IC remains biased which discharges the VCC supply. Once VCC drops below the UVLO threshold, the controller resets itself and then initiates a new soft-start cycle. The controller continues attempting start-up, but does not fully start-up until the fault condition is removed. The output voltage can be high enough to damage the output capacitor when the feedback loop is broken. The iW1692 uses the primary feedback only with no secondary feedback loop. When the VSENSE pin is shorted to GND (by shorting/open sense resistor). The controller will shut off with 6 consecutive pulses after start-up. Figure 9.6.1 Modes of operation 9.7 Variable Frequency Mode The iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes. To avoid operation in continuous conduction (CCM) mode, the iW1692 checks for the falling edge of the VSENSE input on every cycle. If a falling edge of VSENSE is not detected during the normal 25μs period, the switching period is extended until the falling edge VSENSE does occur. If the switching period reaches 75μs without VSENSE being detected, the iW1692 immediately shuts off. 9.8 PFM Mode at Light Load The iW1692 operates in a fixed frequency PWM mode when IOUT is greater than approximately 5% of the specified maximum load current. As the output load IOUT is reduced, the on-time tON is decreased. At the moment that tON drops below tON_MIN, the controller transitions to Pulse Frequency Modulation (PFM) mode. Thereafter, the on-time is modulated by the line voltage and the off-time is modulated by the load current. The device automatically returns to PWM mode when the load current increases. 9.11 Cable Drop Compensation The iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector. A 5 W AC adapter with 5 VDC output has 6% deviation at 1 A load current due to the drop across the DC cable without cable compensation. The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected. The iW1692-30 has 300mV of cable drop compensation at maximum current. The iW1692-00 does not include any cable drop compensation. 9.9 Internal Loop Compensation The iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation. The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and –20dB of gain margin. MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller 10.0 Design Example 10.1 Design Procedure Parameter This design example gives the procedure for a flyback converter using iW1692. Refer to figure 12.0.1 for the application circuit. The design objectives for this adapter are given in table 10.1. It meets UL, IEC, and CEC requirements. Symbol VIN 85 - 264 VRMS Frequency fIN 47 - 64 Hz No Load Input PIN 200 mW Input Voltage Output Voltage VOUT_CABLE 4.95 - 5.05 V Output Current IOUT 1 A Output Ripple VRIPPLE <100 mV POUT 5 W h 65% Power Out Determine the Design Specifications (Vout, Iout_max, Vin_max, Vin_min, efficiency, and ripple) CEC Efficiency Determine Cable Drop Compensation Table 10.1 iW1692 Design Specification Table 10.2 Cable Drop Compensation Determine Rvin Resistors Cable Drop Compensation is an option included in the iW1692-30. This option helps maintain the output voltage at the end of the cable that the power supply is designed for. During CV (constant voltage) mode the output current changes as the voltage remains constant. This is true for the output voltage at the output of the power supply board; however, in certain applications the device to be charged is not directly connected to the power supply, but rather, is connected via a cable. This cable is seen by the power supply as a resistance. So as the output current increases the output voltage at the end of the cable begins to drop. With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance. Determine Turns Ratio Determine Input Bulk Capacitance Determine Current Sensing Resistor Determine Magnetizing Inductance Determine Primary Turns Determine Secondary Turns No Can you wind this transformer ? Yes Determine Vsense Turns and Resistors Determine Bias Turns Is the real cable drop compensation value OK ? Yes Determine Output Capacitance Determine Snubber Network Determine Ton Delay Compensation Finish Figure 3.0.1: Design Flow Chart Figure 10.0.1 iW1692 Design Flow Chart Range No To find the right cable compensation type for a given cable pick the cable drop compensation number that is closest to the voltage drop of the cable under maximum output current. Use equation 10.1 for VOUT, where VFD is the forward voltage of the output diode: VOUT = VOUT _ CABLE + VCABLE _ DROP _ COMPENSATION + V fd (10.1) Using equation 10.1 we know for this design VOUT is 5.5 V, assuming no cable drop compensation is chosen and the forward drop on the output diode (VFD) is 500 mV. 10.3 Input Selection VIN resistors are chosen primarily to scale down the input voltage for the IC. The scale factor for the input voltage in the IC is 0.0043 and the internal impedance of this pin is 20 kΩ. Therefore, the VIN resistors should equate to: MK-4008-B 11/20/07 Page iW1692 Low-Power Off-line Digital PWM Controller RVin = 20k W − 20k W = 4.63M W 0.0043 (10.2) From equation 10.2, ideally RVIN should be 4.63 MΩ because R10 and R11 add up to approximately 4.6 MΩ. By selecting the value of RVIN, the (VIN·TON)MAX_LIMIT and (VIN·TON)PFM are determined: (VIN ⋅ TON )MAX _ LIMIT (VIN ⋅ TON )PFM = 0.0043 × = 0.0043 × 900V ⋅m s 20k W RVin + 20k W (10.3) 185V ⋅m s 20k W RVin + 20k W The product of VIN and TON is typically chosen by equation 10.9 for CC limit performance. For this example we choose 750 V·μs. 700V ⋅ms < (VIN × TON )MAX < 850V ⋅m s Assuming TON_MAX is 9.7µs and TDEAD is typically about 4.8 ms, solving for the minimum turns ratio yields. TRESET _ MAX = 25m s − 9.7m s − 4.8m s TRESET _ MAX = 10.5m s Ntr _ MIN = (10.4) Keep in mind by changing RVIN to be something other than 4.63 MΩ the minimum and maximum input voltage for startup will also change. Since the iW1692 uses the exact scaled value of VIN for its calculations, C6 should be included to filter out any noise that may appear on the VIN signal. This is especially important for line-in surge conditions. Ntr _ MAX = (VIN × TON )PFM TRESET _ MIN × VOUT (10.5) To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD – TON – TDEAD. TPERIOD is given by the PWM switching frequency of 40 kHz. TRESET_MAX is given by: TRESET _ MAX = TPERIOD − TON _ MAX − TDEAD (10.6) Ntr _ MIN = (VIN × TON )MAX TRESET _ MAX × VOUT (10.7) (VIN×TON)PFM is limited by the iW1692 to be 185 V·μs, and TRESET_MIN is required by the IC to be 2.3 μs. Ntr _ MAX 185V ⋅m s = = 15 2.3m s× 5.5V 750V ⋅m sec (10.5m sec )× 5.5V (10.8) = 13 (10.11) 10.5 Input Bulk Capacitor The input bulk capacitance (C1 // C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping. In order for this to be true the minimum total input bulk capacitance must be: C1 + C2 = PIN = VINDC _ MIN arcsin 2 ×VINAC _ MIN 2 × PIN × 2π (10.12) 2 2 (2VINAC _ MIN − VINDC _ MIN ) × f line VOUT × I OUT hPowerSupply VINAC_MIN is the minimum input voltage (rms) to be inputted into the power supply and ƒline is the lowest line frequency for the power supply (in this case 47 Hz). VINDC_MIN is calculated based on the (VIN × TON)MAX product. VINDC _ MIN = Thus, the minimum turns ratio is given by: (10.10) Pick a number between the maximum and minimum turns ratio; in the example the turn ratio is 13. A turns ratio in the range of 11 to 15 is suggested for optimal performance. 10.4 Turns Ratio The maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer, during PFM mode (10.9) (VIN × TON )MAX TON _ MAX (10.13) First we must find TON_MAX to get VINDC_MIN. In order for the power supply to function in discontinuous conduction mode TON_MAX should be smaller than the switching period minus the transformer reset time. Given that the transformer reset time is: TRESET = MK-4008-B 11/20/07 (VIN × TON )MAX Ntr ⋅ VOUT (10.14) Page 10 iW1692 Low-Power Off-line Digital PWM Controller Then the maximum on-time must be: TON _ MAX = TPERIOD − (VIN × TON )MAX Ntr ⋅ VOUT VIsense _ CC = − TDEAD (10.15) TDEAD is about 4.8 μs. Knowing TPERIOD has to be 25 μs, because of the 40 kHz switching frequency: TON _ MAX = 25m s − 750V ⋅m s − 4.8m s = 9.7m s 13 × 5.5V (10.16) From this result we can now get VINDC_MIN from equation 10.13: VINDC _ MIN = (VIN × TON )MAX TON _ MAX = 77.2V (10.17) Substituting VINDC_MIN into equation 10.12 we get: ( C1 + C2 = arcsin 77.2V 2 ×85V 2 × 6.41W × 2π 2 2 (2 × (85V ) − (77.2V ) ) × 47 Hz = 12.06mF (10.18) Increase the value of C1 // C2 to account for efficiency losses. For this example, 13.6 µF is chosen. 10.6 Current Sense Resistor The ISENSE resistor determines the maximum current output of the power supply. The output current of the power supply is determined by: I OUT = 12 × N tr × I PRI _ PK × TRESET TPERIOD (10.19) When the maximum current output is achieved the voltage seen on the ISENSE pin (VISENSE) should reach its maximum. Thus, at constant current limit: I PRI _ PK = VIsense _ CC RIsense (10.20) 2 × I OUT × RIsense TPERIOD × Ntr TRESET (10.21) During constant current mode, where output current is at its maximum, the first term in Equation 10.21 is constant. Therefore, we can call this KC. Substituting this back into equation 10.21 we get: For iW1692 KC is 0.264 V, therefore RIsense depends on the maximum output current by: RIsense = Ntr × K C × hx 2 × I OUT (10.23) Using this equation and Ntr from section 10.4: RIsense = 13 × 0.264V × 87% = 1.5W 2 × 1A (10.24) We recommend using ±1% tolerance resistors for RIsense. 10.7 Magenitizing Inductance Although the constant current limit does not depend on the magnetizing inductance, there are still restrictions on the magnetizing inductance. The maximum LM is limited by the amount of power that needs to come out of the transformer in order for the power supply to regulate. This is given by: (VIN × TON )2MAX × 40kHz LM _ MAX = 2 × PXFMR _ MAX PXFMR _ MAX (10.25) (VOUT − V fd )× IOUT = hX The minimum LM is limited by the maximum allowable primary peak current (IPRI_PK). 0.9 V on the ISENSE pin should correspond to the maximum allowable primary peak current. Therefore, the maximum primary peak current is: I PRI _ PK < 0.9V RIsense (10.26) Thus, LM is limited by: LM _ MIN = Substituting this into equation 10.19 gives: VIsense _ CC = (10.22) A feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve. ) TPERIOD × KC TRESET (VIN ⋅ TON )MAX 0.9V RIsense (10.27) There is also a lower limit on ISENSE signal of 0.2 V. This gives a second maximum value on LM; compare this with the value obtained from equation 10.25 and pick the smaller of the two values. MK-4008-B 11/20/07 Page 11 iW1692 Low-Power Off-line Digital PWM Controller LM _ MAX = 10.8 Primary Winding 2 2 × PXFMR _ MAX × RIsense (0.2V )2 × 40kHz (10.28) Assuming that the efficiency of the transformer is about 87%, we can obtain the amount of power that needs to come out of the transformer as: POUT .87 = 5W .87 = 5.75W (10.29) Substituting this into equation 10.25 we get: LM _ MAX = (750V ⋅m s )2 × 40kHz 2 × 5.75W = 1.96mH (10.30) To get the minimum value of the primary inductance, use the value for RISENSE from equation 10.24. I PRI _ PK < 0.9V = .6 A 1.5W (10.31) Substituting this primary peak current into equation 10.27: LM _ MIN = 750V ⋅m sec = 1.25mH 0.9V 1.5W (10.32) Choose a primary inductance somewhere between 1.91 mH and 1.42 mH; we chose 1.5 mH. Given a nominal LM, we can now find the minimum turns ratio between primary and secondary that ensures the power supply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF. Under VF mode the constant current IOUT may not be as accurate as in pulse width modulation mode. VOUT _ VF × I OUT × 2 × LM × TPERIOD Ntr _ VF = hx VOUT _ VF × TRESET (10.33) VOUT _ SHUTDOWN × I OUT × 2 × LM × TPERIOD hx 2 × VINAC _ MIN N PRI ≥ (VIN ⋅ TON )MAX BMAX × Ae (10.34) Where: BMAX is maximum flux density and Ae is the crosssectional area of the core. Picking (VIN×TON)MAX to be 750 V·μsec and getting the maximum flux density and core area from the transformer datasheet, we can calculate the minimum number of turns for the primary winding. Substitute BMAX as 320mT and the area of the core to be 19.2 mm2 we solve equation 10.35 to get: N PRI ≥ 750V ⋅m s 320mT × 19.2mm 2 = 122.1turns (10.36) To avoid hitting the maximum flux density, pick a value for NPRI to be higher than this. In this example 144 turns is picked. 10.9 Secondary Winding From the primary winding turns, we obtain the secondary winding. N SEC . = N PRI Ntr (10.37) Thus, in our example: N SEC = 144 = 11 turns 13 (10.38) At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind. VCC is the supply to the iW1692 and should be between 12 V and 16 V. Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure it’s within range. The zener, Z1 protects the IC from getting a VCC over voltage. Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V. N BIAS = (10.35) 10.10 Bias Winding and: TON = In order to keep the transformer from saturation, the maximum flux density must not be exceeded. Therefore the minimum primary winding on the transformer must meet: MK-4008-B 11/20/07 ( N SEC × VCC + V fd VOUT ) (10.39) Page 12 iW1692 Low-Power Off-line Digital PWM Controller The number of auxiliary windings can be calculated using equation 10.39. N BIAS = 11turns × 12.5V = 25turns 5.5V (10.40) Here we’ve actually chosen a lower number for the bias winding, 22 turns. 10.11 VSENSE Resistors and Winding The output voltage regulation is mainly determined by the feedback signal VSENSE. VSENSE = VOUT _ PCB × K SENSE (10.41) Where: K SENSE = N R4 × Vsense R4 + R3 N SEC (10.42) Internally, VSENSE is compared to a reference voltage VSENSE(NOM). From equation 10.41 we get: K SENSE 1.538V = = 0.3 5V − 0V (10.43) Solving for R4 in equation 10.42 assuming R3 is 20 kΩ, and NVSENSE is 24 turns we get R4 should be around 3 kΩ. Since the iW1692 uses the VSENSE signal to determine the regulation point, this signal can not be too noisy. Thus, C8 is used to help filter the VSENSE signal. 10.12 Output Capacitors The output capacitors are important for controlling the output voltage ripple of the power supply. This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus, how much charge is lost before the next switching cycle is the ripple on the output voltage. Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then: COUT = QOUT VOUT _ RIPPLE _ PK (10.44) Since charge is equal to current times time. COUT = I OUT × TOFF _ MAX VOUT _ RIPPLE _ PK (10.45) Assuming we want to get under 50 mV of ripple on the output, we substitute this into equation 10.45 to get: COUT = 1A × 25m sec = 500mF 50mV (10.46) In this calculations ESR and ESL are ignored; the reason this calculation is still valid is because of the second stage LC filter, L3 and C11. These two components reduce the ESR and ESL ripple. 10.13 Snubber Network The snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive. The goal is to dissipate the energy from the leakage inductance of the transformer. For simplicity and a more conservative design first assume the energy of the leakage inductance is only dissipated through the snubber. Thus: 1 2 × Llk 2 × I pri _ pk = 1 2 2 × C3 × V pk 2 − Vval (10.47) LLK can be measured from the transformer, IPRI_PK is 0.9 V divided by RISENSE, and VPK is the peak VDS of the MOSFET. Choose C3, keeping in mind that the larger the value of C3 you choose, the lower the voltage stress is that is applied to the MOSFET. However, capacitors are more expensive the larger their capacitance. Choose C3 based on these two criteria and select VPK and VVAL. Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver. The dissipation of this resistor is given by: −Tperiod Vval R5 ⋅C3 =e V pk (10.48) Using equation 10.48 solve for R5. This will give a conservative estimate of what C3 and R5 should be. Included in the snubber network is also a resistor (R6) in series with the diode (D6). D6 directs the current to C3 when the MOSFET is turned off; however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on. This reverse current occurs because there is a short period of time when the diode still conducts after switching from forward biased to reverse biased. This conduction will distort the falling edge of the VSENSE curve and affect the operation of the IC. So, the resistor, R6, is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on. The maximum off-time during maximum output current is 25 μs. MK-4008-B 11/20/07 Page 13 iW1692 Low-Power Off-line Digital PWM Controller 10.14 ON-Time Delay Filter iW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves. The mismatch in high line and low line curves is due to the delay from the IC propagation delay, driver turn on delay, and the MOSFET actually turn on delay. The driver turn on delay is further increased by R9; this is to lessen the amount of stress on the MOSFET during turn on. To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility. R15 and C5 provide extra delay in the circuit to tweak the compensation. To determine values R15 and C5 follow these steps: 1. Measure the difference between high line and low line constant current limit without R15 and C5. 2. Find the curve that best matches this difference from Figure 11.0.7. 3. Find the LM that matches the power supply. Match the tRC. 4. Find R15 and C5 from equation 10.49: t RC = R15 × C5 (10.49) We observe that the difference between high line and low line constant current limit is 20 mA. Matching the primary inductance 2 mH and the curve, we find tRC to be 4.4×10-8 s. We then pick R15 to be 1 kΩ and substitute into equation 10.50. 4.4 ⋅10−8 sec = 1k W × C5 (10.50) Solving for C5 in equation 10.54, we get 44 pF. The result should be a match between high line and low line constant current curves. See figure 11.0.7 for details. 10.15 PCB Layout In the iW1692, there are two signals that are important to control output performance; these are the ISENSE signal and the VSENSE signal. The ISS resistor should be close to the source of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal. Also the ISENSE signal should be placed close to the ISENSE pin. The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal. MK-4008-B 11/20/07 Page 14 iW1692 Low-Power Off-line Digital PWM Controller 11.0 Design Example Performance Characteristics 74 Efficiency (%) 70 66 64 60 56 52 90 VAC 264 VAC 0 250 500 750 Output Current (mA) 1000 Figure 11.0.1 VSENSE Efficiency at 90 VAC and 264 VAC Figure 11.0.4 VSENSE Short Before Start-up (no load) Figure 11.0.2 Regulation without Cable Drop Compensation Figure 11.0.5 ISENSE Short at 90 VAC Figure 11.0.3 Regulation with Cable Drop Compensation Figure 11.0.6 output Short Fault (50% load) MK-4008-B 11/20/07 Page 15 iW1692 Low-Power Off-line Digital PWM Controller 11.0 Design Example Performance Characteristics 180 50 mA 40 mA 30 mA 20 mA 10 mA (R15 x C5), τRC (ns) 150 120 90 60 30 0 0 0.75 1.50 2.25 Magnetizing Inductance LM (mH) 3.0 Figure 11.0.7. TON Compensation Chart 12.0 Application Circuit F1 10 Ω D1 - D4 IN4007 L L1 1 mH N C1 6.8 µF/400 V R9 100 Ω Q1 R15 1 kΩ + + R1 4.7 kΩ C2 6.8 µF/400 V R13 56 kΩ OUTPUT 5 ISENSE R11 2.2 MΩ R6 150Ω C5 47 pF + + R14 560Ω 5V/1A RTN C11 330 µF/10V D6 R16 100 Ω R3 20kΩ VSENSE 1 GND C3 1 nF/500 V C10 650 µF/10 V VCC 4 VIN 6 2 R18 1.5 Ω R5 100 kΩ R10 2.43 MΩ U1 iW1692 3 L3 1µH T1-A Z1 15V R4 C7 C6 C8 3.0 kΩ 68 pF 470 pF 470 nF C9 4.7µF T1-B D5 FR102 R12 1Ω Figure 12.0.1. Typical Application Circuit MK-4008-B 11/20/07 Page 16 iW1692 Low-Power Off-line Digital PWM Controller 13.0 Physical Dimensions 6-Lead Small Outline Transistor Package 5 6 4 Symbol D MIN MAX A - 1.45 A1 0.00 0.15 A2 0.90 1.30 B 0.30 0.50 C D 0.08 0.22 2.90 BSC 2.80 3.00 E E1 1 2 3 e e1 A1 A2 B COPLANARITY 0.10 A SEATING PLANE α E 2.80 BSC E1 1.65 BSC e 0.95 BSC e1 L C Millimeters 1.90 BSC L 0.30 0.60 α 0° 8° Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters Figure 13.0.1. Physical dimensions, 6-lead SOT-23 package 14.0 Ordering Information Part Number Mark Option Package Operating Temp. Range Description iW1692-00 Cxxx Cable Drop Compensation, 0 mV SOT23-6L -40°C ≤ TA ≤ 85°C Tape & Reel1 iW1692-30 Dxxx Cable Drop Compensation, 300 mV SOT23-6L -40°C ≤ TA ≤ 85°C Tape & Reel1 Note 1: Tape & Reel packing quantity is 3,000 units. Note 2: In the mark column, “xxx” represents the lot ID code. Refer to ILG-005 device marking specification for more detailed information. MK-4008-B 11/20/07 Page 17 iW1692 Low-Power Off-line Digital PWM Controller About iWatt iWatt Inc. is a fabless semiconductor company that develops intelligent power management ICs for computer, communication, and consumer markets. The company’s patented pulseTrain™ technology, the industry’s first truly digital approach to power system regulation, is revolutionizing power supply design. Trademark Information © 2007 iWatt, Inc. All rights reserved. iWatt, the iW light bulb, and pulseTrain are trademarks of iWatt, Inc. All other trademarks and registered trademarks are the property of their respective companies. Contact Information Web: http://www.iwatt.com E-mail: [email protected] Phone: 408-374-4200 Fax: 408-341-0455 iWatt Inc. 101 Albright Way Los Gatos CA 95032-1827 Disclaimer iWatt reserves the right to make changes to its products and to discontinue products without notice. The applications information, schematic diagrams, and other reference information included herein is provided as a design aid only and are therefore provided as-is. iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights. Certain applications using semiconductor products may involve potential risks of death, personal injury, or severe property or environmental damage (“Critical Applications”). iWatt SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED, OR WARRANTED TO BE SUITABLE FOR USE IN LIFE‑SUPPORT APPLICATIONS, DEVICES OR SYSTEMS, OR OTHER CRITICAL APPLICATIONS. Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer. Questions concerning potential risk applications should be directed to iWatt, Inc. iWatt semiconductors are typically used in power supplies in which high voltages are present during operation. High-voltage safety precautions should be observed in design and operation to minimize the chance of injury. MK-4008-B 11/20/07 Page 18