MIC2103/04

MIC2103/04
75V, Synchronous Buck Controllers
featuring Adaptive On-Time Control
Hyper Speed Control™ Family
General Description
Features
The
Micrel
MIC2103/04
are
constant-frequency,
synchronous buck controllers featuring a unique adaptive
ON-time control architecture. The MIC2103/04 operates
over an input supply range from 4.5V to 75V and can be
used to supply up to 15A of output current. The output
voltage is adjustable down to 0.8V with a guaranteed
accuracy of ±1%. The device operates with programmable
switching frequency from 200kHz to 600kHz.
®
Micrel’s Hyper Light Load architecture provides the same
high-efficiency and ultra fast transient response as the Hyper
Speed Control architecture under the medium to heavy loads,
but also maintains high efficiency under light load conditions
by transitioning to variable frequency, discontinuous-mode
operation.
The MIC2103/04 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include under-voltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, fold-back current limit, “hiccup” mode shortcircuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
•
•
•
•
•
•
•
•
•
•
•
•
•
Hyper Speed Control™ architecture enables
- High delta V operation (VIN = 75V and VOUT = 1.2V)
- Any Capacitor™ stable
4.5V to 75V input voltage
Adjustable output voltage from 0.8 V to 24V (also
limited by duty cycle)
200kHz to 600kHz, programmable switching frequency
Hyper Light Load Control (MIC2103 only)
Hyper Speed Control (MIC2104 only)
Enable input, Power-Good output
Built-in 5V regulator for single-supply operation
Programmable current limit and fold-back “hiccup”
mode short-circuit protection
5ms internal soft-start, internal compensation, and
thermal shutdown
Supports safe start-up into a pre-biased output
–40°C to +125°C junction temperature range
®
Available in 16-pin 3mm × 3mm MLF package
Applications
•
•
•
Distributed power systems
Networking/Telecom Infrastructure
Printers, scanners, graphic cards and video cards
_________________________________________________________________________________________________________________________
Typical Application
VIN
4.5V to 75V
2.2µF
x3
100µF
Efficiency (VIN = 48V)
vs. Output Current (MIC2103)
100
90
VIN
FREQ
1µF
BST
VDD
DH
AGND
SW
0.1µF
VOUT
5V/10A
6.1µH
95.3k
EN
2.2nF
0.1µF
PG
PG
PGND
FB
ILIM
470µF
DL
10k
100µF
EN
EFFICIENCY (%)
1µF
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
MIC2103/04
PVDD
1.91k
2.21k
70
60
50
40
30
fSW = 200kHz (CCM)
20
10
0
0
1
2
3
4
5
6
7
8
9 10 11 12 13 14
OUTPUT CURRENT (A)
Hyper Speed Control, Hyper Light Load, and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
November 26, 2013
Revision 2.0
Micrel, Inc.
MIC2103/04
Ordering Information
Part Number
Switching
Frequency
Features
Package
Junction
Temperature Range
Lead Finish
MIC2103YML
200kHz to 600kHz
Hyper Light Load
16-pin 3mm x 3mm MLF
–40°C to +125°C
Pb-Free
MIC2104YML
200kHz to 600kHz
Hyper Speed Control
16-pin 3mm x 3mm MLF
–40°C to +125°C
Pb-Free
Pin Configuration
16-Pin 3mm x 3mm MLF (ML)
(TOP VIEW)
Pin Description
Pin Number
Pin Name
1
VDD
Internal +5V linear regulator output. VDD is the internal supply bus for the device. A 1μF ceramic
capacitor from VDD to AGND is required for decoupling. In the applications with VIN<+5.5V, VDD
should be tied to VIN to by-pass the linear regulator.
2
PVDD
5V supply input for the low-side N-channel MOSFET driver, which can be tied to VDD externally.
A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling.
3
ILIM
Current Limit Setting. Connect a resistor from SW to ILIM to set the over-current threshold for the
converter.
DL
Low-Side Drive output. High-current driver output for external low-side MOSFET of a buck
converter. The DL driving voltage swings from ground to VDD. Adding a small resistor between DL
pin and the gate of the low-side N-channel MOSFET can slow down the turn-on and turn-off
speed of the MOSFET.
5
PGND
Power Ground. PGND is the return path for the buck converter power stage. The PGND pin
connects to the sources of low-side N-Channel external MOSFET, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The return path for the power ground
should be as small as possible and separate from the Signal ground (AGND) return path.
6
FREQ
Switching Frequency Adjust input. Tie this pin to VIN to operate at 600kHz and place a resistor
divider to reduce the frequency.
DH
High-Side Drive output. High-current driver output for external high-side MOSFET of a buck
converter. The DH driving voltage is floating on the switch node voltage (VSW). Adding a small
resistor between DH pin and the gate of the high-side N-channel MOSFET can slow down the
turn-on and turn-off speed of the MOSFET.
SW
Switch Node and Current-Sense input. High current output driver return. The SW pin connects
directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be
routed away from sensitive nodes. The SW pin also senses the current by monitoring the voltage
across the low-side MOSFET during OFF time. In order to sense the current accurately, connect
the low-side MOSFET drain to the SW pin using a Kelvin connection.
4
7
8
November 26, 2013
Pin Function
2
Revision 2.0
Micrel, Inc.
MIC2103/04
Pin Description (Continued)
Pin Number
Pin Name
9, 11
NC
No connection.
10
BST
Voltage Supply Pin input for the high-side N-channel MOSFET driver, which can be powered by a
bootstrapped circuit connected between VDD and SW, using a Schottky diode and a 0.1μF
ceramic capacitor. Adding a small resistor at BST pin can slow down the turn-on speed of the
high-side MOSFET.
12
AGND
Signal ground for VDD and the control circuitry, which is connected to Thermal Pad electronically.
The signal ground return path should be separate from the power ground (PGND) return path.
13
FB
Feedback input. Input to the transconductance amplifier of the control loop. The FB pin is
regulated to 0.8V. A resistor divider connecting the feedback to the output is used to set the
desired output voltage.
14
PG
Power Good output. Open Drain Output, an external pull-up resistor to VDD or external power
rails is required.
15
EN
Enable input. A logic signal to enable or disable the buck converter operation. The EN pin is
CMOS compatible. Logic high enables the device, logic low shutdowns the regulator. In the
disable mode, the VDD supply current for the device is minimized to 0.7mA typically.
16
VIN
Supply voltage. The VIN operating voltage range is from 4.5V to 75V. A 1μF ceramic capacitor
from VIN to AGND is required for decoupling.
EP
ePad
November 26, 2013
Pin Function
Exposed Pad. Connect the EPAD to PGND plain on the PCB to improve the thermal
performance.
3
Revision 2.0
Micrel, Inc.
MIC2103/04
Absolute Maximum Ratings(1)
Operating Ratings(3)
VIN ................................................................ –0.3V to +76V
VDD, VPVDD ........................................................ –0.3V to +6V
VFREQ, VILIM, VEN .................................... −0.3V to (VIN +0.3V)
VSW ............................................... (DC) −0.3V to (VIN +0.3V)
VSW ............................................ (Transient ) −5.0V <100ns
VBST to VSW ........................................................ −0.3V to 6V
VBST ................................................................ −0.3V to 82V
VPG ..................................................... −0.3V to (VDD + 0.3V)
VFB .................................................................................. −0.3V to (VDD + 0.3V)
PGND to AGND............................................ −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS) ......................... −65°C to +150°C
Lead Temperature (soldering, 10sec) ........................ 260°C
(2)
ESD Rating ................................................. ESD Sensitive
Supply Voltage (VIN) .......................................... 4.5V to 75V
Enable Input (VEN) .................................................. 0V to VIN
VSW , VFEQ, VILIM, VEN ............................................... 0V to VIN
Junction Temperature (TJ) ........................ −40°C to +125°C
Junction Thermal Resistance
3mm × 3mm MLF-16 (θJA) .................................... 50.8°C/W
3mm × 3mm MLF-16 (θJC) ................................... 25.3°C/W
Electrical Characteristics(4)
VIN = 48V, VOUT = 5V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
75
V
Power Supply Input
(5)
4.5
Input Voltage Range (VIN)
Quiescent Supply Current (MIC2103)
VFB = 1.5V
400
750
µA
Quiescent Supply Current (MIC2104)
VFB = 1.5V
2.1
3
mA
Shutdown Supply Current
SW unconnected, VEN = 0V
0.1
10
µA
VDD Supply
VDD Output Voltage
VIN = 7V to 75V, IDD = 10mA
4.8
5.2
5.4
V
VDD UVLO Threshold
VDD rising
3.8
4.2
4.6
V
VDD UVLO Hysteresis
Load Regulation
400
IDD = 0 to 40mA
mV
0.6
2
3.6
%
TJ = 25°C (±1.0%)
0.792
0.8
0.808
-40°C ≤ TJ ≤ 125°C (±2%)
0.784
0.8
0.816
5
500
Reference
Feedback Reference Voltage
FB Bias Current
VFB = 0.8V
V
nA
Notes:
1.
Exceeding the absolute maximum rating may damage the device.
2.
Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3.
The device is not guaranteed to function outside operating range.
4.
Specification for packaged product only.
5.
The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH.
November 26, 2013
4
Revision 2.0
Micrel, Inc.
MIC2103/04
Electrical Characteristics(4) (Continued)
VIN = 48V, VOUT = 5V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Enable Control
1.8
EN Logic Level High
V
0.6
EN Logic Level Low
EN Hysteresis
EN Bias Current
200
VEN = 48V
V
mV
23
40
600
750
µA
Oscillator
Switching Frequency
400
VFREQ = VIN
VFREQ = 50%VIN
300
Maximum Duty Cycle
Minimum Duty Cycle
VFB > 0.8V
Minimum Off-Time
140
kHz
85
%
0
%
200
260
ns
Soft Start
Soft-Start time
5
ms
Short Circuit Protection
Current-Limit Threshold
VFB = 0.79V
-30
-14
0
mV
Short-Circuit Threshold
VFB = 0V
-23
-7
9
mV
Current-Limit Source Current
VFB = 0.79V
60
80
100
µA
Short-Circuit Source Current
VFB = 0V
27
36
47
µA
0.1
V
FET Drivers
DH, DL Output Low Voltage
ISINK = 10mA
DH, DL Output High Voltage
ISOURCE = 10mA
VPVDD - 0.1V
or
V
VBST - 0.1V
DH On-Resistance, High State
2.1
3.3
Ω
DH On-Resistance, Low State
1.8
3.3
Ω
DL On-Resistance, High State
1.8
3.3
Ω
DL On-Resistance, Low State
1.2
2.3
Ω
50
µA
SW, BST Leakage Current
November 26, 2013
5
Revision 2.0
Micrel, Inc.
MIC2103/04
Electrical Characteristics(4) (Continued)
VIN = 48V, VOUT = 5V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
85
90
95
Units
Power Good
%VOU
Power Good Threshold Voltage
Sweep VFB from Low to High
Power Good Hysteresis
Sweep VFB from High to Low
6
Power Good Delay Time
Sweep VFB from Low to High
100
Power Good Low Voltage
VFB < 90% x VNOM, IPG = 1mA
70
TJ Rising
160
°C
4
°C
T
%VOU
T
µs
200
mV
Thermal Protection
Over-Temperature Shutdown
Over-Temperature Shutdown Hysteresis
November 26, 2013
6
Revision 2.0
Micrel, Inc.
MIC2103/04
Typical Characteristics
1.0%
1.60
1.20
0.80
0.808
0.8%
0.6%
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
VOUT = 5.0V
VOUT = 5V
-0.8%
IOUT = 0A to 10A
IOUT = 0A
-1.0%
0.40
0.00
INPUT VOLTAGE (V)
10 15 20 25 30 35 40 45 50 55 60 65 70 75
INPUT VOLTAGE (V)
Feedback Voltage
vs. Temperature (MIC2103)
5.015
5.010
5.005
5.000
VOUT = 5V
0.808
FEEBACK VOLTAGE (V)
SUPPLY CURRENT (mA)
5.020
1.60
1.20
0.80
VIN = 48V
0.40
IOUT = 0A
-50
-25
0.792
0.3%
0.6%
0.5%
0.4%
0.2%
0.1%
0.0%
VIN = 48V
VOUT = 5.0V
IOUT = 0A to 10A
November 26, 2013
50
75
100
-50
125
-25
0
100
125
25
50
75
100
125
TEMPERATURE (°C)
Feedback Voltage
vs. Output Current (MIC2103)
0.808
0.3%
0.2%
0.1%
0.0%
-0.1%
-0.2%
-0.3%
-0.4%
VIN = 12V to 75V
0.804
0.800
0.796
VIN = 48V
VOUT = 5.0V
-0.5%
-0.6%
-0.3%
TEMPERATURE (°C)
25
FEEDBACK VOLTAGE (V)
0.8%
0.7%
LINE REGULATION (%)
0.4%
75
0
Line Regulation
vs. Temperature (MIC2103)
Load Regulation
vs. Temperature (MIC2103)
50
VIN = 48V
TEMPERATURE (°C)
INPUT VOLTAGE (V)
25
0.796
IOUT = 0A
IOUT = 0A
10 15 20 25 30 35 40 45 50 55 60 65 70 75
0
0.800
VOUT = 5.0V
0.00
-25
0.804
VOUT = 5.0V
4.990
-50
VOUT = 5.0V
0.792
2.00
-0.2%
0.796
VIN Operating Supply Current
vs. Temperature (MIC2103)
5.025
-0.1%
0.800
INPUT VOLTAGE (V)
Output Voltage
vs. Input Voltage (MIC2103)
4.995
0.804
IOUT = 0A
10 15 20 25 30 35 40 45 50 55 60 65 70 75
10 15 20 25 30 35 40 45 50 55 60 65 70 75
OUTPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
TOTAL REGULATION (%)
SUPPLY CURRENT (mA)
2.00
LOAD REGULATION (%)
Feedback Voltage
vs. Input Voltage (MIC2103)
Output Regulation
vs. Input Voltage (MIC2103)
VIN Operating Supply Current
vs. Input Voltage (MIC2103)
VOUT = 5.0V
IOUT = 0A
fSW = 200kHz
0.792
-50
-25
0
25
50
75
TEMPERATURE (°C)
7
100
125
0
1
2
3
4
5
6
7
8
9
10
OUTPUT CURRENT (A)
Revision 2.0
Micrel, Inc.
MIC2103/04
Typical Characteristics (Continued)
Line Regulation
vs. Output Current (MIC2103)
100
100
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
0.2%
80
EFFICIENCY (%)
0.1%
0.0%
-0.1%
60
50
40
30
1
0
3
2
6
5
4
7
9
8
1
2
3
OUTPUT CURRENT (A)
5
6
7
8
9 10 11 12 13 14
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
70
60
50
40
30
20
fSW = 200kHz (CCM)
0
1
2
3
4
5
6
7 8
1
2
9 10 11 12 13 14
70
60
50
40
30
4
5
6
7
8
9 10 11 12 13 14
Efficiency (VIN = 48V)
vs. Output Current (MIC2103)
100
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
70
60
50
40
30
20
10
fSW = 200kHz (CCM)
fSW = 200kHz (CCM)
10
0
0
1
2
3
4
5
6
7
8
9 10 11 12 13 14
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
3
90
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
0
0
fSW = 200kHz (CCM)
OUTPUT CURRENT (A)
20
10
30
0
Efficiency (VIN = 38V)
vs. Output Current (MIC2103)
100
EFFICIENCY (%)
EFFICIENCY (%)
4
90
80
40
OUTPUT CURRENT (A)
Efficiency (VIN = 24V)
vs. Output Current (MIC2103)
90
50
0
0
10
60
10
0
-0.3%
70
20
fSW = 200kHz (CCM)
10
VOUT = 5.0V
100
80
20
VIN = 12V to 75V
-0.2%
70
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
90
EFFICIENCY (%)
LINE REGULATION (%)
90
EFFICIENCY (%)
0.3%
Efficiency (VIN = 18V)
vs. Output Current (MIC2103)
Efficiency (VIN =12V)
vs. Output Current (MIC2103)
0
1
2
3
4
5
6
7
8
9 10 11 12 13 14
OUTPUT CURRENT (A)
Efficiency (VIN = 75V)
vs. Output Current (MIC2103)
100
90
EFFICIENCY (%)
80
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
70
60
50
40
30
20
fSW = 200kHz (CCM)
10
0
0
1
2
3
4
5
6
7
8
9 10 11 12 13 14
OUTPUT CURRENT (A)
November 26, 2013
8
Revision 2.0
Micrel, Inc.
MIC2103/04
Typical Characteristics (Continued)
VIN Operating Supply Current
vs. Input Voltage (MIC2104)
40
30
20
VOUT = 5V
10
IOUT = 0A
0.808
0.804
0.800
VOUT = 5.0V
0.796
fSW = 200kHz
0.4%
0.2%
0.0%
-0.2%
-0.4%
VOUT = 5.0V
-0.6%
IOUT = 0A to 10A
-0.8%
fSW = 200kHz
-1.0%
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
28
24
20
16
VIN = 48V
VOUT = 5.0V
IOUT = 0A
4
0.4%
0.0%
0.3%
-0.2%
0.2%
0.1%
0.0%
VIN = 48V
-0.1%
0
25
50
75
100
IOUT = 0A to 10A
fSW = 200kHz
-0.4%
-0.6%
-0.8%
-1.0%
-1.2%
VIN = 12V to 75V
-1.4%
VOUT = 5.0V
-1.6%
-0.3%
fSW = 200kHz
-25
VOUT = 5.0V
-0.2%
-50
0
-50
Line Regulation
vs. Temperature (MIC2104)
LINE REGULATION (%)
LOAD REGULATION (%)
32
8
INPUT VOLTAGE (V)
Load Regulation
vs. Temperature (MIC2104)
36
12
fSW = 200kHz
10 15 20 25 30 35 40 45 50 55 60 65 70 75
10 15 20 25 30 35 40 45 50 55 60 65 70 75
40
SUPPLY CURRENT (mA)
0.6%
IOUT = 0A
10 15 20 25 30 35 40 45 50 55 60 65 70 75
VIN Operating Supply Current
vs. Temperature (MIC2104)
-25
0
25
50
75
100
125
TEMPERATURE (°C)
125
IOUT = 0A
-1.8%
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
Feedback Voltage
vs. Output Current (MIC2104)
Line Regulation
vs. Output Current (MIC2104)
0.0%
LINE REGULATION (%)
0.808
FEEDBACK VOLTAGE (V)
0.8%
0.792
0
Output Regulation
vs. Input Voltage (MIC2104)
1.0%
TOTAL REGULATION (%)
0.812
FEEDBACK VOLTAGE (V)
SUPPLY CURRENT (mA)
50
Feedback Voltage
vs. Input Voltage (MIC2104)
0.804
0.800
0.796
VIN = 48V
VOUT = 5.0V
-0.1%
-0.2%
-0.3%
-0.4%
VIN = 12V to 75V
-0.5%
VOUT = 5.0V
fSW = 200kHz
fSW = 200kHz
0.792
-0.6%
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
November 26, 2013
9
10
0
1
2
3
4
5
6
7
8
9
10
OUTPUT CURRENT (A)
9
Revision 2.0
Micrel, Inc.
MIC2103/04
Typical Characteristics (Continued)
60
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
50
40
30
20
70
60
50
40
30
10
fSW = 200kHz
0
1
2
3
4
5
6
7
8
9 10 11 12 13 14
30
1
2
3
4
5
6
7
8
0
9 10 11 12 13 14
100
100
60
50
EFFICIENCY (%)
70
40
30
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
70
60
50
40
30
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
DIE TEMPERATURE (°C)
60
40
VIN = 12V
VOUT = 5.0V
20
3
4
5
6
7
OUTPUT CURRENT (A)
40
30
fSW = 200kHz
0
1
2
8
9
3
4
5
6
7
8
Die Temperature* (VIN = 75V)
vs. Output Current
120
100
80
60
40
VIN = 48V
VOUT = 5.0V
20
120
100
80
60
40
VIN = 75V
VOUT = 5.0V
20
fSW = 200kHz
10
9 10 11 12 13 14
OUTPUT CURRENT (A)
fSW = 200kHz
0
2
50
140
fSW = 200kHz
0
1
60
9 10 11 12 13 14
140
80
9 10 11 12 13 14
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
Die Temperature* (VIN = 48V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
100
8
70
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
120
7
0
0
9 10 11 12 13 14
6
10
fSW = 200kHz
0
0
5
20
10
fSW = 200kHz
4
80
20
20
3
90
80
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
1
2
Efficiency (VIN = 75V)
vs. Output Current (MIC2104)
Efficiency (VIN = 48V)
vs. Output Current (MIC2104)
EFFICIENCY (%)
80
0
1
OUTPUT CURRENT (A)
90
10
fSW = 200kHz
OUTPUT CURRENT (A)
90
EFFICIENCY (%)
40
0
0
Efficiency (VIN = 38V)
vs. Output Current (MIC2104)
100
DIE TEMPERATURE (°C)
50
10
fsw = 200kHz
OUTPUT CURRENT (A)
0
60
0
0
140
70
20
20
10
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
EFFICIENCY (%)
70
90
90
DIE TEMPERATURE (°C)
EFFICIENCY (%)
80
Efficiency (VIN = 24V)
vs. Output Current (MIC2104)
100
100
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
90
EFFICIENCY (%)
100
Efficiency (VIN = 18V)
vs. Output Current (MIC2104)
Efficiency (VIN =12V)
vs. Output Current (MIC2104)
0
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
9
10
0
1
2
3
4
5
6
7
8
9
OUTPUT CURRENT (A)
* Case Temperature: The temperature measurement was taken at the hottest point on the MIC2103 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.
November 26, 2013
10
Revision 2.0
10
Micrel, Inc.
MIC2103/04
Typical Characteristics (Continued)
VIN Shutdown Current
vs. Input Voltage
Enable Threshold
vs. Input Voltage
1.50
Rising
540
480
420
360
300
240
180
IDD = 10mA
6
4
IDD = 40mA
120
VOUT = 5.0V
2
60
VEN = 0V
Falling
0.90
0.60
Hyst
0.30
0.00
0
10 15 20 25 30 35 40 45 50 55 60 65 70 75
10 15 20 25 30 35 40 45 50 55 60 65 70 75
10 15 20 25 30 35 40 45 50 55 60 65 70 75
Output Peak Current Limit
vs. Input Voltage
25
CURRENT LIMIT (A)
300
260
220
180
20
15
10
5
VOUT = 5.0V
VOUT = 5.0V
IOUT = 2A
250
25°C
200
-40°C
0
INPUT VOLTAGE (V)
Feedback Voltage
vs. Temperature
Output Peak Current Limit
vs. Temperature
SHUTDOWN CURRENT (uA)
CURRENT LIMIT (A)
15
12
9
6
VIN =48V
VOUT = 5.0V
3
VOUT = 5.0V
-25
0
25
50
75
TEMPERATURE (°C)
November 26, 2013
100
125
240
160
VIN =48V
80
VEN = 0V
IOUT = 0A
0
0
-50
10
320
fSW = 200kHz
IOUT = 0A
0.792
8
400
VIN = 48V
0.796
6
VIN Shutdown Current
vs. Temperature
18
0.800
4
OUTPUT CURRENT (A)
21
0.812
0.804
2
10 15 20 25 30 35 40 45 50 55 60 65 70 75
INPUT VOLTAGE (V)
0.808
VIN = 48V
VOUT = 5.0V
0
10 15 20 25 30 35 40 45 50 55 60 65 70 75
125°C
150
100
fSW = 200kHz
100
Switching Frequency
vs. Output Current
300
SWITCHING FREQUENCY (kHz)
Switching Frequency
vs. Input Voltage
140
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
1.20
fSW = 200kHz
0
FEEBACK VOLTAGE (V)
ENABLE THRESHOLD (V)
8
VDD VOLTAGE (V)
SHUTDOWN CURRENT (uA)
600
VDD Voltage
vs. Input Voltage
10
-50
-25
0
25
50
75
TEMPERATURE (°C)
11
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
Revision 2.0
Micrel, Inc.
MIC2103/04
Typical Characteristics (Continued)
VDD UVLO Threshold
vs. Temperature
4.4
5.5
4.3
5.0
4.2
4.5
4.0
IDD = 40mA
IDD = 10mA
3.5
3.0
2.5
2.0
1.5
1.21
Rising
1.11
1.01
4.1
0.91
4.0
Falling
3.9
0.81
3.8
0.71
3.7
0.61
3.6
0.51
1.0
VIN = 48V
3.5
VIN =48V
0.5
IOUT = 0A
3.4
IOUT = 0A
0.0
0.41
3.3
-50
-25
0
25
50
75
100
125
0.31
-50
0
-25
TEMPERATURE (°C)
50
25
75
100
125
TEMPERATURE (°C)
EN Bias Current
vs. Temperature
-50
-25
0
25
50
75
100
125
TEMPERATURE (C)
Enable Threshold
vs. Temperature
1.5
100
Rising
1.4
80
60
40
20
VIN =48V
VEN = 0V
0
-50
-25
0
25
50
75
TEMPERATURE (°C)
100
125
ENABLE THRESHOLD (V)
EN BIAS CURRENT (µA)
PG Threshold/VREF Ratio vs.
Temperature
PG THRESHOLD (V)
6.0
VDD THRESHOLD (V)
VDD Voltage (V)
VDD Voltage
vs. Temperature
1.3
Falling
1.2
1.1
1.0
0.9
0.8
0.7
VIN = 48V
0.6
0.5
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
November 26, 2013
12
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Characteristics
November 26, 2013
13
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Characteristics (Continued)
November 26, 2013
14
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Characteristics (Continued)
November 26, 2013
15
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Characteristics (Continued)
November 26, 2013
16
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Characteristics (Continued)
November 26, 2013
17
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Diagram
Figure 1. MIC2103/04 Functional Diagram
November 26, 2013
18
Revision 2.0
Micrel, Inc.
MIC2103/04
Functional Description
The MIC2103/04 are adaptive on-time synchronous buck
controllers built for high-input voltage to low-output
voltage conversion applications. They are designed to
operate over a wide input voltage range, from 4.5V to
75V, and the output is adjustable with an external
resistive divider. An adaptive on-time control scheme is
employed to obtain a constant switching frequency and
to simplify the control compensation. Over-current
protection is implemented by sensing low-side
MOSFET’s RDS(ON). The device features internal softstart, enable, UVLO, and thermal shutdown.
The maximum duty cycle is obtained from the 200ns
tOFF(min):
DMAX =
VOUT
VIN × f SW
Eq. 1
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
200ns, the MIC2103/04 control logic will apply the
tOFF(min) instead. TOFF(min) is required to maintain enough
energy in the boost capacitor (CBST) to drive the highside MOSFET.
November 26, 2013
tS
= 1−
200ns
tS
Eq. 2
where tS = 1/fSW . It is not recommended to use
MIC2103/04 with a OFF-time close to tOFF(min) during
steady-state operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC2103/04. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC2103/04 control loop timing
during steady-state operation. During steady-state, the
gm amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple plus injected
voltage ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Theory of Operation
Figure 1 illustrates the block diagram of the MIC2103/04.
The output voltage is sensed by the MIC2103/04
feedback pin FB via the voltage divider R1 and R2, and
compared to a 0.8V reference voltage VREF at the error
comparator through a low-gain transconductance (gm)
amplifier. If the feedback voltage decreases and the
amplifier output is below 0.8V, thenthe error comparator
will trigger the control logic and generate an ON-time
period. The ON-time period length is predetermined by
the “Fixed tON Estimator” circuitry:
t ON(estimated) =
t S − t OFF(MIN)
19
Revision 2.0
Micrel, Inc.
MIC2103/04
Unlike true current-mode control, the MIC2103/04 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC2103/04 feedback voltage ripple should be in phase
with the inductor current ripple and are large enough to
be sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV over full input voltage range. If a low ESR
output capacitor is selected, then the feedback voltage
ripple may be too small to be sensed by the gm amplifier
and the error comparator. Also, the output voltage ripple
and the feedback voltage ripple are not necessarily in
phase with the inductor current ripple if the ESR of the
output capacitor is very low. In these cases, ripple
injection is required to ensure proper operation. Please
refer to “Ripple Injection” subsection in Application
Information for more details about the ripple injection
technique.
Figure 2. MIC2103/04 Control Loop Timing
Figure 3a shows the operation of the MIC2103/04 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC2103/04
converter.
Discontinuous Mode (MIC2103 only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads, the MIC2103
is able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 3b. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC2103 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
The MIC2103 has a zero crossing comparator (ZC
Detection) that monitors the inductor current by sensing
the voltage drop across the low-side MOSFET during its
ON-time. If the VFB > 0.8V and the inductor current goes
slightly negative, then the MIC2103 automatically
powers down most of the IC circuitry and goes into a
low-power mode.
Once the MIC2103 goes into discontinuous mode, both
LSD and HSD are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 3b shows the control loop timing in
discontinuous mode.
Figure 3a. MIC2103/04 Load Transient Response
November 26, 2013
20
Revision 2.0
Micrel, Inc.
MIC2103/04
Figure 4. MIC2103/04 Current Limiting Circuit
In each switching cycle of the MIC2103/04 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage V(ILIM)
is compared with the power ground (PGND) after a
blanking time of 150nS. In this way the drop voltage over
the resistor RCL (VCL) is compared with the drop over the
bottom FET generating the short current limit. The small
capacitor (CCL) connected from ILIM pin to PGND filters
the switching node ringing during the off time allowing a
better short limit measurement. The time constant
created by RCL and CCL should be much less than the
minimum off time.
The VCL drop allows programming of short limit through
the value of the resistor (RCL), If the absolute value of the
voltage drop on the bottom FET is greater than VCL’ in
that case the V(ILIM) is lower than PGND and a short
circuit event is triggered. A hiccup cycle to treat the short
event is generated. The hiccup sequence including the
soft start reduces the stress on the switching FETs and
protects the load and supply for severe short conditions.
The short circuit current limit can be programmed by
using the following formula:
Figure 3b. MIC2103 Control Loop Timing
(Discontinuous Mode)
During discontinuous mode, the bias current of most
circuits are reduced. As a result, the total power supply
current during discontinuous mode is only about 400μA,
allowing the MIC2103 to achieve high efficiency in light
load applications.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC2103/04 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC2103/04 uses the RDS(ON) and external resistor
connected from ILIM pin to SW node to decide the
current limit.
R CL =
(ICLIM − ∆ PP × 0.5) × R DS(ON) + VCL
ICL
Eq. 3
where ISH = Desired Current limit
ΔPP = Inductor current peak to peak
RDS (ON) = On resistance of low-side power MOSFET
VCL = Current limit threshold, the typical value is 14mV in
EC table
ICL = Current Limit source current, the typical value is
80µA in EC table.
November 26, 2013
21
Revision 2.0
Micrel, Inc.
MIC2103/04
In case of hard short, the short limit is folded down to
allow an indefinite hard short on the output without any
destructive effect. It is mandatory to make sure that the
inductor current used to charge the output capacitance
during soft start is under the folded short limit, otherwise
the supply will go in hiccup mode and may not be
finishing the soft start successfully.
The MOSFET RDS(ON) varies 30 to 40% with temperature;
therefore, it is recommended to add a 50% margin to ICL
in the above equation to avoid false current limiting due
to increased MOSFET junction temperature rise. It is
also recommended to connect SW pin directly to the
drain of the low-side MOSFET to accurately sense the
MOSFETs RDS(ON).
MOSFET Gate Drive
The MIC2103/04 high-side drive circuit is designed to
switch an N-Channel MOSFET. Figure 1 shows a
bootstrap circuit, consisting of D1 (a Schottky diode is
recommended) and CBST. This circuit supplies energy to
the high-side drive circuit. Capacitor CBST is charged
while the low-side MOSFET is on and the voltage on the
SW pin is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the high-side MOSFET turns
on, the voltage on the SW pin increases to
approximately VIN. Diode D1 is reverse biased and CBST
floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
less than 10mA so a 0.1μF to 1μF is sufficient to hold
the gate voltage with minimal droop for the power stroke
(high-side switching) cycle, i.e., ΔBST = 10mA x
3.33μs/0.1μF = 333mV. When the low-side MOSFET is
turned back on, CBST is recharged through D1. A small
resistor RG, which is in series with CBST, can be used to
slow down the turn-on time of the high-side N-channel
MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
November 26, 2013
22
Revision 2.0
Micrel, Inc.
MIC2103/04
Application Information
Setting the Switching Frequency
The MIC2103/04 are adjustable-frequency, synchronous
buck controllers featuring a unique adaptive on-time
control architecture. The switching frequency can be
adjusted between 200kHz and 600kHz by changing the
resistor divider network consisting of R19 and R20.
MOSFET Selection
The MIC2103/04 controllers work from input voltages of
4.5V to 75V and have an internal 5V VDD LDO. This
internal VDD LDO provides power to turn the external NChannel power MOSFETs for the high-side and low-side
switches. For applications where VDD < 5V, it is
necessary that the power MOSFETs used are sub-logic
level and are in full conduction mode for VGS of 2.5V. For
applications when VDD > 5V; logic-level MOSFETs,
whose operation is specified at VGS = 4.5V must be
used.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles. In such an application,
the high-side MOSFET is then required to switch as
quickly as possible in order to minimize transition losses,
whereas the low-side MOSFET can switch slower, but
must handle larger RMS currents. When the duty cycle
approaches 50%, the current carrying capability of the
high-side MOSFET starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50 to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2103/04 gate-drive circuit. At 200kHz switching
frequency, the gate charge can be a significant source of
power dissipation in the MIC2103/04. At low output load,
this power dissipation is noticeable as a reduction in
efficiency. The average current required to drive the
high-side MOSFET is:
MIC2103/04
VDD
VDD/PVDD
1µF
AGND
VIN
VIN
BST
SW
CS
R19
2.2µF
x3
FREQ
R20
FB
PGND
Figure 5. Switching Frequency Adjustment
The following formula gives the estimated switching
frequency:
fSW _ ADJ = fO ×
R 20
R19 + R 20
Eq. 4
Where fO = Switching Frequency when R19 is 100k and
R20 being open, fO is typically 550kHz. For a more
precise setting, it is recommended to use the following
graph:
Switching Frequency
600
IG[high- side] (avg) = QG × fSW
R19 = 100k, IOUT =10A
Eq. 5
500
SW FREQ (kHz)
VIN = 48V
where:
IG[high-side](avg) = Average high-side MOSFET gate
current
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VDD.
fSW = Switching Frequency
400
VIN =75V
300
200
100
0
10.00
100.00
1000.00
10000.00
R20 (k Ohm)
Figure 6. Switching Frequency vs. R20
November 26, 2013
23
Revision 2.0
Micrel, Inc.
MIC2103/04
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
For the low-side MOSFET:
IG[low -side] (avg) = C ISS × VGS × f SW
PSW = PCONDUCTION + PAC
Eq. 8
PCONDUCTION = ISW(RMS) 2 × RDS(ON)
Eq. 9
PAC = PAC(off ) + PAC(on)
Eq. 10
where:
RDS(ON) = On-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
Eq. 6
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
Since the current from the gate drive comes from the
VDD, the power dissipated in the MIC2103/04 due to gate
drive is:
tT =
PGATEDRIVE = VDD × (IG[high- side] (avg) + IG[low -side] (avg))
C ISS × VIN + C OSS × VHSD
IG
Eq. 11
Eq. 7
where:
CISS and COSS are measured at VDS = 0
IG = Gate-drive current
A convenient figure of merit for switching MOSFETs is
the on resistance multiplied by the total gate charge;
RDS(ON) × QG. Lower numbers translate into higher
efficiency. Low gate-charge logic-level MOSFETs are a
good choice for use with the MIC2103/04. Also, the
RDS(ON) of the low-side MOSFET will determine the
current-limit value. Please refer to “Current Limit”
subsection is Functional Description for more details.
Parameters that are important to MOSFET switch
selection are:
•
Voltage rating
•
On-resistance
•
Total gate charge
The total high-side MOSFET switching loss is:
PAC = (VHSD + VD ) × IPK × t T × f SW
where:
tT = Switching transition time
VD = Body diode drop (0.5V)
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
November 26, 2013
Eq. 12
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor.
24
Revision 2.0
Micrel, Inc.
MIC2103/04
A good compromise between size, loss and cost is to set
the inductor ripple current to be equal to 20% of the
maximum output current.
The inductance value is calculated by Equation 13:
L=
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × 20% × IOUT(max)
Copper loss in the inductor is calculated by Equation 17:
PINDUCTOR(Cu) = IL(RMS) × RWINDING
2
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
Eq. 13
PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 18
where:
fSW = Switching frequency
20% = Ratio of AC ripple current to DC output current
VIN(max) = Maximum power stage input voltage
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
The peak-to-peak inductor current ripple is:
∆IL(pp) =
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × L
Eq. 14
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
Eq. 15
2
The RMS inductor current is used to calculate the I R
losses in the inductor.
2
IL(RMS) = IOUT(max) +
ΔIL(PP)
12
ESR COUT ≤
2
Eq. 16
ΔVOUT(pp)
Eq. 19
ΔIL(PP)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2103/04 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetic vendor.
November 26, 2013
Eq. 17
25
Revision 2.0
Micrel, Inc.
MIC2103/04
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 20:
2
ΔVOUT(pp)
The input voltage ripple will primarily depend on the
input capacitor’s ESR. The peak input current is equal to
the peak inductor current, so:
ΔVIN = IL(pk) × ESRCIN
ΔIL(PP)


2
 + ΔIL(PP) × ESR C
= 
OUT

C
×
f
×
8
OUT
SW


Eq. 20
(
)
Eq. 23
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
where:
D = duty cycle
COUT = output capacitance value
fsw = switching frequency
ICIN(RMS) ≈ IOUT(max) × D × (1 − D)
Eq. 24
The power dissipated in the input capacitor is:
As described in the “Theory of Operation” subsection in
Functional Description, the MIC2103/04 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide enough feedback
voltage ripple. Please refer to the “Ripple Injection”
subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 21:
ICOUT (RMS) =
ΔIL(PP)
2
PDISS(CIN) = ICIN(RMS) × ESRCIN
Eq. 25
Voltage Setting Components
The MIC2103 requires two resistors to set the output
voltage as shown in Figure 7:
Figure 7. Voltage-Divider Configuration
Eq. 21
12
The output voltage is determined by the equation:
The power dissipated in the output capacitor is:
VOUT = VFB × (1 +
2
PDISS(COUT ) = ICOUT (RMS) × ESR COUT
Eq. 22
Eq. 26
where, VFB = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush
currents
without
voltage
de-rating.
November 26, 2013
R1
)
R2
R2 =
26
VFB × R1
VOUT − VFB
Eq. 27
Revision 2.0
Micrel, Inc.
MIC2103/04
Ripple Injection
The VFB ripple required for proper operation of the
MIC2103/04 gm amplifier and error comparator is 20mV
to 100mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
For a low output voltage, such as a 1V, the output
voltage ripple is only 10mV to 20mV, and the feedback
voltage ripple is less than 20mV. If the feedback voltage
ripple is so small that the gm amplifier and error
comparator cannot sense it, then the MIC2103/04 will
lose control and the output voltage is not regulated. In
order to have some amount of VFB ripple, a ripple
injection method is applied for low output voltage ripple
applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 8a, the converter is stable
without any ripple injection. The feedback voltage
ripple is:
ΔVFB(pp) =
R2
× ESR COUT × ΔIL (pp)
R1 + R2
Figure 8b. Inadequate Ripple at FB
Figure 8c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 8c. The injected ripple
is:
Eq. 28
where ΔIL(pp) is the peak-to-peak value of the
inductor current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feed-forward capacitor Cff in this situation,
as shown in Figure 8b. The typical Cff value is
between 1nF and 100nF. With the feed-forward
capacitor, the feedback voltage ripple is very close
to the output voltage ripple:
ΔVFB(pp) ≈ ESR × ΔIL (pp)
ΔVFB(pp) = VIN × K div × D × (1 - D) ×
K div =
R1//R2
R inj + R1//R2
1
fSW × τ
Eq. 30
Eq. 31
where:
VIN = Power stage input voltage
D = Duty cycle
fSW = Switching frequency
Eq. 29
τ = (R1//R2//Rinj) × Cff
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors:
In Equations 30 and 32, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
= << 1
fSW × τ τ
Eq. 32
Figure 8a. Enough Ripple at FB
November 26, 2013
27
Revision 2.0
Micrel, Inc.
MIC2103/04
If the voltage divider resistors R1 and R2 are in the kΩ
range, then a Cff of 1nF to 100nF can easily satisfy the
large time constant requirements. Also, a 100nF
injection capacitor Cinj is used in order to be considered
as short for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 33:
K div =
ΔVFB(pp)
VIN
×
fSW × τ
D × (1 − D)
Eq. 33
Then the value of Rinj is obtained as:
R inj = (R1//R2) × (
1
K div
− 1)
Eq. 34
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
November 26, 2013
28
Revision 2.0
Micrel, Inc.
MIC2103/04
PCB Layout Guidelines
Warning: To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2103 converter.
Inductor
•
Keep the inductor connection to the switch node
(SW) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
The SW pin should be connected directly to the
drain of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
IC
•
The 1µF ceramic capacitors, which are connected to
the VDD and PVDD pins, must be located right at
the IC. The VDD pin is very noise sensitive and
placement of the capacitor is very critical. Use wide
traces to connect to the VDD and PGND pins.
•
The signal ground pin (GND) must be connected
directly to the ground planes. Do not route the GND
pin to the PGND pin on the top layer.
•
Place the IC close to the point of load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
•
To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
Input Capacitor
•
Place the input capacitor next.
•
Place the input capacitors on the same side of the
board and as close to the MOSFETs as possible.
•
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
MOSFETs
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Low-side MOSFET gate drive trace (DL pin to
MOSFET gate pin) must be short and routed over a
ground plane. The ground plane should be the
connection between the MOSFET source and
PGND.
•
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
Chose a low-side MOSFET with a high CGS/CGD ratio
and a low internal gate resistance to minimize the
effect of dv/dt inducted turn-on.
•
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
Do not put a resistor between the Low-side
MOSFET gate drive output and the gate.
•
Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than a
2.5V or 3.3V rated MOSFET. MOSFETs that are
rated for operation at less than 4.5V VGS should not
be used.
•
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
RC Snubber
•
Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
November 26, 2013
29
Revision 2.0
Micrel, Inc.
MIC2103/04
Evaluation Board Schematic
Figure 9. Schematic of MIC2103/04 Evaluation Board
(J1, J9, J12, R14, and R21 are for testing purposes)
November 26, 2013
30
Revision 2.0
Micrel, Inc.
MIC2103/04
Bill of Materials
Item
C1
Part Number
EEU-FC2A101
C2, C3, C4
C14
Murata
C3225X7R2A225K
TDK
Murata
12106D107MAT2A
AVX
C3225X5ROJ107M
TDK
AVX
C1608X7R1H104K
TDK
AVX
C1608X5R0J105K
TDK
GRM188R72A102KA01D
C11
C12
C13
Murata
AVX
Murata
TDK
AVX
C1608X7R2A102K
TDK
2.2µF/100V Ceramic Capacitor, X7R, Size 1210
3
100µF/6.3V Ceramic Capacitor, X5R, Size 1210
1
0.1µF/50V Ceramic Capacitor, X7R, Size 0603
2
1µF/6.3V Ceramic Capacitor, X7R, Size 0603
3
0.47µF/100V Ceramic Capacitor, X7R, Size 0805
1
0.1µF/100V Ceramic Capacitor, X7R, Size 0603
0.1µF/100V,X7S,0603
1
1nF/100V Cermiac Capacitor, X7R, Size 0603
1
2.2nF/100V Cermiac Capacitor, X7R, Size 0603
1
Murata
06031C222KAT2A
AVX
C1608X7R2A222K
TDK
6SEPC470MX
1
Murata
06031C102KAT2A
GRM188R72A222KA01D
100µF Aluminum Capacitor, 100V
Murata
06036C105KAT2A
C1608X7S2A104K
Qty.
Murata
06035C104KAT2A
GRM188R72A104KA35D
Description
(9)
GRM32ER60J107ME20L
08051C474KAT2A
C10
(8)
AVX
GRM21BR72A474KA73
C9
(6)
(7)
12101C225KAT2A
GRM188R70J105KA01D
C7, C8, C17
Panasonic
GRM32ER72A225K
GRM188R71H104KA93D
C6, C16
Manufacturer
Sanyo
(10)
470µF/6.3V, 7m-ohms, OSCON
1
6SEPC470M
Sanyo
470µF/6.3V, 7m-ohms, OSCON
C15 (OPEN)
6TPB470M
Sanyo
470µF/6.3V, POSCAP
C5 (OPEN)
GRM32ER60J107ME20L
Murata
100µF/6.3V Ceramic Capacitor, X7R, Size 1210
C18
GCM1885C2A100JA16D
06031A100JAT2A
10pF, 100V, 0603, NPO
1
D1
BAT46W-TP
Murata
AVX
(11)
MCC
100V Small Signal Schottky Diode, SOD123
1
6.1µH Inductor, 14.8A RMS Current
1
L1
CDEP147NP-6R1MC-95
Sumida
(12)
Notes:
6.
Panasonic: www.panasonic.com.
7.
Murata: www.murata.com.
8.
TDK: www.tdk.com.
9.
AVX: www.avx.com
10. Sanyo: www.sanyo.com.
11. MCC.: www.mccsemi.com.
12. Sumida: www.sumida.com.
November 26, 2013
31
Revision 2.0
Micrel, Inc.
MIC2103/04
Bill of Materials (Continued)
Item
Part Number
Q1
SIR878DP
Q3
SIR882DP
Manufacturer
(13)
Vishay
Description
Qty
MOSFET, N-CH, Power SO-8
1
Vishay
MOSFET, N-CH, Power SO-8
1
Q2, Q4 (OPEN)
R1
CRCW060310K0FKEA
Vishay Dale
10kΩ Resistor, Size 0603, 1%
1
R2, R23
CRCW08051R21FKEA
Vishay Dale
1.21Ω Resistor, Size 0805, 5%
2
R3
CRCW060395K30FKEA
Vishay Dale
95.3kΩ Resistor, Size 0603, 1%
1
R4
CRCW060380K6FKEA
Vishay Dale
80.6kΩ Resistor, Size 0603, 1%
1
R5
CRCW060340K2FKEA
Vishay Dale
40.2kΩ Resistor, Size 0603, 1%
1
R6
CRCW060320K0FKEA
Vishay Dale
20kΩ Resistor, Size 0603, 1%
1
R7
CRCW060311K5FKEA
Vishay Dale
11.5kΩ Resistor, Size 0603, 1%
1
R8
CRCW06038K06FKEA
Vishay Dale
8.06kΩ Resistor, Size 0603, 1%
1
R9
CRCW06034K75FKEA
Vishay Dale
4.75kΩ Resistor, Size 0603, 1%
1
R10
CRCW06033K24FKEA
Vishay Dale
3.24kΩ Resistor, Size 0603, 1%
1
R11
CRCW06031K91FKEA
Vishay Dale
1.91kΩ Resistor, Size 0603, 1%
1
R12 (OPEN)
CRCW0603715R0FKEA
Vishay Dale
715Ω Resistor, Size 0603, 1%
R13 (OPEN)
CRCW0603348R0FKEA
Vishay Dale
348Ω Resistor, Size 0603, 1%
R14, R15
CRCW06030000FKEA
Vishay Dale
0Ω Resistor, Size 0603, 5%
2
R16
CRCW08052R0FKEA
Vishay Dale
2Ω Resistor, Size 0805, 5%
1
R17
CRCW06032K21FKEA
Vishay Dale
2.21kΩ Resistor, Size 0603, 1%
1
R18, R20
CRCW060349K9FKEA
Vishay Dale
49.9kΩ Resistor, Size 0603, 1%
2
R19, R22
CRCW0603100K0FKEA
Vishay Dale
100kΩ Resistor, Size 0603, 1%
2
R21
CRCW060349R9FKEA
Vishay Dale
49.9Ω Resistor, Size 0603, 1%
1
75V Synchronous Buck DC-DC Controller
1
U1
MIC2103YML
MIC2104YML
(14)
Micrel. Inc.
Notes:
13. Vishay: www.vishay.com.
14. Micrel, Inc.: www.micrel.com.
November 26, 2013
32
Revision 2.0
Micrel, Inc.
MIC2103/04
PCB Layout Recommendations
Figure 10. MIC2103/04 Evaluation Board Top Layer
November 26, 2013
33
Revision 2.0
Micrel, Inc.
MIC2103/04
PCB Layout Recommendations (Continued)
Figure 11. MIC2103/04 Evaluation Board Mid-Layer 1 (Ground Plane)
November 26, 2013
34
Revision 2.0
Micrel, Inc.
MIC2103/04
PCB Layout Recommendations (Continued)
Figure 12. MIC2103/04 Evaluation Board Mid-Layer 2
November 26, 2013
35
Revision 2.0
Micrel, Inc.
MIC2103/04
PCB Layout Recommendations (Continued)
Figure 13. MIC2103/04 Evaluation Board Bottom Layer
November 26, 2013
36
Revision 2.0
Micrel, Inc.
MIC2103/04
Recommended Land Pattern
Red circle indicates Thermal Via. Size should be .300mm − .350mm in diameter
and it should be connected to GND plane for maximum thermal performance.
ALL UNITS ARE IN mm, TOLERANCE ±0.05, IF NOT NOTED
LP # MLF33D-16LD-LP-1
November 26, 2013
37
Revision 2.0
Micrel, Inc.
MIC2103/04
Package Information(15)
16-Pin 3mm × 3mm MLF (ML)
Note:
15. Package information is correct as of the publication date. For updates and most current information, go to www.micrel.com.
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2012 Micrel, Incorporated.
November 26, 2013
38
Revision 2.0