Linear Technology Magazine Circuit Collection, Vol 1

Application Note 52
January 1993
Linear Technology Magazine Circuit Collection, Volume 1
Richard Markell, Editor
Introduction
Over the past several years Linear Technology, the magazine, has come of age. From nothing, the publication has
come into its own, as has its subscriber list. Many innovative circuits have seen the light of day in the pages of our
now hallowed publication.
This Application Note is meant to consolidate the circuits
from the first few years of the magazine in one place.
Circuits herein range from laser diode driver circuits to
data acquisition systems to a 50W high efficiency switcher
circuit. Enough said. I’ll stand aside and let the authors
explain their circuits.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
CIRCUIT INDEX
A-to-D Converters ..................................................................................................................2
LTC1292: 12-BIT DATA ACQUISITION CIRCUITS .....................................................................................................2
Temperature-Measurement System .....................................................................................................................2
Floating, 12-Bit Data Acquisition System .............................................................................................................2
Differential Temperature Measurement System ...................................................................................................2
MICROPOWER SO8 PACKAGED ADC CIRCUITS .....................................................................................................4
Floating 8-Bit Data Acquisition System ................................................................................................................4
0°C – 70°C Thermometer ....................................................................................................................................5
Interface .............................................................................................................................6
LOW DROPOUT REGULATOR SIMPLIFIES ACTIVE SCSI TERMINATORS ...............................................................6
Power ................................................................................................................................7
LT1110 SUPPLIES 6 VOLTS AT 550mA FROM 2 AA NiCad CELLS ...........................................................................7
50 WATT HIGH EFFICIENCY SWITCHER ..................................................................................................................9
Filters .............................................................................................................................. 10
CASCADED 8TH-ORDER BUTTERWORTH FILTERS PROVIDE STEEP ROLL-OFF LOWPASS FILTER......................10
DC-ACCURATE, PROGRAMMABLE-CUTOFF, FIFTH-ORDER BUTTERWORTH LOWPASS FILTER
REQUIRES NO ON-BOARD CLOCK ........................................................................................................................11
Miscellaneous Circuits.......................................................................................................... 12
A SINGLE CELL LASER DIODE DRIVER USING THE LT1110 .................................................................................12
LT1109 GENERATES VPP FOR FLASH MEMORY ...................................................................................................13
RF LEVELING LOOP ...............................................................................................................................................13
HIGH ACCURACY INSTRUMENTATION AMPLIFIER...............................................................................................14
A FAST, LINEAR, HIGH CURRENT LINE DRIVER ....................................................................................................15
an52f
AN52-1
Application Note 52
A-to-D Converters
LTC1292: 12-BIT DATA ACQUISITION CIRCUITS
by Sammy Lum
Temperature-Measurement System
The circuit in Figure 1 shows how a transducer output,
such as a platinum RTD bridge, can be digitized with one
op amp. This circuit is a modification of that found in
Application Note 43.1 The differential input of the LTC1292
removes the common mode voltage. The LT1006 is used
for amplification. The resistor tied between the + input of
the LT1006 and the +IN input of the LTC1292 is to compensate for the loading of the bridge by resistor RS. Full
scale can be adjusted by the 500kΩ trim pot and offset
can be adjusted by the 100Ω trim pot in series with RS.
A lower RPLAT value than that in AN43 is used here to
improve dynamic range. The signal voltage on the +IN
pin must not exceed VREF. The differential voltage range is
VREF minus approximately 100mV. This is enough range to
measure 0°C to 400°C with 0.1°C resolution.
Floating, 12-Bit Data Acquisition System
The circuit in Figure 2 demonstrates how to float the
LTC1292 to make a differential measurement. This circuit
will digitize a 5V range from 10V to 15V with 12 bits of
1 Williams, Jim, “Bridge Circuits, Marrying Gain and Balance,” Application
resolution. The digital I/O has been level translated. The
LT1019-5 is used in shunt mode to create the floating
analog ground for the LTC1292. The digital I/O lines make
use of 4.3V Zeners to clamp the single-transistor inverters.
Opto-isolators can also be used. The floating analog ground
should be laid out as a ground plane for the LTC1292. The
47μF bypass capacitor should be tied from the VCC pin to
the floating ground plane with minimum lead length and
placed as close to the device as possible. Likewise, keep
the lead length from the GND pin to the floating ground
plane at a minimum (a low-profile socket is acceptable).
Differential Temperature Measurement System
The circuit in Figure 3 digitizes the difference in temperature between two locations. The two LM134s are used as
temperature sensors. These are ideally suited for remote
applications because they are current output devices.
This allows long wires to run from the sensor back to
the LTC1292 without any degradation to the signal from
the sensor. Resistor RSET sets the current to 1μA/°K.
The current is converted to a voltage by the resistor R1
connected from V– to ground. The reference voltage and
resistor were selected to give a change of 0.05°C/LSB.
The resolution is given by °C/LSB = VREF / ((4096) (1mA)
(R1)). The maximum temperature at each input is 125°C.
Note 43, Linear Technology Corp.
+5V
+15V
LT1027
100pF
4.7μF
TANTALUM
1μF
1MΩ**
D0
12kΩ*
12.5kΩ*
100Ω
0°C TRIM
1N4148
500kΩ
400°C
TRIM
MPU
(e.g., 68HC11)
0.1μF
–
RS
13kΩ**
LT1006
100Ω*
1MΩ**
VCC
CLK
SCK
–IN
DOUT
MISO
GND
VREF
+IN
+
RPLAT
100Ω RTD
AT 0°C
CS
LTC1292
22μF†
TANTALUM
* TRW-IRC MAR -6 RESISTOR -0.1%
** 1% METLA FILM RESISTOR
= ROSEMOUNT 118MFRTD
R
† PLAT
CERAMICS CAN ALSO BE USED
"/t'
Figure 1. 0° to 400°C Temperature-Measurement System
an52f
AN52-2
Application Note 52
Note that if the temperature on the +IN pin is less than
the temperature on the –IN pin, the output will be
zero. Because the LTC1292 is being driven from a high
source impedance, you should limit the CLK frequency
to 100kHz or less.
The software code for interfacing the LTC1292 to the
Motorola MC68HC11 or the Intel 8051 is found in the
LTC1292 data sheet. The code needs to be modified for the
circuit in Figure 2 to account for the inversion introduced
by the digital level translators.
+15V
OUT
1N5229
1kΩ
+
1kΩ
1N5229
LT1019-5
CS
VCC
+INPUT
+IN
CLK
–INPUT
–IN
DOUT
GND
VREF
10V TO 15V
1N4148
DIODES
LTC1292
1μF
TANTALUM
GND
+
47μF
TANTALUM
+5V
1.5kΩ
1kΩ
1.5kΩ
1kΩ
1kΩ
2N2222
P1.4
2N3906
MPU
(e.g., 8051)
1kΩ
2N2222
P1.3
1N4148
1.5kΩ
"/t'
P1.1
1kΩ
Figure 2. Floating, 12-Bit Data Acquisition System
+5V
+15V
LT1027
1μF
4.7μF**
TANTALUM
LM134
V+
P1.4
V–
228Ω*
R1
10kΩ*
LM134
V+
MPU
(e.g., 8051)
CS
VCC
+IN
CLK
LTC1292
–IN
DOUT
GND
VREF
P1.3
P1.1
14.7kΩ*
V–
22μF**
TANTALUM
10.2kΩ*
22μF**
TANTALUM
228Ω*
R1
10kΩ*
"/t'
*1% METAL FILM RESISTOR
**CERAMICS CAN ALSO BE USED
Figure 3. Differential Temperature-Measurement System
an52f
AN52-3
Application Note 52
MICROPOWER SO8 PACKAGED ADC CIRCUITS
by William Rempfer
Floating 8-Bit Data Acquisition System
Figure 4 shows a floating system that sends data to a
grounded host system. The floating circuitry is isolated
by two opto-isolators and powered by a simple capacitordiode charge pump. The system has very low power
requirements because the LTC1096 shuts down between
conversions and the opto-isolators draw power only when
data is being transferred. The system consumes only
50μA at a sample rate of 10Hz (1ms on-time and 99ms
off-time). This is easily within the current supplied by the
charge pump running at 5MHz. If a truly isolated system
is required, the system’s low power simplifies generating
an isolated supply or powering the system from a battery.
FLOATING SYSTEM
1N5817
47μF
0.001μF
2kV
0.1μF
1N5817
75k
2N3904
VCC
0.022
100k
20k
CS
+IN
1N5817
4N28
100k
CLK
LT1004-2.5
LTC1096
5MHz
300Ω
VREF
–IN
CLK
GND
ANALOG
INPUT
DOUT
1k
10k
DATA
500k
"/t'
Figure 4. Power for this Floating ADC System Is Provided by a Simple Capacitor-Diode
Charge Pump. The Two Opto-Isolators Draw No Current Between Samples, Turning on
Only to Send the Clock and Receive Data
an52f
AN52-4
Application Note 52
0°C – 70°C Thermometer
Figure 5 shows a temperature-measurement system.
The LTC1096 is connected directly to the low cost silicon
temperature sensor. The voltage applied to the VREF pin
adjusts the full scale of the ADC to the output range of
the sensor. The zero point of the converter is matched to
the zero output voltage of the sensor by the voltage on
the LTC1096’s negative input.
Figure 6 shows the operating sequence of the LTC1096.
The converter draws power when the CS pin is low and
shuts itself down when that pin is high. In systems that
convert continuously, the LTC1096/LTC1098 will draw
its normal operating power continuously. A 10μs wake
up time must be provided to the LTC1096 after each
falling CS.
Operating the ADC directly off batteries can eliminate the
space taken by a voltage regulator. Connecting the ADC
directly to sensors can eliminate op amps and gain stages.
The LTC1096/LTC1098 can operate with small, 0.1μF or
0.01μF chip bypass capacitors.
In systems that have significant time between conversions,
lowest power drain will occur with the minimum CS low
time. Bringing CS low, waiting 10μs for the wake up time,
transferring data as quickly as possible, and then bringing
it back high will result in the lowest current drain.
+3V
0.1μF
LM134
75k
VCC
678Ω
+IN
13.5kΩ
CS
LTC1096
CLK
–IN
TO μP
182k
DOUT
VREF
GND
LT1004-1.2
0.01μF
0.01μF
63.4k
"/t'
Figure 5. The LTC1096’s High-Impedance Input Connects Directly to This Temperature Sensor,
Eliminating Signal Conditioning Circuitry in This 0°C–70°C Thermometer
LTC1096 POWER DOWN AND WAKE UP
CS
POWER DOWN
tWAKE UP
(10μSEC MIN)
tCONVERSION
CLK
DOUT
B7
B6
B5
B4
B3
B2
B1
B0
"/t'
Figure 6. The ADC’s Power Consumption Drops to Zero When CS Goes High. 10μs After CS Goes Low,
the ADC Is Ready to Convert. For Minimum Power Consumption Keep CS High for as Much Time as
Possible Between Conversions
an52f
AN52-5
Application Note 52
Interface
LOW DROPOUT REGULATOR SIMPLIFIES ACTIVE SCSI
TERMINATORS
by Sean Gold
characteristic impedance, and the regulator provides a good
AC ground.
The circuit shown in Figure 7 uses an LT1117 low dropout
three terminal regulator to control the terminator’s local
logic supply. The LT1117’s line regulation makes the
output immune to variations in TERMPWR. After accounting for resistor tolerances and variations in the LT1117’s
reference voltage, the absolute variation in the 2.85V
output is only 4% over temperature. When the regulator drops out at TERMPWR-2.85, or 1.25V, the output
linearly tracks the input with a 1V/V slope. The regulator
provides effective signal termination because the 110Ω
series resistor closely matches the transmission line’s
In contrast to a passive terminator, two LT1117s require
half as many termination resistors, and operate at 1/15
the quiescent current or 20mA. At these power levels,
PC traces provide adequate heat sinking for the
LT1117’s SOT-223 package. Beyond solving basic signal
conditioning problems, this LT1117 terminator handles
fault conditions with short circuit current limiting, thermal
shutdown, and on-chip ESD protection.
CONNECTOR
TERMPWR
5V
LOGIC
SUPPLY
1N5817
LT1117
2.85V
SOT-223
10μF
TANTALUM
110Ω
2%
110Ω
0.1μF
CERAMIC
22μF
TANTALUM
SCSI BUS
18 TO 27 LINES
110Ω
2%
110Ω
"/t'
Figure 7. SCSI Active Termination
an52f
AN52-6
Application Note 52
Power
LT1110 SUPPLIES 6 VOLTS AT 550mA FROM 2 AA
NiCad CELLS
by Steve Pietkiewicz
discharged. This allows power to be drawn from the cell at
a far greater rate. The circuit in Figure 8 uses two AA NiCad
cells to supply 6 volts at 550mA. The circuit, developed for
pagers with transmit capability, runs at full output current
for 15 minutes with two Gates Millennium AA NiCad cells.
With a 250mA load, the circuit runs for 36 minutes (see
Figure 9). Less heat is generated with a reduced load,
resulting in the watt-hour difference observable above.
The LT1110 micropower DC-DC converter can provide 5V
at 150mA when operating from two AA alkaline cells. The
internal switch VCE(SAT) sets this power limit. Even with an
external low drop switch, more power is not realistically
possible. The internal impedance (typically 200mΩ fresh
and 500mΩ at end-of-life) of alkaline AA cells limits peak
obtainable battery power. Conversely, nickel-cadmium cells
have a constant internal impedance (35mΩ to 50mΩ) for
AA size) that increases only when the cell is completely
L1
5μH
10T # 18GA
MAGNETICS INC.
55-041-A2 CORE
R5
1k
6
ILIM
AO
Q1
2N4403
2
7
100μF
10V
SANYO
OS-CON
+
C1
1μF
Q4
2N3904
SET
GND
5
R2
1k
220
VIN
3
SW1
LT1110
2 w AA
NiCAD
6VOUT
550mA
220
1
Q3
2N3906
D1
1N5820
FB
SW2
4
8
10Ω
10k
16Ω
Q5
2N3904
R3
15k
1%
Q2
ZETEX
ZTX-849
300pF
R4
394k
1%
+
C2
220μF
10V
SANYO
0S-CON
R1
50mΩ
ZETEX (516) 543-7100
SANYO (619) 661-6322
MAGNETICS INC. (412) 282-8282
"/t5"
Figure 8. Schematic Diagram, 2 AA NiCad to +6 Volt Converter
an52f
AN52-7
BATTERY/OUTPUT VOLTAGE (V)
Application Note 52
7.0
6.5
6.0
5.5
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
ILOAD = 550mA
ILOAD = 250mA
ILOAD = 250mA
ILOAD = 550mA
OUTPUT
BATTERY
0
5
10
15
20
25
30
35
40
TIME (MINUTES)
"/t'
Figure 9. Operating Time at ILOAD = 550mA and 250mA
The circuit uses a micropower LT1110 switching-regulator
IC as a controller. The internal switch of the LT1110
furnishes base drive to Q1 through the 220Ω resistors.
Q1, in turn, supplies base drive to the power switch Q2.
The Zetex ZTX849 NPN device is rated at 5A current and
comes in a TO-92 package. For surface-mount fans, the
FZT-849, also from Zetex, provides the same performance
in an SOT-223 package. The 16Ω resistor provides a turn
off path for Q2’s stored charge. When Q2 is on, current
builds in L1. As Q2 turns off, its collector flies positive until
D1 turns on. L1’s built-up current discharges through D1
into C2 and the load. The voltage at VOUT is divided by R4
and R3 and fed back into the FB pin of the LT1110, which
controls Q2’s cycling action. Switch current limit, which
is necessary to ensure saturation over supply variations,
is implemented by Q3–Q5. Q3, C1, R2, and the auxiliary
gain block inside the LT1110, form a 220mV reference
point at the LT1110’s SET pin. Transistors Q4 and Q5
form a common-base differential amplifier. Q5’s emitter
monitors the voltage across 50mΩ resistor R1. When
the voltage across R1 exceeds 220mV, Q4 turns on hard,
pulling current through R5. When the voltage at the ILIM
pin of the LT1110 reaches a diode drop below the VIN pin,
the internal switch turns off. Thus, maximum switch current is maintained at 220mV/50mΩ, or 4.4A, over input
variations and manufacturing spread in the LT1110’s on
time and frequency.
The circuit’s output ripple measures 200mVP-P, and
efficiency is 78% at full load with a 2.4V input. Output power
can be scaled down for less demanding requirements. To
reduce peak current, increase the value of R1. A 100mΩ
resistor will limit current to 2.2A. L1 should be increased
in value linearly as current is reduced. For a current limit
of 2.2A, L1 should be 10μH. Base drive for Q2 can also be
reduced by increasing the value of the 10Ω resistor. These
lower peak currents are much easier on alkaline cells and
will dramatically increase alkaline battery life.
an52f
AN52-8
Application Note 52
50 WATT HIGH EFFICIENCY SWITCHER
by Milton Wilcox
The high efficiency 10A step-down (buck) switching regulator shown in Figure 10 illustrates how different sized
MOSFETs can be driven by the LT1158 without having to
worry about shoot-through currents. Since 24V is being
dropped down to 5V, the duty cycle for the switch (top
MOSFET) is only 5/24 or 21%. This means that the bottom MOSFET will dominate the RDS(ON) efficiency losses,
because it is turned on nearly four times as long as the
top. Therefore a smaller MOSFET is used on the top, and
the bottom MOSFET is doubled up, all without having to
worry about dead time.
Switching regulator applications can take advantage of
an important protection feature of the LT1158: remote
fault sensing. By sensing the current on the output side
of the inductor and returning the LT1158 FAULT pin to the
PWM soft-start pin, a true current-mode loop is formed.
The Figure 10 circuit regulates maximum current in the
inductor to 15A with no output voltage overshoot upon
recovery from a short circuit.
95
90
EFFICIENCY (%)
The LT1158 uses an adaptive system that maintains dead
time independent of the type, the size, and even the number
of MOSFETs being driven. It does this by monitoring the
gate turn-off to see that it has fully discharged before allowing the opposite MOSFET to turn on. During turn-on,
the hold-off capability of the opposing driver is boosted
to prevent transient shoot-through. In this way, crossconduction is completely eliminated as a design constraint.
85
80
75
0
2
6
4
8
10
OUTPUT CURRENT (A)
The non-critical Schottky diode across the bottom
MOSFETs reduces reverse-recovery losses. Figure 11
shows the operating efficiency for the Figure 10 circuit.
"/t5"
Figure 11. Operating Efficiency for Figure 10 Circuit.
Current Limit is Set at 15A
24V IN
0.1μF
0.1μF
1N4148
+
4.7kΩ
10μF
4.7kΩ
VIN
NI
INV
VREF
+
BST
DR
V+
1N4148
A
BST
+
T FB
B
LT3525
1μF
LT1158
10kΩ
(2) 500μF
LOW ESR
T DR
1N4148
VC
IRFZ34
L1*
70μH
7mΩ
SEN+
+
SEN–
–
BIAS
COMP
+
GND
0.01μF
0.01μF
100Ω
+
(2) 1000μF
LOW ESR
(2) IRFZ44
DISCHG
CT
SOFT-ST
RT
FAULT
+
B DR
5μF
MBR340
B FB
GND
30kΩ
5V/10A
OUT
SRC
30kΩ
2.2nF
UPDATE
For reference only
2.4kΩ
*1 1/4" MP CORE 14GA WIRE
"/t5"
fOSC = 25kHz
Figure 10. 50W High Efficiency Switching Regulator Illustrates the Design Ease Afforded by Adaptive Dead Time Generation
an52f
AN52-9
Application Note 52
Filters
CASCADED 8TH-ORDER BUTTERWORTH FILTERS
PROVIDE STEEP ROLL-OFF LOWPASS FILTER
by Philip Karantzalis and Richard Markell
Sometimes a design requires a filter that exceeds the
specifications of the standard “dash-number” filter. In
this case, the requirement was a low-distortion (–70dB)
filter with roll-off faster than that of an 8th-order Butterworth. An elliptic filter was ruled out because its distortion
specifications are too high. Two low power LTC1164-5s
were wired in cascade to investigate the specifications that
could be achieved with this architecture. The LTC1164-5
is a low power (4 milliamperes with ±5 volt supplies),
clock-tunable, 8th-order filter, which can be configured
for a Butterworth or Bessel response by strapping a pin.
Figure 12 shows the schematic diagram of the two-filter
system. The frequency response is shown in Figure 13,
where it can be seen that the filter’s attenuation is 80dB at
2.3 times the cutoff frequency. The distortion, as shown
in Figure 14, is nothing less than spectacular. From 100Hz
to 1kHz, the two filters have less than –74dB distortion
specifications. At the standard measurement frequency
of 1kHz, the specification is –78dB.
LĀ
2
3
7
5
˜'
7
INV (C)
OUT (C)
R3 SHUNT
VIN
V–
GND
-5$
V+
fCLK
GND
LP (A)
VOUT
RIN (A)
NC
2
˜'
3
V+
9
8
7
V–
-5$
V+
5
˜'
R3 SHUNT
GND
7
OUT (C)
VIN
o7
INV (C)
fCLK
GND
LP (A)
VOUT
RIN (A)
NC
fCLK
˜'
o7
V+
9
8
L
L
"/t'
V–
V–
Figure 12. Schematic Diagram: Low Power, 16th-Order Lowpass Filter (Two 8th-Order Butterworths Cascaded)
10
–40
0
–45
–50
–20
–55
–60
–30
–65
THD (%)
GAIN (dB)
–10
–40
–50
–70
1.4VRMS IN
–75
–80
–60
–85
–70
–90
–80
–90
100
±5V SUPPLY
1k
5k
FREQUENCY (Hz)
–95
–100
100
1k
FREQUENCY (Hz)
"/t'
Figure 13. Frequency Response for fCLK = 20kHz
"/t'
Figure 14. Distortion Performance: Two LTC1164-5s,
fCLK = 60kHz (57:1) Pin 10 Connected to V+
an52f
AN52-10
Application Note 52
DC-ACCURATE, PROGRAMMABLE-CUTOFF,
FIFTH-ORDER BUTTERWORTH LOWPASS FILTER
REQUIRES NO ON-BOARD CLOCK
by Richard Markell
Most users choose to tune the filter with an on-board
microprocessor and/or timer. This is quite convenient if
these components are available. If a clock is not available,
the LTC1063 can be tuned with an external resistor and
capacitor. The scheme shown here allows the filter’s cutoff
frequency to be programmed using an external microprocessor or the parallel port of a personal computer. This
allows the cutoff frequency of the filter to be set before
the product is shipped.
The new LTC1063 is a clock-tunable, monolithic filter with
low-DC output offset (1mV typical with ±5V supplies). The
frequency response of the filter closely approximates a
fifth-order Butterworth polynomial.
20k
+7.5V
1
INPUT
VIN
VOS ADJ
8
20k
2
7
GND
VOUT
LTC1063
3 –
6
V+
V
7.5V
.1μF
–
7
6
LT1007
+7.5V
CLK 5
IN
4 CLK
OUT
2
3
+
VOUT
4
0.1μF
–7.5V
HUGHES
HC2021
6
10
15k
1%
C1
SER CLK
C2
SER IN
SER OUT
8
SD IN
PROG
READ
VDD
PROG
25 PIN
PARALLEL PORT
CONNECTOR
FOR IBM PCs
AND COMPATIBLES
SCLK
GND
READ
1
2
3
16
+7.5
15
14
7.5
7.5
74LS05
74LS14
1k
2
11
10
3
4
5
13
12
1
2
3
5
6
9
8
4
1
2
11
18
1k
+5V
74LS14
SD OUT
GND
51Ω
11
8
74LS05
1k
9
6
5
270Ω
25
0.01μF
+5V
GND
VDD
GND
0.1μF
GND
0.1μF
GND
PROGRAMMER FOR NON-VOLATILE CAPACITOR, HC2021
NOTES:
1. THE HC2021 SHOULD BE LOCATED CLOSE TO THE LTC1063 FOR BEST RESULTS.
2. +3.5 ≤ VDD ≤ +18.8.
3. POSITIVE POWER SUPPLY FOR DEVICES 74LS05 AND 74LS14 IS +5V.
4. HUGHES TELEPHONE NUMBER (714) 759-2665.
"/t5"
Figure 15. Schematic Diagram of LTC1063 with Programmable Cutoff Frequency
an52f
AN52-11
Application Note 52
The tuning scheme makes use of non-volatile, tunable
capacitors available from Hughes Semiconductor. These
capacitors allow approximately a decade of tuning range.
More range could be obtained by using dual devices.
Figure 15 shows the schematic diagram of the application. Be sure to place the variable capacitor as close as
possible to the LTC1063 to minimize parasitic elements.
Figure 16 shows the frequency response of the filter when
the capacitor is varied from minimum to half-value, and
then to maximum capacitance. The programming part of
the circuit may be disconnected once the variable capacitor is set. The capacitor will remember its value until it is
reprogrammed.
10.000
0.0
–10.00
AMPLITUDE
–20.00
–30.00
–40.00
–50.00
o
–70.00
–80.00
–90.00
1k
CAPACITOR AT
MAX VALUE
10k
100k 200k
CAPACITOR AT CAPACITOR AT
1/2 MAX VALUE MIN VALUE
FREQUENCY
"/t5"
Figure 16. LTC1063 Frequency Response
Miscellaneous Circuits
A SINGLE CELL LASER DIODE DRIVER
USING THE LT1110
by Steve Pietkiewicz
1.0A. The output capacitor C2 has been specified for low
ESR, and should not be substituted (damage to the laser
diode may result).
Recently available visible lasers can be operated from 1.5V
supplies, given appropriate drive circuits. Because these
lasers are exceptionally sensitive to overdrive, power to the
laser must be carefully controlled lest it be damaged. Overcurrents as brief as 2 microseconds can cause damage.
The Gain Block output of the LT1110 functions with Q1
as an error amplifier. The differential inputs compare the
photodiode current developed as a voltage across R2
to the 212mV reference. The amplifier drives Q1, which
modulates current into the ILIM pin. This varies oscillator
frequency to control average current.
In the circuit of Figure 17, an LT1110 switching regulator
serves as the controller within the single cell powered laser
diode driver. The LT1110 regulator is a high speed LT1073.
The LT1110 is used here as an FM controller, driving a
PNP power switch Q2, with a typical “ON” time of 1.5
microseconds. Current in L1 reaches a peak value of about
Overall frequency compensation is provided by R1 and C1,
values carefully chosen to eliminate power-up overshoot.
The value of current sense resistor R2 determines the laser
diode power, as shown the 1000 ohms results in about a
0.8 milliwatt output.
LZ1
R1
L
D1
1N4148
1.5V
C1
O'
R3
220Ω
Q1
2N3906
I LIM
AO
Q2
MJE210
V IN
SW1
R4
10Ω
+
˜'
C2
˜'
D2
1N5818
R5
2Ω
LT1110
'#
GND
SET
SW2
-50,0-:',
$4"/:004$0/
-;504)*#"50-%
R1
L
L1
2.2μH
"/t'
Figure 17. LT1110 Laser Diode Driver Operating from a Single Cell
an52f
AN52-12
Application Note 52
LT1109 GENERATES VPP FOR FLASH MEMORY
by Steve Pietkiewicz
Flash memory chips such as the Intel 28F020 2Megabit
device require a VPP program supply of 12 volts at 30mA.
A DC-DC converter may be used to generate 12 volts from
the 5 volt logic supply. The converter must be physically
small, available in surface-mount packaging, and have
logic-controlled shutdown. Additionally, the converter
must have carefully controlled rise time and zero overshoot. VPP excursions beyond 14 volts for 20ns or longer
will destroy the ETOX-process based device.
L1†
33μH
VIN
˜'
Leveling loops are often a requirement for RF transmission
systems. More often than not, low cost is more important
than absolute accuracy. Figure 19 shows such a circuit.
VOUT
12V
N"
SENSE
-5$4
+
SHUTDOWN*
GND
SHUTDOWN
RF LEVELING LOOP
by Jim Williams
MBRS120T3
SW
+VIN
5V
devices. The SHUTDOWN input turns off the converter,
reducing quiescent current to 300μA when pulled to a
logic 0. VPP rises in a controlled fashion, reaching 12 volts
±5% in under 4ms. Output voltage goes to VCC minus a
diode drop when the converter is in shutdown mode. This
is an acceptable condition for Intel flash memories and
does not harm the memory.
˜'
PROGRAM
1*/1"$,"(&0/-:
†
-$0*-530/*$4$590346.*%"$%
$0*-530/*$4
46.*%"
"/t'
Figure 18. All Surface Mount Flash Memory VPP Generator
Figure 18’s circuit is well suited for providing VPP power
for single or multiple flash memory chips. All associated
components, including the inductor, are surface mount
The RF input is applied to A1, an LT1228 operational
transconductance amplifier. A1’s output feeds A2, the
LT1228’s current feedback amplifier. A2’s output, the
output of the circuit, is sampled by the A3-based gain
control configuration. This arrangement closes a gain
control loop back at A1. The 4pF capacitor compensates
rectifier diode capacitance, enhancing output flatness
vs frequency. A1’s ISET input current controls its gain,
allowing overall output level control. This approach to
RF leveling is simple and inexpensive, and provides low
output drift and distortion.
+15V
RF INPUT
0.6VRMS-1.3VRMS
25MHz
10k
+
100Ω
A1
OTA
1/2 LT1228
+
–
300Ω
ISET
A2
CFA
1/2 LT1228
OUTPUT
2Vp-p
–
470Ω
0.01
10k
–15V
10k
4pF
10Ω
0.01
10k
+15V
–
A3
LT1006
+
10k
100k
4.7k
–15V
AMPLITUDE
ADJUST
1N4148’s
COUPLE THERMALLY
–15V
LT1004
1.2V
"/t5"
Figure 19. Simple RF Leveling Loop
an52f
AN52-13
Application Note 52
HIGH ACCURACY INSTRUMENTATION AMPLIFIER
by Dave Dwelley
The LTC1043 and the LTC1047 combine to make a high
performance low frequency instrumentation amplifier as
shown in Figure 20. The LTC1043 switched capacitor block
is configured as a sampling front end, providing exceptional CMRR and rail-to-rail input operation. It works by
attaching a 1μF capacitor across the two inputs, letting it
charge to the input voltage. Once charged, the capacitor
is disconnected from the input terminals and reconnected
to the output terminals, where it transfers its charge to the
1μF capacitor at the LTC1047’s input. Any common mode
voltage present at the inputs is subjected to a capacitive
divider between the 1μF flying cap and the IC’s parasitic
capacitance. With the LTC1043’s parasitics typically below 1pF, this gives AC CMRR above 120dB. The analog
switches in the LTC1043 are purely resistive, so they add
no DC offset to the signal.
The output signal (with the common mode stripped off) is
then amplified by the LTC1047, a precision, micropower
zero-drift op amp. The LTC1047 amplifies the signal by
the desired amount, adding less than 10μV offset and
0.05μV/°C drift. The sampling frequency of the LTC1043
with single 5V supply is about 400Hz, allowing differential
signals below 200Hz to be amplified with no aliasing. Note
that common mode signals are not sampled; thus they
will not alias regardless of frequency until the common
mode/differential mode signal ratio approaches 120dB!
The entire system draws 60μA with a single 5V supply
and provides two independent channels.
+5V
+5V
-5$
+IN
+
8
1/2
-5$
11
C)
˜'
CS˜'
VOUT
–
12
–IN
˜'
˜'
R1
R2
"/t'
GAIN = 1 + R2
R1
VOS˜7
DVOSO7¡$
IS˜"4*%&
$.33E#"5)[
4*/(-&74611-:
*/1653"/(&*/$-6%&4
#05)4611-*&4
Figure 20. High Accuracy Instrumentation Amplifier
an52f
AN52-14
Application Note 52
A FAST, LINEAR, HIGH CURRENT LINE DRIVER
by Walt Jung and Rich Markell
Among linear applications not usually seen are those
which require high speed combined with either very low
DC error, or high load current. Such applications can be
solved by combining the best attributes of two ICs, either
one of which may not be capable by itself of the entire
requirement.
The circuit as shown is configured as a precise gain of 5
non-inverting amplifier by gain set resistors R2 and R1,
with the LT1010 unity gain voltage follower inside the
overall feedback loop. This provides current buffering to
the op amp, allowing it to operate most linearly. Small
signal bandwidth is set by the time constant of R2 and
C1, and is 1MHz as shown, with a corresponding risetime
of about 400ns.
A case in point is the line driver of Figure 21, which uses
an LT1122 JFET input op amp as the gain element combined with an LT1010 buffer. This provides the output
current of the LT1010 (typically 150mA) but with the
basic DC and low level AC characteristics of the LT1122.
The circuit is capable of driving loads as low as 100Ω
with very low distortion. The input referred DC error is
the low DC offset of the LT1122, typically 0.5mV or less.
Large signal characteristics are also very good, due to the
80V/μs symmetrical SR of the LT1122.
Performance with ±18V supplies is shown in Figures 22a
and 22b, with output generally 5VRMS or equivalent, driving 100Ω directly. THD is shown in Figure 22a, with input
level swept up to output clipping level, at a fixed 10kHz
frequency. The distortion is generally well below 0.01%,
and improves substantially for lower frequencies.
V+
1μF
R3
100Ω
1%
+
INPUT
U1
LT1122
R IN
10k
1μF
R BOOST
30
1%
R4
49.9Ω
1%
(OPTIONAL)
U2
LT1010
VOUT
–
1μF
1μF
C1
39pF
V–
R2
4k
1%
"/t5"
R1
1k
Figure 21. Line Driver
an52f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN52-15
Application Note 52
CCIF IM distortion performance of the circuit for similar
loading is shown in Figure 22b, driving a load of 100Ω
at a swept level, again up to output clipping. The LT1122
amplifier is represented by the lower of the two curves,
with distortion around the 0.0001% level. Also shown for
comparison in this plot is the distortion of a type 156 JFET
op amp (also driving the LT1010 buffer with other conditions the same). The 156 op amp uses a design topology
with an intrinsically asymmetric SR. This manifests itself
as rising even order distortion for methods such as this
CCIF test. For this example, the distortion is more than an
order of magnitude higher than that of the faster, symmetric
slewing LT1122 for the same conditions.
Applications of this circuit include low offset linear buffers
such as for A/D inputs, line drivers for instrumentation
use, and audio frequency range buffers such as very high
quality headphone use.
1
CCIF 1M DISTORTION (%)
TOTAL HARMONIC DISTORTION (%)
1
0.1
0.010
1
0.010
0.001
0.0001
0.1
0.001
0.1
0.1
10
INPUT LEVEL (V)
1
5
INPUT LEVEL (V)
"/t'B
"/t'C
Figure 22a. THD vs Input Level
Figure 22b. CCIF IM Distortion vs Input Level
Linear Technology, the magazine, is published 3 times a year.
The magazine features articles, circuits and new product
information from the designers at LTC. To subscribe please call
800-637-5545.
an52f
AN52-16
Linear Technology Corporation
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