® RT6207A/B 5A, 18V, 650kHz, ACOTTM Synchronous Step-Down Converter General Description Features The RT6207A/B is a high-performance 650kHz, 5A stepdown regulator with internal power switches and synchronous rectifiers. It features quick transient response using its Advanced Constant On-Time (ACOTTM) control architecture that provides stable operation with small ceramic output capacitors and without complicated external compensation, among other benefits. The input voltage range is from 4.5V to 18V and the output is adjustable from 0.7V to 8V. The proprietary ACOTTM control improves upon other fast response constant on-time architectures, achieving nearly constant switching frequency over line, load, and output voltage ranges. Since there is no internal clock, response to transients is nearly instantaneous and inductor current can ramp quickly to maintain output regulation without large bulk output capacitance. The RT6207A/B is stable with and optimized for ceramic output capacitors. With internal 60mΩ switches and 22mΩ synchronous rectifiers, the RT6207A/B displays excellent efficiency and good behavior across a range of applications, especially for low output voltages and low duty cycles. Cycle-by-cycle current limit provides protection against shorted outputs, input under-voltage lock-out, externally-adjustable soft-start, output under- and over-voltage protection, and thermal shutdown provide safe and smooth operation in all operating conditions. The RT6207A/B is available in the UQFN-13JL 2x3 (FC) package, with exposed thermal pad. Fast Transient Response Advanced Constant On-Time (ACOTTM) Control 4.5V to 18V Input Voltage Range Adjustable Output Voltage from 0.7V to 8V 5A Output Current 60mΩ Ω Internal High-Side N-MOSFET and 22mΩ Ω Internal Low-Side N-MOSFET Steady 650kHz Switching Frequency Up to 95% Efficiency Optimized for All Ceramic Capacitors Externally-Adjustable, Pre-Biased Compatible SoftStart Cycle-by-Cycle Current Limit Input Under-Voltage Lockout Output Over- and Under-Voltage Protection Power Good Output Externally-Adjustable, Pre-Biased Compatible SoftStart Thermal Shutdown Applications Industrial and Commercial Low Power Systems Computer Peripherals LCD Monitors and TVs Green Electronics/Appliances Point of Load Regulation for High-Performance DSPs, FPGAs, and ASICs Simplified Application Circuit RT6207A/B SW VIN VIN VPVCC Enable CSS BOOT CFF R1 COUT PGOOD EN FB SS PVCC GND Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 VOUT CBOOT CIN RPG L VPVCC CPVCC R2 is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6207A/B Pin Configurations Ordering Information BOOT 12 11 1 10 VIN EN 2 9 VIN FB 3 8 PGOOD NC 4 7 NC A : PSM B : PWM Note : 5 6 PVCC GND UVP Option H : Hiccup Mode L : Latched-Off Mode UQFN-13JL 2x3 (FC) Richtek products are : 13 SS Lead Plating System G : Green (Halogen Free and Pb Free) SW Package Type QUF : UQFN-13JL 2x3 (U-Type) (FC) GND (TOP VIEW) RT6207A/B RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. Suitable for use in SnPb or Pb-free soldering processes. Marking Information RT6207AHGQUF RT6207BHGQUF 0R : Product Code W : Date Code 0RW 0P : Product Code 0PW RT6207ALGQUF RT6207BLGQUF 0Q : Product Code W : Date Code 0QW W : Date Code 0N : Product Code 0NW W : Date Code Functional Pin Description Pin No. 1, 13 Pin Name Pin Function GND Ground. 2 EN Enable Control Input. 3 FB Converter Feedback Input. Connect to output voltage feedback resistor divider. 4, 7 NC No Internal Connection. 5 SS Soft-Start Control. A external capacitor should be connected to GND. 6 PVCC 5V Power Supply Output. A capacitor (typical 1F) should be connected to GND. 8 PGOOD Open Drain Power Good Output. VIN Power Input and Connected to High-Side MOSFET Drain. 11 BOOT Bootstrap Supply for High-Side Gate Driver. This capacitor is needed to drive the power switch's gate abov e the supply voltage. It is connected between SW and BOOT pins to form a floating supply across the power switch driver. A 0.1F capacitor is recommended for use. 12 SW Switch Output. 9, 10 Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Function Block Diagram BOOT PVCC VIN PVCC Reg Min. Off VIBIAS PVCC VIN VREF UGATE Control OC Driver SW LGATE UV & OV PVCC SW 6µA Ripple Gen. SS FB VIN SW GND GND SW + Comparator On-Time Comparator 0.9 VREF FB PGOOD + - EN EN Detailed Description The RT6207A/B is a high-performance 650kHz 5A stepdown regulators with internal power switches and synchronous rectifiers. It features an Advanced Constant On-Time (ACOTTM) control architecture that provides stable operation with ceramic output capacitors without complicated external compensation, among other benefits. The ACOTTM control mode also provide fast transient response, especially for low output voltages and low duty cycles. The input voltage range is from 4.5V to 18V and the output is adjustable from 0.7V to 8V. The proprietary ACOTTM control scheme improves upon other constant on-time architectures, achieving nearly constant switching frequency over line, load, and output voltage ranges. The RT6207A/B are optimized for ceramic output capacitors. Since there is no internal clock, response to transients is nearly instantaneous and inductor current can ramp quickly to maintain output regulation without large bulk output capacitance. Constant On-Time (COT) Control The heart of any COT architecture is the on-time one shot. Each on-time is a pre-determined “fixed” period that is triggered by a feedback comparator. This robust Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 arrangement has high noise immunity and is ideal for low duty cycle applications. After the on-time one-shot period, there is a minimum off-time period before any further regulation decisions can be considered. This arrangement avoids the need to make any decisions during the noisy time periods just after switching events, when the switching node (SW) rises or falls. Because there is no fixed clock, the high-side switch can turn on almost immediately after load transients and further switching pulses can ramp the inductor current higher to meet load requirements with minimal delays. Traditional current mode or voltage mode control schemes typically must monitor the feedback voltage, current signals (also for current limit), and internal ramps and compensation signals, to determine when to turn off the high-side switch and turn on the synchronous rectifier. Weighing these small signals in a switching environment is difficult to do just after switching large currents, making those architectures problematic at low duty cycles and in less than ideal board layouts. Because no switching decisions are made during noisy time periods, COT architectures are preferable in low duty cycle and noisy applications. However, traditional COT is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6207A/B control schemes suffer from some disadvantages that preclude their use in many cases. Many applications require a known switching frequency range to avoid interference with other sensitive circuitry. True constant on-time control, where the on-time is actually fixed, exhibits variable switching frequency. In a step-down converter, the duty factor is proportional to the output voltage and inversely proportional to the input voltage. Therefore, if the on-time is fixed, the off-time (and therefore the frequency) must change in response to changes in input or output voltage. Modern pseudo-fixed frequency COT architectures greatly improve COT by making the one-shot on-time proportional to VOUT and inversely proportional to VIN. In this way, an on-time is chosen as approximately what it would be for an ideal fixed-frequency PWM in similar input/output voltage conditions. The result is a big improvement but the switching frequency still varies considerably over line and load due to losses in the switches and inductor and other parasitic effects. Another problem with many COT architectures is their dependence on adequate ESR in the output capacitor, making it difficult to use highly-desirable, small, low-cost, but low-ESR ceramic capacitors. Most COT architectures use AC current information from the output capacitor, generated by the inductor current passing through the ESR, to function in a way like a current mode control system. With ceramic capacitors the inductor current information is too small to keep the control loop stable, like a current mode system with no current information. ACOTTM Control Architecture Making the on-time proportional to VOUT and inversely proportional to VIN is not sufficient to achieve good constant-frequency behavior for several reasons. First, voltage drops across the MOSFET switches and inductor cause the effective input voltage to be less than the measured input voltage and the effective output voltage to be greater than the measured output voltage. As the load changes, the switch voltage drops change causing a switching frequency variation with load current. Also, at light loads if the inductor current goes negative, the switch dead-time between the synchronous rectifier turn-off and the high-side switch turn-on allows the switching node to Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 rise to the input voltage. This increases the effective on time and causes the switching frequency to drop noticeably. One way to reduce these effects is to measure the actual switching frequency and compare it to the desired range. This has the added benefit eliminating the need to sense the actual output voltage, potentially saving one pin connection. ACOTTM uses this method, measuring the actual switching frequency and modifying the on-time with a feedback loop to keep the average switching frequency in the desired range. To achieve good stability with low-ESR ceramic capacitors, ACOTTM uses a virtual inductor current ramp generated inside the IC. This internal ramp signal replaces the ESR ramp normally provided by the output capacitor's ESR. The ramp signal and other internal compensations are optimized for low-ESR ceramic output capacitors. ACOTTM One-Shot Operation The RT6207A/B control algorithm is simple to understand. The feedback voltage, with the virtual inductor current ramp added, is compared to the reference voltage. When the combined signal is less than the reference and the ontime one-shot is triggered, as long as the minimum offtime one-shot is clear and the measured inductor current (through the synchronous rectifier) is below the current limit. The on-time one-shot turns on the high-side switch and the inductor current ramps up linearly. After the on time, the high-side switch is turned off and the synchronous rectifier is turned on and the inductor current ramps down linearly. At the same time, the minimum off-time one-shot is triggered to prevent another immediate on-time during the noisy switching time and allow the feedback voltage and current sense signals to settle. The minimum off-time is kept short (230ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed. Discontinuous Operating Mode (RT6207A Only) After soft-start, the RT6207A operates in fixed frequency mode to minimize interference and noise problems. The RT6207A uses variable-frequency discontinuous switching at light loads to improve efficiency. During discontinuous switching, the on-time is immediately increased to add is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B “hysteresis” to discourage the IC from switching back to continuous switching unless the load increases substantially. The IC returns to continuous switching as soon as an ontime is generated before the inductor current reaches zero. The on-time is reduced back to the length needed for 650kHz switching and encouraging the circuit to remain in continuous conduction, preventing repetitive mode transitions between continuous switching and discontinuous switching. Current Limit The RT6207A/B current limit is a cycle-by-cycle “valley” type, measuring the inductor current through the synchronous rectifier during the off-time while the inductor current ramps down. The current is determined by measuring the voltage between source and drain of the synchronous rectifier. If the inductor current exceeds the current limit, the on-time one-shot is inhibited (Mask high side signal) until the inductor current ramps down below the current limit. Thus, only when the inductor current is well below the current limit is another on time permitted. This arrangement prevents the average output current from greatly exceeding the guaranteed current limit value, as typically occurs with other valley-type current limits. If the output current exceeds the available inductor current (controlled by the current limit mechanism), the output voltage will drop. If it drops below the output under-voltage protection level, the IC will stop switching (see next section). Output Over-Voltage Protection and Under-Voltage Protection If the output voltage VOUT rises above the regulation level and lower 1.2 times regulation level, the high-side switch naturally remains off and the synchronous rectifier turns on. For RT6207B, if the output voltage remains high the synchronous rectifier remains on until the inductor current reaches the low side current limit. If the output voltage still remains high, then IC's switches remain that the synchronous rectifier turns on and high-side MOS keeps off to operate at typical 500kHz switching protection, again if inductor current reaches low side current, the synchronous rectifier will turn off until next protection Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 clock. If the output voltage exceeds the OVP trip threshold (1.25 times regulation level) for longer than 10μs (typical), then IC's output Over-Voltage Protection (OVP) is triggered. RT6207BL chip enters latch mode. For RT6207A If the output voltage Vo rises above the regulation level and lower 1.2 times regulation level, the high-side switch naturally remains off and the synchronous rectifier turns on until the inductor current reaches zero current. If the output voltage remains high, then IC's switches remain off. If the output voltage exceeds the OVP trip threshold (1.25 times regulation level) for longer than 10μs (typical), the IC's OVP is triggered. RT6207AL chip enters latch mode. The RT6207A/B include output Under-Voltage Protection (UVP). If the output voltage drops below the UVP trip threshold for longer than 250μs (typical) then IC's UVP is triggered. Chip into latch or hiccup mode. (see next section). Hiccup Mode The RT6207AH/BH, use hiccup mode for UVP. When the protection function is triggered, the IC will shut down for a period of time and then attempt to recover automatically. Hiccup mode allows the circuit to operate safely with low input current and power dissipation, and then resume normal operation as soon as UVP is removed. During hiccup mode, the shutdown time is determined by the capacitor at SS. A 2μA current source discharges VSS from its starting voltage (normally VPVCC). The IC remains shut down until VSS reaches 0.2V, about 10ms for a 3.9nF capacitor. At that point the IC begins to charge the SS capacitor at 6μA, and a normal start-up occurs. If the fault remains, UVP protection will be enabled when VSS reaches 2.2V (typical). The IC will then shut down and discharge the SS capacitor from the 2.2V level, taking about 4ms for a 3.9nF SS capacitor. Latch-Off Mode The RT6207AL/BL, use latch-off mode OVP and UVP. When the protection function is triggered the IC will shut down in Latch-Off Mode. The IC stops switching, leaving both switches open, and is latched off. To restart operation, toggle EN or power the IC off and then on again. is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT6207A/B Shut-down, Start-Up and Enable (EN) The enable input (EN) has a logic-low level of 0.4V. When VEN is below this level the IC enters shutdown mode and supply current drops to less than 10μA. When VEN exceeds its logic-high level of 2V the IC is fully operational. Between these 2 levels there are 2 thresholds (1.2V typical and 1.4V typical). When VEN exceeds the lower threshold the internal bias regulators begin to function and supply current increases above the shutdown current level. Switching operation begins when VEN exceeds the upper threshold. Input Under-Voltage Lock-Out In addition to the enable function, the RT6207A/B feature an Under-Voltage Lock-Out (UVLO) function that monitors the internal linear regulator output (VIN). To prevent operation without fully-enhanced internal MOSFET switches, this function inhibits switching when VIN drops below the UVLO-falling threshold. The IC resumes switching when VIN exceeds the UVLO-rising threshold. Soft-Start (SS) The RT6207A/B soft-start uses an external pin (SS) to clamp the output voltage and allow it to slowly rise. After VEN is high and VIN exceeds its UVLO threshold, the IC begins to source 6μA from the SS pin. An external capacitor at SS is used to adjust the soft-start timing. Following below equation to get the minimum capacitance range in order to avoid UV occur. T= COUT VOUT 0.6 1.2 I LIM Load Current 0.8 CSS T 6μA VREF Do not leave SS unconnected. During start-up, while the SS capacitor charges, the RT6207A/B operates in discontinuous switching mode with very small pulses. This prevents negative inductor currents and keeps the circuit from sinking current. Therefore, the output voltage may be pre-biased to some positive level before start-up. Once the VSS ramp charges enough to raise the internal reference above the feedback voltage, switching will begin and the output voltage will smoothly rise from the pre- Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 biased level to its regulated level. After VSS rises above about 2.2V output over- and under-voltage protections are enabled and the RT6207A/B begins continuous-switching operation. Internal Regulator (PVCC) An internal linear regulator (PVCC) produces a 5V supply from VIN. The 5V power supplies the internal control circuit, such as internal gate drivers, PWM logic, reference, analog circuitry, and other blocks. 1μF ceramic capacitor for decoupling and stability is required. PGOOD Comparator PGOOD is an open-drain output controlled by a comparator connected to the feedback signal. If FB exceeds 90% of the internal reference or OVP event is cleared, , PGOOD will be high impedance. Otherwise, the PGOOD output is connected to GND. External Bootstrap Capacitor (CBOOT) Connect a 0.1μF low ESR ceramic capacitor between BOOT and SW. This bootstrap capacitor provides the gate driver supply voltage for the high-side N-channel MOSFET switch. Some of case, such like duty ratio is higher than 65% application or input voltage is lower than 5.5V which are recommended to add an external bootstrap diode between an external 5V and BOOT pin for efficiency improvement The bootstrap diode can be a low cost one such as IN4148 or BAT54. The external 5V can be a 5V fixed input from system or a 5V output of the RT6207A/B. Note that the external boot voltage must be lower than 5.5V Over-Temperature Protection The RT6207A/B includes an Over-Temperature Protection (OTP) circuitry to prevent overheating due to excessive power dissipation. The OTP will shut down switching operation when the junction temperature exceeds 150°C. Once the junction temperature cools down by approximately 20°C the IC will resume normal operation with a complete soft-start. For continuous operation, provide adequate cooling so that the junction temperature does not exceed 150°C. is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Absolute Maximum Ratings (Note 1) Supply Voltage, VIN -----------------------------------------------------------------------------------------------Switch Voltage, SW -----------------------------------------------------------------------------------------------Switch Voltage, <10ns --------------------------------------------------------------------------------------------BOOT Voltage -------------------------------------------------------------------------------------------------------EN to GND ------------------------------------------------------------------------------------------------------------Other Pins ------------------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C UQFN-13JL 2x3 (FC) ----------------------------------------------------------------------------------------------Package Thermal Resistance (Note 2) UQFN-13JL 2x3 (FC), θJA -----------------------------------------------------------------------------------------UQFN-13JL 2x3 (FC), θJC ----------------------------------------------------------------------------------------Junction Temperature Range -------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------Storage Temperature Range -------------------------------------------------------------------------------------ESD Susceptibility (Note 3) HBM (Human Body Model) ---------------------------------------------------------------------------------------- Recommended Operating Conditions −0.3V to 21V −0.3V to (VIN + 0.3V) −3V to (VIN + 0.3V) −0.3V to 27.3V −0.3V to 6V −0.3V to 6V 1.54W 64.8°C/W 10°C/W 150°C 260°C −65°C to 150°C 2kV (Note 4) Supply Voltage, VIN ------------------------------------------------------------------------------------------------ 4.5V to 18V Junction Temperature Range -------------------------------------------------------------------------------------- −40°C to 125°C Ambient Temperature Range -------------------------------------------------------------------------------------- −40°C to 85°C Electrical Characteristics (VIN = 12V, TA = 25°C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit 4.5 -- 18 V 4 4.2 4.4 Supply Voltage VIN Supply Input Operating Voltage VIN Under-Voltage Lockout Threshold VUVLO Under-Voltage Lockout Threshold Hysteresis VUVLO Shutdown Current ISHDN Quiescent Current IQ V -- 0.5 -- VEN = 0V -- 1.5 10 A VEN = 2V, VFB = 0.7V -- 0.8 1.2 mA 1.1 1.2 1.3 V -- 200 -- mV Enable Voltage Enable Voltage Threshold VEN Rising Enable Voltage Hysteresis Feedback Voltage Feedback Voltage Threshold VFB 4.5V VIN 18V 0.693 0.7 0.707 V Feedback Input Current IFB VFB = 0.71V 0.1 -- 0.1 A Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT6207A/B Parameter Symbol Test Conditions Min Typ Max Unit 6V VIN 18V, 0 < IPVCC 5mA -- 5 -- V PVCC Line Regulation 6V VIN 18V, IPVCC = 5mA -- -- 20 PVCC Load Regulation 0 IPVCC 20mA -- -- 100 IPVCC VIN = 6V, VPVCC = 4V -- 150 -- High-Side On-Resistance RDS(ON)_H VBOOT VSW = 5V -- 60 -- Low-Side On-Resistance RDS(ON)_L -- 22 -- ILIM 5.6 6.7 7.9 TSD -- 150 -- -- 20 -- -- 153 -- PVCC Output PVCC Output Voltage PVCC Output Current VPVCC mV mA Internal MOSFET m Current Limit Low-Side Switch Valley Current Limit A Thermal Shutdown Thermal Shutdown Threshold Thermal Shutdown Hysteresis TSD C On-Time Timer Control On-Time tON VOUT = 1.2V Minimum On-Time tON(MIN) -- 60 -- Minimum Off-Time tOFF(MIN) -- 230 -- VSS = 0V 5 6 7 VFB Rising (Good) 85 90 95 VFB Rising (Fault) -- 120 -- ns Soft-Start Internal Charge Current A Power Good PGOOD Good Rising Threshold %VFB PGOOD Good Falling Threshold VFB Falling (Good) -- 112 -- VFB Falling (Fault) -- 80 -- PGOOD Sink Current PGOOD = 0.1V 10 20 -- mA 115 120 125 %VFB -- 10 -- s UVP Detect 55 60 65 Hysteresis -- 10 -- -- 250 -- s -- tSS x 1.7 -- -- Output Under Voltage and Over Voltage Protections OVP Trip Threshold OVP Detect OVP Propagation Delay UVP Trip Threshold UVP Propagation Delay UVP Enable Delay Relative to Soft-Start Time %VFB Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. θJA is measured at TA = 25°C on a highly thermal conductive 4-Layer test board. θJC is measured at the top of the package. Note 3. Devices are ESD sensitive. Handling precaution is recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Typical Application Circuit 9, 10 VIN CIN 10µF x 2 VPVCC RPG Enable CSS RT6207A/B 12 SW VIN BOOT VOUT CBOOT 0.1µF 8 PGOOD 2 EN FB 3 5 SS PVCC 6 GND 1, 13 Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 11 L CFF VPVCC CPVCC R1 COUT 22µF x 3 R2 1µF is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT6207A/B Typical Operating Characteristics Efficiency vs. Output Current 100 90 90 80 80 VIN = 4.5V VIN = 12V VIN = 18V 70 60 Efficiency (%) Efficiency (%) Efficiency vs. Output Current 100 50 40 30 20 VIN = 4.5V VIN = 12V VIN = 18V 70 60 50 40 30 20 10 10 RT6207A, VOUT = 1.2V 0 0 1 2 3 4 RT6207B, VOUT = 1.2V 0 5 0 1 2 Output Current (A) Efficiency vs. Output Current 100 90 90 VOUT = 5V VOUT = 3.3V VOUT = 1.2V 60 50 40 30 20 70 60 50 40 30 20 10 10 RT6207A, VIN = 12V 0 RT6207B, VIN = 12V 0 0 1 2 3 4 5 0 1 2 Output Current (A) 1.23 1.22 1.22 1.21 Output Voltage (V) 1.23 RT6207A 1.20 RT6207B 1.19 4 5 Output Voltage vs. Input Voltage 1.24 1.21 3 Output Current (A) Output Voltage vs. Output Current Output Voltage (V) 5 VOUT = 5V VOUT = 3.3V VOUT = 1.2V 80 Efficiency (%) Efficiency (%) 70 4 Efficiency vs. Output Current 100 80 3 Output Current (A) 1.18 1.17 1.20 1.19 IOUT = 1A IOUT = 3A IOUT = 5A 1.18 1.17 1.16 VIN = 12V, VOUT = 1.2V 1.16 0 1 2 3 4 Output Current (A) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 VOUT = 1.2V 1.15 5 4 6 8 10 12 14 16 18 Input Voltage (V) is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Switching Frequency vs. Output Current Switching Frequency vs. Temperature 850 700 RT6207A 600 RT6207B 800 Switching Frequency (kHz)1 Switching Frequency (kHz)1 800 500 400 300 200 100 750 700 650 600 550 500 IOUT = 1A 0 450 0 1 2 3 4 5 -50 -25 0 Output Current (A) 75 100 125 Feedback Voltage vs. Temperature 0.73 0.73 0.72 0.72 Feedback Voltage (V) Feedback Voltage (V) 50 Temperature (°C) Feedback Voltage vs. Input Voltage 0.71 0.70 0.69 0.68 0.67 0.71 0.70 0.69 0.68 0.67 0.66 0.66 4 6 8 10 12 14 16 18 -50 -25 0 Input Voltage (V) 25 50 75 100 125 Temperature (°C) Inductor Valley Current Limit vs. Input Voltage Inductor Valley Current Limit vs. Temperature 7.0 7.0 Inductor Valley Current Limit (A)1 Inductor Valley Current Limit (A)1 25 6.5 6.0 5.5 5.0 4.5 4.0 6.5 6.0 5.5 5.0 4.5 4.0 4 6 8 10 12 14 16 Input Voltage (V) Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 18 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT6207A/B 3.5 3.0 2.5 2.0 1.5 1.0 VEN = 0V 0.5 4 6 8 10 12 14 16 Shutdown Quiescent Current vs. Temperature Shutdown Quiescent Current (μA)1 Shutdown Quiescent Current (μA)1 Shutdown Quiescent Current vs. Input Voltage 18 15 12 9 6 3 VEN = 0V 0 18 -50 -25 0 Input Voltage (V) 800 800 700 600 500 400 8 10 12 14 16 600 500 400 300 18 -50 -25 0 Input UVLO vs. Temperature 50 75 100 125 Enable Threshold vs. Temperature 1.5 4.4 1.4 Enable Threshold (V) Input UVLO (V) 25 Temperature (°C) 4.6 4.2 Rising 3.8 3.6 125 700 Input Voltage (V) 4.0 100 VEN = 2V, VFB = 0.7V VEN = 2V, VFB = 0.7V 6 75 Quiescent Current vs. Temperature 900 Quiescent Current (μA) Quiescent Current (μA) Quiescent Current vs. Input Voltage 4 50 Temperature (°C) 900 300 25 Falling 3.4 1.3 Rising 1.2 1.1 1.0 Falling 0.9 3.2 0.8 -50 -25 0 25 50 75 100 Temperature (°C) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 125 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Load Transient Response VOUT (100mV/Div) IOUT (2A/Div) Load Transient Response VOUT (100mV/Div) RT6207A, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 5A IOUT (2A/Div) RT6207B, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 5A Time (100μs/Div) Time (100μs/Div) Voltage Ripple Voltage Ripple VSW (10V/Div) VSW (10V/Div) VOUT (10mV/Div) VOUT (10mV/Div) VIN = 12V, VOUT = 1.2V, IOUT = 5A VIN = 12V, VOUT = 1.2V, IOUT = 1A Time (1μs/Div) Time (1μs/Div) Power On from Input Voltage Power Off from Input Voltage VIN (20V/Div) VIN (20V/Div) VSW (10V/Div) VSW (10V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (5A/Div) IOUT (5A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 5A Time (25ms/Div) Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 VIN = 12V, VOUT = 1.2V, IOUT = 5A Time (25ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6207A/B Power On from Enable Voltage Power Off from Enable Voltage VEN (5V/Div) VEN (5V/Div) VPGOOD (5V/Div) VPGOOD (5V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (5A/Div) IOUT (5A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 5A Time (25ms/Div) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 VIN = 12V, VOUT = 1.2V, IOUT = 5A Time (2.5ms/Div) is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Application information Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (ΔIL) about 20% to 50% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (fSW), the maximum output current (IOUT(MAX)) and estimating a ΔIL as some percentage of that current. L= VOUT VIN VOUT VIN fSW IL Once an inductor value is chosen, the ripple current (ΔIL) is calculated to determine the required peak inductor current. VOUT VIN VOUT I IL = and IL(PEAK) = IOUT(MAX) L VIN fSW L 2 To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and short circuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. IL(PEAK)should not exceed the minimum value of IC's upper current limit level or the IC may not be able to meet the desired output current. If needed, reduce the inductor ripple current (ΔIL) to increase the average inductor current (and the output current) while ensuring that IL(PEAK) does not exceed the upper current limit level. Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Considering the Typical Operating Circuit for 1.2V output at 5A and an input voltage of 12V, using an inductor ripple of 1A (20%), the calculated inductance value is : L= 1.2 12 1.2 = 1.66μH 12 650kHz 1A The ripple current was selected at 1A and, as long as we use the calculated 1.8μH inductance, that should be the actual ripple current amount. The ripple current and required peak current as below : IL = 1.2 12 1.2 = 0.923A 12 650kHz 1.8μH and IL(PEAK) = 5A 0.923A = 5.4615A 2 For the 1.8μH value, the inductor's saturation and thermal rating should exceed at least 5.4615A. For more conservative, the rating for inductor saturation current must be equal to or greater than switch current limit of the device rather than the inductor peak current. Input Capacitor Selection The input filter capacitors are needed to smooth out the switched current drawn from the input power source and to reduce voltage ripple on the input. The actual capacitance value is less important than the RMS current rating (and voltage rating, of course). The RMS input ripple current (IRMS) is a function of the input voltage, output voltage, and load current : V IRMS = IOUT(MAX) OUT VIN VIN 1 VOUT Ceramic capacitors are most often used because of their low cost, small size, high RMS current ratings, and robust surge current capabilities. However, take care when these capacitors are used at the input of circuits supplied by a wall adapter or other supply connected through long, thin wires. Current surges through the inductive wires can induce ringing at the RT6207A/B input which could is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT6207A/B potentially cause large, damaging voltage spikes at VIN. If this phenomenon is observed, some bulk input capacitance may be required. Ceramic capacitors (to meet the RMS current requirement) can be placed in parallel with other types such as tantalum, electrolytic, or polymer (to reduce ringing and overshoot). Output Transient Undershoot and Overshoot Choose capacitors rated at higher temperatures than required. Several ceramic capacitors may be paralleled to meet the RMS current, size, and height requirements of the application. The typical operating circuit uses two 10μF and one 0.1μF low ESR ceramic capacitors on the input. However, the combination of small ceramic output capacitors (with little capacitance), low output voltages (with little stored charge in the output capacitors), and low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 650kHz switching frequency. Output Capacitor Selection The RT6207A/B are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output Ripple Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE = VRIPPLE(ESR) VRIPPLE(C) VRIPPLE(ESR) = IL RESR VRIPPLE(C) = IL 8 COUT fSW For the Typical Operating Circuit for 1.2V output and an inductor ripple of 0.923A, with 3 x 22μF output capacitance each with about 5mΩ ESR including PCB trace resistance, the output voltage ripple components are : VRIPPLE(ESR) = 0.923A 5m = 4.615mV VRIPPLE(C) = 0.923A = 2.689mV 8 66μF 650kHz VRIPPLE = 4.615mV+2.689mV = 7.304mV Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. But some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. The output voltage transient undershoot and overshoot each have two components : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value. The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR _STEP = ΔIOUT x RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOTTM control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasites) and maximum duty cycle for a given input and output voltage as : is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B tON = VOUT tON and DMAX = VIN fSW tON tOFF(MIN) The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increase compensates for the voltage losses. Calculate the output voltage sag as : L (IOUT )2 VSAG = 2 COUT VIN(MIN) DMAX VOUT The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR = L (IOUT )2 2 COUT VOUT For the Typical Operating Circuit for 1.2V output, the circuit has an inductor 1.8μH and 3 x 22μF output capacitance with 5mΩ ESR each. The ESR step is 5A x 1.67mΩ = 8.35mV which is small, as expected. The output voltage sag and soar in response to full 0A-5A-0A instantaneous transients are : tON = 1.2V = 153ns 12V 650kHz and DMAX = 153ns = 0.399 153ns 230ns where 230ns is the minimum off time VSAG = 1.8μH (5A)2 = 95mV 2 66μF 12V 0.399 1.2V VSOAR = 1.8μH (5A)2 = 284mV 2 66μF 1.2V The sag is about 7.92% of the output voltage and the soar is a full 23.7% of the output voltage. The ESR step is negligible here but it does partially add to the soar, so keep that in mind whenever using higher-ESR output capacitors. The soar is typically much worse than the sag in high input, low-output step-down converters because the high input voltage demands a large inductor value which stores lots of energy that is all transferred into the output if the load stops drawing current. Also, for a given inductor, the soar for a low output voltage is a greater voltage change and an even greater percentage of the output voltage. Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 Any sag is always short-lived, since the circuit quickly sources current to regain regulation in only a few switching cycles. With the RT6207B, any overshoot transient is typically also short-lived since the converter will sink current, reversing the inductor current sharply until the output reaches regulation again. The RT6207A discontinuous operation at light loads prevents sinking current so, for that IC, the output voltage will soar until load current or leakage brings the voltage down to normal. Most applications never experience instantaneous full load steps and the RT6207A/B high switching frequency and fast transient response can easily control voltage regulation at all times. Also, since the sag and soar both are proportional to the square of the load change, if load steps were reduced to 1A (from the 5A examples preceding) the voltage changes would be reduced by a factor of almost ten. For these reasons sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. In any application with large quick transients, always calculate soar to make sure that over-voltage protection will not be triggered. Under-voltage is not likely since the threshold is very low (60%), that function has a long delay (250μs), and the IC will quickly return the output to regulation. Over-voltage protection has a minimum threshold of 120% and short delay of 10μs and can actually be triggered by incorrect component choices, particularly for the RT6207A which does not sink current. Feed-forward Capacitor (Cff) The RT6207A/B are optimized for ceramic output capacitors and for low duty cycle applications. However for high-output voltages, with high feedback attenuation, the circuit's response becomes over-damped and transient response can be slowed. In high-output voltage circuits (VOUT > 3.3V) transient response is improved by adding a is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT6207A/B small “feed-forward” capacitor (Cff) across the upper FB divider resistor (Figure 1), to increase the circuit's Q and reduce damping to speed up the transient response without affecting the steady-state stability of the circuit. Choose a suitable capacitor value that following below step. Get the BW the quickest method to do transient response form no load to full load. Confirm the damping frequency. The damping frequency is BW. VOUT Cff EN VIN FB RT6207A/B For automatic start-up the low-voltage EN pin can be connected to VIN through a 100kΩ resistor. Its large hysteresis band makes EN useful for simple delay and timing circuits. EN can be externally pulled to VIN by adding a resistor-capacitor delay (REN and CEN in Figure 2). Calculate the delay time using EN's internal threshold where switching operation begins. An external MOSFET can be added to implement digital control of EN when no system voltage above 2V is available (Figure 3). In this case, a 100kΩ pull-up resistor, REN, is connected between VIN and the EN pin. MOSFET Q1 will be under logic control to pull down the EN pin. To prevent enabling circuit when VIN is smaller than the VOUT target value or some other desired voltage level, a resistive voltage divider can be placed between the input voltage and ground and connected to EN to create an additional input under voltage lockout threshold (Figure 4). BW R1 Enable Operation (EN) R2 REN EN RT6207A/B CEN GND GND Figure 1. Cff Capacitor Setting Cff can be calculated base on below equation : 1 Cff 2 3.1412 R1 BW 0.8 Figure 2. External Timing Control VIN REN 100k EN Q1 Enable RT6207A/B Soft-Start (SS) The RT6207A/B soft-start uses an external capacitor at SS to adjust the soft-start timing according to the following equation : CSS nF 0.7V t ms ISS μA Following below equation to get the minimum capacitance range in order to avoid UV occur. COUT VOUT 0.6 1.2 T (ILIM Load Current) 0.8 T 6μA CSS VREF GND Figure 3. Digital Enable Control Circuit VIN REN1 REN2 EN RT6207A/B GND Figure 4. Resistor Divider for Lockout Threshold Setting Do not leave SS unconnected. Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 18 is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Output Voltage Setting External BOOT Capacitor Series Resistance Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected to FB. The output voltage is set according to the following equation : The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, but also slow enough to reduce EMI. Switch turn-on is when most EMI occurs since VSW rises rapidly. During switch turn-off, SW is discharged relatively slowly by the inductor current during the dead time between high-side and low-side switch on-times. In some cases it is desirable to reduce EMI further, at the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (<47Ω) resistance between BOOT and the external bootstrap capacitor. This will slow the high-side switch turn-on and VSW's rise. To remove the resistor from the capacitor charging path (avoiding poor enhancement due to undercharging the BOOT capacitor), use the external diode shown in Figure 6 to charge the BOOT capacitor and place the resistance between BOOT and the capacitor/diode connection. VOUT = 0.7V x (1 + R1 / R2) VOUT R1 FB RT6207A/B R2 GND Figure 5. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose R2 between 10kΩ and 100kΩ to minimize power consumption without excessive noise pick-up and calculate R1 as follows : R2 (VOUT VREF ) VREF For output voltage accuracy, use divider resistors with 1% or better tolerance. R1 External BOOT Bootstrap Diode When the input voltage is lower than 5.5V it is recommended to add an external bootstrap diode between VIN (or VINR) and the BOOT pin to improve enhancement of the internal MOSFET switch and improve efficiency. The bootstrap diode can be a low cost one such as 1N4148 or BAT54. 5V 0.1µF SW Figure 6. External Bootstrap Diode Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 Decouple PVCC to GND with a 1μF ceramic capacitor. High grade dielectric (X7R, or X5R) ceramic capacitors are recommended for their stable temperature and bias voltage characteristics. Thermal Considerations For continuous operation, do not exceed absolute maximum junction temperature. The maximum power dissipation depends on the thermal resistance of the IC package, PCB layout, rate of surrounding airflow, and difference between junction and ambient temperature. The maximum power dissipation can be calculated by the following formula : PD(MAX) = (TJ(MAX) − TA) / θJA BOOT RT6207A/B PVCC Capacitor Selection where TJ(MAX) is the maximum junction temperature, TA is the ambient temperature, and θJA is the junction to ambient thermal resistance. For recommended operating condition specifications, the maximum junction temperature is 125°C. The junction to ambient thermal resistance, θJA, is layout dependent. For UQFN-13JL 2x3 (FC) package, the thermal resistance, θJA, is 64.8°C/W on a standard four-layer thermal test is a registered trademark of Richtek Technology Corporation. www.richtek.com 19 RT6207A/B board. The maximum power dissipation at TA = 25°C can be calculated by the following formula : Layout Consideration PD(MAX) = (125°C − 25°C) / (64.8°C/W) = 1.54W for UQFN-13JL 2x3 (FC) package The maximum power dissipation depends on the operating ambient temperature for fixed T J(MAX) and thermal resistance, θJA. The derating curve in Figure 7 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation. Maximum Power Dissipation (W)1 2.4 Follow the PCB layout guidelines for optimal performance of the device. Keep the traces of the main current paths as short and wide as possible. Put the input capacitor as close as possible to VIN and VIN pins. SW node is with high frequency voltage swing and should be kept at small area. Keep analog components away from the SW node to prevent stray capacitive noise pickup. Connect feedback network behind the output capacitors. Keep the loop area small. Place the feedback components near the device. Connect all analog grounds to common node and then connect the common node to the power ground behind the output capacitors. An example of PCB layout guide is shown in Figure 8 for reference. Four-Layer PCB 2.0 1.6 1.2 0.8 0.4 0.0 0 25 50 75 100 125 Ambient Temperature (°C) Figure 7. Derating Curve of Maximum Power Dissipation Place the input and output capacitors as close to the IC as possible. SW should be connected to inductor by wide and short trace, and keep sensitive components away from this trace. VOUT CBOOT L R2 SW BOOT 12 11 CIN GND 1 10 VIN EN 2 9 VIN FB 3 8 PGOOD NC 4 7 NC CSS 5 6 PVCC Place the feedback as close to the IC as possible. 13 SS R1 GND COUT GND GND CPVCC Figure 8. PCB Layout Guide Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 20 is a registered trademark of Richtek Technology Corporation. DS6207A/B-01 July 2015 RT6207A/B Outline Dimension Symbol Dimensions In Millimeters Dimensions In Inches Min. Max. Min. Max. A 0.500 0.600 0.020 0.024 A1 0.000 0.050 0.000 0.002 A3 0.100 0.152 0.004 0.006 b 0.200 0.300 0.008 0.012 b1 0.370 0.470 0.015 0.019 D 1.900 2.100 0.075 0.083 E 2.900 3.100 0.114 0.122 K 0.750 0.030 e 0.500 0.020 e1 0.585 0.023 e2 0.500 0.020 L 0.400 0.500 0.016 0.020 L1 0.950 1.050 0.037 0.041 L2 0.325 0.425 0.013 0.017 L3 1.325 1.425 0.052 0.056 U-Type 13JL QFN 2x3 (FC) Package Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6207A/B-01 July 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 21 RT6207A/B Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. www.richtek.com 22 DS6207A/B-01 July 2015