AN607 Vishay Siliconix DC-to-DC Design Guide Serge Jaunay, Jess Brown INTRODUCTION Manufacturers of electronic systems that require power conversion are faced with the need for higher-density dc-to-dc converters that perform more efficiently, within a smaller footprint, and at lower cost—despite increasing output loads. To meet these demands, Siliconix has combined advanced TrenchFETR and PWM-optimized process technologies, along with innovative new packages, to provide: D lowest on-resistance for minimum power dissipation D lowest gate charge for minimum switching losses D dV/dt shoot-through immunity1 D improved thermal management. Breakthroughs in thermal management for increasing power density are being achieved with Vishay Siliconix packaging technologies such as the PowerPAKt (Si7000 Series), the thick leadframe D2PAK (SUM Series), and ChipFETt (Si5000 Series). D The PowerPAK SO-8 offers the steady-state thermal resistance of a DPAK in an SO-8 footprint. D The PowerPAK 1212-8 is approximately half the size of a TSSOP-8 while decreasing the thermal resistance by an order of magnitude. D The SUM Series reduces thermal resistance by 33% over standard D2PAK packaging. D ChipFET is 40% smaller than a TSOP-6 package while offering lower on-resistance and lower thermal resistance. It should be noted that lower thermal resistance results in higher possible maximum current and power dissipation. The complete array of Vishay Siliconix MOSFET-packaged products ranges from the D2PAK (SUM or SUB series), DPAK (SUD Series), and PowerPAK (Si7000 Series) types of packages to the LITTLE FOOTR packages. These small outline devices range from the SO-8 down to the tiniest MOSFET available - the LITTLE FOOT SC-89. BACKGROUND MATERIAL Switching Characteristics The basic characteristics of a MOSFET are key to understanding how these devices work in switchmode power supplies. In reality the freewheel diode will have some form of reverse recovery effect (¢a and b, Figure 2), and as a result, the current through the drain source of the MOSFET (Q1, Figure 1) will increase. To accommodate the extra drain-source current, VGS must increase above the value necessary to support the load current. The gate voltage keeps rising until the device is carrying the combined load and recovery current (period ¢). Therefore, the recovery current of the freewheel diode adds to the load current seen by the controlling MOSFET (Q1). At the end of period ¢a, the reverse recovery current falls, along with the gate-source voltage. This is because the diode has recovered. The recovery current in turn will decay to zero, resulting in the gate voltage reducing to the original value required to support the load current (period ¢b). During this period, the freewheel diode starts to support voltage, and the VDS voltage falls, and the Miller Plateau begins. As with the ideal-recovery diode explanation, this continues until the voltage falls to its on-state value (end of ¢) and the gate-source voltage is unclamped and continues to the applied gate-voltage value. Turn-off is effectively the reverse of turn-on, apart from that there is no limitation by the freewheeling diode (in this particular circuit). For turn-off the Miller Plateau indicates the start of the rise of the drain-source voltage, and the voltage of the Miller Plateau will represent the required VGS to sustain the load current. The turn-off delay is the period from when the gate voltage falls from its on-state value to when it reaches the Miller Plateau value (i.e. load-current value). A simple buck converter, shown in Figure 1, shows the behavior of the MOSFET during turn-on and turn-off when switching an inductive load. During these periods, a positive step input is applied to turn the device on, and a step transition, from positive to zero, is applied to turn the MOSFET off. With a positive step-input voltage on the gate, the voltage across the gate-source of the MOSFET (VGS) ramps up according to the time constant formed by the gate resistance (Rg) and input capacitance (Ciss), as shown in Figure 2a (period ¡). Once VGS reaches the threshold voltage (Vth), the channel is turned on, and the current through the device starts to ramp up (period ©). At the end of period ©, there are two possible switching transients that VGS could follow. In the first case, the freewheel diode (D1, Figure 1) is assumed to have an ideal reverse recovery, represented by the solid waveforms in Figure 2. Once the channel is supporting the full-load current, the voltage across the device can begin to decay (the end of point ©) because the diode is now able to support voltage. As the drain-source voltage falls, the gate-source voltage stays approximately constant. This phenomenon is called the ”Miller Plateau,” and it continues until the voltage a) Specifically designed to prevent spurious turn-on during high rates of dV/dt b) SUM is an improved D2PAK package, with lower rDS(on) and thermal resistance Document Number: 71917 10-Oct-02 www.vishay.com 1 AN607 Vishay Siliconix falls to its on-state value. At the end of period ¢ (Figure 2), the gate-source voltage is unclamped and continues to the applied gate-voltage value. This additional gate voltage fully enhances the MOSFET channel and reduces the rDS(on). In reality the freewheel diode will have some form of reverse recovery effect (¢a and b, Figure 2), and as a result, the current through the drain source of the MOSFET (Q1, Figure 1) will increase. To accommodate the extra drain-source current, VGS must increase above the value necessary to support the load current. The gate voltage keeps rising until the device is carrying the combined load and recovery current (period ¢). Therefore, the recovery current of the freewheel diode adds to the load current seen by the controlling MOSFET (Q1). At the end of period ¢a, the reverse recovery current falls, along with the gate-source voltage. This is because the diode has recovered. The recovery current in turn will decay to zero, resulting in the gate voltage reducing to the original value required to support the load current (period ¢b). During this period, the freewheel diode starts to support voltage, and the VDS voltage falls, and the Miller Plateau begins. As with the ideal-recovery diode explanation, this continues until the voltage falls to its on-state value (end of ¢) and the gate-source voltage is unclamped and continues to the applied gate-voltage value. Turn-off is effectively the reverse of turn-on, apart from that there is no limitation by the freewheeling diode (in this particular circuit). For turn-off the Miller Plateau indicates the start of the rise of the drain-source voltage, and the voltage of the Miller Plateau will represent the required VGS to sustain the load current. The turn-off delay is the period from when the gate voltage falls from its on-state value to when it reaches the Miller Plateau value (i.e. load-current value). L Q1 VIN D1 C VOUT FIGURE 1. Typical circuit of a buck converter FIGURE 2. Switching waveforms for a typical MOSFET in a buck converter Note: The solid line shows an idealized curve with no recovery of the anti-parallel diode. The dotted line shows the effect of www.vishay.com 2 reverse recovery of the freewheel diode on the gate waveform and the corresponding switching waveforms. Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix Driving MOSFETs: N- and P-Channel. a) use an isolated supply with the 0 V referenced to the source voltage to ensure that the applied VGS is the same as the voltage driving the gate; There are two fundamental types of MOSFETs: n-channel and p-channel. An n-channel device needs a positive gate voltage with respect to the source voltage, whereas a p-channel MOSFET requires the gate voltage to be negative with respect to the source. Due to these criteria, each device sometimes appears to be geared for specific applications, such as p-channels for load switches and n-channels for low-side switches. In reality it is only the drive circuits that need to be different. b) use a charge-pump circuit that generates a voltage higher than the dc-link voltage to drive the gate; or D VGS G V S S FIGURE 3. Schematic of an n-channel MOSFET S S V G VGS c) use a bootstrap circuit that again generates a voltage higher than the dc-link voltage, but which requires a switching circuit to charge up the bootstrap capacitor after the top device is turned off. Another method is to use a p-channel device in situations in which the gate voltage does not need to be higher than the dc-link voltage. This is appropriate when the drain voltage of the MOSFET is less than 20 V because the gate signal can be derived directly from the input signal. With dc-link voltages >20 V and with limitations of » 20 V on maximum gate voltages, it is necessary to level-shift the applied gate voltage to ensure that the gate voltage does not exceed the maximum value. Therefore, with a dc-link voltage of 50 V, the applied gate voltage must be level-shifted to at least 30 V. It should also be noted that the performance characteristics of a p-channel generally are inferior to those of an n-channel due to the physical structure of the device. For low-side devices, it is generally accepted that n-channel devices are used because the source connection of the MOSFET is connected to power ground. As such, the n-channel MOSFET only will require a positive signal referenced to power ground, whereas a p-channel device would require a negative signal to ground to keep the device turned on. Synchronous Rectification D FIGURE 4. Schematic of a p-channel MOSFET For the high-side switch portrayed in the buck converter of Figure 1, it would be possible to use either a p- or n-channel device; however, the operating conditions of the buck converter will determine which is to be used. Again, consider the high-side MOSFET as shown in Figure 1. Once it is turned on, the source voltage will tend towards the drain voltage (minus the voltage across the device Vs»Vd). Therefore, if the gate drive is generated from the input voltage (Vin) as the MOSFET turns on, VGS will reduce as the source pin (Vs) goes to Vin (Vd). For an n-channel device, this means that the gate voltage must be higher than the drain voltage to maintain VGS above the Miller Plateau voltage to ensure that the MOSFET stays fully on. To achieve this there are three common strategies or circuits: Document Number: 71917 10-Oct-02 Improvements in efficiency can be made by replacing the rectifying diodes, or freewheel diodes, with MOSFETs. This is because the MOSFET has the capability to conduct current in both directions, and reductions in conduction loss can be achieved due to the I2R losses of the MOSFET being lower than the IV losses associated with the diode. However, the circuit and load conditions will determine whether the increase in efficiency offsets the extra cost, and sometimes additional circuitry, demanded by synchronous MOSFETs. It should be noted that the freewheel diode (D1, Figure 1), or rectifying diode, is still required to prevent both MOSFETs conducting at the same time -- the necessity of dead time between Q1 and Q2 results in a short period of diode conduction -- and causing shoot-through conditions. However, with the inclusion of a MOSFET, it is possible to use the inherent body diode, though this typically demonstrates performance inferior to that of an external Schottky diode. As a result, it is sometimes beneficial to use a Schottky diode as the anti-parallel diode bypassing the inherent body diode and resulting in an improvement of the conduction and recovery performance of the freewheel diode. www.vishay.com 3 AN607 Vishay Siliconix NON-ISOLATED TOPOLOGIES di L V − Vout = in L dt Non-isolated Buck Converter Basic operation [1] L Q1 The buck (or step-down) converter, shown in Figure 5, is used to convert a positive dc voltage to a lower positive dc voltage. It can be a bi-directional converter, but for simplicity’s sake, consider only the power flow from the higher voltage to the lower voltage. Q2 Sch2 VIN VOUT D2 IL L Q1 FIGURE 7. Turn-off of Q1 Q2 VIN Sch2 D2 C1 VOUT Once Q1 is turned off (Figure 7), the current flowing through the inductor cannot be reduced to zero instantaneously. Rather, the current requires a freewheel path, which will be Q2, D2, or Sch2, depending on the circuit topology. The current decays through the freewheel path according to: FIGURE 5. Basic circuit schematic for a buck converter Note: Q2 is the MOSFET channel, D2 is the body diode of the MOSFET, and Sch2 is an external Schottky diode. The input voltage has to be greater than the output voltage for energy to flow from the input through to the output. L di L V = out L dt Table 1 shows the approximate voltage and current stresses for the buck converter based on continuous-conduction mode. Table 1. Voltage and current stresses for the buck converter Controlling Switch Voltage (ideal) Q2 Vin IL Vout Voltage (practical) Figure 6 shows the turn-on of Q1. Because there is a positive voltage difference between Vin and Vout, there is a current build-up in the inductor according to: 4 Freewheel Element V in V in V in − Ir DS(on) V in + V out or V in + Ir DS(on) Current pk (ideal) Io Current pk (practical) Io + Current rms I o δ FIGURE 6. Turn-on of Q1 www.vishay.com [2] Io ∆I o 2 Io + ∆I o 2 I o 1 − δ Figure 8 shows the idealized waveforms for the buck converter under continuous-current-mode operation. Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix V out = Io2L δ 2T V 2in + V in [5] Because the discontinuous-current mode of operation is dependent on load current, the buck converter normally is operated in continuous-current mode. The high-side switch, Q1, can either be an n-channel or p-channel. If an n-channel MOSFET is used, then some form of charge pump or bootstrap is required to drive the gate of the MOSFET. If a p-channel MOSFET is used, it is sometimes necessary to use a level-shift circuit, depending on the dc-link voltage (Vin). Synchronous rectification in a buck converter FIGURE 8. Current and voltage waveforms for the buck converter in constant-current operation As shown in Figure 8, the input current will be the same as the current though the MOSFET Q1. This means that it will consist of a high-frequency current-square wave. If the input voltage is supplied by a battery, then the switched current will have a degrading effect on the battery life when compared with a continuous-current demand. There are two modes of operation in a buck converter: continuous-current mode and discontinuous-current mode. In continuous mode the inductor current stays above zero for all load conditions, and the output voltage is directly related to duty cycle by the following equation: V out = Vin t on = V inδ T [3] In discontinuous-current operation, the inductor’s minimum current reaches zero. The boundary condition is defined as: Vin(1 − δ)δT Io > 2L [4] Therefore, to maintain continuous current, the load current (Io) must remain above a minimum value determined by the switching period (T) and the inductance (L). During discontinuous-current mode, the output voltage is defined as: Document Number: 71917 10-Oct-02 The simplest method of providing the freewheel path is to use a diode (usually a Schottky diode) that has a low saturation voltage. Recent topologies implement synchronous rectification, where a MOSFET is used to conduct the current during the freewheel period. The MOSFET is turned on just after the freewheel diode goes into conduction, resulting in the current being transferred from the diode to the active region of the MOSFET, and it is turned off just before the controlling MOSFET, Q1, is turned on. The synchronous MOSFET is used because the I2R power losses due to the rDS(on) of the MOSFET will be less than the IV power losses associated with the saturation voltage of the diode. However, even with synchronous rectification there is a small percentage of the switching time that the diode is in conduction, brought about by the necessity to ensure that both MOSFETs are not turned on at the same time. For that reason, in some topologies, a Schottky diode is used for this small period of diode conduction. Because there is little or no voltage present across the MOSFET during turn-off and turn-on, the switching losses of the synchronous MOSFET are reduced considerably. Losses in a buck converter1 The power loss of the MOSFET in the buck converter can be split into four categories: switching losses, on-state losses, off-state losses, and gate losses. However, the leakage currents in power semiconductor devices during the off state are several orders of magnitude smaller than the rated current and hence can be assumed to be negligible. Conduction losses The generic conduction losses can be equated to the product of the saturation voltage of the device (VDS(sat)) under consideration, the current (I) through it, and the time the device is on (ton) of the switching waveform: P con = Vds(sat)It on [6] The conduction voltage across a saturated power semiconductor junction consists of a constant component www.vishay.com 5 AN607 Vishay Siliconix (VTO), plus a component that depends linearly upon current (kTO), as described by Equation 7. [7] Therefore, the conduction power loss of the switching device at a constant duty cycle operation is: t on PT = 1 Tc V ds(sat)I [8] r DS(on) -- On-Resistance (Ω) (Normalized) V ds(sat) = VTO + kTOI 1.8 1.6 On-Resistance vs. Junction Temperature VGS = 10 V ID = 23 A 1.4 1.2 1.0 0.8 0 where Tc is the period of the carrier frequency or: 0.6 --50 --25 0 25 50 75 100 125 150 TJ -- Junction Temperature (_C) P con = (VTO + kTOI)It on Tc FIGURE 9. Normalized rDS(on) versus temperature for a typical device [9] Because a MOSFET is purely a resistive element, Equation 9 can be expressed as: P con = kTOI 2t on = kTOI 2δ Tc [10] = rDS(on)I 2δ Likewise, the conduction power loss of the freewheeling diode is: P D = (VDO + kDOI)(1 − δ)I [11] Conduction losses with synchronous rectification Switching losses Switching losses are difficult to predict accurately and model because the parameters that make up switching transients vary greatly not only with temperature, but also with parasitic elements in the circuit. Furthermore, the gate-drive capability, gate-drive parasitics, and the operating conditions such as current and voltage influence the switching times, which are greatly dependent on individual circuit designs. Therefore, the following expressions for switching losses should be used to obtain an approximation of the performance of the device and should not be used as a definitive model. To develop a loss model for the switching loss, consider an idealized switching waveform as shown in Figure 10. The switching losses can be separated into turn-on (from t1 to t2), turn-off (from t5 to t6), and recovery (t2 to t4) components. With synchronous rectification there will be a time when either the body drain diode or external Schottky diode will be in conduction. This period can be approximated to the dead time (tdeadtime). Hence, Equation 11 should be replaced by the following two equations for synchronous rectification. P con = rDS(on)I 2(1 − δ − t deadtimef sw) P D = (VDO + kDOI)(t deadtimef sw)I [12] [13] The values for the rDS(on) should be taken at a realistic value by estimating the junction temperature of the device and using the normalized curve for the MOSFET. A typical graph is shown in Figure 9. www.vishay.com 6 FIGURE 10. Idealized switching waveform for a MOSFET Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix idealized near-triangular current waveform with Irr being the peak recovery current, the switching device current may be expressed as the following function over the period t4: Therefore, the energy dissipated during turn-on will be: tr E on = V It t dt in o [14] r i = I rr t + Io ta 0 Integrating the instantaneous power over ta (t2 to t3) to get the recovery loss gives: Hence: E on = 1 Vint r Io 2 [15] P on = 1 Vint r I o fs 2 The energy dissipated during turn-off is: [17] The instantaneous loss in the device is Vceic, while the instantaneous loss in the diode is (Vin-Vce)ic. Hence, if the losses can be treated as one, the total power loss in the freewheel diode and device is the same as during ta, which results in the total recovery loss being: And the corresponding power loss is: [18] The recovery losses due to the freewheeling diode occur from time t2 to t4 in Figure 10. At time t2, the current in the switching device increases beyond the load current, owing to the stored charges in the freewheeling diode. At time t3, a depletion region is formed in the freewheel diode. Then, the diode begins to support voltage, the stored charge disappears by recombination, and the collector voltage begins to fall. At time t4, the recovery current can be assumed to be zero because it is within 10% of Irr. The supply voltage, Vin, is completely supported by the switching device from t2 to the end of t3, and hence the majority of the losses are generated in the switching device during this period. Assuming an Table 2. [20] From t3 to t4, the losses are generated in both the freewheeling diode and the main device, and the voltage across the device reaches its on-state value at about the same time as the full recovery of the freewheel diode. [16] E off = 1 VinI ot f 2 Irr E rra = Vint a 2 = Io While the power can be found by: P off = 1 VinI o t f fs 2 [19] I2 + I rr E rr = Vint rr o [21] And hence the power is: I2 + I f rr P rr = Vint rr o s ]22] A summary of a simple power-loss model for the buck and synchronous-buck converters described in the text above, is shown in Table 2. Summary of the generic loss equations for a buck converter Buck P con = r Q1 Q2 D2 or Sch2 Q1 & (D2 or Sch2) Synchronous Buck Iδ P con = rDS(on)I 2oδ 2 DS(on) o P sw = 1 VinI ot f + t 4fs P sw = 1 VinI ot f + t rfs 2 2 P gate = QgV gf sw P gate = QgV gf sw P con = rDS(on)I 2o(1 − δ − δbbm) − P sw ≈ 0Pgate = Q gV gfsw P con = VsatI oδ bbm P con = VsatI o(1 − δ) I2 + I f P rr = Vint rr rr o s I2 + I f P rr = Vint rr rr o s The appropriate Vishay power ICs for non-isolated buck converters are shown in Appendix A. Document Number: 71917 10-Oct-02 www.vishay.com 7 AN607 Vishay Siliconix Non-Isolated Boost Converter L Basic operation of a boost converter IL VIN The boost (or step-up) converter shown in Figure 11 is used to convert a positive dc voltage to a higher positive dc voltage. As with the buck converter, it can have a bidirectional power flow, but for simplicity’s sake, consider only the power flow from the lower voltage to the higher voltage. VVOUT out FIGURE 12. Turn-on of Q1 di L V = in L dt Q2 L Q1 L D2 [23] Q2 D2 Sch2 Q1 Sch2 VIN VVOUT out IL VIN VVOUT out FIGURE 11. Basic circuit schematic for a boost converter FIGURE 13. Turn-off of Q1 The input voltage must be less than the output voltage, otherwise the freewheel diode will be forward-biased, and uncontrolled power will flow. Figure 12 shows the turn-on of Q1, which builds up the current in the inductor according to Equation 23. During this period, the output capacitor will have to support the load current. Table 3. Once Q1 is turned off (Figure 13), the current flowing through the inductor is passed to the freewheel component, this being either the body drain diode, the Schottky diode, or the synchronous MOSFET, depending on the control strategy and the topology implemented. The current through the inductor decays according to Equation 24. di L V − Vin = out L dt [24] Voltage and current stresses for the boost converter Voltage (ideal) Voltage (practical) Controlling Switch V out V out − IR ind Freewheel Element V out V out + V dsat or V out + Ir DS(on) Current pk (ideal) Current pk (practical) Current rms www.vishay.com 8 Io 1−δ Io ∆I + o 2 1−δ I o δ (I − δ) Io 1−δ Io ∆I + o 2 1−δ Io 1 − δ Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix Again, as with the buck converter, there are two modes of operation. In continuous-current mode, the output voltage is related to the duty cycle determined by: V out = Vin 1 1−δ [25] The boundary between continuous and discontinuous operation is given by: I o Vin Tδ(1 − δ) 2L [26] And for discontinuous operation the output voltage can be determined by: V out = Vin V2LIδ T + 1 in 2 [27] o Once more, this is not an ideal solution, as the output voltage is dependent on load current. Synchronous rectification in a boost converters As with the buck converter, there is a freewheel path required for the inductor current during the off-time of Q1. This current path can be provided with a Schottky diode, but with synchronous rectification the MOSFET provides the freewheel path. This reduces the conduction losses of the converter, as described in section 2.3. FIGURE 14. Voltage and current-switching waveforms for a boost converter The controlling MOSFET in this case is referenced to power ground, and therefore the simplest device to use is an n-channel. An advantage to using this topology is the fact that the input current consists of a continuous-current demand with a slight ripple, rather than a switched current. Document Number: 71917 10-Oct-02 Losses in a boost converter Losses The simple loss model for a boost converter is provided in Table 4. These equations are similar to those derived for the buck converter. However, in this case the inductor current is not the same as the load current, and as such, the rms current through the controlling MOSFET Q1 will be: I rms = Io 1−δ [28] www.vishay.com 9 AN607 Vishay Siliconix Table 4. Summary of the generic loss equations for a boost converter Boost P con = rDS(ON) Synchronous Boost I 2oδ I 2oδ P con = rDS(ON) 2 2 (1 − δ) (1 − δ) I o t + trfs P sw = 1 Vin I o tf + trfs P sw = 1 Vin 2 (1 − δ) f 2 (1 − δ) P gate = QgV gf sw P gate = QgV gf sw I 2o (1 − δ − δ bbm P con = rDS(on) 2 − (1 − δ) Q1 Q2 P sw ≈ 0Pgate = Q gV gfsw P con = VsatI o D2 Q1 & D2 P con = Vdsat I2 + (1 −I δ)f P rr = Vint rr rr o s I2 + (1 −I δ)f P rr = Vint rr Non-Isolated Buck-Boost Converter Basic operation of a buck-boost converter rr o s Q1 The buck-boost, or inverting, converter is shown in Figure 15. As its name suggests, this converter either steps up or steps down the input voltage. The voltage output is negative with respect to the input voltage, due to the nature of the operation of the circuit. VIN L IL VOUT Sch2 Q1 D2 FIGURE 16. Turn-on of Q1 Q2 VIN Io δ (1 − δ) bbm L di L V = in L dt VOUT [29] D2 Sch2 FIGURE 15. Basic circuit schematic for the buck-boost converter During turn-on of Q1, the current in the inductor will ramp up according to Equation 29. IL L VIN Q2 VOUT FIGURE 17. Turn-off of Q1 Once Q1 is turned off, the current in the inductor will decay according to Equation 30. However, the current in the inductor will force the output to be negative with respect to the input voltage. didt = VL L o [30] The voltage output for continuous operation is: www.vishay.com 10 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix V o = − V in δ 1−δ Io > [31] Therefore, for @<0.5 the magnitude of the output is smaller than the magnitude of the input, and for @>0.5 the magnitude of the output is greater than the magnitude of the input. [32] For discontinuous operation the output is: V o = Vin The boundary between continuous and discontinuous operation is given by: Table 5. Vin δT(1 − δ) 2L V inTδ2 2LI o [33] Which again is dependent on load current. Voltage and current stresses for the buck converter Controlling Switch V out + |V in| Voltage (Ideal) V out − Ir DS(on) Voltage (practical) Current pk (ideal) Current pk (practical) Freewheel Element V out + |V in| V out + V dsat or − V out + IR Io 1−δ Io 1−δ Io I +∆ o 2 1−δ Io I +∆ o 2 1−δ I o δ 1 − δ) I o δ 1 − δ) Current rms Synchronous rectification in a buck-boost converter conduction losses dissipated in the converter. (See section 2.3.) As with the other non-isolated converters, there is a freewheel path required for the inductor current during the off-time of Q1. Synchronous rectification is provided by a MOSFET as shown in Figure 15. This has the advantage of reducing the Losses in a buck-boost converter Table 6. The generic losses in a buck-boost converter are given below in Table 6. Summary of the generic loss equations for a buck-boost converter Buck-Boost P con = rDS(ON) Q1 Q2 Sch2 or D2 Q1 Sch2 or D2 Document Number: 71917 10-Oct-02 Synchronous Buck-Boost Iδ 2 o P con = rDS(ON) 2 I 2oδ 2 (1 − δ) (1 − δ) Io P sw = 1 Vin t + t rf s P sw = 1 Vin I o t f + t rf s 2 (1 − δ) f 2 (1 − δ) P gate = QgV gf sw P gate = QgV gf sw I 2o (1 − δ − δ bbm) P con = rDS(on) 2 (1 − δ) P sw ≈ 0Pgate = Q gV gfsw P con = VsatI o Io P con = Vdsat δ (1 − δ) bbm I Io I Io P rr = Vint rr rr + f P rr = Vt rr rr + f 2 1−δ s 2 (1 − δ) s www.vishay.com 11 AN607 Vishay Siliconix NON-ISOLATED CONVERTER APPLICATIONS Transformer Isolated Flyback Converter D1 VOUT Point of Load (POL) Converters As required core voltages decrease to levels of 2.5 V and below, point of load converters are becoming more common in power-supply systems. These converters are placed at the point of use and are normally non-isolated because the isolation usually has been achieved by the front-end converter. There is a distributed voltage architecture - at present this is typically between 12 V and 8 V -- and the point of load converters are generally synchronous-buck converters. However, there are trends for this voltage to be as low as 3.3 V. As these distributed voltages go lower, there is more need for boost converters and buck-boost converters. POL converters enable designers to overcome the problems caused by the high peak-current demands and low-noise margins of the latest high-speed digital devices by situating individual, non-isolated, dc sources near their point of use. This helps to minimize voltage drops and noise pick-up/emission, and ensures tight regulation under dynamic load conditions. ISOLATED TOPOLOGIES For some applications galvanic isolation is required to provide high-voltage isolation between the input voltage and output voltage. Therefore, isolated-converter topologies provide galvanic isolation and, due to the presence of a transformer, are able to convert practically any voltage level to another via the medium of the transformer. The transformer ratio is a key element, with the larger number-of-turns ratio providing the greater voltage change, but a badly designed transformer can lead to large inefficiencies in the converter. FIGURE 18. Schematic of an isolated flyback converter The flyback transformer is the simplest of the isolated topologies, and usually it is used for low power levels in the region of 5 W to 100 W. These converters can provide either single or several outputs by the addition of secondary windings. The energy acquired by the transformer during the on-time of the primary MOSFET (Figure 18) is delivered to the output in the non-conducting period of the primary switch. Basic operation of a flyback converter During the conduction period of the primary MOSFET, the current flows from the positive terminal (+) of the primary winding through the switch to ground. A voltage in the opposing direction is generated in the primary and secondary windings. Because the secondary winding is connected with reverse polarity, there can be no current flow to the output due to the blocking diode (D1). VIN One main trend in dc-to-dc converters is the increase in switching frequencies, which results in smaller magnetic components. However, without the introduction of resonant converters, the maximum switching frequency is highly dependent on the maximum available switching speed of the controlling or primary MOSFET. The higher the switching frequency, the higher the switching losses. If switched too quickly, this increased power dissipation could result in catastrophic failure of the MOSFET. www.vishay.com 12 Q1 VIN Iprimary + -- -- + D1 VOUT Q1 FIGURE 19. Schematic showing turn-on of Q1 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix When the primary MOSFET ceases to conduct, the induced voltage is reversed by the collapse of the magnetic field, and the output capacitor is charged through the diode (D1). -+ + D1 -- Isecondary VOUT VIN FIGURE 20. Schematic showing turn-off of Q1 FIGURE 21. Current and voltage waveforms for a flyback converter in discontinuous mode In reality the isolation transformer (shown in Figure 18) is really a storage medium or coupled inductor. Energy is transferred to the output during the blocking or non-conducting period of the switching primary MOSFET. Energy is stored in the inductor when the transistor is turned on, and then it is delivered to the output when the transistor is turned off again. In reality the isolation transformer (shown in Figure 18) is really a storage medium or coupled inductor. Energy is transferred to the output during the blocking, or non-conducting period of the switching primary MOSFET. Energy is stored in the inductor when the transistor is turned on, and then delivered to the output when the transistor is again turned off. Flyback converters are more suitable than forward converters for relatively low power levels because of their lower circuit complexity resulting from the elimination of the output inductor and freewheel diode, which would be present in the secondary stage of a forward converter. Table 7. Voltage and current stresses for the flyback converter Controlling switch Voltage Current rms Document Number: 71917 10-Oct-02 V inNs Np Np V N s out V out + I pk δ 3 I pkNp 0.8 − δ Ns 3 V in + Note: This assumes the converter is discontinuous for 20%. In continuous mode only a portion of the stored energy is transferred to the output. This results in lower peak currents in output capacitors, but because it’s more difficult to stabilize, it is not as common as the discontinuous flyback converter. In continuous mode the output voltage depends on the duty cycle in an identical manner to that of a buck-boost converter, and it is: V o = Vin There are two modes of operation for a flyback converter: discontinuous and continuous. In discontinuous mode, all of the energy stored in the inductor is transferred to the output. This results in a smaller transformer and a feedback loop that is easier to stabilize. Freewheel element Ns δ Np 1 − δ [34] The average diode current also has a similar relationship as the buck-boost: Id = Io 1−δ [35] www.vishay.com 13 AN607 Vishay Siliconix And therefore, the average primary current is: Ip = Ns I o Np 1 − δ [36] Losses in a flyback converter Table 8. Summary of the generic loss equations for a flyback converter Flyback Discontinuous P con = rDS(on) D1 D1 I 2pk δ 3 I pk δ Np t f P sw = 1 Vin + V out 3 f s Ns 2 P gate = QgV g f sw N p (0.8 − δ) P com = Vdsat Ipk Ns 3 Flyback Continuous NN (1 −I δδ) N N Iδ = 1 V + V f N N (1 − δ) 2 2 P con = rDS(on) Q1 P sw 2 o s 2 p p in s out s o p s P gate = QgV g f sw P con = Vdsat I o D1 Transformer Isolated Forward Converter FIGURE 22. Current and voltage waveforms for a flyback converter in continuous mode Synchronous Although synchronous rectification could be implemented in a flyback converter, usually it is not, due to the lower power levels and cost constraints associated with the flyback topology. The forward converter is very similar to the step-down dc-to-dc converter, with the transformer providing galvanic isolation and not being used to store energy. For the topology investigated in this paper, a simple reset winding is included to reset the magnetizing current in the transformer to prevent core saturation2. The circuit used is a self-resonant reset circuit, which resets the magnetizing current and also recovers this magnetizing energy by charging it back to the input. This topology also allows for large ratios of input-to-output voltages. Reset Circuit VIN L D1 D2 Controller VOUT Q1 Feedback FIGURE 23. Schematic of an isolated forward converter www.vishay.com 14 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix Basic operation of a forward converter Reset Circuit VIN The forward converter does come in various configurations and is generally used for power levels from 10 W to 250 W. During the on-time, the power is transferred to the output via the diode D1. This is because D1 is forward-biased and D2 is reverse-biased, and the voltage across the output inductor can be determined by: VL = N2 V − V out N1 in Q3 VOUT Q2 Controller Q1 Feedback [37] During the off-time, the output inductor current circulates via D2 according to: V o = L di dt L FIGURE 25. Self-driven synchronous rectification in a forward converter [38] Reset Circuit L * Q1 VIN Q3 Q2 VOUT FIGURE 26. Synchronous rectification in a forward converter using a discrete driver Table 9. Voltage and current stresses for the forward converter Controlling switch Voltage FIGURE 24. Current and voltage waveforms for a forward converter Current pk Synchronous rectification in a forward converter Synchronous rectification in a forward converter can be relatively easy to achieve, but there are several circuit configurations that can be used. One such circuit is the self-driven topology where the synchronous MOSFETs are driven directly from the secondary side of the transformer (Figure 25). One disadvantage of this circuit is the fact that the gate voltage for the synchronous MOSFETs is not constant. As a result, it is sometimes preferable to opt for a discrete driver solution as shown in Figure 26. Document Number: 71917 10-Oct-02 Current rms Freewheel element Ns Np 2V in 2V in Ns Np Io Io I oNs δ Np I o δ for D1 I o 1 − δ for D2 Losses in a forward converter. The simple loss model for the forward converter is shown in Table 6. This does not take into account synchronous rectification, as the losses will depend on which circuit topology is used. However, a simple approximation would be to substitute the saturation voltage of the diode with the IR product of the MOSFET. www.vishay.com 15 AN607 Vishay Siliconix two large, equal capacitors connected in series across the dc input, providing a constant potential of 1/2 Vin at their junction, as shown in Figure 3. The MOSFET switches SW1 and SW2 are turned on alternatively and are subjected to a voltage stress equal to that of the input voltage. Due to the capacitors providing a mid-voltage point, the transformer sees a positive and negative voltage during switching. The result is twice the desired peak flux value of the core because the transformer core is operated in the first and third quadrant of the B-H loop and it experiences twice the flux excursion as a similar forward-converter core. Table 10. Summary of the generic loss equations for a forward converter Forward NN I δ 2 P con = rDS(on) Q1 s 2 o p N P sw = 1 t f + t rV inI o s f s Np 2 P gate = QgV gf sw D1 D2 P con = VdsatI oδ P con = VdsatI o(1 − δ) Half-Bridge Isolated Converter The half-bridge dc-to-dc converter configuration consists of Isolation Error Amplifier Vin 2nd SW1 SW1 C 1/2 VIN Si9122 SW2 2nd SW2 C GND Synchronous Isolation Drivers Interface FIGURE 27. Schematic of a half-bridge converter www.vishay.com 16 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix References and further reading 1. ”An assessment of the efficiencies of soft and hard switched inverters for applications in powered electric vehicles.” A J Brown, PhD Thesis, December 2000, Department of Electronic and Electrical Engineering, The University of Sheffield, United Kingdom. Document Number: 71917 10-Oct-02 2. AN707. ”Designing a High Frequency, Self--Resonant Reset Single Switch Forward Converter Using Si9118/Si9119 PWM/PSM.” http://www.vishay.com/document/70824/70824.pdf 3. ”Power Electronics, Converters, Applications and Design.” Mohan, Undeland and Robbins, 2nd edition Wiley. ISBN 0--471--58408--8. www.vishay.com 17 AN607 Vishay Siliconix Alphanumeric Index Part Number Si1302DL Si1553DL Si1553DL Si1900DL Si2301DS Si2305DS Si2306DS Si2308DS Si2320DS Si2328DS Si3420DV Si3422DV Si3430DV Si3443DV Si3446DV Si3454ADV Si3454DV Si3456DV Si3458DV Si3460DV Si3552DV Si3552DV Si3812DV Si3850DV Si3850DV Si4300DY Si4308DY Si4308DY Si4356DY Si4362DY Si4364DY Si4366DY Si4376DY Si4376DY Si4404DY Si4406DY Si4408DY Si4412ADY Si4426DY Si4433DY Si4442DY Si4450DY Si4466DY Si4470EY SI4480EY Si4482DY Si4484EY Si4486EY Si4488DY Si4490DY Si4496DY Si4724CY Si4724CY Si4732CY Si4732CY Si4736DY Si4738CY Si4738CY Si4800DY Si4804DY Si4808DY Si4810DY www.vishay.com 18 VDS (V) 30 20 --20 30 --20 --8 30 60 200 100 200 200 100 --20 20 30 30 30 60 20 --30 30 20 20 --20 30 30 30 30 30 30 30 30 30 30 30 20 30 20 --20 30 60 20 60 80 100 100 100 150 200 100 30 30 30 30 30 20 20 30 30 30 30 VGS (V) 20 12 12 20 8 8 20 20 20 20 20 20 20 12 12 20 20 20 20 8 20 20 20 12 12 20 12 20 12 12 16 12 20 12 20 20 20 20 12 8 12 20 12 20 20 20 20 20 20 20 20 12 20 12 25 20 20 20 VGS = 10V 0.4800 0.4800 0.0570 0.1600 7.0000 0.2500 3.7000 5.0000 0.1700 rDS(on) 9 VGS = VGS = 6V 4.5V 0.7000 0.3850 0.9950 0.7000 0.1300 0.0520 0.0940 0.2200 0.1850 0.0600 0.0650 0.0450 0.1000 0.2000 0.1050 0.0185 0.0100 0.0120 0.0060 0.0045 0.0045 0.0048 0.02 0.019 0.0040 0.0045 0.0045 0.0240 0.0045 0.0240 0.0110 0.0350 0.0600 0.0340 0.0250 0.0500 0.0800 0.0250 0.0100 0.0180 0.0220 0.0220 0.0130 0.0300 0.0130 0.0400 0.0800 0.0400 0.0280 0.0900 0.0310 0.1850 0.0650 0.0450 0.0850 0.0950 0.0650 0.1300 0.0270 0.3600 0.1750 0.1250 0.5000 1.0000 0.0330 0.0110 0.0180 0.0075 0.0055 0.0055 0.0055 0.0275 0.023 0.0080 0.0055 0.0068 0.0350 0.0250 0.1100 0.0050 0.0300 0.0090 0.0130 0.0400 0.0800 0.0400 0.0280 0.0900 0.0310 0.0375 0.029 0.0080 0.0240 0.0110 0.009 0.006 0.0330 0.0300 0.0300 0.0200 VGS = 2.5V 0.6300 1.8000 0.1900 0.0710 0.1000 0.0650 0.0320 0.2000 0.0350 0.1600 0.0075 0.0130 Qg (nC) VGS = VGS = 10V 4.5V 0.9 0.5 0.8 1.2 0.9 0.5 5.8 10.0 8.5 4.2 4.8 2.3 1.1 3.3 2.0 2.2 2.1 5.5 3.0 8.5 10.0 9.0 4.5 8.0 3.5 12.0 5.7 8.0 3.9 13.5 4.2 2.4 3.7 2.1 2.1 0.8 1.1 15.0 8.7 40.0 20.0 11.5 30.0 40.0 48.0 48.0 20.0 9.0 28.0 12.5 75.0 36.0 34.0 21.0 16.0 6.5 25.0 4.4 80.0 36.0 31.0 50.0 46.0 25.0 30.0 15.0 30.0 14.0 24.0 14.0 36.0 20.0 30.0 16.5 34.0 20.0 29.0 6.5 46 10 19 38 8.7 13.0 13.0 36.0 QGS (nC) 0.2 0.1 0.5 0.2 0.9 2.0 1.9 0.8 0.3 0.5 0.7 0.5 1.5 2.8 2.5 2.5 1.8 2.8 4.0 2.3 0.9 0.7 0.3 0.3 0.5 2.3 10.0 3.0 7.2 12.8 16.0 17.0 3.8 4.0 15.0 15.0 8.9 3.0 6.5 1.4 8.0 7.7 13.0 11.5 9.0 7.5 7.6 10.0 8.5 7.5 9.9 QGD (nC) 0.1 0.3 0.3 0.1 1.7 2.0 1.4 1.0 0.4 1.5 1.0 0.9 1.4 1.7 2.2 1.5 1.3 1.6 2.0 2.2 0.8 0.7 0.4 0.2 0.2 4.2 8.8 4.5 6.7 7.7 11.0 10.0 3.1 3.2 12.0 10.0 6.4 1.5 4.0 0.7 10.5 8.3 9.0 11.5 5.6 7.0 5.4 8.6 8.5 12.0 10.3 Rg Typ (9) 1.2 Vth (V) 1.0 0.6 0.6 1.0 0.5 0.5 1.0 1.5 2.0 2.0 2.0 2.0 2.0 0.6 0.6 1.0 1.0 1.0 1.0 0.5 1.0 1.0 0.6 0.6 0.6 0.8 0.8 0.8 0.6 0.6 0.8 1.3 1.0 0.8 1.0 1.0 1.0 1.0 0.6 0.5 0.6 2.0 0.6 2.0 2.0 2.0 2.0 2.0 2.0 2.0 2.0 3.5 7.0 37.0 10.0 8.8 0.8 1.0 13.0 PD (W) 0.3 0.3 0.3 0.3 1.3 1.3 1.3 1.3 1.3 1.3 2.1 2.1 2.0 2.0 2.0 2.0 2.0 2.0 2.0 2.0 1.2 1.2 1.2 1.3 1.3 2.5 3.0 2.0 3.0 3.5 3.5 3.5 2.0 2.0 3.5 3.5 3.5 2.5 2.5 2.5 3.5 2.5 2.5 3.8 3.0 2.5 3.8 3.8 3.1 3.1 3.1 1.2 1.2 4.0 4.0 3.1 15.0 7.0 7.0 20.0 2.3 2.0 2.0 8.0 4.2 2.7 2.7 7.0 1.3 1.3 0.8 0.8 0.8 1.0 9.0 7.5 7.5 10.0 2.5 2.0 2.0 2.5 1.9 0.8 1.5 2.0 1.3 1.1 1.3 1.3 1.3 1.3 1.4 ID (A) 0.6 0.7 0.4 0.6 2.3 3.5 3.5 2.0 0.3 1.5 0.5 0.4 2.4 4.4 5.3 4.5 4.2 5.1 3.2 6.8 1.8 2.5 2.4 1.2 0.9 9.0 13.5 9.6 17.0 20.0 20.0 20.0 7.5 7.5 23.0 20.0 21.0 8.0 8.5 3.9 22.0 7.5 13.2 12.7 6.2 4.6 6.9 7.9 5.0 4.0 7.7 5.1 6.5 Package SC70-3 SC70-6 SC70-6 SC70-6 SOT-23 SOT-23 SOT-23 SOT-23 SOT-23 SOT-23 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 TSOP-6 SO-8 SO-14 SO-14 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-16 SO-16 SO-16 SO-16 SO-8 SO-16 SO-16 SO-8 SO-8 SO-8 SO-8 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix Alphanumeric Index (cont’d.) Part Number Si4812DY Si4814DY Si4814DY Si4816DY Si4816DY Si4818DY Si4818DY Si4820DY Si4824DY Si4824DY Si4826DY Si4826DY Si4830ADY Si4830DY Si4832DY Si4834DY Si4835DY Si4836DY Si4837DY Si4838DY Si4840DY Si4842DY Si4848DY Si4850EY Si4852DY Si4854DY Si4856DY Si4858DY Si4860DY Si4862DY Si4864DY Si4866DY Si4876DY Si4884DY Si4888DY Si4892DY Si4894DY Si4896DY Si4924DY Si4924DY Si4926DY Si4926DY Si4942DY Si4946EY Si4980DY Si4982DY VGS = 10V 0.0180 0.0200 0.0210 0.0220 0.0130 0.0220 0.0155 0.0130 0.0175 0.0400 0.0150 0.0220 0.0220 0.0220 0.0180 0.0220 0.0190 rDS(on) 9 VGS = VGS = 6V 4.5V 0.0280 0.0265 0.0325 0.0300 0.0185 0.0300 0.0205 0.0200 0.0270 0.0650 0.0200 0.0300 0.0290 0.0300 0.0280 0.0300 0.0330 0.0030 0.0300 0.0030 0.0120 0.0060 0.0950 0.0950 0.0310 0.0175 0.0300 0.0085 0.0070 0.0110 0.0033 0.0035 0.0055 0.0050 0.0160 0.0100 0.0200 0.0180 0.0220 0.0220 0.0140 0.0300 0.0170 0.0300 0.0280 0.0750 0.0950 0.0950 0.1800 0.1800 VDS (V) 30 30 30 30 30 30 30 30 30 30 30 30 30 30 30 30 --30 12 --30 12 40 30 150 60 30 30 30 30 30 16 20 12 20 30 30 30 30 80 30 30 30 30 40 60 80 100 VGS (V) 20 20 20 20 20 20 20 20 20 20 20 20 12 20 20 20 25 8 20 8 20 20 20 20 20 12 20 20 20 8 8 8 12 20 20 20 20 20 20 20 20 20 20 20 20 20 Si5515DC --20 8 Si5515DC 20 8 Si6410DQ Si6434DQ Si6466DQ Si6802DQ Si6820DQ 30 30 20 20 20 20 20 12 12 20 0.0140 0.0280 Si7358DP 30 20 0.0053 Si7370DP 60 20 0.0110 Si7388DP 30 20 0.007 0.010 Si7404DN 30 12 0.0130 0.0150 Document Number: 71917 10-Oct-02 0.0200 0.0090 0.0045 0.0850 0.0220 0.0120 0.0260 0.0060 0.0053 0.0080 0.0100 0.0070 0.0120 0.0120 0.0165 0.0100 0.0220 0.0125 0.0220 0.0210 0.0550 0.0750 0.1500 VGS = 2.5V 0.0040 0.0040 0.0410 0.0055 0.0047 0.0080 0.0075 0.1310 0.1850 0.0760 0.1030 0.0210 0.0420 0.0140 0.0750 0.1600 Qg (nC) VGS = VGS = 10V 4.5V 27.5 16.0 19.0 9.7 12.0 6.5 14.0 8.0 29.0 15.0 14.0 8.0 29.0 15.0 37.0 20.0 31.0 17.5 11.0 6.5 29.0 15.0 14.0 8.0 5.0 13.0 7.5 27.5 16.0 13.0 7.5 37.0 21.0 56.0 40.0 22.0 40.0 35.0 18.5 55.0 25.0 17.0 10.0 18.0 9.5 41.0 23.0 20.0 9.0 21.0 65.0 30.5 13.0 48.0 47.0 21.0 55.0 55.0 30.0 15.3 32.0 16.3 16.0 8.7 20.0 11.0 34.0 21.0 43.0 25.5 14.0 8.0 30.0 18.0 14.0 8.0 21.0 11.0 19.0 9.0 15.0 15.0 7.0 0.0130 0.0220 QGD (nC) 6.0 3.8 2.7 3.2 4.6 3.2 4.6 7.0 6.5 2.5 4.6 3.2 1.5 2.7 6.0 2.7 8.0 10.5 6.6 9.2 7.5 9.7 6.0 5.3 7.2 2.6 7.2 9.5 4.0 8.9 13.4 3.5 11.0 4.8 5.9 3.5 4.5 11.0 11.5 3.2 7.8 3.2 5.8 3.0 4.0 2.7 Rg Typ (9) 1.8 1.6 1.3 1.3 1.6 1.5 1.4 1.7 1.3 1.5 2.3 2.2 1.0 1.1 3.1 4.3 Vth (V) 1.0 0.8 0.8 0.8 1.0 0.8 1.0 1.0 1.0 1.0 1.0 0.8 1.2 0.8 1.0 0.8 1.0 0.4 1.0 0.6 1.0 1.0 2.0 1.0 1.0 0.6 1.0 1.0 1.0 1.2 0.6 0.6 0.6 1.0 0.8 0.8 0.8 2.0 0.8 0.8 0.8 0.8 1.0 1.0 2.0 2.0 ID (A) 9.0 7.4 7.0 6.3 10.0 6.3 9.5 10.0 9.0 4.7 9.5 6.3 PD (W) 2.5 2.0 1.9 1.4 2.4 1.4 2.4 2.5 2.3 1.4 2.4 1.4 7.5 9.0 7.5 8.0 25.0 8.3 25.0 14.0 23.0 3.7 8.5 11.0 6.9 17.0 20.0 16.0 25.0 25.0 17.0 21.0 12.0 16.0 12.4 12.5 9.5 11.5 6.3 10.5 6.3 7.4 4.5 3.7 2.6 2.0 2.5 2.0 2.5 3.5 2.5 3.5 3.1 3.5 3.0 3.3 2.5 2.0 3.0 3.5 3.5 3.5 3.5 3.0 3.0 3.0 3.5 3.1 3.0 3.1 2.4 1.4 2.4 1.4 2.1 2.0 2.0 2.0 0.6 2.9 2.1 0.6 4.2 2.1 1.0 1.0 0.6 0.6 0.6 7.8 5.6 7.8 3.3 1.9 1.5 1.5 1.5 1.5 1.2 40.0 18.0 22.5 9.0 34.0 4.5 2.1 9.0 3.3 8.1 1.0 0.4 7.0 2.6 6.7 0.7 0.3 65.0 30.5 13.5 9.5 1.4 1.0 23.0 5.4 46.0 24.0 11.5 11.5 0.9 2.0 15.8 5.2 31 16.3 4 5.9 0.8 19 5 47.0 20.0 5.8 7.1 0.6 13.3 3.8 0.0210 0.0070 QGS (nC) 6.0 2.6 1.5 1.8 5.3 1.8 5.3 8.0 7.5 3.0 5.3 1.8 2.0 2.0 6.0 2.0 6.5 8.0 9.0 6.7 6.0 6.7 3.2 3.4 8.6 2.1 8.0 13.5 5.0 11.8 10.0 4.6 13.0 5.8 4.0 2.4 3.0 7.5 4.5 1.8 3.6 1.8 3.3 4.0 3.2 4.0 Package SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 SO-8 1206-8 ChipFET 1206-8 ChipFET TSSOP-8 TSSOP-8 TSSOP-8 TSSOP-8 TSSOP-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK 1212-8 www.vishay.com 19 AN607 Vishay Siliconix Alphanumeric Index (cont’d.) rDS(on) 9 VGS = VGS = 6V 4.5V VDS (V) VGS (V) Si7414DN 60 20 0.0250 0.0360 16.0 Si7415DN --60 20 0.0650 0.1100 Si7440DP 30 20 0.0065 0.0080 Si7445DP --20 8 Si7446DP 30 20 Si7448DP 20 12 Si7450DP 200 20 0.0800 0.0900 0.0900 Si7454DP 100 20 0.0340 0.0400 Si7456DP 100 20 0.0250 0.0280 Si7458DP 20 12 0.0045 0.0075 Si7540DP 12 8 0.0210 0.0260 0.7 Si7540DP --12 8 0.0340 0.0470 0.7 Si7806DN 30 20 0.0110 Si7810DN 100 20 0.0620 Si7840DP 30 20 Si7842DP 30 Si7844DP Part Number QGS (nC) QGD (nC) Rg Typ (9) Vth (V) ID (A) 8.0 2.7 4.4 1.0 1.0 8.7 3.8 15.0 7.5 4.0 3.2 1.0 5.7 3.8 10.0 29.0 10.5 1.4 1.0 21.0 5.4 92.0 19.0 16.5 2.0 0.5 19.0 5.4 36.0 14.0 12.0 2.4 1.0 19.0 5.2 38.0 8.0 8.5 0.9 0.6 22.0 5.2 34.0 20.0 7.5 12.0 2.0 5.3 5.2 0.0400 24.0 14.0 7.6 5.4 2.0 7.8 4.8 0.0280 36.0 20.0 10.0 8.6 2.0 9.3 5.2 38.0 8.0 8.5 0.6 22.0 5.2 1.0 14.4 3.8 2.0 5.4 3.8 0.0077 0.0075 VGS = 2.5V Qg (nC) VGS = VGS = 10V 4.5V VGS = 10V 0.0094 0.0100 0.0065 76.0 0.0090 1.3 0.0175 19.0 8.5 3.6 3.0 0.0840 13.0 7.8 3.0 4.6 0.0095 0.0140 29.0 15.5 3.8 6.0 0.8 1.0 18.0 5.0 20 0.0220 0.0300 13.0 7.0 2.0 2.7 1.2 0.8 10.0 3.5 30 20 0.0220 0.0300 13.0 7.0 2.0 2.7 1.2 0.8 10.0 3.5 Si7846DP 150 20 0.0500 30.0 18.0 8.5 8.5 2.0 6.7 5.2 Si7848DP 40 20 0.0090 0.0120 35.0 18.5 6.0 7.5 0.8 1.0 17.0 5.0 Si7850DP 60 20 0.0220 0.0310 18.0 9.5 3.4 5.3 1.4 1.0 10.3 4.5 Si7852DP 80 20 0.0165 0.0220 34.0 22.0 7.5 11.0 0.9 2.0 12.5 5.2 Si7856DP 30 20 0.0045 34.0 15.0 10.0 1.3 1.0 25.0 5.4 Si7858DP 12 8 1.0 0.6 Si7860DP 30 20 Si7862DP 16 8 0.0033 Si7864DP 20 8 0.0035 Si7866DP 20 20 0.0025 Si7868DP 20 16 Si7880DP 30 20 Si7882DP 12 8 Si7884DP 40 20 0.0070 0.0095 Si7886DP 30 12 0.0045 0.0055 Si7888DP 30 20 0.0120 0.0200 www.vishay.com 20 0.0840 0.0220 0.0055 0.0030 0.0080 0.0040 0.0110 2.0 PD (W) 13.0 5.0 4.0 1.7 1.0 18.0 5.0 0.0055 48.0 11.8 8.9 1.3 0.6 29.0 5.4 0.0047 47.0 10.0 13.4 1.5 0.6 29.0 5.4 0.0033 40.0 15.0 11.0 1.2 0.8 29.0 5.4 0.0023 0.0028 50.0 12.0 11.0 1.2 0.6 29.0 5.4 0.0030 0.0043 40.5 18.0 10.5 1.2 1.0 29.0 5.4 21.0 4.6 3.5 0.6 22.0 5.0 18.5 6.0 7.5 0.8 1.0 20.0 5.2 42.0 12.8 7.7 1.3 0.6 25.0 5.4 8.7 2.4 3.5 1.0 0.8 15.7 5.0 0.0055 88.0 0.0080 35.0 16.0 Package PowerPAK 1212-8 PowerPAK 1212-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK 1212-8 PowerPAK 1212-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix Alphanumeric Index (cont’d.) rDS(on) 9 VGS = VGS = 6V 4.5V VDS (V) VGS (V) VGS = 10V Si7892DP 30 20 0.0045 0.0060 Si7894DP 30 12 0.0045 0.0055 Si7898DP 150 20 0.0850 0.0950 Si7922DN 100 20 0.1700 0.2060 Part Number VGS = 2.5V Qg (nC) VGS = VGS = 10V 4.5V 55.0 17.0 QGS (nC) QGD (nC) Rg Typ (9) Vth (V) ID (A) 25.0 6.7 9.7 1.9 1.0 25.0 5.4 48.0 17.0 10.0 1.3 0.6 25.0 5.4 10.0 3.2 6.0 0.9 2.0 4.8 5.0 0.2060 2.0 Si7940DP 12 8 Si9420DY Si9422DY Si9428DY 200 200 20 20 20 8 1.0000 0.4200 11.5 3.2 2.5 0.6 11.8 3.5 5.0 21.0 1.5 4.5 6.5 3.2 3.5 2.9 2.0 2.0 0.6 1.0 1.7 6.0 2.5 2.5 2.5 SUB40N06-25L 60 20 0.0220 0.0250 40.0 18.0 9.0 10.0 1.0 40.0 90.0 SUB70N03-09BP 30 20 0.0090 0.0130 26.0 15.5 5.0 6.0 0.8 70.0 93.0 SUB75N06-07L 60 20 0.0070 0.0080 75.0 50.0 18.0 27.0 1.0 75.0 250.0 SUB75N06-08 60 20 0.0080 28.0 26.0 2.0 75.0 250.0 SUB85N02-03 20 8 0.0030 0.0034 140.0 18.0 24.0 0.5 85.0 250.0 SUB85N02-06 20 12 0.0060 0.0090 135.0 65.0 13.0 14.0 0.6 85.0 120.0 SUB85N03-04P 30 20 0.0040 0.0070 71.0 35.0 15.0 16.0 1.0 85.0 166.0 SUB85N03-07P 30 20 0.0070 0.0100 60.0 26.0 13.0 10.0 1.0 85.0 107.0 SUB85N06-05 60 20 0.0050 0.0070 155.0 70.0 28.0 44.0 1.0 85.0 250.0 SUB85N10-10 100 20 0.0100 0.0120 105.0 55.0 17.0 23.0 1.0 85.0 250.0 SUD15N15-95 150 20 0.0950 0.1000 0.1000 15.0 62.0 SUD19N20-90 200 20 0.0900 0.1050 0.1050 34.0 7.5 8.0 12.0 2.0 19.0 100.0 SUD25N15-52 150 20 0.0520 0.0600 0.0600 33.0 7.5 9.0 12.0 2.0 25.0 100.0 SUD30N04-10 40 20 0.0100 0.0140 50.0 23.0 9.0 11.0 1.0 30.0 97.0 SUD40N02-08 20 12 54.0 26.0 5.0 7.0 0.6 40.0 71.0 SUD40N06-25L 60 20 0.0220 40.0 18.0 9.0 10.0 1.0 20.0 75.0 SUD40N08-16 80 20 0.0160 42.0 22.0 7.0 13.0 2.0 40.0 100.0 SUD40N10-25 100 20 0.0250 11.0 9.0 1.0 40.0 33.0 SUD50N02-06 20 12 65.0 13.0 14.0 0.6 30.0 100.0 SUD50N03-07 30 20 0.0070 0.0100 70.0 35.0 16.0 10.0 1.0 20.0 83.0 SUD50N03-07AP 30 20 0.0070 0.0100 60.0 28.0 12.0 10.0 1.8 1.0 25.0 88.0 SUD50N03-06P 30 20 0.0065 0.0095 48.0 22.0 10.0 7.5 1.9 1.0 25.0 88.0 SUD50N03-09P 30 20 0.0095 0.0140 31.0 13.5 7.5 5.0 1.5 1.0 21.0 65.2 SUD50N03-10BP 30 20 0.0100 0.0140 27.0 15.5 5.0 6.0 0.8 20.0 71.0 SUD50N03-10CP 30 20 0.0100 0.0120 38.0 13.0 4.5 4.0 1.0 15.0 71.0 100 20 0.0095 110.0 55.0 24.0 24.0 2.0 110.0 437.5 SUM110N10-09 Document Number: 71917 10-Oct-02 0.0170 PD (W) 0.0250 8.6 13.0 0.0300 0.0400 85.0 0.0085 0.0140 0.0250 0.0280 0.0060 40.0 0.0090 1.7 Package PowerPAK SO-8 PowerPAK SO-8 PowerPAK SO-8 PowerPAK 1212-8 PowerPAK SO-8 SO-8 SO-8 SO-8 D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) DPAK (TO-252) D2PAK (TO-263) www.vishay.com 21 AN607 Vishay Siliconix Alphanumeric Index (cont’d.) VDS (V) VGS (V) VGS = 10V 200 20 0.0300 SUM85N03-06P 30 20 0.0060 SUM85N03-08P 30 20 0.0075 SUM85N15-19 150 20 0.0190 SUP18N15-95 SUP70N03-09BP SUP85N02-03 SUP85N03-04P SUP85N03-07P SUP85N10-10 SUU15N15-95 SUY50N03-10CP TN0201T 150 30 20 30 30 100 150 30 20 20 20 8 20 20 20 20 20 20 0.0950 0.0090 Part Number SUM65N20-30 www.vishay.com 22 0.0040 0.0070 0.0100 0.0950 0.0100 0.7500 rDS(on) 9 VGS = VGS = 6V 4.5V 0.1000 0.1000 VGS = 2.5V Qg (nC) VGS = VGS = 10V 4.5V QGS (nC) QGD (nC) Rg Typ (9) Vth (V) ID (A) PD (W) 2.0 65.0 375.0 90.0 17.0 23.0 34.0 0.0090 48.0 22.0 10.0 7.5 1.9 1.0 85.0 100.0 0.0105 37.5 13.0 4.5 4.0 1.9 1.0 85.0 100.0 76.0 17.0 21.0 26.0 2.0 85.0 375.0 0.8 0.5 1.0 1.0 1.0 2.0 1.0 1.0 18.0 70.0 85.0 85.0 85.0 85.0 15.0 15.0 0.4 88.0 93.0 250.0 166.0 107.0 250.0 62.0 71.0 0.4 0.1000 0.0130 0.0030 0.0070 0.0100 0.0120 0.1000 0.0120 1.0000 26.0 0.0034 71.0 60.0 105.0 20.0 38.0 1.4 15.5 140.0 35.0 26.0 55.0 5.0 13.0 0.8 5.0 18.0 15.0 13.0 17.0 5.5 4.5 0.3 6.0 24.0 16.0 10.0 23.0 7.0 4.0 0.2 1.7 Package D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) D2PAK (TO-263) TO-220 TO-220 TO-220 TO-220 TO-220 TO-220 TO-251 TO-251 SOT-23 Document Number: 71917 10-Oct-02 AN607 Vishay Siliconix PWM Converters and Controllers, and MOSFET Drivers Part Number Distributed Power Si9117* Si9118 Si9119 Si9121-5* Si9121-3.3* Si9138 SSOP-28 Si9102* Si9104* Si9105* Si9108 Si9110 Si9111 Si9112 Si9113 Si9114A Reference Voltage (V) Maximum Supply Current (mA) 10 - 70 Current 1 4 1 Buck, Flyback, Forward 10 - 120 Current 1 4 1 Buck, Flyback, Forward 10 - 120 Current 1 4 1 Buck, Flyback, Forward 10 - 120 Current 1 4 0.5 Buck, Flyback, Forward 10 - 120 Current 1 4 0.5 Buck, Flyback, Forward 10 - 120 Current 1 4 1 Buck, Flyback, Forward 10 - 120 Current 1 4 1 Buck, Flyback, Forward 10 - 120 Current 1 4 1 Buck, Flyback, Forward 23 - 200 Current 0.5 1.3 1.4 Buck, Flyback, Forward 15 - 200 Current 1 4 3 Buck, Flyback, Forward Boost, Flyback, Forward Boost, Flyback, Forward Buck/Boost Converter Buck/Boost Converter Triple outut, individual On/Off Control Power Supply Controller 15 - 200 10 - 200 10 - 200 -10 to -60 -10 to -60 Current Current Current Current Current 1 1 1 0.11 0.11 4 4 4 1.25 1.25 4.5 2.5 2.5 1.5 1.5 5.5 - 30 Current 0.33 3.3 1.8 Buck, Flyback, Forward 15 - 450 Current 1 4 1.5 Buck Buck Buck, ISHARE Buck, Boost, Flyback, Forward 2.7 - 8 4.75 - 13.2 4.75 - 13.2 Voltage Voltage Voltage 2 1 1 1.5 1.3 1.3 1 1.2 1.2 2.7 - 8 Voltage 2 1.5 1.4 Dual Synchronous Buck Dual Synchronous Buck Triple Output, SMBus Triple Output Triple output, sequence selectable controller 5.5 - 30 5.5 - 30 5.5 - 30 5.5 - 30 Current Current Current Current 0.3 0.3 0.3 0.3 3.3 3.3 3.3 3.3 1.6 1.6 1.8 1.8 5.5 - 30 Current 0.33 3.3 1.8 Topology Buck, Flyback, Forward Package PDIP-14 PLCC-20 PDIP-14 PLCC-20 SO-16WB SO-16WB PDIP-14 PLCC-20 SO-16WB PDIP-14 PLCC-20 SO-14 PDIP-14 SO-14 PDIP-14 SO-14 PDIP-14 SO-14 SO-14 PDIP-14 SO-16 SO-16 SO-16 SO-8 SO-8 Si9100* Mode Maximum Oscillator Frequency (MHz) Input Voltage (V) Offline SO-16 PDIP-16 Computer Point-of-Use Si9140 SO-16 Si9142 SO-20 Si9143 SSOP-24 SO-16 Si9145 TSSOP-16 Portable Computer Si786 SSOP-28 Si9130 SSOP-28 Si9135 SSOP-28 Si9136 SSOP-28 Si9120 Si9137 SSOP-28 Mosfet Drivers Part Number Si9910 Si9912 Si9913 Function High Voltage Mosfet Driver Half Bridge Mosfet driver Half Bridge Mosfet driver Supply Voltage 11 - 16 for Driver 4.5 - 30V 4.5 - 30V Output Drive Capacity Drives 1 N-Ch. MOSFET Drives 2 N-Ch. MOSFET Drives 2 N-Ch. MOSFET Input Drive Requirements 12-V Logic 5 V , TTL/CMOS 5 V , TTL/CMOS Features Package dv/dt, di/dt Control DIP-8, SO-8 Shutdown Quiescent current Synchronous switch enable SO-8 SO-8 Protection Short Circuit, Under Voltage Undervoltage. Shoot Through Undervoltage. Shoot Through *Converters, with integrated MOSFET Document Number: 71917 10-Oct-02 www.vishay.com 23