INTERSIL ISL8112

ISL8112
®
Data Sheet
November 21, 2006
High Light-Load Efficiency, Dual-Output,
Main Power Supply Controllers
ISL8112 is a dual-output Synchronous Buck controller with 2A
integrated driver. It features high light load efficiency which is
especially preferred in systems concerned with high efficiency
in wide load range, like the battery powered system. ISL8112
includes two constant on-time PWM controllers. Either of the
two outputs can operate in output fixed mode or adjustable
mode. In fixed mode, one output can be 5V or 3.3V and the
other can output 1.5V or 1.05V. In output adjustable mode,
one output can be 0.7V to 5.5V, and the other output can
range from 0V to 2.5V (sensing output voltage directly) or up
to 5V (using resistor divider voltage for voltage sensing). This
device also features a linear regulator providing 3.3V/5V, or
adjustable from 0.7V to 4.5V via LDOREF. The linear
regulator provides up to 100mA output current with automatic
linear-regulator bootstrapping to the BYP input. When in
switch over, the LDO output can source up to 200mA.
ISL8112 includes on-board power-up sequencing, the powergood (PGOOD_) outputs, digital soft-start, and internal softstop output discharge that prevents negative voltages on
shutdown.
ISL8112 is implemented with constant on-time PWM control
scheme which need no sense resistors and provides 100ns
response to load transients while maintaining a relatively
constant switching frequency. The unique ultrasonic pulseskipping mode maintains the switching frequency above
25kHz, eliminating undesired audible noises in low frequency
operation at light load. Other features include pulse skipping
which maximizes efficiency in light-load applications, and
fixed-frequency PWM mode which reduces RF interference in
sensitive applications.
Ordering Information
PART
NUMBER
ISL8112IRZ
(Note)
PART
MARKING
TEMP. (°C)
FN6396.0
Features
• Wide Input Voltage Range 5.5V to 25V
• Constant ON-TIME Control with 100ns Load-Step
Response
• Dual Fixed Outputs of 1.05V (3.3V) and 1.5V (5.0V), or
Adjustable Outputs of 0.7V to 5.5V (SMPS1) and 0V to
2.5V/5V (SMPS2), ±1.5% Accuracy
• Adjustable Switching Frequency: 400/500kHz,
300/400kHz, 200/300kHz
• Very High Light Load Efficiency (Skip Mode)
• 5mW Quiescent Power Dissipation
• ±1.5% (LDO): 100mA, 200mA (Switch Over)
• 3.3V Reference Voltage ±2.0%: 5mA
• 2.0V Reference Voltage ±1.0%: 50µA
• Temperature Compensated rDS(ON) Current Sensing
• Programmable Current Limit with Foldback Capability
• Selectable PWM, Skip or Ultrasonic Mode
• Independent PGOOD1 and PGOOD2 Comparators
• Soft-Start with Pre-Biased Output and Soft-Stop
• 1.7ms Digital Soft-Start and Independent Shutdown
• Independent ENABLE
• Thermal Shutdown
• Extremely Low Components Count
• Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
• Power Supply for Telecom/Datacom and POL
PACKAGE
ISL8112 IRZ -40 to +100 32 Ld QFN
(Pb-free)
PKG.
DWG. #
L32.5x5B
• System Requiring High Efficiency in Wide Load Range
• Compact Design with Minimum Components Count
• PDAs and Mobile Communication Devices
L32.5x5B
ISL8112IRZ-T ISL8112 IRZ -40 to +100 32 Ld QFN
(Note)
(Pb-free)
(Tape and Reel)
• 3- and 4-Cell Li+ Battery-Powered Devices
• DDR1, DDR2, and DDR3 Applications
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100% matte
tin plate termination finish, which are RoHS compliant and compatible
with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL8112
Pinout
OUT2REF
ILIM2
VSEN2
MODE
PGOOD2
EN2
UG2
PH2
ISL8112
(32 LD 5X5 QFN)
TOP VIEW
32
31
30
29
28
27
26
25
VCC
3
22
PGND
EN_LDO
4
21
GND
VREF2
5
20
NC
VIN
6
19
PVCC
LDO
7
18
LG1
LDOREF
8
17
BOOT1
9
10
11
12
13
14
15
16
PH1
LG2
UG1
23
EN1
2
PGOOD1
FS
ILIM1
BOOT2
FB1
24
VSEN1
1
BYP
VREF1
2
FN6396.0
November 21, 2006
ISL8112
Absolute Voltage Ratings
Thermal Information
VIN, EN_LDO to GND . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +27V
BOOT_ to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT_ to PH_ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
VCC, EN_, MODE, FS,
PVCC, PGOOD_ to GND . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
LDO, FB1, OUT2REF, LDOREF to GND . . . . -0.3V to (VCC+0.3V)
VSEN_, VREF2, VREF1 to GND . . . . . . . . . . . -0.3V to (VCC+0.3V
UG_ to PH_ . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V)
ILIM_ to GND . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VCC + 0.3V)
LG_, BYP to GND . . . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V)
PGND to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to + 0.3V
LDO, VREF1, VREF2 Short Circuit to GND . . . . . . . . . . Continuous
VCC Short Circuit to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1s
LDO Current (Internal Regulator) Continuous . . . . . . . . . . . . 100mA
LDO Current (Switched Over to VSEN1) Continuous . . . . . +200mA
Thermal Resistance (Typical)
θJA (°C/W)
θJC (°CW)
32 Ld QFN (Notes 1, 2) . . . . . . . . . . . . 32
3.0
Operating Temperature Range . . . . . . . . . . . . . . . .-40°C to +100°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Storage Temperature Range . . . . . . . . . . . . . . . . . .-65°C to +150°C
CAUTION: Stress above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational section of this specification is not implied.
NOTE:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Circuit of Figure 17, and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted.
Typical values are at TA = +25°C.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
MAIN SMPS CONTROLLERS
VIN Input Voltage Range
LDO in regulation
5.5
25
V
VIN = LDO, VSEN1 < 4.43V
4.5
5.5
V
3.3V Output Voltage in Fixed Mode
VIN = 5.5V to 25V, OUT2REF > (VCC - 1V), MODE = 5V
3.285
3.330
3.375
V
1.05V Output Voltage in Fixed Mode
VIN = 5.5V to 25V, 3.0 < OUT2REF < (VCC - 1.1V),
MODE = 5V
1.038
1.05
1.062
V
1.5V Output Voltage in Fixed Mode
VIN= 5.5V to 25V, FB1 = VCC, MODE = 5V
1.482
1.500
1.518
V
5V Output Voltage in Fixed Mode
VIN= 5.5V to 25V, FB1 = GND, MODE = 5V
4.975
5.050
5.125
V
FB1 in Output Adjustable Mode
VIN = 5.5V to 25V
0.693
0.700
0.707
V
OUT2REF in Output Adjustable Mode
VIN = 5.5V to 25V
0.7
2.50
V
SMPS1 Output Voltage Adjust Range
SMPS1
0.70
5.50
V
SMPS2 Output Voltage Adjust Range
SMPS2
0.50
2.50
V
SMPS2 Output Voltage Accuracy
(Referred for OUT2REF)
OUT2REF = 0.7V to 2.5V, MODE = VCC
-1.0
1.0
%
DC Load Regulation
Either SMPS, MODE = VCC, 0 to 5A
-0.1
%
Either SMPS, MODE = VREF1, 0 to 5A
-1.7
%
Either SMPS, MODE = GND, 0 to 5A
-1.5
%
0.005
%/V
Line Regulation
Either SMPS, 6V < VIN < 24V
Current-Limit Current Source
Temperature = +25°C
ILIM_ Adjustment Range
4.75
5
0.2
Current-Limit Threshold (Positive, Default)
3
ILIM_ = VCC, GND - PH_
(No temperature compensation)
93
100
5.25
µA
2
V
107
mV
FN6396.0
November 21, 2006
ISL8112
Electrical Specifications
Circuit of Figure 17, and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted.
Typical values are at TA = +25°C. (Continued)
PARAMETER
CONDITIONS
Current-Limit Threshold
(Positive, Adjustable)
GND - PH_
MIN
TYP
MAX
UNITS
VILIM_ = 0.5V
40
50
60
mV
VILIM_ = 1V
93
100
107
mV
VILIM_ = 2V
185
200
215
mV
Zero-Current Threshold
MODE = GND, VREF1, or OPEN, GND - PH_
Current-Limit Threshold (Negative, Default)
MODE = VCC, GND - PH_
Soft-Start Ramp Time
Zero to full limit
Operating Frequency
(VFS = GND), MODE = VCC
On-Time Pulse Width
3
mV
-120
mV
1.7
ms
SMPS 1
400
kHz
SMPS 2
500
kHz
(VFS = VREF1 or OPEN),
MODE = VCC
SMPS 1
400
kHz
SMPS 2
300
kHz
(VFS = VCC), MODE = VCC
SMPS 1
200
kHz
SMPS 2
300
kHz
VFS = GND (400kHz/500kHz)
VSEN1 = 5.00V
0.895
1.052
1.209
µs
VSEN2 = 3.33V
0.475
0.555
0.635
µs
VFS = VREF1 or OPEN
(400kHz/300kHz)
VSEN1 = 5.05V
0.895
1.052
1.209
µs
VSEN2 = 3.33V
0.833
0.925
1.017
µs
VFS = VCC (200kHz/300kHz)
VSEN1 = 5.05V
1.895
2.105
2.315
µs
VSEN2 = 3.33V
0.833
0.925
1.017
µs
200
300
400
ns
Minimum Off-Time
Maximum Duty Cycle
VFS = GND
VFS = VREF1 or OPEN
VFS = VCC
Ultrasonic SKIP Operating Frequency
VSEN1 = 5.05V
88
%
VSEN2 = 3.33V
85
%
VSEN1 = 5.05V
88
%
VSEN2 = 3.33V
91
%
VSEN1 = 5.05V
94
%
VSEN2 = 3.33V
91
%
25
37
kHz
MODE = VREF1 or OPEN
INTERNAL REGULATOR AND REFERENCE
LDO Output Voltage
BYP = GND, 5.5V < VIN < 25V, LDOREF < 0.3V,
0 < ILDO < 100mA
4.925
5.000
5.075
V
LDO Output Voltage
BYP = GND, 5.5V < VIN < 25V, LDOREF > (VCC-1V),
0 < ILDO < 100mA
3.250
3.300
3.350
V
LDO Output in Adjustable Mode
VIN = 5.5V to 25V, VLDO = 2 x VLDOREF
4.5
V
LDO Output Accuracy in Adjustable Mode
VIN = 5.5V to 25V, VLDOREF = 0.35V to 0.5V
±2
%
VIN = 5.5V to 25V, VLDOREF = 0.5V to 2.25V
±1.5
%
2.25
V
0.7
LDOREF Input Range
VLDO = 2 x VLDOREF
LDO Output Current
BYP = GND, VIN = 5.5V to 25V, Guaranteed by design
100
mA
LDO Output Current During Switch Over
BYP = 5V, VIN = 5.5V to 25V, LDOREF < 0.3V
200
mA
LDO Output Current During Switch Over to
3.3V
BYP = 3.3V, VIN = 5.5V to 25V, LDOREF > (VCC-1V)
100
mA
LDO Short-Circuit Current
LDO = GND, BYP = GND
400
mA
4
0.35
200
FN6396.0
November 21, 2006
ISL8112
Electrical Specifications
Circuit of Figure 17, and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted.
Typical values are at TA = +25°C. (Continued)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
4.35
4.05
4.5
V
3.9
Rising edge at BYP regulation point
LDOREF = GND
4.53
4.68
4.83
V
LDO 3.3V Bootstrap Switch Threshold to BYP Rising edge at BYP regulation point
LDOREF = VCC
3.0
3.1
3.2
V
Undervoltage-Lockout Fault Threshold
LDO 5V Bootstrap Switch Threshold to BYP
Rising edge of PVCC
Falling edge of PVCC
LDO 5V Bootstrap Switch Equivalent
Resistance
LDO to BYP, BYP = 5V, LDOREF > (VCC-1V),
Guaranteed by design
0.7
1.5
Ω
LDO 3.3V Bootstrap Switch Equivalent
Resistance
LDO to BYP, BYP = 3.3V, LDOREF < 0.3V, Guaranteed
by design
1.5
3.0
Ω
VREF2 Output Voltage
No external load, VCC > 4.5V
3.235
3.300
3.365
V
No external load, VCC < 4.0V
3.220
3.300
3.380
V
VREF2 Load Regulation
0 < ILOAD < 5mA
10
VREF2 Current Limit
VREF2 = GND
10
17
mA
VREF1 Output Voltage
No external load
2.000
2.020
V
VREF1 Load Regulation
0 < ILOAD < 50µA
VREF1 Sink Current
VREF1 in regulation
VIN Operating Supply Current
Both SMPSs on, FB1 = MODE = GND,
OUT2REF = VCC
VSEN1 = BYP = 5.3V, VSEN2 = 3.5V
25
50
µA
VIN Standby Supply Current
VIN = 5.5V to 25V, both SMPSs off, EN_LDO = VCC
180
250
µA
VIN Shutdown Supply Current
VIN = 4.5V to 25V, EN1=EN2=EN_LDO=0V
20
30
µA
Quiescent Power Consumption
Both SMPSs on, FB1 = MODE = GND,
OUT2REF = VCC,
VSEN1 = BYP = 5.3V, VSEN2 = 3.5V
5
7
mW
1.980
mV
10
mV
10
µA
FAULT DETECTION
Overvoltage Trip Threshold
FB1 with respect to nominal regulation point
+8
+11
+14
%
OUT2REF with respect to nominal regulation point
+12
+16
+20
%
Overvoltage Fault Propagation Delay
FB1 or OUT2REF delay with 50mV overdrive
PGOOD_ Threshold
FB1 or OUT2REF with respect to nominal output, falling
edge, typical hysteresis = 1%
PGOOD_ Propagation Delay
Falling edge, 50mV overdrive
PGOOD_ Output Low Voltage
ISINK = 4mA
PGOOD_ Leakage Current
High state, forced to 5.5V
10
-12
-9
µs
-6
10
Thermal-Shutdown Threshold
%
µs
0.2
V
1
µA
+150
°C
Output Undervoltage Shutdown Threshold
FB1 or OUT2REF with respect to nominal output voltage
65
70
75
%
Output Undervoltage Shutdown Blanking
Time
From EN_ signal
10
20
30
ms
0.3
V
INPUTS AND OUTPUTS
FB1 Input Voltage
Low level
High level
5
VCC-1.0
V
FN6396.0
November 21, 2006
ISL8112
Electrical Specifications
Circuit of Figure 17, and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted.
Typical values are at TA = +25°C. (Continued)
PARAMETER
CONDITIONS
OUT2REF Input Voltage
LDOREF Input Voltage
MIN
TYP
0.5
2.50
V
Fixed VSEN2 = 1.05V
3.0
VCC1.1
V
Fixed VSEN2 = 3.3V
VCC-1.0
VSEN2 Dynamic Range, VLDO = 2 x VLDOREF
Fixed LDO = 3.3V
V
0.35
Float level (ULTRASONIC SKIP)
1.7
High level (PWM)
2.4
Input Leakage Current
2.25
V
Float level
1.7
High level
2.4
0.8
V
2.3
V
V
0.8
V
2.3
V
V
Clear fault level/SMPS off level
EN_LDO Input Voltage
V
V
Low level
EN1, EN2 Input Voltage
0.30
VCC-1.0
Low level (SKIP)
FS Input Voltage
UNITS
VSEN2 Dynamic Range, VSEN2= VOUT2REF
Fixed LDO = 5V
MODE Input Voltage
MAX
0.8
V
2.3
V
Delay start level
1.7
SMPS on level
2.4
Rising edge
1.2
1.6
2.0
V
Falling edge
0.94
1.00
1.06
V
-1
+1
µA
-0.1
+0.1
µA
-1
+1
µA
VFB1 = 0V or 5V
-0.2
+0.2
µA
VREFIN = 0V or 2.5V
-0.2
+0.2
µA
VLDOREF = 0V or 2.75V
-0.2
+0.2
µA
0.8
V
500
nA
VFS = 0 or 5V
VEN_ = VEN_LDO = 0 or 5V
VMODE = 0V or 5V
V
INTERNAL BOOT DIODE
VD Forward Voltage
PVCC - VBOOT, IF = 10mA
IBOOT_LEAKAGE Leakage Current
VBOOT = 30V, PH = 25V, PVCC = 5V
0.65
MOSFET DRIVERS
UG_ Gate-Driver Sink/Source Current
UG1, UG2 forced to 2V
2
A
LG_ Gate-Driver Source Current
LG1 (source), LG2 (source), forced to 2V
1.7
A
LG_ Gate-Driver Sink Current
LG1 (sink), LG2 (sink), forced to 2V
3.3
A
UG_ Gate-Driver On-Resistance
BST_ - PH_ forced to 5V, Guaranteed by design
1.5
4.0
Ω
LG_ Gate-Driver On-Resistance
LG_, high state (pull-up), Guaranteed by design
2.2
5.0
Ω
LG_, low state (pull-down), Guaranteed by design
0.6
1.5
Ω
Dead Time
VSEN1, VSEN2 Discharge On Resistance
6
LG_ Rising
15
20
35
ns
UG_ Rising
20
30
50
ns
25
40
Ω
FN6396.0
November 21, 2006
ISL8112
Pin Descriptions
PIN
NAME
FUNCTION
1
VREF1
2
FS
3
VCC
4
EN_LDO
LDO Enable Input. The LDO is enabled if EN_LDO is within logic high level and disabled if EN_LDO is less than the
logic low level.
5
VREF2
3.3V Reference Output. VREF2 can source up to 5mA for external loads. Bypass to GND with a 0.01µF capacitor if
loaded. Leave open if there is no load.
6
VIN
Power-Supply Input. VIN is used for the constant-on-time PWM on-time one-shot circuits. VIN is also used to power the
linear regulators. The linear regulators are powered by SMPS1 if VSEN1 is set greater than 4.78V and BYP is tied to
VSEN1. Connect VIN to the battery input and bypass with a 1µF capacitor.
7
LDO
Linear-Regulator Output. LDO can provide a total of 100mA external loads. The LDO regulate at 5V If LDOREF is
connected to GND. When the LDO is set at 5V and BYP is within 5V switch over threshold, the internal regulator shuts
down and the LDO output pin connects to BYP through a 0.7Ω switch. The LDO regulate at 3.3V if LDOREF is
connected to VCC. When the LDO is set at 3.3V and BYP is within 3.3V switch over threshold, the internal regulator
shuts down and the LDO output pin connects to BYP through a 1.5Ω switch. Bypass LDO output with a minimum of
4.7µF ceramic.
8
LDOREF
LDO Reference Input. Connect LDOREF to GND for fixed 5V operation. Connect LDOREF to VCC for fixed 3.3V
operation. LDOREF can be used to program LDO output voltage from 0.7V to 4.5V. LDO output is two times the voltage
of LDOREF. There is no switch over in adjustable mode.
9
BYP
BYP is the switch over source voltage for the LDO when LDOREF connected to GND or VCC. Connect BYP to 5V if
LDOREF is tied GND. Connect BYP to 3.3V if LDOREF is tied to VCC.
10
VSEN1
SMPS1 Output Voltage-Sense Input. Connect to the SMPS1 output. VSEN1 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS1 feedback input in fixed-voltage mode.
11
FB1
12
ILIM1
SMPS1 Current-Limit Adjustment. The GND-PH1 current-limit threshold is 1/10th the voltage seen at ILIM1 over a 0.2V
to 2V range. There is an internal 5µA current source from VCC to ILIM1. Connect ILIM1 to VREF1 for a fixed 200mV
threshold. The logic current limit threshold is default to 100mV value if ILIM1 is higher than VCC - 1V.
13
PGOOD1
SMPS1 Power-Good Open-Drain Output. PGOOD1 is low when the SMPS1 output voltage is more than 10% below the
normal regulation point or during soft-start. PGOOD1 is high impedance when the output is in regulation and the softstart circuit has terminated. PGOOD1 is low in shutdown.
14
EN1
SMPS1 Enable Input. The SMPS1 is enabled if EN1 is greater than the logic high level and disabled if EN1 is less than
the logic low level. If EN1 is connected to VREF1, the SMPS1 starts after the SMPS2 reaches regulation (delay start).
Drive EN1 below 0.8V to clear fault level and reset the fault latches.
15
UG1
High-Side MOSFET Floating Gate-Driver Output for SMPS1. UG1 swings between PH1 and BOOT1.
16
PH1
Inductor Connection for SMPS1. PH1 is the internal lower supply rail for the UG1 high-side gate driver. PH1 is the
current-sense input for the SMPS1.
17
BOOT1
18
LG1
19
PVCC
20
NC
21
GND
22
PGND
2V Reference Output. Bypass to GND with a 0.1µF (min) capacitor. VREF1 can source up to 50A for external loads.
Loading VREF1 degrades FB and output accuracy according to the VREF1 load-regulation error.
Frequency Select Input. Connect to GND for 400kHz/500kHz operation. Connect to VREF1 (or leave OPEN) for
400kHz/300kHz operation. Connect to VCC for 200kHz/300kHz operation (5V/3.3V SMPS switching frequencies,
respectively).
Analog Supply Voltage Input for PWM Core. Bypass to GND with a 1µF ceramic capacitor.
SMPS1 Feedback Input. Connect FB1 to GND for fixed 5V operation. Connect FB1 to VCC for fixed 1.5V operation
Connect FB1 to a resistive voltage-divider from VSEN1 to GND to adjust the output from 0.7V to 5.5V.
Boost Flying Capacitor Connection for SMPS1. Connect to an external capacitor according to the typical application
circuits (Figure 17 and Figure 18). See “MOSFET Gate Drivers (UG_, LG_)” on page 19.
SMPS1 Synchronous-Rectifier Gate-Drive Output. LG1 swings between GND and PVCC.
PVCC is the supply voltage for the low-side MOSFET driver LG_. Connect a 5V power source to the PVCC pin (bypass
with 1µF MLCC capacitor to PGND if necessary). There is internal 10Ω connecting PVCC to VCC. Make sure that both
VCC and PVCC are bypassed with 1µF MLCC capacitors.
No connection pin. Externally connect it to ground.
Analog Ground for both SMPS_ and LDO. Connect externally to the underside of the exposed pad.
Power Ground for SMPS_ controller. Connect PGND externally to the underside of the exposed pad.
7
FN6396.0
November 21, 2006
ISL8112
Pin Descriptions (Continued)
PIN
NAME
23
LG2
24
BOOT2
25
PH2
Inductor Connection for SMPS2. PH2 is the internal lower supply rail for the UG2 high-side gate driver. PH2 is the
current-sense input for the SMPS2.
26
UG2
High-Side MOSFET Floating Gate-Driver Output for SMPS2. UG1 swings between PH2 and BOOT2.
27
EN2
SMPS2 Enable Input. The SMPS2 is enabled if EN2 is greater than the logic high level and disabled if EN2 is less than
the logic low level. If EN2 is connected to VREF1, the SMPS2 starts after the SMPS1 reaches regulation (delay start).
Drive EN2 below 0.8V to clear fault level and reset the fault latches.
28
PGOOD2
SMP2 Power-Good Open-Drain Output. PGOOD2 is low when the SMPS2 output voltage is more than 10% below the
normal regulation point or during soft-start. PGOOD2 is high impedance when the output is in regulation and the softstart circuit has terminated. PGOOD2 is low in shutdown.
29
MODE
Low-Noise Mode Control. Connect MODE to GND for normal Idle-Mode (pulse-skipping) operation or to VCC for PWM
mode (fixed frequency). Connect to VREF1 or leave floating for ultrasonic skip mode operation.
30
VSEN2
SMPS2 Output Voltage-Sense Input. Connect to the SMPS2 output. VSEN2 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS2 feedback input in fixed-voltage mode.
31
ILIM2
SMPS2 Current-Limit Adjustment. The GND-PH1 current-limit threshold is 1/10th the voltage seen at ILIM2 over a 0.2V
to 2V range. There is an internal 5µA current source from VCC to ILIM2. Connect ILIM2 to VREF1 for a fixed 200mV.
The logic current limit threshold is default to 100mV value if ILIM2 is higher than VCC - 1V.
32
FUNCTION
SMPS2 Synchronous-Rectifier Gate-Drive Output. LG2 swings between GND and PVCC.
Boost Flying Capacitor Connection for SMPS2. Connect to an external capacitor according to the typical application
circuits (Figure 17 and Figure 18). See “MOSFET Gate Drivers (UG_, LG_)” on page 19.
OUT2REF Output voltage control for SMPS2. Connect OUT2REF to VCC for fixed 3.3V. Connect OUT2REF to VREF2 for fixed
1.05V. OUT2REF can be used to program SMPS2 output voltage from 0.5V to 2.50V. SMPS2 output voltage is 0V if
OUT2REF<0.5V.
Typical Performance Curves
12 VIN ULTRA SKIP MODE
25 VIN SKIP MODE
25 VIN PWM MODE
25 VIN ULTRA SKIP MODE
1.0
1.0
0.9
0.9
0.8
0.8
0.7
0.7
0.6
0.6
EFFICIENCY
EFFICIENCY
7 VIN SKIP MODE
7 VIN PWM MODE
7 VIN ULTRA SKIP MODE
12 VIN SKIP MODE
12 VIN PWM MODE
Circuit of Figure 17 and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1,
VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C,
unless otherwise noted. Typical values are at TA = +25°C.
0.5
0.4
0.3
0.1
1.000
10.000
FIGURE 1. VOUT2 = 1.05V EFFICIENCY vs LOAD (300kHz)
8
1.000
0.3
0.2
0.100
OUTPUT LOAD (A)
0.010
0.4
0.1
0.010
12 VIN ULTRA SKIP MODE
25 VIN SKIP MODE
25 VIN PWM MODE
25 VIN ULTRA SKIP MODE
0.5
0.2
0
0.001
7 VIN SKIP MODE
7 VIN PWM MODE
7 VIN ULTRA SKIP MODE
12 VIN SKIP MODE
12 VIN PWM MODE
0
0.001
0.100
OUTPUT LOAD (A)
10.000
FIGURE 2. VOUT1 = 1.5V EFFICIENCY vs LOAD (200kHz)
FN6396.0
November 21, 2006
ISL8112
Typical Performance Curves
0.9
0.8
0.8
0.7
0.7
0.6
0.5
0.4
0.3
0.5
0.4
0.3
0.2
0.1
0.1
0.010
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
0.010
1.000
10.000
FIGURE 4. VOUT1 = 5V EFFICIENCY vs LOAD (400kHz)
300
50
45
250
40
35
RIPPLE (mV)
200
PWM
150
100
ULTRA-SKIP
50
30
PWM
25
ULTRA-SKIP
20
15
10
0
0.001
0.010
0.100
SKIP
5
SKIP
1.000
0
0.001
10.000
0.010
OUTPUT LOAD (A)
0.100
1.000
10.000
OUTPUT LOAD (A)
FIGURE 5. VOUT2 = 1.05V FREQUENCY vs LOAD
FIGURE 6. VOUT2 = 1.05V RIPPLE vs LOAD
250
50
45
PWM
200
40
PWM
35
RIPPLE (mV)
FREQUENCY (kHz)
0.100
OUTPUT LOAD (A)
FIGURE 3. VOUT2 = 3.3V EFFICIENCY vs LOAD (500kHz)
FREQUENCY (kHz)
0.6
0.2
0
0.001
12 VIN ULTRA SKIP MODE
25 VIN SKIP MODE
25 VIN PWM MODE
25 VIN ULTRA SKIP MODE
7 VIN SKIP MODE
7 VIN PWM MODE
7 VIN ULTRA SKIP MODE
12 VIN SKIP MODE
12 VIN PWM MODE
1.0
0.9
EFFICIENCY
EFFICIENCY
12 VIN ULTRA SKIP MODE
25 VIN SKIP MODE
25 VIN PWM MODE
25 VIN ULTRA SKIP MODE
7 VIN SKIP MODE
7 VIN PWM MODE
7 VIN ULTRA SKIP MODE
12 VIN SKIP MODE
12 VIN PWM MODE
1.0
Circuit of Figure 17 and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1,
VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C,
unless otherwise noted. Typical values are at TA = +25°C. (Continued)
150
100
ULTRA-SKIP
50
30
25
SKIP
20 ULTRA-SKIP
15
10
SKIP
0
0.001
5
0.010
0.100
1.000
OUTPUT LOAD (A)
FIGURE 7. VOUT1 = 1.5V FREQUENCY vs LOAD
9
10.000
0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 8. VOUT1 = 1.5V RIPPLE vs LOAD
FN6396.0
November 21, 2006
ISL8112
Typical Performance Curves
Circuit of Figure 17 and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1,
VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C,
unless otherwise noted. Typical values are at TA = +25°C. (Continued)
14
600
PWM
PWM
12
10
400
RIPPLE (mV)
FREQUENCY (kHz)
500
300
200
SKIP
6
2
SKIP
0
0.001
ULTRA-SKIP
4
ULTRA-SKIP
100
8
0.010
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 10. VOUT2 = 3.3V RIPPLE vs LOAD
FIGURE 9. VOUT2 = 3.3V FREQUENCY vs LOAD
450
40
400
35
PWM
30
PWM
300
RIPPLE (mV)
FREQUENCY (kHz)
350
250
200
150
ULTRA-SKIP
20
SKIP
15
10
ULTRA-SKIP
100
25
5
50
SKIP
0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
FIGURE 11. VOUT1 = 5V FREQUENCY vs LOAD
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 12. VOUT1 = 5V RIPPLE vs LOAD
5.04
3.35
BYP = 0V
5.00
3.30
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
5.02
4.98
4.96
4.94
4.92
4.90
BYP = 5V
4.88
BYP = 0V
3.20
3.15
BYP = 3.3V
3.10
3.05
4.86
4.84
0
3.25
50
100
OUTPUT LOAD (mA)
150
FIGURE 13. LDO OUTPUT 5V vs LOAD
10
200
3.00
0
50
100
OUTPUT LOAD (mA)
150
200
FIGURE 14. LDO OUTPUT 3.3V vs LOAD
FN6396.0
November 21, 2006
ISL8112
Circuit of Figure 17 and Figure 18, no load on LDO, VSEN1, VSEN2, VREF2, and VREF1,
VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C,
unless otherwise noted. Typical values are at TA = +25°C. (Continued)
177.5
26.5
177.0
26.0
176.5
25.5
INPUT CURRENT (µA)
INPUT CURRENT (µA)
Typical Performance Curves
176.0
175.5
175.0
174.5
174.0
173.5
173.0
7
25.0
24.5
24.0
23.5
23.0
22.5
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
22.0
7
25
FIGURE 15. STANDBY INPUT CURRENT vs VIN
(EN = EN2 = 0, EN_LDO = VCC)
Typical Application Circuits
The typical application circuits are shown in Figures 17, 18
and 19. In Figure 17, the power supply system generates
1.25V/5A and dynamic voltage/10A. Figure 18 shows
system having1.5V/5A and 1.05V/5A output. The input
supply range is 5.5V to 25V. Figure 19 shows system
having1.2V/15A and 2.5V/5A output. The input supply range
is 5.5V to 25V and 4.5V to 5.5V respectively.
Detailed Description
The ISL8112 dual-buck, BiCMOS, switch-mode powersupply controller generates logic supply voltages for
notebook computers. The ISL8112 is designed primarily for
battery-powered applications where high efficiency and lowquiescent supply current are critical. The ISL8112 provides a
pin-selectable switching frequency, allowing operation for
200kHz/300kHz, 400kHz/300kHz, or 400kHz/500kHz on the
SMPSs.
Light-load efficiency is enhanced by automatic Idle-Mode
operation, a variable-frequency pulse-skipping mode that
reduces transition and gate-charge losses. Each step-down,
power-switching circuit consists of two n-channel MOSFETs,
a rectifier, and an LC output filter. The output voltage is the
average AC voltage at the switching node, which is
regulated by changing the duty cycle of the MOSFET
switches. The gate-drive signal to the n-channel high-side
MOSFET must exceed the battery voltage, and is provided
by a flying-capacitor boost circuit that uses a 100nF
capacitor connected to BOOT_.
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 16. SHUTDOWN INPUT CURRENT vs VIN
(EN = EN2 = EN_LDO = 0)
undervoltage and overvoltage conditions. A power-on
sequence block controls the power-up timing of the main
PWMs and monitors the outputs for undervoltage faults. The
ISL8112 includes an adjustable low drop-out linear regulator.
The bias generator blocks include the linear regulator, 3.3V
precision reference, 2V precision reference and automatic
bootstrap switch over circuit.
The synchronous-switch gate drivers are directly powered
from PVCC, while the high-side switch gate drivers are
indirectly powered from PVCC through an external capacitor
and an internal Schottky diode boost circuit.
An automatic bootstrap circuit turns off the LDO linear
regulator and powers the device from BYP if LDOREF is set
to GND or VCC. See Table 1.
TABLE 1. LDO OUTPUT VOLTAGE TABLE
LDO VOLTAGE
CONDITIONS
COMMENT
VOLTAGE at BYP
LDOREF < 0.3V,
BYP > 4.63V
Internal LDO is
disabled.
VOLTAGE at BYP
LDOREF > VCC - 1V,
BYP > 3V
Internal LDO is
disabled.
5V
LDOREF < 0.3V,
BYP < 4.63V
Internal LDO is
active.
3.3V
LDOREF > VCC - 1V,
BYP < 3V
Internal LDO is
active.
2 x LDOREF
0.35V <LDOREF < 2.25V
Internal LDO is
active.
Both SMPS1 and SMPS2 PWM controllers consist of a
triple-Mode feedback network and multiplexer, a multi-input
PWM comparator, high-side and low-side gate drivers and
logic. In addition, SMPS2 can also use OUT2REF to track its
output from 0.5V to 2.50V. The ISL8112 contains faultprotection circuits that monitor the main PWM outputs for
11
FN6396.0
November 21, 2006
ISL8112
For loads above the critical conduction point, the actual
switching frequency is:
FREE-RUNNING, CONSTANT ON-TIME PWM
CONTROLLER WITH INPUT FEED-FORWARD
The constant on-time PWM control architecture is a
pseudo-fixed-frequency, constant on-time, current-mode
type with voltage feedforward. The constant on-time PWM
control architecture relies on the output ripple voltage to
provide the PWM ramp signal; thus the output filter
capacitor's ESR acts as a current-feedback resistor. The
high-side switch on-time is determined by a one-shot whose
period is inversely proportional to input voltage and directly
proportional to output voltage. Another one-shot sets a
minimum off-time (300ns typ). The on-time one-shot triggers
when the following conditions are met: the error
comparator's output is high, the synchronous rectifier current
is below the current-limit threshold, and the minimum off time
one-shot has timed out.
V OUT + V DROP1
f = ------------------------------------------------------t ON ( V IN + V DROP2 )
where:
• VDROP1 is the sum of the parasitic voltage drops in the
inductor discharge path, including synchronous rectifier,
inductor, and PC board resistances
• VDROP2 is the sum of the parasitic voltage drops in the
charging path, including high-side switch, inductor, and PC
board resistances
• tON is the on-time calculated by the ISL8112.
TABLE 2. APPROXIMATE K-FACTOR ERRORS
ON-TIME ONE-SHOT (FS)
Each PWM core includes a one-shot that sets the high-side
switch on-time for each controller. Each fast, low-jitter,
adjustable one-shot includes circuitry that varies the on-time
in response to battery and output voltage. The high-side
switch on-time is inversely proportional to the battery voltage
as measured by the VIN input and proportional to the output
voltage. This algorithm results in a nearly constant switching
frequency despite the lack of a fixed-frequency clock
generator. The benefit of a constant switching frequency is
that the frequency can be selected to avoid noise-sensitive
frequency regions:
K ( V OUT + I LOAD ⋅ r DSON ( LOWERQ ) )
t ON = -----------------------------------------------------------------------------------------------------V IN
(EQ. 2)
SMPS
APPROXIMATE
SWITCHING
K-FACTOR
FREQUENCY K-FACTOR
ERROR (%)
(kHz)
(µs)
(FS = GND,
VREF1, or OPEN),
VSEN1
400
2.5
±10
(FS = GND),
VSEN2
500
2.0
±10
(FS = VCC),
VSEN1
200
5.0
±10
(FS = VCC,
VREF1, or OPEN),
VSEN2
300
3.3
±10
(EQ. 1)
See Table 2 for approximate K- factors. Switching frequency
increases as a function of load current due to the increasing
drop across the synchronous rectifier, which causes a faster
inductor-current discharge ramp. On-times translate only
roughly to switching frequencies. The on-times guaranteed
in the Electrical Characteristics are influenced by switching
delays in the external high-side power MOSFET. Also, the
dead-time effect increases the effective on-time, reducing
the switching frequency. It occurs only in PWM mode
(MODE = VCC) and during dynamic output voltage
transitions when the inductor current reverses at light or
negative load currents. With reversed inductor current, the
inductor's EMF causes PH_ to go high earlier than normal,
extending the on-time by a period equal to the UG-rising
dead time.
12
FN6396.0
November 21, 2006
ISL8112
VIN: 5.5V to 25V
5V
C5
1µF
C8
1µF
PVCC
VCC
VIN
C10
10µF
OUT1 – PCI-e
L1: 3.3µH
1.25V/5A
GND
LDOREF
BOOT1
Q3a
SI4816BDY
NC
LDO
C1
10µF
10
BOOT2
UG1
UG2
PH1
PH2
LG1
LG2
C9
0.1µF
C4
0.22µF
Q3b
C11
330µF
9mΩ
6.3V
BYP
AGND
OUT2REF
VREF2
MODE
EN_LDO
VCC
NC
VCC
FS
2 BITS
DAC
+
R5
200kΩ
ILIM2
ILIM1
GND
C2
2 x 330µF
4mΩ
6.3V
OUT2REF: DYNAMIC 0 TO 2.5V
OUT2REF TIED TO VREF2 = 1.05V
OUT2REF TIED TO VCC = 3.3V
R3
200kΩ
R2
10kΩ
VCC
EN2
ISL8112
FB1
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
Q2
IRF7832
VSEN2
EN1
5V
OUT2-GFX
L2: 2.2µH TRACK OUT2REF/10A
PGND
VSEN1
VCC
R1
7.87kΩ
Q1
IRF7821
C3
OPEN
C7
0.1µF
VREF1
VCC
+
DROOP
+
VCC
R4
200kΩ
R6
200kΩ
PGOOD1
PGOOD2
PAD
FREQUENCY-DEPENDENT COMPONENTS
1.25V/1.05V SMPS
SWITCHING
FREQUENCY
FS = VCC
200kHz/300kHz
L1
3.3µH
L2
2.7µH
C2
2 x 330µF
C11
330µF
FIGURE 17. ISL8112 TYPICAL DYNAMIC GFX APPLICATION CIRCUIT
13
FN6396.0
November 21, 2006
ISL8112
VIN: 5.5V to 25V
5V
C5
1µF
LDOREF TIED TO GND = 5V
LDOREF TIED TO VCC = 3.3V
LDO
C8
1µF
PVCC
VCC
LDO
VIN
C10
10µF
SI4816BDY
VCC
LDOREF
BOOT1
C1
10µF
10
BOOT2
Q3a
OUT1
1.5V/5A
L1: 3.3µH
PH1
PH2
LG1
LG2
Q3b
C11
33µF
9mΩ
6.3V
Q1a
UG2
UG1
C9
0.1µF
C4
0.22µF
OUT2
L2: 2.2µF 1.05V/5A
Q1b SI4816BDY
PGND
VSEN1
C6
F
4.7µF
C2
330µF
4mΩ
6.3V
VCC
VSEN2
EN1
3.3V
VCC
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
BYP
ISL8112
VCC
EN2
OUT2REF: DYNAMIC 0 TO 2.5V
OUT2REF TIED TO VREF2 = 1.05V
VREF2 OUT2REF TIED TO VCC = 3.3V
FB1
R3
200kΩ
OUT2REF
AGND
R5
200kΩ
ILIM2
ILIM1
MODE
VREF2
EN_LDO
VREF1
ON
OFF
VCC
NC
VCC
FS
C3
0.01µF
C7
0.1µF
VCC
R4
200kΩ
VCC
R6
200kΩ
PGOOD1
PGOOD2
PAD
FREQUENCY-DEPENDENT COMPONENTS
1.5V/1.05V SMPS
SWITCHING
FREQUENCY
FS = VCC
200kHz/300kHz
L1
3.3µH
L2
2.7µH
C2
330µF
C11
330µF
FIGURE 18. ISL8112 TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT
14
FN6396.0
November 21, 2006
ISL8112
VIN: 4.5V to 5.5V
R7
Ω
1O
C5
1µF
C8
1µF
PVCC
VCC
VIN
LDO
NC
GND
LDOREF
LDOREF
C1
C10
BOOT1
10 µ F
10 µ F
BOOT2
Q3a
SI4816BDY
2.5V/5A
UG2
UG1
PH1
PH2
LG1
LG2
Q3b
C11
IRF7821
C4
C9
0.1µF
L2: 1.5µH
Q1
VSEN2
EN1
GND
BYP
ISL8112
AGND
OUT2REF
ILIM 1
ILIM2
R5
Ω
225kO
VREF2
MODE
VCC
GND
EN_LDO
VCC
NC
R8
Ω
73kO
OUT2REF: DYNAMIC 0 TO 2.5V
OUT2REF tied to VREF2
VREF3=1.05V
OUT2REF tied to VCC=3.3V
R3
200k Ω
R2
Ω
43kO
VREF1
REF
VCC
EN2
FB1
FB1 tied to VCC=1.5V
GND=5V
FB1 tied to VCC=1.5V
C2
3 x 330µF
Ω
4mO
6.3V
IRF7832
PGND
VSEN1
VCC
R1
Ω
110kO
1.2V/15A
Q2
330µF
Ω
9mO
6.3V
L1: 1.5µH
0.22µF
VREF1
R9
Ω
110kO
C3
OPEN
C7
0.1µF
VCC
VCC
R4
Ω
225kO
R6
Ω
225kO
PGOOD1
FS
PGOOD2
PAD
FREQUENCY-DEPENDENT COMPONENTS
1.2V/2.5V SMPS
SWITCHING
FREQUENCY
FS = GND
400kHz/500kHz
L1
1.5µH
L2
1.5µH
C2
3X330µF
C11
330µF
FIGURE 19. ISL8112 TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT
15
FN6396.0
November 21, 2006
ISL8112
FS
MODE
BOOT2
BOOT1
UG1
UG2
PH1
PH2
PVCC
PVCC
LG1
SMPS1
SMPS2
SYNCH.
SYNCH.
PWM BUCK
PWM BUCK
CONTROLLER
CONTROLLER
GND
ILIM1
EN1
FB1
PGOOD1
VSEN1
PGND
ILIM2
EN2
OUT2REF
PGOOD2
VSEN2
VSEN2
VSEN1
BYP
LG2
PGOOD2
+
-
SW THRES.
PGOOD1
LDO
LDO
VCC
INTERNAL
LOGIC
LDOREF
10 Ω
VIN
PVCC
EN_LDO
POWER-ON
POWER-ON
VREF2
SEQUENCE
SQUENCE
EN1
VREF2
CLEAR
FAULT
CLEAR
FAULT
LATCH
LATCH
EN2
THERMAL
THERMAL
SHUTDOWN
SHUTDOWN
VREF1
VREF1
FIGURE 20. DETAIL FUNCTIONAL DIAGRAM ISL8112
16
FN6396.0
November 21, 2006
ISL8112
FS
Min. tOFF
Q
TRIG
ONE SHOT
VIN
+
TO UG_DRIVER
Q
R Q
VSEN_
S Q
Q
OUT2REF (SMPS2)
VREF
COMP
SLOPE COMP
+
+
+
++
ILIM_
+
5µA
VCC
BOOT_
BOOT
UV
DETECT
+
TO LG_ DRIVER
Â
S
+
PH_
Q
S Q
+
VSEN_
R Q
Q
MODE
+
PGOOD_
+
FB
DECODER
OV_LATCH_
0.9VREF
FB_
1.1VREF
UV_LATCH_
+
0.7VREF
FAULT
FAULT
LATCH
LATCH
LOGIC
20ms
BLANKING
FIGURE 21. PWM CONTROLLER (ONE SIDE ONLY)
Automatic Pulse-Skipping Switch Over
(Idle Mode)
K ⋅ V OUT V IN – V OUT
I LOAD ( SKIP ) = ------------------------ -------------------------------2⋅L
V IN
=
VIN-VOUT
L
IPEAK
ILOAD= IPEAK /2
(EQ. 3)
where K is the on-time scale factor (see “On-Time One-Shot
(FS)” on page 12). The load-current level at which
PFM/PWM crossover occurs, ILOAD(SKIP), is equal to half
the peak-to-peak ripple current, which is a function of the
inductor value (Figure 22). For example, in the ISL8112
typical application circuit with VOUT1 = 5V, VIN = 12V,
L = 7.6µH, and K = 5µs, switch over to pulse-skipping
operation occurs at ILOAD = 0.96A or about on-fifth full load.
The crossover point occurs at an even lower value if a
swinging (soft-saturation) inductor is used.
17
t
INDUCTOR CURRENT
In Idle Mode (MODE = GND), an inherent automatic switch
over to PFM takes place at light loads. This switch over is
affected by a comparator that truncates the low-side switch
on-time at the inductor current's zero crossing. This
mechanism causes the threshold between pulse-skipping
PFM and non-skipping PWM operation to coincide with the
boundary between continuous and discontinuous
inductor-current operation (also known as the critical
conduction point):
ΔII
0
ON-TIME
TIME
FIGURE 22. ULTRASONIC CURRENT WAVEFORMS
The switching waveforms may appear noisy and
asynchronous when light loading causes pulse-skipping
operation, but this is a normal operating condition that
results in high light-load efficiency. Trade-offs in PFM noise
vs. light-load efficiency are made by varying the inductor
value. Generally, low inductor values produce a broader
efficiency vs. load curve, while higher values result in higher
full-load efficiency (assuming that the coil resistance remains
fixed) and less output voltage ripple. Penalties for using
FN6396.0
November 21, 2006
ISL8112
higher inductor values include larger physical size and
degraded load-transient response (especially at low
input-voltage levels).
DC output accuracy specifications refer to the trip level of the
error comparator. When the inductor is in continuous
conduction, the output voltage has a DC regulation higher
than the trip level by 50% of the ripple. In discontinuous
conduction (MODE = GND, light load), the output voltage
has a DC regulation higher than the trip level by
approximately 1.0% due to slope compensation.
40µs (MAX)
INDUCTOR
CURRENT
ZERO-CROSSING
DETECTION
DETECTION
0A
FB<REG.POINT
FB<REG.POINT
Forced-PWM Mode
The low-noise, forced-PWM (MODE = VCC) mode disables
the zero-crossing comparator, which controls the low-side
switch on-time. Disabling the zero-crossing detector causes
the low-side, gate-drive waveform to become the
complement of the high-side, gate-drive waveform. The
inductor current reverses at light loads as the PWM loop
strives to maintain a duty ratio of VOUT/VIN. The benefit of
forced-PWM mode is to keep the switching frequency fairly
constant, but it comes at a cost: the no-load battery current
can be 10mA to 50mA, depending on switching frequency
and the external MOSFETs.
Reference and Linear Regulators (VREF2,
VREF1, and LDO)
Forced-PWM mode is most useful for reducing
audio-frequency noise, improving load-transient response,
providing sink-current capability for dynamic output voltage
adjustment, and improving the cross-regulation of
multiple-output applications that use a flyback transformer or
coupled inductor.
The 2V reference (VREF1) is accurate to ±1% over
temperature, also making VREF1 useful as a precision
system reference. Bypass VREF1 to GND with a 0.1µF (min)
capacitor. VREF1 can supply up to 50µA for external loads.
Enhanced Ultrasonic Mode
(25kHz (min) Pulse Skipping)
Leaving MODE unconnected or connecting MODE to
VREF1 activates a unique pulse-skipping mode with a
minimum switching frequency of 25kHz. This ultrasonic
pulse-skipping mode eliminates audio-frequency modulation
that would otherwise be present when a lightly loaded
controller automatically skips pulses. In ultrasonic mode, the
controller automatically transitions to fixed-frequency PWM
operation when the load reaches the same critical
conduction point (ILOAD(SKIP)).
An ultrasonic pulse occurs when the controller detects that
no switching has occurred within the last 20µs. Once
triggered, the ultrasonic controller pulls LG high, turning on
the low-side MOSFET to induce a negative inductor current.
After FB drops below the regulation point, the controller turns
off the low-side MOSFET (LG pulled low) and triggers a
constant on-time (UG driven high). When the on-time has
expired, the controller re-enables the low-side MOSFET until
the controller detects that the inductor current dropped
below the zero-crossing threshold. Starting with a LG pulse
greatly reduces the peak output voltage when compared to
starting with a UG pulse, as long as VFB < VREF, LG is off
and UG is on, similar to pure SKIP mode.
18
ON-TIME (TON
ON)
FIGURE 23. ULTRASONIC CURRENT WAVEFORMS
The 3.3V reference (VREF2) is accurate to ±1.5% over
temperature, making VREF2 useful as a precision system
reference. VREF2 can supply up to 5mA for external loads.
Bypass VREF2 to GND with a 0.01µF capacitor. Leave open
if there is no load.
An internal regulator produces a fixed 5V (LDOREF < 0.2V)
or 3.3V (LDOREF > VCC - 1V). In an adjustable mode, the
LDO output can be set from 0.7V to 4.5V. The LDO output
voltage is equal to two times the LDOREF voltage. The LDO
regulator can supply up to 100mA for external loads. Bypass
LDO with a minimum 4.7µF ceramic capacitor. When the
LDOREF < 0.2V and BYP voltage is 5V, the LDO bootstrapswitch over to an internal 0.7Ω p-channel MOSFET switch
connects BYP to LDO pin while simultaneously shutting
down the internal linear regulator. These actions bootstrap
the device, powering the loads from the BYP input voltages,
rather than through internal linear regulators from the
battery. Similarly, when the BYP = 3.3V and
LDOREF = VCC, the LDO bootstrap-switch over to an
internal 1.5Ω P-Channel MOSFET switch connects BYP to
LDO pin while simultaneously shutting down the internal
linear regulator. No switch over action in adjustable mode.
Current-Limit Circuit (ILIM_) with rDS(ON)
Temperature Compensation
The current-limit circuit employs a "valley" current-sensing
algorithm. The ISL8112 uses the on-resistance of the
synchronous rectifier as a current-sensing element. If the
magnitude of the current-sense signal at PH_ is above the
current-limit threshold, the PWM is not allowed to initiate a
new cycle. The actual peak current is greater than the
FN6396.0
November 21, 2006
ISL8112
current-limit threshold by an amount equal to the inductor
ripple current. Therefore, the exact current-limit
characteristic and maximum load capability are a function of
the current-limit threshold, inductor value and input and
output voltage.
INDUCTOR CURRENT
I PEAK
I LOAD
ΔI
I LIMIT
I LOAD(MAX)
A negative current limit prevents excessive reverse inductor
currents when VOUT sinks current. The negative
current-limit threshold is set to approximately 120% of the
positive current limit and therefore tracks the positive current
limit when ILIM_ is adjusted. The current-limit threshold is
adjusted with an external resistor for ISL8112 at ILIM_. The
current-limit threshold adjustment range is from 20mV to
200mV. In the adjustable mode, the current-limit threshold
voltage is 1/10th the voltage at ILIM_. The voltage at ILIM
pin is the product of 5µA * RILIM. The threshold defaults to
100mV when ILIM_ is connected to VCC. The logic
threshold for switch-over to the 100mV default value is
approximately VCC - 1V.
The PC board layout guidelines should be carefully
observed to ensure that noise and DC errors do not corrupt
the current-sense signals at PH_.
I LIM( VAL ) = ILOAD - Δ I
2
MOSFET Gate Drivers (UG_, LG_)
TIME
FIGURE 24. “VALLEY” CURRENT LIMIT THRESHOLD POINT
For lower power dissipation, the ISL8112 uses the
on-resistance of the synchronous rectifier as the
current-sense element. Use the worst-case maximum value
for rDS(ON) from the MOSFET data sheet. Add some margin
for the rise in rDS(ON) with temperature. A good general rule
is to allow 0.5% additional resistance for each °C of
temperature rise. The ISL8112 controller has a built-in 5µA
current source as shown in Figure 25. Place the hottest
power MOSEFTs as close to the IC as possible for best
thermal coupling. The current limit varies with the onresistance of the synchronous rectifier. When combined with
the undervoltage-protection circuit, this current-limit method
is effective in almost every circumstance.
The UG_ and LG_ gate drivers sink 2.0A and 3.3A
respectively of gate drive, ensuring robust gate drive for
high-current applications. The UG_ floating high-side
MOSFET drivers are powered by diode-capacitor charge
pumps at BOOT_. The LG_ synchronous-rectifier drivers are
powered by PVCC.
The internal pull-down transistors that drive LG_ low have a
0.6Ω typical on-resistance. These low on-resistance
pulldown transistors prevent LG_ from being pulled up
during the fast rise time of the inductor nodes due to
capacitive coupling from the drain to the gate of the low-side
synchronous-rectifier MOSFETs. However, for high-current
applications, some combinations of high- and low-side
MOSFETs may cause excessive gate-drain coupling, which
leads to poor efficiency and EMI-producing shoot-through
currents. Adding a 4.7Ω resistor in series with BOOT_
increases the turn-on time of the high-side MOSFETs at the
expense of efficiency, without degrading the turn-off time
(Figure 26).
ILIM_
+
5V
BOOT_
5µA
4.7Ω
+
R ILIM
VILIM
9R
VCC
TO CURRENT
LIMIT LOGIC
VIN
UG_
Q1
R
C BOOT
OUT_
PH_
ISL8112
FIGURE 25. CURRENT LIMIT BLOCK DIAGRAM
FIGURE 26. REDUCING THE SWITCHING-NODE RISE TIME
19
FN6396.0
November 21, 2006
ISL8112
Adaptive dead-time circuits monitor the LG_ and UG_
drivers and prevent either FET from turning on until the other
is fully off. This algorithm allows operation without shootthrough with a wide range of MOSFETs, minimizing delays
and maintaining efficiency. There must be low-resistance,
low-inductance paths from the gate drivers to the MOSFET
gates for the adaptive dead-time circuit to work properly.
Otherwise, the sense circuitry interprets the MOSFET gate
as "off" when there is actually charge left on the gate. Use
very short, wide traces measuring 10 to 20 squares (50 mils
to 100 mils wide if the MOSFET is 1” from the device).
Boost-Supply Capacitor Selection (Buck)
The boost capacitor should be 0.1µF to 4.7µF, depending on
the input and output voltages, external components, and PC
board layout. The boost capacitance should be as large as
possible to prevent it from charging to excessive voltage, but
small enough to adequately charge during the minimum
low-side MOSFET conduction time, which happens at
maximum operating duty cycle (this occurs at minimum input
voltage). The minimum gate to source voltage (VGS(MIN)) is
determined by:
C BOOT
V GS ( MIN ) = PVCC ⋅ --------------------------------------C BOOT + C GS
(EQ. 4)
where:
• PVCC is 5V
Fault Protection
The ISL8112 provides overvoltage/undervoltage fault
protection in the buck controllers. Once activated, the
controller continuously monitors the output for undervoltage
and overvoltage fault conditions.
• Out-of-bound Condition
When the output voltage is 5% above the set voltage, the
out-of-bound condition activates. LG turns on until output
reaches within regulation. Once the output is within
regulation, the controller will operate as normal. It is the
"first line of defense" before OVP.
• Overvoltage Protection
When VSEN1 is 11% (16% for for VSEN2) above the set
voltage, the overvoltage fault protection activates. This
latches on the synchronous rectifier MOSFET with 100%
duty cycle, rapidly discharging the output capacitor until
the negative current limit is achieved. Once negative
current limit is met, UG is turned on for a minimum ontime, followed by another LG pulse until negative current
limit. This effectively regulates the discharge current at the
negative current limit in an effort to prevent excessively
large negative currents that cause potentially damaging
negative voltages on the load. Once an overvoltage fault
condition is set, it can only be reset by toggling SHDN#,
EN_, or cycling VIN (POR).
• Undervoltage Protection
• CGS is the gate capacitance of the high-side MOSFET
POR, UVLO, and Internal Digital Soft-Start
Power-on reset (POR) occurs when VIN rises above
approximately 3V, resetting the undervoltage, overvoltage,
and thermal-shutdown fault latches. PVCC undervoltage
lockout (UVLO) circuitry inhibits switching when PVCC is
below 4V. LG_ is low during UVLO. The output voltages begin
to ramp up once PVCC exceeds its 4V UVLO and VREF1 is in
regulation. The internal digital soft-start timer begins to ramp
up the maximum-allowed current limit during start-up. The
1.7ms ramp occurs in five steps of positive current limit and
the step size is 20%, 40%, 60%, 80% and 100%.
Power-Good Output (PGOOD_)
The PGOOD_ comparator continuously monitors both output
voltages for undervoltage conditions. PGOOD_ is actively
held low in shutdown, standby, and soft-start. PGOOD1
releases and digital soft-start terminates when VSEN1 reach
the error-comparator threshold. PGOOD1 goes low if VOUT1
output turns off or is 10% below its nominal regulation point.
PGOOD1 is a true open-drain output. Likewise, PGOOD2 is
used to monitor VSEN2.
20
When the output voltage drops below 70% of its regulation
voltage for at least 100µs, the controller sets the fault latch
and begins the discharge mode (see the Shutdown and
Output Discharge section). UVP is ignored for at least
20ms (typical), after start-up or after a rising edge on EN_.
Toggle EN_ or cycle VIN (POR) to clear the undervoltage
fault latch and restart the controller. UVP only applies to
the buck outputs.
• Thermal Protection
The ISL8112 has thermal shutdown to protect the devices
from overheating. Thermal shutdown occurs when the die
temperature exceeds +150°C. All internal circuitry shuts
down during thermal shutdown. The ISL8112 may trigger
thermal shutdown if LDO_ is not bootstrapped from
VSEN_ while applying a high input voltage on VIN and
drawing the maximum current (including short circuit) from
LDO_. Even if LDO_ is bootstrapped from VSEN_,
overloading the LDO_ causes large power dissipation on
the bootstrap switches, which may result in thermal
shutdown. Cycling EN_, EN_LDO, or VIN (POR) ends the
thermal-shutdown state.
FN6396.0
November 21, 2006
ISL8112
Discharge Mode (Soft-Stop)
Power-Up Sequencing and On/Off Controls (EN_)
When a transition to standby or shutdown mode occurs, or
the output undervoltage fault latch is set, the outputs
discharge to GND through an internal 25Ω switch.
The reference remains active to provide an accurate
threshold and to provide overvoltage protection.
EN1 and EN2 control SMPS power-up sequencing. EN1 or
EN2 rising above 2.4V enables the respective outputs. EN1
or EN2 falling below 1.6V disables the respective outputs.
Connecting EN1 or EN2 to VREF1 will force its outputs off
while the other output is below regulation. The sequenced
SMPS will start once the other SMPS reaches regulation.
The second SMPS remains on until the first SMPS turns off,
the device shuts down, a fault occurs or PVCC goes into
undervoltage lockout. Both supplies begin their power-down
sequence immediately when the first supply turns off. Driving
EN_ below 0.8V clears the overvoltage, undervoltage and
thermal fault latches.
Shutdown Mode
The ISL8112 SMPS1, SMPS2 and LDO have independent
enabling control. Drive EN1, EN2 and EN_LDO below the
precise input falling-edge trip level to place the ISL8112 in its
low-power shutdown state. The ISL8112 consumes only
20µA of quiescent current while in shutdown. When
shutdown mode activates, the 3.3V VREF2 remain on. Both
SMPS outputs are discharged to 0V through a 25Ω switch.
TABLE 3. OPERATING-MODE TRUTH TABLE
MODE
CONDITION
COMMENT
Power-Up
PVCC < UVLO threshold.
Transitions to discharge mode after a VIN POR and after VREF1 becomes valid.
LDO, VREF2, and VREF1 remain active.
Run
EN_LDO = high, EN1 or EN2
enabled.
Normal operation
Overvoltage
Protection
Either output > 111% (VSEN1) or
116% (VSEN2) of nominal level.
LG_ is forced high. LDO, VREF2 and VREF1 active. Exited by a VIN POR, or by
toggling EN1 or EN2.
Undervoltage
Protection
Either output < 70% of nominal after
20ms time-out expires and output is
enabled.
The internal 25Ω switch turns on. LDO, VREF2 and VREF1 are active. Exited by a
VIN POR or by toggling EN1 or EN2.
Discharge
Either SMPS output is still high in
either standby mode or shutdown
mode
Discharge switch (25Ω) connects VSEN_ to GND. One output may still run while the
other is in discharge mode. Activates when PVCC is in UVLO, or transition to UVLO,
standby, or shutdown has begun. LDO, VREF2 and VREF1 active.
Standby
EN1, EN2 < startup threshold,
EN_LDO= High
LDO, VREF2 and VREF1 active.
Shutdown
EN1, EN2, EN_LDO = low
Discharge switch (25Ω) connects VSEN_ to PGND. All circuitry off except VREF2.
Thermal Shutdown
TJ > +150°C
All circuitry off. Exited by VIN POR or cycling EN_. VREF2 remain active.
TABLE 4. SHUTDOWN AND STANDBY CONTROL LOGIS
VEN_LDO
VEN1 (V)
VEN2 (V)
LDO
SMPS1
SMPS2
Low
Low
Low
Off
Off
Off
“>2.5” → High
Low
Low
On
Off
Off
“>2.5” → High
High
High
On
On
On
“>2.5” → High
High
Low
On
On
Off
“>2.5” → High
Low
High
On
Off
On
“>2.5” → High
High
VREF1
On
On
On (after SMPS1 is up)
“>2.5” → High
VREF1
High
On
On (after SMPS2 is up)
On
21
FN6396.0
November 21, 2006
ISL8112
Adjustable-Output Feedback (Dual-Mode FB)
Connect FB1 to GND to enable the fixed 5V or tie FB1 to
VCC to set the fixed 1.5V output. Connect a resistive
voltage-divider at FB1 between output and GND to adjust
the respective output voltage between 0.7V and 5.5V
(Figure 27). Choose R2 to be approximately 10k and solve
for R1 using Equation 5.
⎛ V OUT1
⎞
R1 = R2 ⋅ ⎜ ------------------- – 1⎟
⎝ V FB1
⎠
(EQ. 5)
VIN
Q3
ISL88732
ISL88733
ISL8112
ISL6236
ISL88734
ISL88734
OUT1
LGATE_
LGATE1
LG1
Q4
OUT1
VOUT_
VOUT_
VSEN1
(EQ. 6)
OUT2
where:
• VR = 2V nominal (if tied to VREF1)
or
• VR = 3.3V nominal (if tied to VREF2)
Design Procedure
Establish the input voltage range and maximum load current
before choosing an inductor and its associated ripple-current
ratio (LIR). The following four factors dictate the rest of the
design:
where VFB1 = 0.7V nominal.
UG1
UGATE_
UGATE1
VR
R3 = R4 ⋅ ⎛ ------------------- – 1⎞
⎝V
⎠
1. Input Voltage Range. The maximum value (VIN(MAX))
must accommodate the maximum AC adapter voltage.
The minimum value (VIN(MIN)) must account for the
lowest input voltage after drops due to connectors, fuses
and battery selector switches. Lower input voltages result
in better efficiency.
2. Maximum Load Current. The peak load current
(ILOAD(MAX)) determines the instantaneous component
stress and filtering requirements and thus drives output
capacitor selection, inductor saturation rating and the
design of the current-limit circuit. The continuous load
current (ILOAD) determines the thermal stress and drives
the selection of input capacitors, MOSFETs and other
critical heat-contributing components.
R1
FB1
FB_
FB1
R2
3. Switching Frequency. This choice determines the basic
trade-off between size and efficiency. The optimal
frequency is largely a function of maximum input voltage
and MOSFET switching losses.
FIGURE 27. SETTING VOUT1 WITH A RESISTOR DIVIDER
Likewise, connect OUT2REF to VCC to enable the fixed
3.3V or tie OUT2REF to VREF2 to set the fixed 1.05V
output. Set OUT2REF from 0 to 2.50V for SMPS2 tracking
mode (Figure ).
VIN
UG2
UGATE_
UGATE2
Q1
ISL88732
ISL88733
ISL6236
ISL8112
ISL88734
OUT2
LG2
LGATE_
LGATE2
Q2
VSEN2
VOUT_
OUT2
OUT2REF
FB_
REFIN2
VR
R3
R4
FIGURE 28. SETTING VOUT2 WITH A VOLTAGE DIVIDER FOR
TRACKING
22
4. Inductor Ripple Current Ratio (LIR). LIR is the ratio of the
peak-peak ripple current to the average inductor current.
Size and efficiency trade-offs must be considered when
setting the inductor ripple current ratio. Low inductor
values cause large ripple currents, resulting in the
smallest size, but poor efficiency and high output noise.
The minimum practical inductor value is one that causes
the circuit to operate at critical conduction (where the
inductor current just touches zero with every cycle at
maximum load). Inductor values lower than this grant no
further size-reduction benefit.
The ISL8112 pulse-skipping algorithm (MODE = GND)
initiates skip mode at the critical conduction point, so the
inductor's operating point also determines the load
current at which PWM/PFM switch over occurs. The
optimum point is usually found between 20% and 50%
ripple current.
Inductor Selection
The switching frequency (on-time) and operating point (%
ripple or LIR) determine the inductor value as follows:
V OUT_ ( V IN + V OUT_ )
L = --------------------------------------------------------------------V IN ⋅ f ⋅ LIR ⋅ I LOAD ( MAX )
(EQ. 7)
FN6396.0
November 21, 2006
ISL8112
Output Capacitor Selection
Example: ILOAD(MAX) = 5A, VIN = 12V, VOUT2 = 5V,
f = 200kHz, 35% ripple current or LIR = 0.35:
5V ( 12V – 5V )
L = ----------------------------------------------------------------- = 8.3μH
12V ⋅ 200kHz ⋅ 0.35 ⋅ 5A
(EQ. 8)
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite cores
are often the best choice. The core must be large enough
not to saturate at the peak inductor current (IPEAK):
IPEAK = I LOAD ( MAX ) + [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ]
(EQ. 9)
The inductor ripple current also impacts transient response
performance, especially at low VIN - VSEN_ differences. Low
inductor values allow the inductor current to slew faster,
replenishing charge removed from the output filter capacitors
by a sudden load step. The peak amplitude of the output
transient (VSAG) is also a function of the maximum duty
factor, which can be calculated from the on-time and
minimum off-time:
The output filter capacitor must have low enough equivalent
series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR to
satisfy stability requirements. The output capacitance must
also be high enough to absorb the inductor energy while
transitioning from full-load to no-load conditions without
tripping the overvoltage fault latch. In applications where the
output is subject to large load transients, the output
capacitor's size depends on how much ESR is needed to
prevent the output from dipping too low under a load transient.
Ignoring the sag due to finite capacitance:
V DIP
R SER ≤ ---------------------------------I LOAD ( MAX )
(EQ. 14)
where VDIP is the maximum-tolerable transient voltage drop.
In non-CPU applications, the output capacitor's size depends
on how much ESR is needed to maintain an acceptable level
of output voltage ripple:
VP – P
R ESR ≤ ----------------------------------------------L IR ⋅ I LOAD ( MAX )
⎞⎞
⎛ ⎛ V OUT_
2
( ΔI LOAD ( MAX ) ) ⋅ L ⎜ K ⎜ ------------------- + t OFF ( MIN )⎟ ⎟
V
⎠⎠
⎝ ⎝
IN
VSAG = ---------------------------------------------------------------------------------------------------------------------------–
V
V
⎛ IN
OUT⎞
2 ⋅ C OUT ⋅ V OUT K ⎜ --------------------------------⎟ - t
V IN
⎝
⎠ OFF ( MIN )
(EQ. 10)
where minimum off-time = 0.35µs (max) and K is from
Table 2.
Determining the Current Limit
(EQ. 15)
where VP-P is the peak-to-peak output voltage ripple. The
actual capacitance value required relates to the physical size
needed to achieve low ESR, as well as to the chemistry of the
capacitor technology. Thus, the capacitor is usually selected
by ESR and voltage rating rather than by capacitance value
(this is true of tantalum, OS-CON, and other electrolytic-type
capacitors).
The minimum current-limit threshold must be great enough
to support the maximum load current when the current limit
is at the minimum tolerance value. The valley of the inductor
current occurs at ILOAD(MAX) minus half of the ripple
current; therefore:
When using low-capacity filter capacitors such as polymer
types, capacitor size is usually determined by the capacity
required to prevent VSAG and VSOAR from tripping the
undervoltage and overvoltage fault latches during load
transients in ultrasonic mode.
I LIMIT ( LOW ) > I LOAD ( MAX ) – [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ]
For low input-to-output voltage differentials (VIN/ VOUT < 2),
additional output capacitance is required to maintain stability
and good efficiency in ultrasonic mode. The amount of
overshoot due to stored inductor energy can be calculated as:
(EQ. 11)
where: ILIMIT(LOW) = minimum current-limit threshold
voltage divided by the rDS(ON) of Q2/Q4.
Use the worst-case maximum value for rDS(ON) from the
MOSFET Q2/Q4 data sheet and add some margin for the
rise in rDS(ON) with temperature. A good general rule is to
allow 0.2% additional resistance for each °C of temperature
rise. Examining the 5A circuit example with a maximum
rDS(ON) = 5mΩ at room temperature. At +125°C reveals the
following:
I LIMIT ( LOW ) = ( 25mV ) ⁄ ( ( 5mΩ × 1.2 ) > 5A – ( 0.35 ⁄ 2 )5A )
(EQ. 12)
4.17A > 4.12A
(EQ. 13)
4.17A is greater than the valley current of 4.12A, so the circuit
can easily deliver the full-rated 5A using the 30mV nominal
current-limit threshold voltage.
23
2
I PEAK ⋅ L
V SOAR = -----------------------------------------------2 ⋅ C OUT ⋅ V OUT_
(EQ. 16)
where IPEAK is the peak inductor current.
Input Capacitor Selection
The input capacitors must meet the input-ripple-current
(IRMS) requirement imposed by the switching current. The
ISL8112 dual switching regulator operates at different
frequencies. This interleaves the current pulses drawn by
the two switches and reduces the overlap time where they
add together. The input RMS current is much smaller in
comparison than with both SMPSs operating in phase. The
input RMS current varies with load and the input voltage.
The maximum input capacitor RMS current for a single
SMPS is given by:
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ISL8112
⎛ V OUT ( V IN – V OUT_ )⎞
I RMS ≈ I LOAD ⎜ ------------------------------------------------------------⎟
V IN
⎝
⎠
(EQ. 17)
When V IN = 2 ⋅ V OUT_ ( D = 50% ) , IRMS has maximum
current of I LOAD ⁄ 2 .
The ESR of the input-capacitor is important for determining
capacitor power dissipation. All the power (IRMS2 x ESR)
heats up the capacitor and reduces efficiency. Nontantalum
chemistries (ceramic or OS-CON) are preferred due to their
low ESR and resilience to power-up surge currents. Choose
input capacitors that exhibit less than +10°C temperature
rise at the RMS input current for optimal circuit longevity.
Place the drains of the high-side switches close to each
other to share common input bypass capacitors.
adequate rDS(ON) at low battery voltages if it becomes
extraordinarily hot when subjected to VIN(MAX).
Calculating the power dissipation in NH (Q1/Q3) due to
switching losses is difficult since it must allow for quantifying
factors that influence the turn-on and turn-off times. These
factors include the internal gate resistance, gate charge,
threshold voltage, source inductance, and PC board layout
characteristics. The following switching-loss calculation
provides only a very rough estimate and is no substitute for
bench evaluation, preferably including verification using a
thermocouple mounted on NH (Q1/Q3):
2 ⎛ C RSS ⋅ f SW ⋅ I LOAD⎞
PD ( Q H Switching ) = ( V IN ( MAX ) ) ⎜ -----------------------------------------------------⎟
I GATE
⎝
⎠
(EQ. 19)
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability (>5A)
when using high-voltage (>20V) AC adapters. Low-current
applications usually require less attention.
Choose a high-side MOSFET (Q1/Q3) that has conduction
losses equal to the switching losses at the typical battery
voltage for maximum efficiency. Ensure that the conduction
losses at the minimum input voltage do not exceed the
package thermal limits or violate the overall thermal budget.
Ensure that conduction losses plus switching losses at the
maximum input voltage do not exceed the package ratings
or violate the overall thermal budget.
Choose a synchronous rectifier (Q2/Q4) with the lowest
possible rDS(ON). Ensure the gate is not pulled up by the
high-side switch turning on due to parasitic drain-to-gate
capacitance, causing cross-conduction problems. Switching
losses are not an issue for the synchronous rectifier in the
buck topology since it is a zero-voltage switched device
when using the buck topology.
Worst-case conduction losses occur at the duty-factor
extremes. For the high-side MOSFET, the worst-case power
dissipation (PD) due to the MOSFET's rDS(ON) occurs at the
minimum battery voltage:
(EQ. 18)
Generally, a small high-side MOSFET reduces switching
losses at high input voltage. However, the rDS(ON) required
to stay within package power-dissipation limits often limits
how small the MOSFET can be. The optimum situation
occurs when the switching (AC) losses equal the conduction
(rDS(ON)) losses.
Switching losses in the high-side MOSFET can become an
insidious heat problem when maximum battery voltage is
applied, due to the squared term in the CV2f switching-loss
equation. Reconsider the high-side MOSFET chosen for
24
For the synchronous rectifier, the worst-case power
dissipation always occurs at maximum battery voltage:
V OUT ⎞
⎛
2
PD ( Q L ) = ⎜ 1 – --------------------------⎟ I LOAD ⋅ r DS ( ON )
V IN ( MAX )⎠
⎝
(EQ. 20)
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
ILOAD(MAX) but are not quite high enough to exceed the
current limit and cause the fault latch to trip. To protect
against this possibility, "overdesign" the circuit to tolerate:
I LOAD = I LIMIT ( HIGH ) + ( ( LIR ) ⁄ 2 ) ⋅ I LOAD ( MAX )
(EQ. 21)
where ILIMIT(HIGH) is the maximum valley current allowed
by the current-limit circuit, including threshold tolerance and
resistance variation.
Rectifier Selection
MOSFET Power Dissipation
⎛ V OUT_ ⎞
2
PD ( Q H Resistance ) = ⎜ ------------------------⎟ ( I LOAD ) ⋅ r DS ( ON )
⎝ V IN ( MIN )⎠
where CRSS is the reverse transfer capacitance of QH
(Q1/Q3) and IGATE is the peak gate-drive source/sink
current.
Current circulates from ground to the junction of both
MOSFETs and the inductor when the high-side switch is off.
As a consequence, the polarity of the switching node is
negative with respect to ground. This voltage is
approximately -0.7V (a diode drop) at both transition edges
while both switches are off (dead time). The drop is
I L ⋅ r DS ( ON ) when the low-side switch conducts.
The rectifier is a clamp across the synchronous rectifier that
catches the negative inductor swing during the dead time
between turning the high-side MOSFET off and the
synchronous rectifier on. The MOSFETs incorporate a
high-speed silicon body diode as an adequate clamp diode if
efficiency is not of primary importance. Place a Schottky
diode in parallel with the body diode to reduce the forward
voltage drop and prevent the Q2/Q4 MOSFET body diodes
from turning on during the dead time. Typically, the external
diode improves the efficiency by 1% to 2%. Use a Schottky
diode with a DC current rating equal to one-third of the load
FN6396.0
November 21, 2006
ISL8112
current. For example, use an MBR0530 (500mA-rated) type
for loads up to 1.5A, a 1N5817 type for loads up to 3A, or a
1N5821 type for loads up to 10A. The rectifier's rated
reverse breakdown voltage must be at least equal to the
maximum input voltage, preferably with a 20% derating
factor.
Applications Information
Dropout Performance
The output voltage-adjust range for continuous-conduction
operation is restricted by the nonadjustable 350ns (max)
minimum off-time one-shot. Use the slower 5V SMPS for the
higher of the two output voltages for best dropout
performance in adjustable feedback mode. The duty-factor
limit must be calculated using worst-case values for on-times
and off-times, when working with low input voltages.
Manufacturing tolerances and internal propagation delays
introduce an error to the FS K-factor. Also, keep in mind that
transient-response performance of buck regulators operated
close to dropout is poor, and bulk output capacitance must
often be added (see Equation 10 on page 23).
The absolute point of dropout occurs when the inductor
current ramps down during the minimum off-time (ΔIDOWN)
as much as it ramps up during the on-time ( ΔIUP). The ratio
h = ΔIUP/ΔIDOWN indicates the ability to slew the inductor
current higher in response to increased load, and must
always be greater than 1. As h approaches 1, the absolute
minimum dropout point, the inductor current is less able to
increase during each switching cycle and VSAG greatly
increases unless additional output capacitance is used.
A reasonable minimum value for h is 1.5, but this can be
adjusted up or down to allow trade-offs between VSAG,
output capacitance and minimum operating voltage. For a
given value of h, the minimum operating voltage can be
calculated as:
( V OUT_ + V DROP )
V IN ( MIN ) = --------------------------------------------------- + V DROP2 – V DROP1
t OFF ( MIN ) ⋅ h
1 – ⎛ ------------------------------------⎞
⎝
⎠
K
(EQ. 22)
where VDROP1 and VDROP2 are the parasitic voltage drops
in the discharge and charge paths (see “On-Time One-Shot
(FS)” on page 12), tOFF(MIN) is from the Electrical
Specifications table on page 3 and K is taken from Table 2.
The absolute minimum input voltage is calculated with h = 1.
Operating frequency must be reduced or h must be
increased and output capacitance added to obtain an
acceptable VSAG if calculated VIN(MIN) is greater than the
required minimum input voltage. Calculate VSAG to be sure
of adequate transient response if operation near dropout is
anticipated.
Dropout Design Example:
ISL8112: With VOUT2 = 5V, fsw = 400kHz, K = 2.25µs,
tOFF(MIN) = 350ns, VDROP1 = VDROP2 = 100mV, and h = 1.5,
the minimum VIN is:
25
( 5V + 0.1V )
V IN ( MIN ) = ---------------------------------------------- + 0.1V – 0.1V = 6.65V
0.35μs ⋅ 1.5
1 – ⎛ -------------------------------⎞
⎝ 2.25μs ⎠
(EQ. 23)
Calculating with h = 1 yields:
( 5V + 0.1V )
V IN ( MIN ) = ----------------------------------------- + 0.1V – 0.1V = 6.04V
0.35μs ⋅ 1
1 – ⎛ --------------------------⎞
⎝ 2.25μs ⎠
(EQ. 24)
Therefore, VIN must be greater than 6.65V. A practical input
voltage with reasonable output capacitance would be 7.5V.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve minimal
switching losses and clean, stable operation. This is
especially true when multiple converters are on the same PC
board where one circuit can affect the other. Refer to the
ISL8112 Evaluation Kit data sheet for a specific layout
example.
Mount all of the power components on the top side of the
board with their ground terminals flush against one another,
if possible. Follow these guidelines for good PC board
layout:
• Isolate the power components on the top side from the
sensitive analog components on the bottom side with a
ground shield. Use a separate PGND plane under the
VSEN1 and VSEN2 sides (called PGND1 and PGND2).
Avoid the introduction of AC currents into the PGND1 and
PGND2 ground planes. Run the power plane ground
currents on the top side only, if possible.
• Use a star ground connection on the power plane to
minimize the crosstalk between VSEN1 and VSEN2.
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
• Keep the power traces and load connections short. This
practice is essential for high efficiency. Using thick copper
PC boards (2oz vs. 1oz) can enhance full-load efficiency
by 1% or more. Correctly routing PC board traces must be
approached in terms of fractions of centimeters, where a
single mΩ of excess trace resistance causes a
measurable efficiency penalty.
• PH_ (ISL8112) and GND connections to the synchronous
rectifiers for current limiting must be made using Kelvinsense connections to guarantee the current-limit accuracy
with 8-pin SO MOSFETs. This is best done by routing
power to the MOSFETs from outside using the top copper
layer, while connecting PH_ traces inside (underneath) the
MOSFETs.
• When trade-offs in trace lengths must be made, it is
preferable to allow the inductor charging path to be made
longer than the discharge path. For example, it is better to
allow some extra distance between the input capacitors
and the high-side MOSFET than to allow distance
between the inductor and the synchronous rectifier or
between the inductor and the output filter capacitor.
FN6396.0
November 21, 2006
ISL8112
• Ensure that the VSEN_ connection to COUT_ is short and
direct. However, in some cases it may be desirable to
deliberately introduce some trace length between the
VSEN_ connector node and the output filter capacitor (see
the Stability Considerations section).
• Route high-speed switching nodes (BOOT_, UG_, PH_,
and LG_) away from sensitive analog areas (VREF1,
ILIM_, and FB_). Use PGND1 and PGND2 as an EMI
shield to keep radiated switching noise away from the IC's
feedback divider and analog bypass capacitors.
• Make all pin-strap control input connections (MODE,
ILIM_, etc.) to GND or VCC of the device.
Layout Procedure
Place the power components first with ground terminals
adjacent (Q2/Q4 source, CIN_, COUT_). If possible, make
all these connections on the top layer with wide, copper-filled
areas.
Mount the controller IC adjacent to the synchronous rectifier
MOSFETs close to the hottest spot, preferably on the back
side in order to keep UG_, GND, and the LG_ gate drive
lines short and wide. The LG_ gate trace must be short and
wide, measuring 50 mils to 100 mils wide if the MOSFET is
1” from the controller device.
On the board's top side (power planes), make a star ground
to minimize crosstalk between the two sides. The top-side
star ground is a star connection of the input capacitors and
synchronous rectifiers. Keep the resistance low between the
star ground and the source of the synchronous rectifiers for
accurate current limit. Connect the top-side star ground
(used for MOSFET, input, and output capacitors) to the small
island with a single short, wide connection (preferably just a
via). Create PGND islands on the layer just below the
top-side layer (refer to the ISL8112 EV kit for an example) to
act as an EMI shield if multiple layers are available (highly
recommended). Connect each of these individually to the
star ground via, which connects the top side to the PGND
plane. Add one more solid ground plane under the device to
act as an additional shield, and also connect the solid
ground plane to the star ground via.
Connect the output power planes (VCORE and system
ground planes) directly to the output filter capacitor positive
and negative terminals with multiple vias.
Group the gate-drive components (BOOT_ capacitor, VIN
bypass capacitor) together near the controller device.
Make the DC/DC controller ground connections as follows:
1. Near the device, create a small analog ground plane.
2. Connect the small analog ground plane to GND and use
the plane for the ground connection for the VREF1 and
VCC bypass capacitors, FB dividers and ILIM resistors (if
any).
3. Create another small ground island for PGND and use
the plane for the VIN bypass capacitor, placed very close
to the device.
4. Connect the GND and PGND planes together at the
metal tab under device.
26
FN6396.0
November 21, 2006
ISL8112
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L32.5x5B
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHD-2 ISSUE C
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
A3
b
0.18
D
0.23
9
0.30
5,8
5.00 BSC
D1
D2
9
0.20 REF
-
4.75 BSC
3.15
3.30
9
3.45
7,8
E
5.00 BSC
-
E1
4.75 BSC
9
E2
3.15
e
3.30
3.45
7,8
0.50 BSC
-
k
0.25
-
-
-
L
0.30
0.40
0.50
8
L1
-
-
0.15
10
N
32
2
Nd
8
3
Ne
8
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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27
FN6396.0
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