V17N1 - MARCH

LINEAR TECHNOLOGY
MArch 2007
IN THIS ISSUE…
Cover Article
Easy Drive Delta-Sigma ADCs Deliver
Powerful Features and Reduce Design
Effort...................................................1
Mark Thoren
Linear Technology in the News…..........2
Design Features
Current Mode Flyback DC/DC Controller
Provides Tremendous Design Flexibility
............................................................8
Arthur Kelley
Isolated Forward Controllers Offer
Buck Simplicity and Performance......10
Charles Hawkes and Arthur Kelley
Rugged 3.3V RS485/RS422
Transceivers with Integrated
Switchable Termination.....................14
Steven Tanghe and Ray Schuler
Tiny High Efficiency 2A Buck Regulator
Directly Accepts Automotive, Industrial
and Other Wide Ranging Inputs.........18
Kevin Huang
36V Dual 1.4A Monolithic
Step-Down Converter has Start-Up
Tracking and Sequencing..................21
VOLUME XVII NUMBER 1
Easy Drive Delta-Sigma
ADCs Deliver Powerful
Features and Reduce
Design Effort
by Mark Thoren
Introduction
Easy Drive™ delta-sigma ADCs are
rich in features but easy to use. The
Easy Drive feature simplifies or eliminates active amplification or filtering
at the inputs. Even the software interface is significantly less complicated
than other ADCs (see sidebar on page
6). Overall, much of the traditional
complexity around an ADC, such as
external components and software
timing, is simply gone, saving significant design time.
Table 1 lists the features of the 18
available Easy Drive devices, including
1-, 4- or 16-channel versions with I2C
or SPI interfaces. The 24-bit devices
suit very high performance applications, while 16-bit devices are more
general-purpose. A programmable
gain amplifier (PGA) is available on
the 16-bit devices for intermediate
requirements or where several input
ranges need to be accommodated.
Easy Drive Technology
Simplifies Measurement of
High Impedance Sensors
Delta-Sigma ADCs, with their high
accuracy and high noise immunity,
are ideal for directly measuring many
continued on page Keith Szolusha
5V
Theo Phillips and Teo Yang Long
DESIGN IDEAS
.....................................................30–41
(complete list on page 30)
New Device Cameos............................42
Design Tools.......................................43
5V
R1
51.1k
3-Phase Buck Controller Governs
One, Two or Three Outputs................26
C4
0.1µF
12
IIN+ = 0
10µF
0.1µF
R4
51.1k
IIN– = 0
13
REF+
14
REF–
8
5V
9
10
5V
102k
11
+
Sales Offices......................................44
0.1µF
10k TO 100k
LT1494
–
1
FO
7
1k
CH0
2
SDI
3
SCK
SDO
= EXTERNAL
OSCILLATOR
= INTERNAL
OSCILLATOR
LTC2492
R3
10k TO 100k
C3
0.1µF
VCC
4-WIRE
SPI INTERFACE
5
CH1
CH2
CH3
COM
4
CS
GND
6
0.1µF
1k
0.1µF
Figure 1. Easy Drive ADCs simplify measurement of high impedance sensors.
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM,
UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L EDITOR’S PAGE
Linear Technology in the News…
EDN Innovation Award Finalists
EDN magazine announced that Linear Technology’s
LTC®6908 Resistor Set Oscillator with Spread Spectrum
Modulation is a finalist for their annual EDN Innovation Awards in the Analog ICs category. According to
EDN, “Linear Technology developed the LTC6908 silicon
oscillator for the reduction of EMI by intelligently controlling the switching regulator clock. It accomplishes
this goal in three ways. First, by synchronizing multiple
switching regulators out of phase with each other, it
decreases the peak switching currents… The second
benefit of the LTC6908 in controlling EMI is that its
high-accuracy clock frequency allows you to move the
switcher EMI to a frequency that is out of band for the
receiving electronics. The third and most dramatic improvement to EMI is achievable by continuously varying
the switcher’s clock frequency.”
EE Times ACE Award Finalist
EE Times announced the selection of Linear Technology’s
LTC3035 300mA VLDO™ Linear Regulator with Charge
Pump Bias Generator as a finalist for the EE Times ACE
Awards in the Analog ICs category. EE Times stated, “The
LTC3035’s tight output voltage ±2 percent accuracy, low
quiescent and shutdown currents of 100 microamps and
1 microamp, respectively, combined with fast transient
response and small solution footprint with few external
components, make it practical for battery-powered
handheld devices such as Bluetooth-enabled devices,
cellular phones, media players, handheld medical and
industrial instruments.”
AnalogZone Products of the Year
helps you retain your existing users; it also brings you
many more. The LTC3836 will do famously well.”
EE Times China ACE Award Finalists
Four Linear Technology products were selected as finalists for EE Times China ACE Awards:
qLTC2208 16-Bit, 130Msps ADC in the AD/DA
Converter ICs category
qLTC2859/61 20Mbps RS485 Transceivers with
Integrated Switchable Termination in Interface ICs
qLT®6003/04/05 1.6V, 1µA Precision Rail-to-Rail
Input and Output Op Amps in Amplifier ICs
qLTM4600 10A High Efficiency DC/DC µModule™
in the Power category
Linear Unveils Solutions for 3G and
WiMAX Basestations
Two new devices from Linear Technology enhance performance and reduce solution cost of 3G and WiMAX
basestations. The LT5575 High Linearity Direct Conversion I/Q Demodulator was featured in a cover article in
High Frequency magazine and in numerous publications
worldwide. The device provides an operating frequency
range of 800MHz to 2.7GHz, covering all cellular and
3G infrastructure, WiMAX and RFID bands with a
single part. Its capability to convert from RF directly to
baseband at DC or low frequency results in simplified
receiver designs, reduced component count and the
ability to use lower cost, low frequency components.
The LT5557 3.3V High Linearity Downconverting
Active RF Mixer was also announced worldwide. The
device provides the broadest bandwidth of any high
performance active mixer on the market, enabling a
cost-effective, easy to use solution for 3G wireless and
WiMAX basestations, covering frequencies from 400MHz
to 3.8GHz. L
AnalogZone has awarded two Linear Technology products
as Products of the Year: LT5560 RF Active Mixer and
LTC3836 Dual Output 2-Phase Synchronous Controller.
AnalogZone stated, “With the LT5560 Linear has
taken its own product, the LT5525/26, and made
it better in almost every way… Conversion gain is
up, noise is down, quiescent is down… Another
superb product in this Linear family keeping the
company well ahead in the industry…”
AnalogZone commented on the LTC3836, “Linear Technology introduced the LTC3836, a dual
output 2-phase, low input voltage current mode
synchronous step-down switching controller.
Operation from an input voltage from 2.75V to
4.5V makes the device ideal for 3.3V, single cell
Li-Ion, multi-cell Alkaline or NiMH input sources.
… This is a nice next-generation part obviously
The LT5575 High Linearity Direct Conversion I/Q Demodulator converts
resulting from a lot of listening on Linear’s side from RF directly to baseband at DC or low frequency to reduce component
of the fence. Keeping up with the market not only count, and allow the use of lower cost, low frequency components.
Linear Technology Magazine • March 2007
DESIGN FEATURES L
Easy Drive, continued from page types of sensors. Nevertheless, input
sampling currents can overwhelm high
source impedances or low bandwidth,
micropower signal conditioning circuits. Easy Drive solves this problem
by balancing the input currents, thus
simplifying or eliminating the need for
signal conditioning circuits.
A common application for a deltasigma ADC is thermistor measurement.
Figure 1 shows two examples of thermistor digitization benefiting from Easy
Drive technology. The first circuit (applied to input channels CH0 and CH1)
uses two equal reference resistors that
set the input common mode voltage
equal to the reference common mode
voltage and balance the differential
input source resistance. If reference
resistors R1 and R4 are exactly equal,
the input current is zero and no er-
rors result. If these resistors have a
1% tolerance, the maximum error in
measured resistance is 1.6Ω due to
a shift in common mode voltage, far
less than the 1% error of the reference
resistors themselves. No amplifier is
Input sampling currents
can overwhelm high
source impedances or lowbandwidth, micropower
signal conditioning circuits.
Easy Drive solves this
problem by balancing
the input currents, thus
simplifying or eliminating
the need for signal
conditioning circuits.
required, making this an ideal solution
in micropower applications.
Easy Drive also enables very low
power, low bandwidth amplifiers to
drive the input of the LTC2492. As
shown in Figure 1, CH2 is driven by
an LT1494. The LT1494 has excellent
DC specs for an amplifier with 1.5µA
supply current, offering maximum
offset voltage of 150µV and an open
loop gain of 100,000. However, its
2kHz bandwidth makes it unsuitable
for driving conventional delta-sigma
ADCs. Adding a 1kΩ, 0.1µF filter solves
this problem by providing a charge
reservoir that supplies the LTC2492
instantaneous current, while the 1kΩ
resistor isolates the capacitive load
from the LT1494. The input sampling
current of conventional delta-sigma
ADCs leads to DC errors as a result
Table 1. Complete Easy Drive delta-sigma family
Part Number
# Inputs
Interface
Bits
Temp
PGA
2×
Package
LTC2480
1
SPI
16
L
L
L
3mm × 3mm DFN
LTC2481
1
I2C
16
L
L
L
3mm × 3mm DFN
LTC2482
1
SPI
16
3mm × 3mm DFN
LTC2483
1
I2C
16
3mm × 3mm DFN
LTC2484
1
SPI
24
L
L
3mm × 3mm DFN
LTC2485
1
I2C
24
L
L
3mm × 3mm DFN
LTC2486
2/4
SPI
16
L
L
L
3mm × 4mm 14DFN
LTC2487
2/4
I2C
16
L
L
L
3mm × 4mm 14DFN
LTC2488
2/4
SPI
16
3mm × 4mm 14DFN
LTC2489
2/4
I2C
16
3mm × 4mm 14DFN
LTC2492
2/4
SPI
24
L
L
3mm × 4mm 14DFN
LTC2493
2/4
I2C
24
L
L
3mm × 4mm 14DFN
LTC2494
8/16
SPI
16
L
L
L
5mm × 7mm QFN
LTC2495
8/16
I2C
16
L
L
L
5mm × 7mm QFN
LTC2496
8/16
SPI
16
5mm × 7mm QFN
LTC2497
8/16
I2C
16
5mm × 7mm QFN
LTC2498
8/16
SPI
24
L
L
5mm × 7mm QFN
LTC2499
8/16
I2C
24
L
L
5mm × 7mm QFN
Linear Technology Magazine • March 2007
L DESIGN FEATURES
LTC2498
MUXOUTP
INPUT
MUX
MUXOUTN
ANALOG 17
INPUTS
SDI
∆Σ ADC
WITH
EASY DRIVE
INPUTS
SCK
SDO
CS
2
–
1
1/2 LTC6078
3
+
6
–
5
1k
0.1µF
7
1/2 LTC6078
1k
0.1µF
+
Figure 2. External buffers provide high impedance inputs and amplifier offsets are automatically cancelled.
CS
1
2
3
4
5
1
0
EN
SGL
ODD
EOC
“0”
SIG
MSB
6
7
8
9
10
A2
A1
A0
EN2
11
12
13
14
32
SCK
(EXTERNAL)
SDI
DON'T CARE
SDO
Hi-Z
IM
FA
FB
SPD
DON'T CARE
Hi-Z
BIT 31 BIT 30 BIT 29 BIT 28 BIT 27 BIT 26 BIT 25 BIT 24 BIT 23 BIT 22 BIT 21 BIT 20 BIT 19 BIT 18 BIT 17
CONVERSION
BIT 0
CONVERSION
DATA INPUT/OUTPUT
SLEEP
Figure 3. SPI interface, configuration and data output timing
S
R/W
7-BIT ADDRESS
CONVERSION
ACK
DATA
SLEEP
Sr
DATA TRANSFERRING
P
DATA INPUT/OUTPUT
CONVERSION
Figure 4. I2C conversion sequence
1
SCL
2
…
7
8
9
1
2
3
4
5
6
7
8
9
1
2
3
4
5
FA
FB
SPD
6
7
8
9
SDA
7-BIT ADDRESS
START BY
MASTER
1
W
ACK BY
LTC2499
SLEEP
0
EN
SGL ODD
A2
A1
A0
EN2
ACK
LTC2499
IM
(OPTIONAL 2ND BYTE)
ACK
LTC2499
DATA INPUT
Figure 5. I2C configuration and data output timing
Linear Technology Magazine • March 2007
DESIGN FEATURES L
of incomplete settling in the external
RC network. Linear Technology’s Easy
Drive technology cancels the differential input current. By balancing the
negative input (CH3) with a 1kΩ-0.1µF
RC network, errors due to the common
mode input current are cancelled.
5V
3.35kΩ
CH15
MUXOUT/
ADCIN
0.1µF
VCC
10µF
100Ω
CH14
Complete Easy Drive
Delta-Sigma Family
•
•
•
CH3
100Ω
CH2
100Ω
REF+
IN+
16-CHANNEL
MUX
16-BIT ΔΣ ADC
WITH EASY-DRIVE
–
IN
CH1
REF–
SDI
SCK
SDO
CS
4-WIRE
SPI INTERFACE
100Ω
CH0
100Ω
COM
TEMPERATURE
SENSOR
MUXOUT/
ADCIN
FO
OSC
Figure 6. Use this setup to quickly sort out which
SDI word is associated with each input channel.
5V
4.15kΩ
CH15
MUXOUT/
ADCIN
0.1µF
VCC
10µF
100Ω
CH14
•
•
•
CH3
100Ω
CH2
16-CHANNEL
MUX
REF+
IN+
16-BIT ΔΣ ADC
WITH EASY-DRIVE
IN–
CH1
REF–
SDI
SCK
SDO
CS
CH0
100Ω
COM
TEMPERATURE
SENSOR
MUXOUT/
ADCIN
FO
OSC
Figure 7. Use this setup to quickly sort out which SDI word
is associated with each differential input channel.
Linear Technology Magazine • March 2007
4-WIRE
SPI INTERFACE
Easy Drive ADCs are at home in a
vast array of applications. The 24-bit,
16-channel LTC2498 with integrated
temperature sensor is ideal for high
performance data acquisition systems.
It can directly digitize thermocouples
without any signal conditioning and
provide cold junction compensation.
It can also directly measure low level
strain gage outputs. At the same time
it can handle industrial sensor voltages
with the addition of a simple resistive
divider—no active circuitry required.
The 16-bit, 16-channel devices are
suitable for measuring voltages and
currents on large circuit boards that
have several high current supplies.
Up to 16 ground referred measurements can be taken if the COM pin is
grounded to a common point for all
supplies. Using the inputs differentially (up to 8 differential input channels)
allows high side sensing of current
shunts as long as the shunt common
mode voltage is less than or equal to
the ADCs’ supply voltage. Differential
measurements also allow voltages to
be sensed remotely, eliminating errors
due to large ground currents.
Another big advantage of using a
delta-sigma ADC for power supply
measurements is the very strong rejection of noise and switching transients.
The ADC’s internal SINC4 filter, in
conjunction with a simple 1-pole filter
at the ADC input, is adequate to attenuate switching power supply noise
below the ADC noise floor. What is left
is an extremely accurate measurement
of the DC value of the power supply
voltage or current.
The single channel LTC2482 is
ideal for cost sensitive applications
such as portable medical devices and
consumer products. Don’t be fooled by
its relatively low cost—it is essentially
a perfect 16-bit ADC that shares the
L DESIGN FEATURES
same 600nV input noise floor as the
24-bit parts. This means it would also
be ideal for a 4½ digit handheld or
bench-top voltmeter with a ±1 count
linearity specification.
Automatic Offset Calibration
of External Buffers/Amplifiers
In addition to the Easy Drive input current cancellation, the 16-channel Easy
Drive ADCs allow an external amplifier
to be inserted between the multiplexer
output and the ADC input (see Figure 2). This is useful in applications
where balanced source impedances
are not possible or where the source
impedance is very high. One pair of
external buffers/amplifers can be
shared between all 17 analog inputs.
The LTC2498 performs an internal
offset calibration every conversion
cycle in order to remove the offset and
drift of the ADC. This calibration is performed through a combination of front
end switching and digital processing.
Since the external amplifier is placed
between the multiplexer and the ADC,
it is inside this correction loop. This
results in automatic removal of the
offset and offset drift of the external
amplifer.
The LTC6078 is an excellent amplifier for this function. It operates with
supply voltages as low as 2.7V and its
voltage noise level is a low 18nV⁄√Hz. The
LTC2498’s Easy Drive inputs allow an
RC network to be added directly to the
output of the LTC6078. The capacitor
reduces the magnitude of the current
spikes seen at the input to the ADC
and the resistor isolates the capacitor
load from the op amp output enabling
stable operation.
Software Interface
The simplicity of the analog interfacing
requirements of Linear Technology’s
Easy Drive ADCs is matched by the
simplicity of their serial interface. The
No Latency architecture eliminates
the annoyance of having to discard
readings after switching channels on
the multichannel devices. The start
of conversion is directly controlled
by the serial interface, so external
signal conditioning or sensor excitation can be switched in at the proper
Sample Code Drivers for
Basic Communications with the LTC2448 and LTC2449
// Make sure this
// context of the
struct fourbytes
{
int8 te0;
int8 te1;
int8 te2;
int8 te3;
}
structure applies in the
following functions.
// Define structure of four consecutive bytes
// To allow byte access to a 32-bit int or float.
//
// The make32() function in some compilers will
// also work, but a union of 4 bytes and a 32-bit int
// is more portable because it is standard C.
// Some defines for I2C communication
#define READ
0x01
// bitwise OR with address for read or write
#define WRITE
0x00
/***************************************************************
Blocking version of read_LTC2498() function. When called,
it will wait for the LTC2498 to finish converting and then
read data. The longest this function should ever take to return
is the maximum conversion time of the LTC2498. It is a good
idea to use a watchdog when your program has blocking functions
like this.
The spi_readwrite() function simultaneously reads and writes
an 8-bit byte from the SPI port. Most compilers that support
processors that have a hardware SPI port have a similar function.
As a starting point, configure the SPI port for data transitions
on the falling clock edge, valid on the rising edge.
Arguments:
channel - channel to program for the next conversion
config - configuration bits for next conversion
Returns:
32 bit word from the LTC2498 when the conversion finishes
****************************************************************/
Int32 read_LTC2498(char channel, char config);
{
// Create a union of the four-byte structure and a 32-bit
// signed integer.
union// adc_code.bits32
all 32 bits
{
// adc_code.by.te0
byte 0
signed int32 bits32;
// adc_code.by.te1
byte 1
struct fourbytes by;
// adc_code.by.te2
byte 2
} adc_code;
// adc_code.by.te3
byte 3
output_low(CS_);
// Lower Chip Select, enabling serial port
while(input(SDO)); // Wait for SDO to go low. You can also put a
// timeout here in case something bad happens
adc_code.by.te3 = spi_readwrite(channel);
adc_code.by.te2 = spi_readwrite(config);
adc_code.by.te1 = spi_readwrite(0);
adc_code.by.te0 = spi_readwrite(0);
return adc_code.bits32;
} // end of read_LTC2498()
/***************************************************************
Non-blocking version of read_LTC2498() function. When called,
it will see if the LTC2498 has finished converting. If so,
data will be read and returned. If not, zero will be returned.
Since all zeros is NOT a valid code from the LTC2498, the calling
program can ignore the return result if zero.
Arguments:
channel - channel to program for the next conversion
config - configuration bits for next conversion
Returns: 32 bit word from the LTC2498 if conversion is done,
zero if not.
****************************************************************/
Int32 read_LTC2498(char channel, char config);
{
// Create a union of the four byte structure and a 32 bit
// signed integer.
union// adc_code.bits32
all 32 bits
{
// adc_code.by.te0
byte 0
signed int32 bits32;
// adc_code.by.te1
byte 1
struct fourbytes by;
// adc_code.by.te2
byte 2
} adc_code;
// adc_code.by.te3
byte 3
Linear Technology Magazine • March 2007
DESIGN FEATURES L
output_low(CS_);
// Lower Chip Select, enabling serial port
while(input(SDO)); // Wait for SDO to go low. You can also put a
// timeout here in case something bad happens
adc_code.by.te3 = spi_readwrite(channel);
adc_code.by.te2 = spi_readwrite(config);
adc_code.by.te1 = spi_readwrite(0);
adc_code.by.te0 = spi_readwrite(0);
return adc_code.bits32;
} // end of read_LTC2498()
/***************************************************************
Non-blocking read_LTC2499() function.
the i2c_xxxx() functions do the following:
void i2c_start(void):
generate an i2c start or repeat start condition
void i2c_stop(void):
generate an i2c stop condition
char i2c_read(boolean): return 8 bit i2c data while generating
an ack or nack
boolean i2c_write():
send 8 bit i2c data and return ack or
nack from slave device
These functions are very compiler specific, and can use either a
hardware i2c port or software emulation of an i2c port. This example
uses software emulation.
A good starting point when porting to other processors is to write
your own i2c functions. Note that each processor has its own way of
configuring the i2c port, and different compilers may or may not have
built-in functions for the i2c port.
Arguments:
addr - LTC2499 I2C address
channel - channel to program for the next conversion
config - configuration bits for next conversion
Returns:
32 bit word from the LTC2499 if conversion is done,
zero if not.
*******************************************************************/
signed int32 read_LTC2499(char addr, char channel, char config)
{
union
// adc_code.bits32
all 32 bits
{
// adc_code.by.te0
byte 0
signed int32 bits32;
// adc_code.by.te1
byte 1
struct fourbytes by;
// adc_code.by.te2
byte 2
} adc_code;
// adc_code.by.te3
byte 3
// Start communication with LTC2481:
i2c_start();
if(i2c_write(addr | WRITE))// If no acknowledge, return zero
{
i2c_stop();
return 0;
}
i2c_write(channel);
i2c_write(config);
i2c_start();
i2c_write(addr | READ);
adc_code.by.te3 = i2c_read();
adc_code.by.te2 = i2c_read();
adc_code.by.te1 = i2c_read();
adc_code.by.te0 = i2c_read();
i2c_stop();
return adc_code.bits32;
} // End of read_LTC2499()
/***************************************************************
Note: you can create a non-blocking version of this function
by repeatedly attempting to write the LTC2499 address, sending
a stop condition if there is no acknwoledge to keep the bus free.
When the LTC2499 acknowledges, read the data and return.
****************************************************************/
Linear Technology Magazine • March 2007
time. The implicit offset and gain
calibration that takes place in every
conversion eliminates the need for
complicated internal register set or
calibration cycles. Communication for
both the SPI and I2C interface parts is
a simple read/write operation where
data from one conversion is read as
the configuration for the next channel
is programmed into the ADC.
Figure 3 shows the data input/output operation for the LTC2498. This is
the SPI-interface ADC with the most
channels and features—other SPI
parts have similar interfaces.
Figure 4 shows the data input/
output operation for the LTC2499.
Likewise, this is the most feature-laden
I2C part—other I2C parts have similar
interfaces. Figure 5 shows the details
of writing the channel and configuration to the input registers.
To help the software/firmware designer get started, see the sidebar for C
code drivers for basic communications
with the LTC2448 and LTC2449. These
functions can be easily ported to any
C compiler and can easily be adapted
to the other Easy Drive ADCs.
Try this Trick!
While the Easy Drive serial interface is
easy to program—just read the data for
sample N while programming the channel for sample N+1—it can still be tricky
to figure out what was just read when
looking at a microcontroller’s registers
through a debugger. Here is a hardware
trick that can significantly reduce code
design headaches. Figure 6 shows a
simple circuit that applies a known
voltage to each single-ended input.
With the values shown, CH0 has a
voltage of 101mV, CH1 202mV, and
so on up to CH15, which produces
1.616V. Figure 7 shows the equivalent
circuit for differential inputs. Use this
setup to quickly sort out which SDI
word is associated with each input
channel. L
Want to know more? Visit:
www.linear.com
or call
1-800-4-LINEAR
L DESIGN FEATURES
Current Mode Flyback DC/DC
Controller Provides Tremendous
by Arthur Kelley
Design Flexibility
Introduction
By its nature, a flyback DC/DC converter is one of the most versatile power
converter topologies. Because it uses
a transformer, it can step up or step
down voltages and provide DC isolation
if needed. Applications include power
supplies for networking equipment,
Power-over-Ethernet (PoE), automotive, consumer and general system
house keeping. The LTC3805 has been
designed to enhance the flexibility of
the basic flyback converter, making
it possible to optimize a single design
for diverse applications. The converter
input and output voltage is limited only
by the rating of external components
such as the power MOSFET and the
transformer. The LTC3805 can be
programmed for frequency, slope
compensation, soft-start, input voltage RUN/STOP thresholds (including
programmable hysteresis), synchronization to an external frequency source,
and overcurrent protection to protect
the converter from faults.
36V–72V to 3.3V at 3A
Non-Isolated Flyback
Figure 1 shows the LTC3805 in a
non-isolated flyback converter with
an input voltage range of 36V to 72V
and an output voltage of 3.3V at 3A.
VIN
36V TO
72V
221k
2.2µF
×2
MMBTA42
221k
221k
PDZ6.8B
6.8V
BAS516
1µF
8.66k
RUN
UPS840
VCC
LTC3805
SSFLT
SYNC
FS
0.1µF
470pF
118k
FB
GND
100µF
6.3V
×3
FDC2512
GATE
42.2k
ITH
20k
VOUT
3.3V
AT 3A
OC
ISENSE
1.33k
3.01k
68mΩ
13.7k
Figure 1. Non-isolated 36V to 72V to 3.3V 3A flyback converter
The remainder of this section details
the design decisions made in creating
this converter and describes methods
for altering the design for various
applications. An isolated version of
the converter is described in the next
section.
VCC Power and Start-Up
In this design, start-up VCC power for
the LTC3805 is provided by an external
pre-regulator using an NPN transistor, a zener diode and two resistors.
Once the converter begins operation,
a winding on the transformer provides a bias supply which turns off
the NPN transistor to save power and
increase efficiency. Alternately, since
the LTC3805 has an ultralow shutdown current of 40µA, a simple trickle
charger could be used to eliminate the
NPN pre-regulator. The LTC3805 has
a VCC rising threshold of 8.5V and a
falling threshold of 4V so there is plenty
of hysteresis to implement a trickle
charger. In either case, note that VCC
is not connected to VIN so that almost
any input supply above 8.5V can be
accommodated by proper selection of
external components and that, once
started, the LTC3805 can run with
input supplies down to 4V.
Programming VOUT
The FB pin monitors the output voltage by comparing it—via a resistive
divider—to the 0.8V internal reference
of the LTC3805. Since the FB pin is
not connected directly to the output,
the LTC3805 can accommodate any
output voltage down to 0.8V simply by
adjustment of the resistor values.
Figure 2. Isolated 36V to 72V to 3.3V 3A flyback converter
Selecting Frequency
The 200kHz operating frequency is
programmed by the 118kΩ resistor
on the FS pin. By changing this resistor, the operating frequency can
Linear Technology Magazine • March 2007
DESIGN FEATURES L
Programming the VIN Thresholds
The rising threshold on VIN, which is
independent of the thresholds on VCC,
is set by the 221kΩ and 8.86kΩ resistors connected to the RUN pin. The
rising threshold on the RUN pin is 1.2V
while its absolute maximum voltage is
18V—a 15:1 ratio. Therefore the RUN
pin accommodates designs with a wide
range of input voltages and still has a
high enough voltage rating to survive
a transient overvoltage on VIN. Once
started, the LTC3805 sources a 5µA
current from the RUN pin. Multiplied
by the 221kΩ resistor, this current
sets the hysteresis on VIN to 1.1V. A
different hysteresis, with the same
rising threshold, can be selected by
changing the values of the 221kΩ and
8.86kΩ resistors while keeping their
ratio constant.
Setting the Soft-Start
The rate of change of VOUT at start-up
is programmed by the capacitor on the
SSFLT pin—0.1µF in this case. A major
consideration in the selection of the
SSFLT capacitor is the filter capacitor used to bypass VOUT. Generally, a
larger output filter capacitor requires
a slower soft-start to limit the inrush
current caused by the charging filter
capacitor. Conversely, if the converter
has a small output filter capacitor, the
SSFLT capacitor can be omitted and
the LTC3805 internal soft-start ramps
up the output voltage in 1.8ms.
Programming Slope Compensation
and Overcurrent Operation
The 68mΩ resistor monitors the current through the main NMOS switch
and implements both current mode
control and overcurrent protection via
the ISENSE and OC pins, respectively.
The ISENSE pin monitors the current
through the main switch and turns it
off when the current exceeds a level
Linear Technology Magazine • March 2007
set by the voltage on the ITH pin. The
3.01kΩ resistor sets the amount of
slope compensation using a ramp
of current that is sourced by the
LTC3805.
The overcurrent protection level is
set by the 1.33kΩ resistor in series
with the OC pin using a constant 10µA
current sourced by the OC pin. Several
behaviors can be programmed using
this resistor. This particular design
is set to regulate output voltage up
to 3A and then overcurrent trip just
above that. An alternate strategy,
using a smaller resistor, would be to
allow the output voltage to sag as the
converter goes into current limiting
and then trip on overcurrent only
to prevent damage. In either case,
once there is an overcurrent trip the
LTC3805 shuts down, waits for a time
out interval determined by discharging the capacitor on the SSFLT pin
and then restarts if the overcurrent
fault has been removed. If the fault
is not removed, the LTC3805 enters
a hiccup mode in which it periodically
tries to restart with the period determined by the capacitor on the SSFLT
pin. Thusly, the LTC3805 completely
protects a flyback converter from short
circuits on the output.
Frequency Synchronization
to an External Source
Although shown grounded in Figure 1,
the SYNC pin is used to synchronize
the frequency of operation of the
LTC3805 to an external source. The
synchronization signal can be applied
and removed without any particular
sequencing requirement—it can be
100
90
80
EFFICIENCY (%)
be set anywhere between 70kHz and
700kHz. High power designs tend to
use lower frequencies while low power
designs tend to use higher frequencies. The frequency programmability
of the LTC3805 allows selection of
the optimum frequency for any given
design.
70
60
50
40
VIN
30
36V
48V
60V
72V
20
10
0
0
1
2
IOUT (A)
3
4
Figure 3. Efficiency for isolated and nonisolated 36V–72V to 3.3V 3A flyback converter
present before the LTC3805 begins
operation or it can be applied after
the LTC3805 has begun operation
using the frequency programmed by
the resistor on the FS pin. When the
synchronization signal is applied,
the LTC3805 locks on to the signal
within two cycles of operation. When
the synchronization signal is removed,
the LTC3805 takes no more than two
cycles to jump back to the frequency
programmed by the FS pin.
Isolated Converter Design
The basic design shown in Figure 1
can be modified to provide DC isolation between the input and output by
the addition of a reference, such as
the LT4430, on the secondary side of
the transformer and an optoisolator
to provide feedback from the isolated
secondary to the LTC3805. Figure 2
shows a photo of the DC1045 demonstration circuit, which is an isolated
converter with the same basic design
and performance as the converter
in Figure 1, and is representative of
the size of both the isolated and nonisolated designs. Figure 3 shows the
efficiency of the isolated converter
and is also representative of the nonisolated converter.
Modifications for Different
Input or Output Voltages
The two applications described above
represent typical non-isolated and
isolated 10W flyback converters. It is
fairly easy to take this basic design
and change the input or output voltage
by scaling the external components in
direct proportion to the change in voltage. These changes are transparent to
the LTC3805 and can be accomplished
with a circuit no more complex than
that of Figure 1 and a board no bigger
than that shown in Figure 2.
A decrease of the input voltage, and
increase of the input current, mainly
involves selecting a NMOS power
switch with a lower voltage and higher
current rating and selecting a transformer primary winding with a reduced
number of turns and a proportionally
larger wire size. For the input filter
capacitor, the voltage rating can be
continued on page 17
L DESIGN FEATURES
Isolated Forward Controllers Offer
Buck Simplicity and Performance
by Charles Hawkes and Arthur Kelley
Buck converter designers have long
benefited from the simplicity, high
efficiency and fast transient response
made possible by the latest buck
controller ICs, which feature synchronous rectification and PolyPhase®
operation. Unfortunately, these
same features have been difficult or
impossible to implement in the buck
converter’s close relative, the forward
converter. That is, until now. The
LTC3706/26 secondary-side synchronous controller and its companion
smart gate driver, the LTC3705/25,
make it possible to create an isolated
forward converter with the simplicity
and performance of the familiar buck
converter.
rectifier timing and optoisolator feedback to control the output (secondary).
This architecture is commonly known
as primary-side control. By contrast,
secondary-side control places the
controller IC on the secondary side,
and uses a gate-drive transformer
to directly control the primary-side
MOSFETs. This approach eliminates
the need for an optoisolator and
puts the controller where it is really
needed: with the load. This results in
a significantly faster response, taming
large-signal overshoot and reducing
output capacitance requirements.
In addition, secondary-side control
simplifies the design of the loop compensation to that of a simple buck
converter.
With the apparent advantages of
secondary-side control, why is it not
used in more isolated applications?
This is primarily because of the need
for a separate bias supply to power
The Benefits of SecondarySide Control Made Accessible
Many isolated supplies place the
controller IC on the input (primary)
side and rely on indirect synchronous
VIN+
1µF
100V
×2
1µF
100V
VGATE
Si7450DP
92
36V
48V
72V
90
88
86
84
10
5
20
15
LOAD CURRENT (A)
365k
VCC, PRI
2•
4
• 10
3•
5
•7
2.2nF
200V
L2
0.85µH
11
100µF
6.3V
×2
1.2Ω
1/4W
9
HAT2165H
×2
0.0012Ω
2W
1nF
Q2
FCX491A
UVLO
VSLMT PGND
33nF
8
FS/IN–
3,4
T2
•
LTC3725
SSFLT
1µF
1
FB/IN+
VCC
220µF
6.3V
VOUT–
VCC, SEC
2.2µF
2.74k
1µF
0.1µF
IS
+
10µF
5.1k
NDRV GATE
30
VOUT+
3.3V
30A
D1
CMPSH1-4
HAT2165H
×2
25
Figure 2. Efficiency of the converter
shown in Figure 1
0.03Ω
1W
100k
Q1
FDC2512
15k
94
T1
23.4mm × 20.1mm × 9.4mm
PLANAR
L1
1µH
36V
TO
72V
VIN–
up the controller on the secondary
side, since there is initially no voltage
present there. With the introduction
of the LTC3706/26 and LTC3705/25,
however, this barrier has now been
completely eliminated. All of the complex issues associated with start-up
and fault monitoring in a secondaryside control forward converter have
EFFICIENCY (%)
Introduction
•
FG SW IS–
IS+
SG
NDRV MODE VCC
PT+
FB
LTC3706
5,6
PT–
FS
GND PGND REGSD PHASE RUN/SS
GND
VIN
SLP
ITH
162k
L1: VISHAY IHLP2525CZER0M01 T1: PULSE PA0815 (6:6:2:1)
L2: PULSE PA1294.910
T2: PULSE PA0297 (2:1:1)
100k
33nF
470pF
3.3k
604Ω
47nF
Figure 1. Complete 100W single-switch high efficiency, low cost, minimum part count, isolated
telecom converter. Other output voltages and power levels require only simple component changes.
10
Linear Technology Magazine • March 2007
DESIGN FEATURES L
been seamlessly integrated into these
powerful new products. Moreover, a
proprietary scheme is used to multiplex gate drive signals and DC bias
power across the isolation barrier
through a single, tiny pulse transformer. This eliminates the primary-side
bias winding that is otherwise needed.
The result is an isolated supply that
has been architected from the ground
up to achieve unprecedented simplicity
and performance. Figure 1 illustrates
how this remarkable new architecture
is used to make a complete 100W forward converter with minimal design
effort and complexity.
Family of Products Supports
Single or Dual Switch
Topologies
Table 1 summarizes how the
LTC3706/26 and LTC3705/25 products can be combined to cover a broad
range of applications. The LTC3706
is a full-featured product available
in a 24-lead SSOP package. For high
precision applications, the LTC3706
includes a 1% accuracy output voltage,
a remote-sense differential amplifier
and a power good output voltage monitor. The high voltage linear regulator
controller simplifies the design of the
bias supply, and PLL frequency synchronization with selectable phase
angle enables PolyPhase operation
with up to twelve phases. In addition,
the flexible current-sense inputs allow
the LTC3705 greatly facilitates the use
of the simple and robust dual switch
forward converter topology. Figure 3
shows a typical dual-switch converter
application using the LTC3705 and
the LTC3706.
Table 2 highlights some of the relative merits of using either single or dual
switch forward converter topologies.
In general, for applications that have
a limited input voltage variation, or
where a robust and simple design is
a priority, the dual-switch forward
converter may be preferred. For a wide
input voltage application (greater than
2:1), or whenever a lower cost or size
justifies the complication of the transformer reset design, a single-switch
forward should be used.
Table 1. LTC3705/06/25/26 combinations
LTC3705
LTC3725
LTC3706
LTC3726
Dual-Switch,
PolyPhase
Dual-Switch,
Single Phase
Single-Switch, Single-Switch,
PolyPhase
Single Phase
for the use of either resistive or current transformer sensing techniques.
Protection features include an output
overvoltage crowbar as well as currentlimiting and over-current protection.
The 16-lead LTC3726 does not include
the remote voltage sensing or linear
regulator features, so it is more suitable for a single phase application.
Both the LTC3706 and the LTC3726
have a selectable maximum duty cycle
limit of either 75% or 50% to support a
single or dual-switch forward converter
application, respectively.
The LTC3725 primary driver is
intended for use in single-switch
forward converter. The LTC3725 includes a start-up linear regulator and
an integrated bridge rectifier for bias
generation. Protection features include
volt-second limit, over-current protection and a fault monitoring system that
detects a loss of encoded gate-drive
signal from the signal transformer.
The LTC3705 is a dual-switch forward
driver, and includes an 80V (100V
transient) high side gate driver. The
integration of this high side driver into
VIN+
Bringing the Power of
PolyPhase to Isolated Supplies
The LTC3706/26 defies typical forward
converter limits by allowing simple
implementation of a PolyPhase current
share design. PolyPhase operation
allows two or more phase-interleaved
power stages to accurately share the
load. The advantages of PolyPhase
current sharing are numerous, including much improved efficiency, faster
transient response and reduced input
and output ripple.
The LTC3706/26 supports standard output voltages such as 5V, 12V,
28V and 52V as well as low voltages
down to 0.6V. Figure 4 shows how
T1
•
Si7852DP
1µF
100V
x3
•
1.2Ω
MURS120
Si7852DP
Si7336ADP
×2
Si7336ADP
VOUT+
L1
1.2µH
330µF
6.3V
×3
CMPSH1-4
MURS120
2mΩ
2W
30mΩ
1W
VIN–
10µF
25V
VOUT–
CZT3019
100k
FQT7N10
365k
1%
L1: COILCRAFT SER2010-122
T1: PULSE PA0807
T2: PULSE PA0297
BAS21 0.22µF
NDRV
BOOST TG
TS BG IS
UVLO
15k
1%
2.2µF
25V
33nF
VCC
SS/FLT
LTC3705
GND PGND VSLMT
FB/IN+
1µF
T2
•
•
FS/IN–
162k
33nF
2.2µF
16V
IS–
IS+
PT+
FG
SW SG
VIN
NDRV
102k
1%
VCC
FS/SYNC
LTC3706
FB
ITH
PT–
RUN/SS GND PGND PHASE SLP MODE REGSD
680pF
20k
22.6k
1%
Figure 3. Isolated forward converter for 36V–72V input to 3.3V/20A out
Linear Technology Magazine • March 2007
11
L DESIGN FEATURES
easy it is to parallel two 1.2V supplies
to achieve a 100A supply. Figure 5
shows excellent output inductor current tracking during a 0A to 100A load
current step and the smooth handoff
during start-up to secondary-side control at approximately VOUT = 0.25V.
Table 2. Single and dual switch forward converter relative merits
+
LTC3705/LTC3706 VOUT
36V-72VIN TO 1.2VOUT
50A SUPPLY
VIN–
VOUT–
SSP VBIAS SYNC ITH
VIN+
Requires Design
Transformer Reset Circuit
to Prevent Saturation
+
75% Max Duty
(>2:1)
VIN+
VIN–
Simple Design
Wide Input Supply Range
The circuit of Figure 1 shows a
complete 100W, one-switch forward
converter. In this example, the
LTC3706 controller is used on the
secondary and the LTC3725 driver
with self-starting capability is used
on the primary. This design features
off-the-shelf magnetics and high efficiency (see Figure 2). The start-up
behavior of this supply is illustrated
in Figure 6. When input voltage is
first applied, the LTC3725 uses Q1
to generate a bias voltage VCC,PRI, and
begins a controlled soft-start of the
output voltage. As the output voltage
begins to rise, the LTC3706 secondary controller is quickly powered up
by using T1, D1 and Q2 to generate
VCC,SEC. As shown in Figure 6, the
VCC,SEC voltage rises very quickly as
compared with the output voltage
VOUT of the converter. The LTC3706
VIN+
Single-Switch
–
Anatomy of a Start-Up:
A Simple Isolated 3.3V,
30A Forward Converter
SSP VBIAS SYNC ITH
Requirement
SSS
+
High Efficiency
–
+
Two FETs
+
One FET and Better
Transformer Utilization
then assumes control of the output
voltage by sending encoded PWM gate
pulses to the LTC3725 primary driver
via signal transformer T2. As soon as
the LTC3725 begins decoding these
PWM gate pulses, it shuts down the
linear regulator by tying NDRV to VCC
and begins extracting bias power for
VCC,PRI from the signal transformer T2.
This complete transition from primary
to secondary control occurs seamlessly
at a fraction of the output voltage. From
PRIMARYSIDE MODE
+
Limited to VIN
–
One FET
Small Size
–
50% Max Duty
Good
Can be 2 × VIN or Greater
Low Cost
+
Reset Circuit not
Required—Can’t Saturate
+
Good
Low Switch Voltage Stress
Dual-Switch
–
Two FETs and 50%
Transformer Utilization
that point on, operation and design
simplifies to that of a simple buck
converter. Even the design and optimization of the feedback loop makes use
of the familiar and proven OPTI-LOOP®
compensation techniques.
A 10V–30V Input, 15V Output
at 5A Forward Converter
Figure 7 highlights the flexibility of
the LTC3706 and LTC3725 by illustrating a 12V/24V input application.
SECONDARY-SIDE MODE
VOUT+
1.2V/100A
VOUT–
SSS
VIN
+
LTC3705/LTC3706 VOUT
36V-72VIN TO 1.2VOUT
–
50A SUPPLY
VIN
VOUT–
VCCPRI SUPPLIED BY Q1
VCCPRI SUPPLIED BY
TRANSFORMER T2
VCCPRI
VGATE CONTROLLED BY LTC3706
Figure 4. Paralleling supplies for higher power operation
VGATE
VGATE CONTROLLED BY LTC3725
ILOUT1
ILOUT2
10A/DIV
10µs/DIV
VOUT
VCC,SEC
VOUT
0.5V/DIV
2ms/DIV
VPT+,VPT –
Figure 5. 1.2V, 100A load current step
(top trace) and start-up (bottom trace)
12
Figure 6. Anatomy of a start-up
Linear Technology Magazine • March 2007
DESIGN FEATURES L
L1
13µH
VOUT+
VIN+
10Ω
C3
0.5W
2.2nF
220pF
100V
200V
T1
C1
220µF
50V
×2
VIN
C2
10µF
35V
×5
Si7852DP
2.2nF
250VAC
Q4
100Ω
68pF
C3
10µF
25V
×2
R3
33k
0.5W
Q6
150Ω
IRF6648
×2
5mΩ
2W
Q5
Q7
1:3
–
D1
ES1C
174Ω
C4
180µF
16V
C6
10µF
200V
6mΩ
1W
VOUT–
470pF
383k
Q2
301Ω
100Ω
Q3
R1
10k
1nF
5.1kΩ
Q1
NDRV GATE
VCC
0.1µF
•
2:1
68pF
162k
75k
•
470pF
FS/IN–
SS/FLT GND PGND VSLMT
68pF
68nF
L1: PULSE PA1961.133
T1: PULSE PA0810
T2: PULSE PA0297
SW
SG
VIN NDRV
25.5k
1%
D2 BAT54
LTC3706
ITH
C5
0.1µF
GND PGND PHASE SLP MODE REGSD
33pF
1.07k
1%
330pF
33nF
C1: NIPPON CHIMICON EMZA500ADA221MUA0G
C2: TAIYO YUDEN GMK325BJ106MN
C3: TAIYO YUDEN TMK325BJ106MM
C4: SANYO OSCON 16SVP180MX
VCC
R2
8.66k
FB
1µF
PT–
RUN/SS
10µF
16V
FS/SYNC
PT+
T2
100Ω
LTC3725
1µF
25V
FG
IS–
IS+
IS
FB/IN+
UVLO
FCX1051A
100Ω
1nF
Q1: FMMT38C
Q2: MMBFJ201
Q3: ZVN3320F
Q4: FDMS2572 ×2
100k
43.2k
Q5: FMMT 618
Q6: FMMT 718
Q7: MMBT 2907A
Figure 7. Isolated forward converter for 10V–30V input to 15V/5A out
Linear Technology Magazine • March 2007
gate of Q2) during normal operation
when VCC = VNDRV = 12V and VIN is
less than 12V.
On the secondary side, the output
voltage is used directly as a source
of bias voltage for the LTC3706. This
is possible for output voltages of 9V
or greater. Q3 is used to limit the
peak voltage seen by the SW pin on
the LTC3706, while still allowing the
detection circuits in the LTC3706 to
function normally. Capacitor C3 is
used to establish the resonant reset
of the main transformer T1 during the
off-time of the primary-side switches.
In order to reduce the inrush current
during start-up, D2, R2 and C5 are
continued on page 39
95
VIN = 12V
90
VOUT
200mV/DIV
EFFICIENCY (%)
In this circuit, the main transformer
T1 is used to step up the voltage so
that the output can be either higher
or lower than the input. This circuit
is an excellent alternative to a flyback
converter where higher efficiency or
lower noise is a priority.
The UVLO on the LTC3725 has been
set to turn on at VIN = 9.5V and off at
VIN = 7.5V, and a linear regulator (Q1)
is used to establish bias for start-up.
Note that the LTC3725 requires that
the NDRV pin be at least 1V above
the VCC pin for proper linear regulator
operation. To meet this requirement,
while providing the lowest possibly
dropout voltage, a darlington transistor is used (Q1). JFET Q2 is used to
provide adequate bias current for the
NDRV pin at low input voltage, while
limiting the maximum current seen at
high input voltage. R11 is needed to
prevent back-feeding of current from
the NDRV pin into base of Q1 (and
VIN = 24V
85
80
IOUT
5A/DIV
20µs/DIV
VIN = 12V
VOUT = 15V
LOAD STEP = 0A TO 5A
Figure 8. Transient response
of the circuit in Figure 7
75
0
2
4
LOAD CURRENT (A)
6
Figure 9. Efficiency of
the circuit in Figure 7
13
L DESIGN FEATURES
Rugged 3.3V RS485/RS422
Transceivers with Integrated
Switchable Termination
by Steven Tanghe and Ray Schuler
Introduction
Medium and high speed RS485 networks must be terminated to avoid
data-corrupting reflections. This
means a termination resistor is placed
at each end of the bus. Of course, if
the network is expanded or reconfigured, the termination resistors must
also move. The 3.3V LTC2854 and
LTC2855 transceivers eliminate the
cumbersome task of shuffling termination resistors. These devices have
an integrated termination resistor
connected across the receiver inputs
that can be enabled or disabled with
simple logical control of an input
pin, making network configuration
and reconfiguration a snap. These
devices come in tiny packages and are
extremely robust, withstanding ESD
strikes of up to ±25kV HBM (LTC2854)
on the line I/O pins—the industry’s
highest protection level for an RS485
transceiver.
Other features of the LTC2854
and LTC2855 include a receiver with
balanced thresholds for excellent
duty cycle performance, high input
Block diagrams for the LTC2854 and
LTC2855 are shown in Figure 2.
Switchable Termination
Figure 1. Photograph of the (left to right)
LTC2854 3mm × 3mm DFN, LTC2855
4mm × 3mm DFN, and the LTC2855 SSOP
resistance allowing as many as 256
devices to be connected to one bus,
and a full failsafe output. The driver
offers low power operation, which in
conjunction with the receiver and
integrated termination resistor, provide a single die impedance-matched
network solution. Parts are available
in half- and full-duplex configurations
in tiny packages including 10- and
12-pin DFN as well as 16-lead SSOP
(see Table 1 and photo in Figure 1).
Differential signals propagating down
a twisted pair transmission line are
partially reflected when an impedance mismatch is encountered. The
reflected signal causes constructive
and/or destructive interference on the
line that can corrupt data. To prevent
this condition and optimize system
performance, transmission lines
should be terminated at each end with
a resistor matching the characteristic
impedance of the cable.
The LTC2854 and LTC2855 transceivers integrate this termination
resistor so that it can be selectively
included or excluded simply by controlling the Termination Enable pin
(TE). The resistor is effectively connected across the receiver input pins
by setting TE high and disconnected
when TE is low or the device is unpowered. This arrangement is nearly
ideal from a system management
VCC
RE
DE
VCC
SLEEP/SHUTDOWN
LOGIC AND DELAY
120Ω
RTERM
DE
SLEEP/SHUTDOWN
LOGIC AND DELAY
120Ω
RTERM
RECEIVER
B
(25kV)
B
(15kV)
125k
RIN
DI
DRIVER
A
(15kV)
TE
125k
RIN
RO
RECEIVER
125k
RIN
DI
RE
TE
125k
RIN
RO
A
(25kV)
Z
(15kV)
DRIVER
Y
(15kV)
GND
LTC2854
GND
LTC2855
Figure 2. Block diagrams of the LTC2854 and LTC2855
14
Linear Technology Magazine • March 2007
DESIGN FEATURES L
200 FEET
CAT 5 CABLE
100 FEET
CAT 5 CABLE
LTC2854
LTC2854
120Ω
LTC2854
120Ω
R
120Ω
R
R
D
RO RE TE DE
D
DI
RO RE TE DE
NODE 1 - Tx
D
DI
RO RE TE DE
NODE 2 - Rx
NODE 2
NODES 1 AND 2 PRESENT;
TE ON AT NODES 1 AND 2
DI
NODE 3 - Rx
NODE 2
NODE 2
NODE 3
NODE 3
NODES 1, 2 AND 3 PRESENT;
TE ON AT NODES 1 AND 3
NODES 1, 2 AND 3 PRESENT;
TE ON AT NODES 1 AND 2
Figure 3. Effects of termination placement with network expansion
standpoint, especially under conditions where a network configuration
changes and the termination resistor
needs to be moved to the new end of
the bus. In this case, manual removal
and placement of a discrete resistor
is not necessary; rather the change
is controlled digitally with the appropriate selection of TE pins on the
LTC2854 or LTC2855.
To illustrate the importance of
termination placement, consider the
configuration shown in Figure 3 where
the effects of network expansion are
presented. The initial configuration
consists of nodes 1 and 2, made up of
LTC2854 transceivers connected with
200 feet of Cat 5 cable. The waveforms
in the lower left of the figure show
the signal received at node 2, driven
from node 1. Both ends of the cable
are terminated by setting the TE
pins high on both transceivers. The
received signal looks clean because
the bus is properly terminated. A
small impedance mismatch between
the cable characteristic impedance of
100Ω and the termination resistor of
120Ω, results in a slight bump in the
waveform. This effect is minor and the
figure serves to illustrate that the termination resistor in the LTC2854 and
LTC2855 is compatible with popular
low cost 100Ω cables.
The second set of waveforms on the
bottom of Figure 3 show the results of
introducing a third node to the system through 100 feet of added cable
but without moving the termination
resistor to the new end location. The
Table 1. Product selection
PART NUMBER
DUPLEX
PACKAGE
ESD on Line I/O
(HBM)
LTC2854
HALF
DFN-10
±25kV
LTC2855
FULL
SSOP-16, DFN-12
±15kV
Linear Technology Magazine • March 2007
waveforms at node 3 and node 2 are
both severely distorted from reflections
caused by the improper termination.
In the third set of waveforms, the
termination placement has been corrected by setting TE high at nodes 1
and 3 only, thereby cleaning up the
signals received at nodes 2 and 3. The
logic-selectable termination resistors
in the LTC2854 permit this correction with no physical intervention
required.
The termination resistance is well
maintained over temperature, common mode voltage and frequency (as
illustrated in Figure 4). Furthermore,
the termination network adds only
insignificant capacitive loading to the
receiver pins. The input capacitance
on the LTC2855’s A and B pins is approximately 9pF measured to ground
and 3.5pF differentially.
Balanced Threshold Receiver
with Full Failsafe
The LTC2854 and LTC2855 feature
a low power receiver that draws
only 450µA. The single-ended input
resistance to ground on each of the
15
L DESIGN FEATURES
135
150
VAB = 2V
185
140
120
115
110
130
MAGNITUDE (Ω)
RESISTANCE (Ω)
125
120
110
105
100
–40 –20
0
20
40
60
80
100
–10
100 120
30
170
TEMPERATURE (˚C)
(a)
–5
5
10
0
COMMON MODE VOLTAGE (V)
15
15
PHASE
155
140
125
0
–15
MAGNITUDE
–30
110
– 45
95
– 60
80
10 –1
PHASE (°)
RESISTANCE (Ω)
130
– 75
101
10 0
FREQUENCY (MHz)
(b)
(c)
Figure 4. LTC2855 termination resistance vs (a) temperature, (b) common mode voltage, and (c) frequency.
receiver inputs is greater than 96kΩ
when the termination is disabled. This
is eight times higher than the requirements specified in the TIA/EIA-485-A
standard and thus this receiver represents a one-eighth unit load. This,
in turn, means that 8× the standard
number of receivers, or 256 total, can
be connected to a line without loading it beyond what is called out in the
standard.
The receiver implements a full failsafe design that drives RO high when
the inputs to the receiver are shorted,
left open, or terminated (externally or
internally) but not driven.
A key element of the LTC2854/
LTC2855 receiver is that it uses a
window comparator with two voltage
thresholds balanced around zero for
excellent duty cycle performance. As
illustrated in Figure 5, for a differential
signal approaching from a negative
direction, the threshold is +65mV.
When approaching from the positive
direction, the threshold is –65mV.
Each of these thresholds has 20mV of
hysteresis (not shown in the figure).
This windowing around 0V preserves
duty cycle for small inputs with heavily slewed edges. This performance
is highlighted in Figure 6, where a
RECEIVER
OUTPUT HIGH
–200mV
–65mV
0V
65mV
FAILSAFE THRESHOLD
(DELAYED)
Figure 5. Receiver input
threshold characteristics
16
200mV
detect the shorted failsafe condition
that preserves normal signal integrity. In normal operation, the two
thresholds shown in Figure 5 are
used to determine the receiver output
state. However, if the receiver inputs
remain between thresholds for more
than about 3µs, the receiver output
is driven high, reflecting this failsafe
condition.
Driver
The differential driver of the LTC2854
and LTC2855 easily delivers RS485/
RS422 signals at data rates up to
20Mbps. Figure 7 shows the clean
edges and excellent zero crossings of
the LTC2854 driver running at 20Mbps
into a 54Ω load. Figure 8 shows a single
50ns pulse (equivalent to one bit at
20Mbps) delivered through 100 feet
of standard unshielded Cat 5 cable
and received by a second LTC2854
transceiver.
Driver outputs have current limiting that offers protection from short
circuits to any voltage within the absolute maximum range of (VCC–15V)
DI
A, B
100mV/DIV
A
(A-B)
100mV/DIV
RO
RECEIVER
OUTPUT LOW
signal is driven through 4000 feet of
Cat 5e cable at 3Mbps. The top set of
traces show the signals coming into
the receiver after traveling down the
long cable. The middle trace is the difference of the top two signals and the
bottom trace is the resulting waveform
out of the receiver at the RO pin. It is
clear that even though the differential
signal peaks at just over ±100mV and
is heavily slewed, the output maintains
a nearly perfect signal with almost no
duty cycle distortion.
Few devices can match this level
of performance because the balanced
receiver thresholds are at odds with
shorted failsafe requirements. Other
parts typically include a negative
threshold in the receiver so that when
the inputs are shorted together (i.e., 0V
differential) the receiver output drives
high, indicating a failsafe condition.
Unfortunately, the negative offset can
cause severe duty cycle distortion for
small, slow-edge rate signals like those
presented in Figure 6.
The LTC2854 and LTC2855 avoid
this problem by using a method to
B
2V/DIV
A-B
RO
2V/DIV
VAB
200ns/DIV
Figure 6. A 3Mbps signal driven down 4000
feet of Cat 5e cable. Top traces: received
signals after transmission through cable;
middle trace: math showing difference of top
signals; bottom trace: receiver output.
20ns/DIV
Figure 7. The LTC2854 driver toggling at the
maximum data rate of 20Mbps into 54Ω. A and
B are the driver outputs.
Linear Technology Magazine • March 2007
DESIGN FEATURES L
DI
B
2V/DIV A
RO
100ns/DIV
Figure 8. The LTC2854 driver delivering a
single 50ns pulse through 100ft of Cat 5 cable,
which is received by another LTC2854. Both
parts have their on-chip termination enabled.
Top trace is the input to the transmitting
device and the middle and bottom traces are
observed at the receiving part.
to +15V, with typical peak current
not exceeding 180mA. Additionally,
thermal shutdown protection disables
the driver, receiver, and terminator if
excessive power dissipation causes
the device to heat to temperatures
above 160°C. When the temperature
drops below 140°C, normal operation
resumes.
Extreme ESD Protection
The driver output pins and receiver
input pins on the LTC2854 are protected to ESD levels of ±25kV HBM
with respect to ground or VCC. The fullduplex LTC2855 withstands ±15kV
ESD. These protection levels exist for
all modes of device operation including
power-down, standby, receive, transmit, termination and all combinations
of these. Furthermore, the protection
level is valid whether VCC is on, shorted
to ground, or disconnected.
When a line I/O pin on the
LTC2854/LTC2855 is hit with an
LTC3805, continued from page reduced and the capacitance increased
in proportion. Also, the resistor divider
connected to the RUN pin must be
adjusted for the new input voltage.
Finally, the 68mΩ current sense resistor should be reduced in value to
account for the higher input current.
For an increase in input voltage, everything is changed proportionally in
the opposite direction.
Similarly, a change in the output
voltage involves a change in the diode,
Linear Technology Magazine • March 2007
Figure 9. The LTC2854 sending data (see scope traces in background)
while hit with multiple 30kV ESD strikes on the ‘A’ pin.
ESD strike during operation, the part
undergoes a short disturbance of duration similar to the ESD event and
then fully recovers. The device does
not latch up and there is no need to
toggle states or cycle the supply to
recover. This is true whether the part
is in a static state or sending/receiving
data and for the full range of ground
common mode voltages called out in
the RS485 standard. The photo in Figure 9 shows the LTC2854 absorbing
the energy from an ESD gun (configured for IEC air discharge) delivering
repeated 30kV strikes to the ‘A’ pin
while transmitting data. The oscilloscope traces in the background show
data toggling happily on the A and B
pins before and after a strike, with a
positive glitch only during the ESD
event. This device can handle many
such strikes without damage.
Conclusion
The LTC2854 and LTC2855 break
new ground in the world of 3.3V
RS485/RS422 transceivers. The inclusion of a selectable termination
resistor provides a complete solution
to RS485 networking with the ability
to remotely configure the network
for optimal data transfer. Unparalleled ESD performance provides
outstanding ruggedness while a balanced-threshold receiver with full
failsafe capability makes this family
of small-footprint devices a natural
choice for modern RS485/RS422
systems. L
Conclusion
the number of turns in the secondary
winding of the transformer and the
voltage rating and value of the output
filter capacitor along with the appropriate change to the voltage divider
that senses the output voltage. If the
output voltage is between 4V and 9V,
the design of non-isolated converters
is very simple because VCC can be provided by a diode connected directly to
the output instead of the third winding
on the transformer.
Because of its flexibility, the flyback
converter is the most widely used
transformer-based converter. The
LTC3805 maximizes the flexibility of
the flyback converter by making it possible to use the same basic circuit for a
wide range of converter input and output voltages. Simply scale component
values to match voltage and current
conditions, greatly simplifying board
design and updates. L
17
L DESIGN FEATURES
Tiny High Efficiency 2A Buck
Regulator Directly Accepts
Automotive, Industrial and
Other Wide Ranging Inputs
Introduction
2.4MHz by using a resistor tied from
the RT pin to ground. This allows a
trade off between component size and
efficiency. The switching frequency
can be synchronized to an external
clock for noise sensitive applications.
An external resistor divider programs
Automotive batteries, industrial power
supplies, distributed supplies and
wall transformers are all sources of
wide-ranging, high voltage inputs. The
easiest way to step down these sources
is with a high voltage monolithic stepdown regulator that can directly accept
a wide input range and produce a
well-regulated output. The LT3480 is a
new step-down regulator that accepts
input from up to 38V (60V transient)
while providing excellent line and load
regulation and dynamic response. The
LT3480 offers high efficiency solutions
over wide load range and keeps the
output ripple low during Burst Mode®
operation.
The LT3480 is a new
step-down regulator that
accepts input from up to
38V (60V transient).
the output voltage to any value above
the part’s 0.8V reference.
The LT3480 offers soft-start via a
resistor and capacitor on the RUN/SS
pin, thus reducing maximum inrush
currents during start-up. The LT3480
can withstand a shorted output. A
cycle-by-cycle internal current limit
protects the circuit in overload and
limits output power; when the output
voltage is pulled to ground by a hard
short, the LT3480 reduces its operating frequency to limit dissipation
LT3480 Features
Available in either a 10-pin MSOP
or a 3mm × 3mm DFN package, the
LT3480 offers an integrated 3.5A
power switch and external compensation for design flexibility. The LT3480
employs a constant frequency, current
mode architecture. The switching frequency can be set between 250kHz and
by Kevin Huang
and peak switch current. This lower
frequency allows the inductor current
to safely discharge, thus preventing
current runaway.
The high side bootstrapping boost
diode is integrated into the IC to minimize solution size and cost. When the
output voltage above 2.5V, the anode
of the boost diode can be connected to
output. For output voltages lower than
2.5V, the boost diode can be tied to a
separate rail or to the input (<28V). For
systems that rely on a well-regulated
power source, the LT3480 provides
a power good flag that signals when
VOUT reaches 90% of the programmed
output voltage.
Modes of Operation:
Low Ripple Burst and
Forced Continuous
Two modes of operation can be selected through the SYNC pin. Applying
a logic low to the SYNC pin enables
low ripple Burst Mode operation,
which maintains high efficiency at
light load while keeping the output
voltage ripple low. During Burst Mode
L1*
10µH
D1
DFLS240L-7
C1
0.47µF
1
2
3
VIN E1
6.3V TO 38V
TRANSIENT
TO 60V
5V
0V
4
5
C3
2.2µF
50V
BD
LT3480
BOOST
RT
VC
SW
FB
VIN
PGOOD
RUN/SS
SYNC
10
R1
56.2k
9
R2
110k
8
7
R3
590k
6
GND
11
R6
100k
R4
100k
C6
680pF
E5 V
OUT
5V, 2A
E7
R5
20k
C7
0.1µF
50V
C4
22µF
10V
PGOOD
C8
100pF
* L1: SUMIDA CDR7D43MNNP-100NC
Figure 1. A 600kHz 6.3V–38V input DC/DC Converter using the LT3480 delivers 2A at 5V output.
18
Linear Technology Magazine • March 2007
DESIGN FEATURES L
100
VIN = 12V
EFFICIENCY (%)
90
VOUT
10mV/DIV
80
70
IL
0.2A/DIV
60
50
0
500
1000
1500
LOAD CURRENT (mA)
2000
5µs/DIV
Figure 2. Efficiency for circuit in Figure 1
Figure 3. LT3480 Burst Mode operation at 10mA load
operation, the LT3480 delivers single
cycle bursts of current to the output
capacitor followed by sleep periods
when the output power is delivered to
the load by the output capacitor. Between bursts, all circuitry associated
with controlling the output switch is
shut down, reducing the input supply
current and BD quiescent current to
30µA and 80µA respectively. As the
load current decreases to a no load
condition, the percentage of time
that LT3480 operates in sleep mode
increases and the average input current is greatly reduced, resulting in
high efficiency. The LT3480 has a very
low (less than 1µA) shutdown current
which significantly extends battery
life in applications that spend long
periods of time in sleep or shutdown
mode. For applications that require
constant frequency operation even at
no load, the LT3480 can be put into
VOUT
2V/DIV
IL
2A/DIV
200µs/DIV
Figure 4. Soft-start of LT3480
forced continuous mode operation by
tying the SYNC pin above 2.5V.
6.3V–38V to 5V, 2A DC/DC
Converter with All Ceramic
Capacitors
Figure 1 shows the LT3480 producing 5V at 2A from an input of 6.3V to
38V with 60V transient. The circuit is
programmed for a 600kHz switching
frequency. Figure 2 shows the circuit
efficiency at 12V input. The efficiency
peaks at 90% and remains high across
the entire load range. The SYNC pin
is tied to the ground to enable Burst
Mode operation and achieve high efficiency at light load. Figure 3 shows the
inductor current and output voltage
ripple under single pulse Burst Mode
operation at 10mA load. The output
L1*
2.2µH
D1
DFLS240L-7
C1
0.47µF
1
2
3
VIN E1
9V TO 22V
60V TRANSIENT
4
5
C3
2.2µF
50V
BD
LT3480
BOOST
VC
SW
FB
VIN
PGOOD
RUN/SS
SYNC
0V
10
R1
11.5k
9
R2
110k
8
7
R3
590k
6
GND
11
R6
100k
5V
RT
R4
100k
E7
R5
20k
C7
0.1µF
C6
680pF
E5 V
OUT
5V, 2A
C4
10µF
10V
PGOOD
C8
100pF
* L1: SUMIDA CDRH4D22/HP-2R2NC
Figure 5. High operating frequency allows the use of small inductors and capacitors.
This 2MHz, 9V–22V input DC/DC converter using the LT3480 delivers 2A at 5V output.
Linear Technology Magazine • March 2007
19
L DESIGN FEATURES
L1*
2.2µH
D1
DFLS240L-7
C1
0.47µF
1
2
3
VIN E1
9V TO 22V
60V TRANSIENT
4
5
C3
2.2µF
50V
BD
LT3685
RT
BOOST
VC
SW
FB
VIN
PGOOD
RUN/SS
SYNC
R1
11.5k
10
9
R2
110k
8
7
R3
590k
6
GND
11
R4
100k
R6
100k
5V
E5 V
OUT
5V, 2A
C4
10µF
10V
E7
R5
20k
0V
C7
0.1µF
C6
680pF
PGOOD
C8
100pF
* L1: SUMIDA CDRH4D22/HP-2R2NC
Figure 6. A 2MHz 9V–22V input DC/DC converter using the LT3685 delivers 2A at 5V output.
voltage ripple VP–P is less than 10mV
as a result of low ripple Burst Mode
operation.
An external signal can drive the
RUN/SS pin through a resistor and
capacitor to program the LT3480’s
soft-start, reducing maximum inrush
current during start-up. Figure 4
shows the start-up waveform.
2MHz, 9V–22V to 5V, 2A
DC/DC Converter with All
Ceramic Capacitors
Figure 5 shows a step-down DC/DC
converter using all ceramic capacitors.
This circuit provides a regulated 5V
output at up to 2A from an input of
9V to 22V. The high 2MHz switching
frequency allows the use of small
inductor and capacitors.
In typical automotive batteryvoltage applications, high voltage
line transients, such as during a
load-dump condition, must be accommodated. The circuit shown in Figure 5
can operate through intermittent high
voltage excursions to 60V. This converter is an ideal choice for operation
near an AM radio receiver because it
operates above the broadcast band
and the switching noise can be filtered
in a predictable manner. The SYNC
pin is tied to output to disable Burst
Mode operation in order to eliminate
AM band interference. The efficiency
of this circuit reaches 85%.
The LT3685, similar to the LT3480
without Burst Mode operation, is also
20
a good candidate for this application.
Figure 6 shows the circuit using the
LT3685 for this application.
negative output tracks the positive
output within 5%. For a more complete
description of this circuit, see Linear
Technology Design Note 100.
Dual Output Converter
Conclusion
Dual output supplies are required
for many applications. The circuit in
Figure 7 uses an LT3480 to generate
both positive and negative 5V supplies.
The two inductors shown are actually
two windings on a coupled inductor.
The load current on the positive output
should be larger than the load on the
negative output. With this restriction
satisfied, the voltage magnitude of the
The wide input range, low quiescent
current, small size and robustness
of the LT3480 make it an easy fit in
automotive, industrial and distributed
power applications. It is highly efficient
over the entire load range. Its unique
low ripple Burst Mode operation helps
to save battery power life while maintaining low output ripple. L
L1A*
10µH
D1
DFLS240L-7
C1
0.47µF
1
2
3
VIN E1
6.3V TO 38V
5V
0V
4
C3
2.2µF
50V
5
LT3480
BD
RT
BOOST
VC
SW
FB
VIN
PGOOD
RUN/SS
SYNC
10
R1
60.4k
R2
110k
9
8
7
R3
590k
6
GND
11
R6
100k
R4
100k
R5
20k
C7
0.1µF
C9
10µF
D2
10V DFLS240L
C6
680pF
C10
22µF
10V
* L1: COOPER ELECTRONIC DRQ74-100
VOUT1
5V, 1A
C4
22µF
10V
PGOOD
C8
100pF
VOUT2
–5V, 0.5A
L1B*
10µH
Figure 7. A ±5V dual output DC/DC converter. As long as the load on the negative channel is less
than the load on the positive channel, the voltage magnitude of the negative output tracks the
positive output within 5%.
Linear Technology Magazine • March 2007
DESIGN FEATURES L
36V Dual 1.4A Monolithic Step-Down
Converter has Start-Up Tracking and
by Keith Szolusha
Sequencing
Introduction
The LT3508 simplifies the design
of dual output, wide-input-range
power converters—especially those
that require power supply tracking
and sequencing. It is a dual output
current mode PWM step-down DC/DC
converter with internal power switches
capable of generating a pair of 1.4A
outputs. Its wide 3.6V to 36V input
range makes it suitable for regulating
power from a wide variety of sources,
including automotive batteries, 24V
industrial supplies and unregulated
wall adaptors. Both converters are
synchronized to a single oscillator
programmable from 250kHz up to
2.5MHz and run with opposite phases,
reducing input ripple current. The high
operating frequency allows the use of
small, low cost inductors and ceramic
capacitors, resulting in low, predictable output ripple. Each regulator has
independent tracking and soft-start
circuits and generates a power good
signal when its output is in regulation,
making power supply sequencing and
VIN
6V TO 36V
95
SHDN
1N4448W
VIN1
BOOST1
VIN2
80
75
65
0
0.5
1
LOAD CURRENT (A)
Figure 2. Efficiency for circuit of Figure 1
1N4448W
10µH
SW2
LT3508EFE
B240A
35.7k
FB1
VC1
VC2
TRACK/SS1
PG1
RT/SYNC
1000pF
B240A
CDRH6D28
56.2k
10µF
10V
OUT2
5V
1.4A
FB2
TRACK/SS2
20k
1000pF
1.5
0.22µF
SW1
CDRH5D28
1500pF
VOUT1 = 3.3V
85
BOOST2
0.22µF
11.5k
VOUT2 = 5V
70
6.2µH
10µF
6.3V
VIN = 12V
90
10µF
50V
SHDN
OUT1
3.3V
1.4A
interfacing with microcontrollers and
DSPs easy.
Cycle-by-cycle current limit, frequency foldback and thermal shutdown
provide protection against shorted outputs, and soft-start eliminates input
current surge during start-up. The
low current (<2µA) shutdown mode
enables easy power management in
battery-powered systems.
EFFICIENCY (%)
The latest DSPs and microcontrollers
found in automotive electronics,
industrial supplies, and even walltransformers typically require power
supplies with output voltages of both
1.8V and 3.3V and output current
capability of 1A or greater. DSL and
cable modems also require multiple
supplies, usually a combination of
a single 5V supply rail and either a
3.3V or 1.8V rail. PCI Express and
motherboard interconnect devices
supply 3.3V or 5V in addition to a
12V intermediary source. In all of
these cases, the supplies must follow
a specific start-up sequence or track
each other to avoid system latch up
or worse.
One common challenge in these applications is producing well-regulated
outputs from wide ranging inputs. For
instance, a 12V automotive battery
produces a voltage range from a low
of 4V to a high of 36V. 24V industrial supplies and rectified 12V wall
transformers produce similarly wide
voltage ranges.
52.3k
PG2
43k
10.7k
GND
330pF
f = 700kHz
Figure 1. Dual 1.4A monolithic step-down converter with 3.3V and 5V outputs
Linear Technology Magazine • March 2007
21
L DESIGN FEATURES
VOUT2
200mV/DIV
VOUT1
200mV/DIV
ILOAD1
500mA/DIV
ILOAD2
500mA/DIV
VIN = 6.8V
20µs/DIV
VOUT2 = 5V
LOAD STEP = 700mA TO 1400mA
VIN = 12V
20µs/DIV
VOUT1 = 3.3V
LOAD STEP = 700mA TO 1400mA
(a)
(b)
Figure 3. Dual step-down 5V (a) and 3.3V (b) output voltage transient response
VIN
5V TO 16V
4.7µF
25V
SHDN
SHDN
1N4448W
VIN1
VIN2
BOOST1
0.1µF
0.1µF
OUT1
1.8V
1A
1N4448W
BOOST2
3.0µH
22µF
6.3V
4.7µH
CDRH3D18
SW1
SW2
DFLS130
LT3508EUF
CDRH5D18C
DFLS130
12.4k
35.7k
FB1
10.0k
FB2
VC1
VC2
TRACK/SS1
PG1
TRACK/SS2
9.1k
RT/SYNC
1200pF
1000pF
1000pF
10µF
6.3V
OUT2
3.3V
1A
PG2
11k
11.5k
GND
16.9k
680pF
f = 1.6MHz
Figure 4. Small dual step-down 1.8V and 3.3V schematic with output sequencing
Versatility Comes from
Independent Control of
Two 1.4A Channels
Each channel has its own power good,
track/soft-start and, unlike most dual
channel converters, each has its own
VIN pin (more about this below). The
boost pin for each channel can be
tied to the higher of the two outputs,
one to each output (if the channels
are turned on and off separately), the
input, or an external source. The boost
pin voltage must be at least 3V above
the switch pin voltage for saturation
of the internal power switch.
Individual track/soft-start and
power good pins offer a variety of
supply tracking and sequencing options. The channels can track each
other coincidentally or ratiometrically.
The power good pins can be used
for sequencing the two channels or
22
simply interfacing with an external
microcontroller.
The unique, separate VIN pins for
each channel offer uncommon design
flexibility. For instance, the converter
can satisfy high VIN/VOUT ratio applications that might be otherwise limited by
a single converter’s typical minimum
duty cycle constraints. Simply cascade
the two converters by attaching the
output of one channel to the input
of the other channel. This allows the
input voltage to be twice as high for
a given output voltage and switching
frequency without violating minimum
duty cycle constraints. In some cases,
VOUT1
500mV/DIV
VIN
2V/DIV
VOUT2
500mV/DIV
VTRACK/SS2
500mV/DIV
VIN1 = VIN2 = 12V
VOUT1 = 1.8V
VOUT2 = 3.3V
ILOAD1 = ILOAD2 = 1A
500µs/DIV
Figure 5. Dual step-down 1.8V and 3.3V start-up with output sequencing
Linear Technology Magazine • March 2007
DESIGN FEATURES L
VIN
8.5V TO 28V
CIN1
2.2µF
35V
CIN2
1µF
10V
SHDN
SHDN
1N4448W
VIN1
VIN2
BOOST1
1N4448W
BOOST2
0.1µF
COUT1
4.7µF
10V
0.1µF
33µH
7.7V
2.4µH
SW1
CDRH3D18LP
SW2
LT3508EUF
DFLS130
86.6k
FB1
10.0k
1%
1000pF
12.4k
VC2
TRACK/SS1
PG1
OUT2
1.8V
1.4A
PG2
10.0k
18k
GND
680pF
RT
9.76k
2200pF
COUT2
22µF
6.3V
FB2
RT/SYNC
2200pF
DFLS220L
VC1
TRACK/SS2
30k
CDRH3D14/HP
f = 2.2MHz
Figure 6. 2.2MHz 28V to 1.8V step-down with cascaded channels and output sequencing
VIN
2V/DIV
VOUT1
2V/DIV
VOUT2
500mV/DIV
VTRACK/SS2
500mV/DIV
VIN1 = 12V
VIN2 = VOUT1 = 7.7V
VOUT2 = 1.8V
ILOAD2 = 1.4A
500µs/DIV
Figure 7. 2.2MHz 28V to 1.8V step-down start-up with output sequencing
the separate VIN pins also allow the two
channels to be run from two separate
current-limited sources that may not
have enough power alone to provide full
power to both channels’ outputs.
The LT3508’s two channels run
180° out of phase to minimize input
current ripple and voltage ripple, thus
limiting EMI and reducing the required
size of the input capacitor.
High VIN, Low VOUT
and Adjustable
Switching Frequency
The wide input range of 3.6V to 36V
makes the LT3508 suitable for regulating power from a wide variety of
sources, including automotive batteries, 24V industrial supplies and
unregulated wall adaptors. The operating frequency for the converters can
be programmed by a single resistor
Linear Technology Magazine • March 2007
or synchronized to an external clock
ranging from 250kHz to 2.5MHz. High
operating frequency allows the use of
VIN
12V
C1
4.7µF
40.2k
OUT1
5V
0.9A
VIN1
C3
0.1µF
6.8µH
BOOST1
C6
10µF
BOOST2
SW1
D3
52.3k
10.0k
VIN2
C2
4.7µF
D2
VIN2
3.3V
SHDN
D1
14.7k
small, low cost inductors and ceramic
capacitors, resulting in low, predictable output ripple. However, selecting
SW2
100pF
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
C1 TO C6: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B140
D4: DIODES INC. B120
47k
33.2k
C5
47µF
15.0k
RT/SYNC
0.1µF
OUT2
1.8V
1.4A
18.7k
FB2
GND
3.3µH
D4
LT3508
FB1
43k
C4
0.1µF
100k
330pF
fSW = 1MHz
POWER
GOOD
Figure 8. PCI Express power supply with separate inputs
23
L DESIGN FEATURES
a low operating frequency makes it
possible to produce high input voltage, low output voltage applications by
reducing the duty cycle. The LT3508’s
low minimum switch on time of 130ns
offers the benefits of high frequency
even in high input-output ratio applications. For instance, a frequency
of 700kHz is low enough to provide
6V to 36V input voltage range for both
5V and 3.3V outputs at full 1.4A load
current (see Figure 1).
The output voltage for the LT3508
can be set as low as the 0.8V refer-
ence voltage. With 130ns minimum
on-time, the maximum input voltage
is calculated by:
VIN(MAX ) =
( VOUT + VF )
t ON(MIN) • fOSC
– VF + VSW
VF is the forward voltage of the catch
diode, VSW is the internal switch saturation voltage, and fOSC is the oscillator
frequency. For 36VIN to 3.3VOUT, fOSC
must be below 790kHz. To achieve
36VIN to 1.8VOUT, fOSC must be 470kHz
or less. Likewise, a simple 12V to
Independent Start-Up
Ratiometric Start-Up
Coincident Start-Up
VOUT1
VOUT1
VOUT2
VOUT1
VOUT2
1V/DIV
VOUT2
1V/DIV
1V/DIV
20ms/DIV
0.1µF
TRACK/SS1 VOUT1
20ms/DIV
5V
0.22µF
TRACK/SS2 VOUT2
20ms/DIV
5V
TRACK/SS1 VOUT1
LT3508
0.047µF
3.3V step-down ratio is possible with
a switching frequency of 2.3MHz. An
application converting 12VIN to 5VOUT
and 3.3VOUT can take advantage of a
high switching frequency of 2.2MHz
and remain above the AM band for
automotive electronics.
In cases where both a high switching frequency and a high step down
ratio are required (as in the case of
an automotive power supply that requires a 2.2MHz switching frequency
to keep interference outside of the
AM band), a cascaded solution can
0.1µF
LT3508
3.3V
5V
TRACK/SS1 VOUT1
LT3508
3.3V
TRACK/SS2 VOUT2
3.3V
TRACK/SS2 VOUT2
R1
28.7k
(9a)
R2
10.0k
(9b)
Output Sequencing
(9c)
Controlled Power Up and Down
VOUT1
VOUT1
VOUT2
VOUT2
1V/DIV
1V/DIV
EXTERNAL SOURCE
20ms/DIV
0.1µF
20ms/DIV
TRACK/SS1 VOUT1
5V
EXTERNAL
SOURCE
LT3508
PG1
0.047µF
TRACK/SS2 VOUT2
+
–
3.3V
TRACK/SS1 VOUT1
5V
LT3508
TRACK/SS2 VOUT2
3.3V
R1
28.7k
R2
10.0k
(9d)
(9e)
Figure 9. Tracking and soft-start options
24
Linear Technology Magazine • March 2007
DESIGN FEATURES L
be used. As shown in Figure 6, 28VIN
to 1.8VIN is possible if one output is
set for 7.7V and tied to the VIN pin
of the 1.8V channel. Higher switching frequency reduces inductor and
capacitor sizes and achieves faster
transient response.
Fast Transient Response
The current mode architecture of the
LT3508 control loop yields fast transient response with small, ceramic
output capacitors and simple compensation. Small 0805 and 1206 case size
10µF and 22µF 6.3V ceramic output
capacitors are typical for up to 1.4A
output applications. High temperature
coefficient capacitors such as X5R and
X7R ceramics are recommended for
most designs.
Figure 3 shows the transient
response for a typical LT3508 application. Transient response ripple is
about 200mVP–P for both the 3.3V output and the 5V output. The response
time is about 20µs to 40µs, excellent
for 1.4A outputs. This is an important
feature when the power supply is used
with DSPs and microcontrollers that
are sensitive to voltage ripple.
Low Dropout
The LT3508 features low dropout for
output voltages above 3V. The minimum operating voltage of the device
is determined either by the LT3508’s
undervoltage lockout or by its maximum duty cycle. If VIN1 and VIN2 are
tied together, the undervoltage lockout
is at 3.7V or below. If the two inputs
are used separately, then VIN1 has an
undervoltage lockout of 3.7V or below
and VIN2 has an undervoltage lockout
of 3V or below. Because the internal
supply runs off VIN1, channel 2 will
not operate unless VIN1 is above its
undervoltage lockout. The dropout of
the 5VOUT circuit shown in Figure 1 is
less than 1V, with start-up occurring at
a minimum of 5.9V and the converter
running down to 5.5V before dropout
occurs.
Unlike many fixed frequency regulators, the LT3508 can extend its duty
cycle by turning on for multiple cycles.
The LT3508 will not switch off at the
end of each clock cycle if there is sufLinear Technology Magazine • March 2007
ficient voltage on the boost capacitor.
Eventually, the voltage on the boost
capacitor falls and requires refreshing.
A bigger boost capacitor allows for
higher maximum duty cycle. Circuitry
detects a depleted boost capacitor
and forces the switch to turn off, allowing the inductor current to charge
up the boost capacitor. This places a
limitation on the maximum duty cycle.
The minimum input voltage can be
calculated as:
VIN(MIN) =


 – VF + VSW
SW 
( VOUT + VF )  1+ β 1
βSW is the switch current to boost
current ratio. Refer to the data sheet
section “Minimum Operating Voltage”
for details.
Track/Soft-Start and
Power Good Pins Simplify
Supply Sequencing
DSPs and microcontrollers require
power supply sequencing and tracking. Both LT3508 channels have
independent tracking and soft-start
circuits and each generates a power
good signal when its output is in
regulation. Most start-up/shut-down
scenarios are possible by combining
the function of the track/soft-start
(TRACK/SS) with the power good (PG)
pins. Figure 9 shows how easy it is to
implement independent channel softstart, ratiometric start-up, coincident
start-up, output sequencing, and
externally controlled power up and
power down.
Soft-start prevents inrush current
spikes, which can drag down the
source voltage upon start-up and
cause other system problems. Simple
soft-start of each channel requires only
a capacitor on the pin (Figure 9a). The
rate of soft-start is determined by the
size of capacitor and by the capabilities
of the power source.
As the name suggests, the TRACK/
SS pins also facilitate supply tracking,
including ratiometric, coincident and
externally controlled start-up and
shut-down. Figure 9d shows how to
connect the PG pin of one channel to
the track pin of another channel to
sequence the two—one channel is held
off until the other channel is good.
The track/soft-start function can
also be used to power a channel down,
but to minimize current draw, shut
down the regulator via the shutdown
(SHDN) pin as described below.
Low Shutdown Current
When the shutdown pin is pulled low,
both channels turn off and the part
consumes a very low quiescent current (<2µA), saving battery energy and
extending lifetime. The shutdown pin
can also be used as a 2.63V accurate
undervoltage lockout (UVLO) with a
resistor divider from VIN. In shutdown,
the power good comparator is disabled
and not valid and the soft-start capacitors are reset.
TSSOP-16 and QFN Packages
The LT3508 is available in two types
of thermally-enhanced packages. The
UF package is a 4mm × 4mm 24-pin
QFN. The FE package is a 16-pin thermally-enhanced TSSOP surface mount
with an exposed thermal pad. Both
packages have equally low 40°C/W
junction-to-ambient thermal impedance and 10°C/W junction-to-case
impedance, important for applications
that require a high input voltage, high
switching frequency and high load
current, all of which raise the junction
temperature.
Conclusion
The LT3508 is a wide input voltage
36V dual 1.4A monolithic step-down
converter with tracking/soft-start
pins and power good pins for power
supply sequencing and simple diagnostic interface with DSPs and
microcontrollers. It has adjustable
switching frequency from 250kHz to
2.5MHz, either set by a resistor or
synchronized to an external source. Its
thermally enhanced packages and Eand I-grade temperature ratings allow
it to be used in thermally demanding
environments. Separate VIN pins for
each channel provide the capability
of cascading channels and achieving
extreme VIN to VOUT ratios by using the
output of one channel as the input for
the other. L
25
L DESIGN FEATURES
3-Phase Buck Controller Governs
One, Two or Three Outputs
by Theo Phillips and Teo Yang Long
+
10µF
0.1µF
Switching frequency can be phaselocked to an external source from
160kHz to 700kHz, or can be set with
a DC voltage on the PLLFLTR pin.
Typical pin-selectable frequencies of
220kHz, 400kHz and 560kHz are also
available. In either case, the CLKOUT
pin expresses the operating frequency
at zero, 60, or 180 degrees with respect
to channel 1’s switching frequency, a
useful feature where multiple controller ICs operate from the same set of
input capacitors.
D4
PLLIN/FC
PGOOD
VOUT1
2.5V/15A
COUT1
330µF
4V × 2
0.003Ω
2.2µH
+
SW2
BOOST1
0.1µF
M4
+
D6
BOOST2
BOOST3
70
50
40
SW2
VIN = 12V
VCC = 5V
fSW = 220kHz
VOUT1 = 2.5V
CHANNELS 2, 3: SHUTDOWN
30
20
10
0.01
0.1
1
10
CHANNEL 1 LOAD CURRENT (A)
L2
1.5µH
M2
M5
0.003Ω
+
D2
BG2
31.6k
68pF
0.01µF
10k
SENSE2+
SENSE2–
LTC3773
TRACK2
TRACK3
VFB1
TG3
SW3
ITH1
150pF
2200pF
M6
VIN
L3
1.2µH
D3
0.003Ω
+
VOUT2
1.8V/15A
COUT2
330µF
4V × 2
10µF
25V
×6
+
VIN
4.5V TO 22V
56µF
25V
×5
VOUT3
1.2V/15A
COUT3
330µF
4V × 2
PGND
5.6k
680pF
47pF
PLLFLTR
M3
BG3
ITH2
15k
10
100
D1, D2, D3: DIODES, INC. B340B
D4, D5, D6: CENTRAL SEMI CMDSH-3
L1: SUMIDA CDEP145-2R2MC
L2: SUMIDA CDEP145-1R5MC
L3: SUMIDA CDEP145-1R2MC
COUT1, COUT2, COUT3: SANYO POSCAP 4TPD330M
M1, M2, M3: RENESAS HAT2168H
M4, M5, M6: RENESAS HAT2165H
10µF
+
SENSE1
SENSE1–
TRACK1
100
Figure 2. Efficiency in Burst Mode operation
for the circuit of Figure 1.
VDR
BG1
1k
POWER LOSS
60
VCC
4.5V TO 6V
TG2
SW1
EFFICIENCY
80
SW3
TG1
M1
D1
VCC
0.1µF
D5
10k
90
Where three 15A outputs are required
in the smallest possible footprint,
the LTC3773 is the obvious choice.
Figure 1 shows a single-controller
schematic delivering three low voltage, high current outputs from a
single, loosely regulated supply. Each
output reference is guaranteed to
remain within ±1% over temperature.
SW1
PGOOD
VIN
100
Three Outputs,
One Controller
0.1µF
10k
During start-up, ratiometric tracking
holds the feedback references of VOUT2
and VOUT3 to 0.6V × (VOUT1/2.5), so
that the three outputs reach their
nominal operating levels at the same
time (Figure 3). TRACK1 ramps up by
charging the 0.01µF capacitor with an
internal 1µA source. Where tracking
is not required, all TRACK pins can
CHANNEL 1 EFFICIENCY (%)
The LTC3773 is an efficient, 3-phase
DC/DC controller capable of handling
inputs as high as 36V and supporting
one, two, or three output voltages from
0.6V to 5V with currents in excess of
15A per phase. Two channels may be
tied together for a 30A output, or three
channels for a single 45A output. In
all cases the channels are operated
out of phase to minimize stress on the
input capacitors.
Each channel provides for ratiometric or coincident tracking of any
supply, and sequencing requires just
an external capacitor. When all three
channels are disabled, the controller
typically draws just 18µA in shutdown
mode. Three light load operating
modes satisfy the priorities of various
applications. Burst Mode operation
yields maximum efficiency while
forced continuous mode sacrifices
some efficiency for low, predictable
current ripple. Pulse skipping mode is
a compromise between the two.
POWER LOSS (mW)
Introduction
SGND
SDB1
SDB2
SDB3
330pF
SENSE3+
SENSE3–
VFB3
VFB2
ITH3
47pF
POWER UP/SHUTDOWN
20k
20k
3.9k
20k
330pF
10k
1500pF
Figure 1. The LTC3773 regulates three high current outputs with ratiometric tracking, providing 2.5V, 1.8V, and 1.2V from a 4.5V–22V supply.
26
Linear Technology Magazine • March 2007
DESIGN FEATURES L
TRACK 1
0.5V/DIV
TRACK 1
1V/DIV
SDB
1V/DIV
2.5V VOUT1
1.8V VOUT2
1.2V VOUT3
1V/DIV
2.5V VOUT1
1.8V VOUT2
1.2V VOUT3
1V/DIV
2.5V VOUT1
1.8V VOUT2
1.2V VOUT3
1V/DIV
0.1s/DIV
0.1s/DIV
Figure 4. Coincident Tracking
Figure 5. Supply sequencing
implemented by applying a single
external ramp to all three SDB pins.
The power-up thresholds for SDB1,
2, and 3 are 1.2V, 1.8V, and 2.4V.
0.1s/DIV
Figure 3. Ratiometric tracking in
action. Channel 1’s reference does not
exceed the lesser of VTRACK1 or 0.6V.
The same is true for channels 2 and 3,
except that their track voltages follow
channel 1’s reference. This ensures that
channels 2 and 3 follow channel 1.
10Ω
10Ω
POWER DOWN VOUT1
1nF
POWER DOWN VOUT2
1500pF
4
5
330pF
6
6.8k
7
15k
8
20k
9
10
20k
11
VOUT2
12
10k
PGOOD
PHASEMD
SENSE1+
SDB1
SENSE1–
VFB1
TG1
ITH1
SW1
SGND
SW2
LTC3773
ITH2
TG2
ITH3
BOOST2
VFB2
BOOST3
VFB3
TG3
TRACK2
SW3
TRACK3
BG1
–
SENSE2
BG2
SENSE2+
VDR
SENSE3–
10k
BOOST1
13
14
15
16
17
18
31
30
10µF
25V
x6
CMDSH-3
0.1µF
29
HAT2168H
28
27
26
24
VIN
4.5V TO 22V
+
CIN
56µF
25V
x5 L1
1µH
3mΩ
+
B340B
0.1µF
CMDSH-3
HAT2165H
VIN
25
0.1µF
CMDSH-3
HAT2168H
HAT2165H
L2
0.6µH
VOUT1
2.5V/15A
COUT1
330µF
4V
x2
47µF
3mΩ
VOUT2
1.8V/30A
B340B
23
22
VIN
+
21
20
HAT2168H
L3
0.6µH
19
1nF
1nF
V5V
4.5V TO 6V
PGOOD
32
BG3
100pF
33
CLKOUT
3
34
PLLIN/FC
15k
35
PLLFLTR
2
1500pF
VCC
47.5k
36
TRACK1
SENSE3+
1
37
SDB2
PGND
VOUT1
0.01µF
38
SDB3
39
10k
COUT2
330µF
2.5V
x4
47µF
x2
3mΩ
B340B
CLKOUT
HAT2165H
CLKIN
10Ω
10Ω
10Ω
10k
0.1µF
10Ω
+
10µF
1µF
2Ω
CONTINUOUS
MODE FOR
TRACKING
+
10µF
L1: PULSE PG0006.102
L2, L3: PULSE PG0006.601
COUT1: SANYO POSCAP 4TPD330M
COUT2: SANYO POSCAP 2R5TPE330M9
Figure 6. 3-phase, dual output converter with coincident tracking. The 1.8V output is operated in an antiphase configuration, using channels 2
and 3 of the LTC3773. This is implemented by tying PHASEMD high, which causes TG1 to lead TG2 and TG3 by 90° and 270°, respectively, and by
connecting together the TRACK, SDB, VFB and ITH pins of channels 2 and 3.
Linear Technology Magazine • March 2007
27
L DESIGN FEATURES
be connected to external capacitors,
so that they soft-start their respective
channels without regard to external
voltage sources.
What happens to channels 2 and
3 if channel 1’s output is shorted?
Pulling the positive node of a TRACK
divider to zero doesn’t always produce
zero volts at the respective output; the
minute pull-up current in the TRACK
pins could create offsets in the voltage
dividers to which they are connected,
producing unwanted low output voltages or hiccupping on channels 2 and
3. But the LTC3773 uses a 30mV
offset in its tracking circuits, disabling
each channel’s driver until its TRACK
pin sees at least 30mV. This offset
disappears as the TRACK level rises
to 100mV, so that channels 2 and 3
can track predictably when they are
anywhere near their final values.
The LTC3773 also allows simultaneous ramping of output voltages
(coincident tracking). Just connect
TRACK2 and TRACK3 to resistor dividers of the same ratio as their respective
feedback networks and tie these
7
10.0k 1%
8
9
0.01µF
10
11
12
10.0k
1%
PGOOD
BOOST1
ITH1
SW1
SGND
SW2
TG2
LTC3773
ITH3
BOOST2
VFB2
BOOST3
VFB3
TG3
TRACK2
SW3
TRACK3
BG1
SENSE2–
BG2
SENSE2+
14
15
16
20.0k
1%
1000pF
+
1000pF
L1, L2, L3: TDK RLF7030T
10Ω
10Ω
17
18
10Ω
2.2µF
0.1µF
19
VIN
4.5V TO 14V
47µF
16V
+
10k
4.7µF
16V
TG1
ITH2
VCC
4.5V TO 6V
32
PHASEMD
SENSE1+
SENSE1–
SDB1
SDB2
33
VFB1
13
10Ω
34
BG3
6
TRACK1
PGOOD
CLKOUT
5
35
10Ω
PLLIN/FC
100pF
100pF
36
PLLFLTR
4
37
VCC
100pF
0.01µF
SDB3
PGND
1
3
20.0k 1%
38
SENSE3–
10.0k 1%
dividers between VOUT1 and ground.
Figure 4 shows the clean results. If
sequencing is required, the LTC3773’s
shutdown (SDB) pins offer the simplest
solution. Higher numbered channels
have higher enable thresholds at SDB,
so that applying a single rising voltage
ramp to all of them will cause them to
turn on consecutively. The configuration can be as simple as the SDB pins’
10Ω
39.2k 1%
1nF
Figure 7. Waveforms for the circuit of Figure 6
show excellent current sharing between
channels 2 and 3, and minimal output voltage
ripple at twice the switching frequency of each
channel.
1000pF
2
1nF 39.2k 1%
1µs/DIV
POWER DOWN VOUT3
0.01µF
1nF
SW2
10V/DIV
POWER DOWN VOUT1
POWER DOWN VOUT2
39
10.0k
1%
IL2, IL3,
5A/DIV
SENSE3+
31.6k
1%
internal current sources charging up
a single external capacitor with 1.5µA.
An externally controlled ramp can be
applied where needed (Figure 5).
Accurate current limiting is provided by monitoring the sense resistor
in series with the inductor. If a small
increase in efficiency is needed, the
LTC3773 can be configured for DCR
sensing across the inductor. The
controller protects against excessive
inrush current during start-up and
limits current through the inductor
and main MOSFET during short-circuits on the output. It pulls the output
down by turning on the synchronous
MOSFET whenever the feedback pin
VFB is 3.75% above the 0.6V reference voltage, protecting the output
capacitors and the load. It shuts off
whenever the bias supply VCC drops
below 3.94V, ensuring that the external MOSFETs operate at safe gate drive
levels. When the feedback voltage of
any channel is not within ±10% of the
0.600V internal reference for 100µs,
the open drain power good output
PGOOD pulls low.
VOUT2
2.5V
20mV/DIV
AC COUPLED
VDR
31
30
CMDSH-3
0.1µF
8 Si4816BDY
1
5
29
6, 7
L1
2.2µH
7mΩ
4
47µF
X5R
×2
28
27
26
2, 3
0.1µF
CMDSH-3
25
CMDSH-3
24
0.1µF
8 Si4816BDY
1
23
5
22
4
6, 7
VIN
4.7µF
16V
L2
2.2µH
7mΩ
2, 3
4.7µF
+
8 Si4816BDY
1
5
6, 7
4
10Ω
2, 3
VOUT2
1.8V/5A
47µF
X5R
×2
21
20
VOUT1
2.5V/5A
VIN
4.7µF
16V
L3
1.5µH
7mΩ
VOUT3
1.2V/5A
47µF
X5R
×2
Figure 8. Schematic for a 3-output, all ceramic COUT regulator
28
Linear Technology Magazine • March 2007
DESIGN FEATURES L
Compared to single phase switching regulators, 2-phase converters
impose lower ripple current on the
input capacitors, reducing their size
and cost. This technique interleaves
the current pulses coming from
the switches, greatly reducing the
amount of time when they overlap
and add together. Lower ripple current means less power dissipated and
higher efficiency, as well as reduced
electromagnetic interference. 2-phase
converters also double the effective
switching frequency, lowering the
output ripple voltage.
To fully realize these benefits, the
two channels should be operated 180°
out of phase. The LTC3773 allows
channels 2 and 3 to be operated out
of phase, a useful option when they
are tied together as a single, high
current output. Figure 6 shows the
schematic for such a converter. Channel 1’s output is 2.5V/15A and the
combined 2-phase channel’s output
is 1.8V/30A. The 2-phase channel
exhibits excellent current sharing, no
channel-to-channel interaction, and
minimal output ripple (at twice the
switch nodes’ operating frequency),
as Figure 7 demonstrates.
A Low Ripple, 3-Output
Supply with All Ceramic
Output Capacitors
Figure 8 shows the schematic for a
triple output supply which converts
12V to three 5A outputs: 2.5V, 1.8V,
and 1.2V. Each channel is stable with
just two 47µF ceramic capacitors at
its output, providing very low ripple
at moderate to heavy loads and the
fastest possible transient response.
With current-mode operation, the
converter responds quickly to input
voltage transients, correcting the
pulse width cycle-by-cycle as the input voltage swings widely (Figure 9).
Channel-to-channel interaction is
practically nonexistent during a substantial load step on one channel, as
Figure 10 shows.
With the PLLIN/FC tied to ground,
the LTC3773 enters Burst Mode
continued on page 35
Linear Technology Magazine • March 2007
3A
ILOAD1
2A/DIV
300mA
VIN
5V TO 12V
VOUT1
AC COUPLED
50mV/DIV
VOUT1
AC COUPLED
100mV/DIV
VOUT2
AC COUPLED
50mV/DIV
VOUT2
AC COUPLED
100mV/DIV
VOUT3
AC COUPLED
50mV/DIV
VOUT3
AC COUPLED
50mV/DIV
20µs/DIV
10µs/DIV
LINE TRANSIENT, VIN = 5V TO 12V
3A LOAD ON EACH CHANNEL
VIN = 12V
3A LOAD ON VOUT2, VOUT3
Figure 9. Line transient
for the circuit of Figure 8
Figure 10. Load transient for the
circuit of Figure 8, showing a negligible
coupling effect on the two other outputs
100
INDUCTOR
CURRENT
CHANNEL 3
5A/DIV
90
EFFICIENCY (%)
A Better Alternative
for Two Outputs
VSW3
20V/DIV
VOUT3
1V/DIV
80
70
60
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.2V
50
20µs/DIV
Figure 11. Short circuit for circuit of Figure 8.
The LTC3773’s typical maximum current
comparator threshold is 75mV/RSENSE,
resulting in a peak inductor current (or
maximum MOSFET current) of just over 10A
in this application. This limit protects the
components during the initial short circuit.
Subsequently, current foldback initiates cycleskipping, which limits short-circuit ripple
current even further.
(a)
40
10
100
1000
LOAD (mA)
10000
Figure 12. Efficiency curve
for the circuit of Figure 8
(b)
Figure 13. One of several evaluation boards available for the LTC3773, which is offered in 5mm
× 7mm QFN and small 36-lead SSOP packages. With (a) top and (b) bottom layers populated, the
core of this dual output, 15A application occupies just 17mm2.
29
L DESIGN IDEAS
Use an Ideal Diode to Combine Low
Voltage Supplies for High Current
A common method of delivering high
currents at low voltages to power
microprocessors is to combine the
outputs of several lower current
DC/DC converters. Although simple
in principle, the implementation can
be complicated. The outputs cannot
just be soldered together directly;
low voltage, high current converters
use synchronous topologies, so some
means of preventing back feeding from
one converter output to another is
necessary. Diodes come to mind, but
the losses, even with low forward voltage Schottky diodes, are prohibitive at
the necessary currents, suggesting an
active solution is necessary.
An unlikely candidate is the
LTC4354 negative voltage diode-OR
controller. Intended for –48V telecom
applications, this device functions
equally well in positive low voltage
applications. The circuit shown in
Figure 1 combines the outputs of
Design Ideas
Use an Ideal Diode to Combine Low
Voltage Supplies for High Current......30
Mitchell Lee
White LED Driver in
3mm × 2mm DFN Drives Ten LEDs......31
Molly Zhu
Shrink MP3 Players and
Digital Cameras with Two New
Dual Input USB/Wall-Adapter Linear
Li-Ion/Li-Polymer Battery Chargers.....32
Alfonso Centuori
Four Rails from One
Small Footprint Regulator.................34
Kevin Soch
Synchronous Buck Controller Regulates
from Input Voltages as Low as 2.2V
..........................................................36
David Ng
Dual High Speed Amplifier Doubles as
Differential 100Mbps Line Receiver
..........................................................37
Cheng Wei Pei and Mitchell Lee
1A Synchronous Boost Converters for
Portable Applications up to 7.5V........40
Eddy Wells
30
by Mitchell Lee
12V
470Ω
VCC
LTC4354
1µF
VEE
1.2V
100A
INPUT
GA,GB
DA,DB
HAT2165 ×6
240Ω*
12V
470Ω
1.2V, 200A
OUTPUT BUS
VCC
LTC4354
1µF
VEE
1.2V
100A
INPUT
240Ω*
GA,GB
DA,DB
HAT2165 ×6
*OPTIONAL PRELOAD
Figure 1. Positive low voltage diode-OR combines multiple switching converters.
multiple high current switching converters, without concern about back
feeding or supply failure shorting out
the common bus. Each diode “channel” comprises the LTC4354 and six
parallel MOSFETs, supplying 100A
to a 1.2V load. The circuit is easily
adapted to any supply voltage between
0V and 5V, provided there is a path
for up to 4mA VEE current to ground
at either the input or the output. Most
high current switching converters
can easily sink 4mA and no preload
is necessary. No circuit changes are
necessary for operation over a range
of nearly zero to 5V.
The circuit features two notable
improvements beyond serving as an
ideal diode. First, the forward drop
across the MOSFET is regulated at a
low level, about 30mV. By regulating
the forward drop, any tendency to
oscillate—a problem associated with
hysteretic systems—is eliminated,
without compromising forward losses.
Second, the LTC4354 has two levels of
turn-off when blocking reverse current
flow. The LTC4354 responds slowly to
small reverse overdrives, and responds
quickly whenever the reverse voltage
exceeds 120mV. Thus, the circuit accommodates a moderate amount of
reverse current on a transient basis,
preserving the dynamic performance
of synchronous DC/DC converters,
which sink current to achieve good
load step regulation.
Dissipation in the ORing MOSFETs
is about 500mW each at 100A, exclusive of distribution losses. While
two power stages are combined in
Figure 1, the technique is extensible
to three or more stages with no circuit
changes. L
Authors can be contacted
at (408) 432-1900
Linear Technology Magazine • March 2007
DESIGN IDEAS L
White LED Driver in 3mm × 2mm DFN
by Molly Zhu
Drives Ten LEDs
Introduction
SHUTDOWN AND
DIMMING CONTROL
The LT3591 is a step-up white LED
driver that drives up to ten LEDs
from a single Li-Ion battery. Its high
level of integration minimizes solution
size, board space and the number of
external components—a complete LED
driver requires only 30mm2. The high
side sense feature of the LT3591 allows for a 1-wire current source, and
makes the LT3591 applicable in buck
and buck-boost circuits as well.
CTRL
VIN
3V TO 5V
VIN
CAP
L1
22µH
RSENSE
10Ω
LT3591
SW
LED
C2
2.2µF
GND
C1
1µF
Driving up to Ten LEDs
High Side Sense
The high side sense of LT3591 brings
unique benefits for a wide range of applications. First, it allows for a 1-wire
current source, meaning that the end
of the string can be connected directly
to ground instead of returning to the
driver, as is required with many LED
drivers. This simplifies the wiring design when the driver and the LEDs are
on separate boards, which is often the
case in cell phones, or the LEDs and
driver are some distance apart.
continued on page 41
Linear Technology Magazine • March 2007
C1: TAIYO YUDEN EMK107BJ105MA
C2: MURATA GRM31CR71H225KA88
L1: TAIYO YUDEN NR4018T220M
Figure 1. Li-Ion powered driver for ten white LEDs
C2
10µF
PVIN
12V
100
RSENSE
0.545Ω
C3
1µF
90
D1, 350mA
VIN
3V
SHUTDOWN
AND DIMMING
CONTROL
CAP
C1
1µF
LED
L1
22µH
VIN
LT3591
CTRL
SW
GND
70
50
L1
22µH
0
50
100
150
VIN
LT3591
CAP
0V
250
300
350
RSENSE
10Ω
LED
C2
2.2µF
CTRL
5V
200
Figure 3. Efficiency of the circuit in Figure 2
SW
GND
VIN=12V
2 LEDs
LOAD CURRENT (mA)
Figure 2. A buck converter drives
two LEDs at 350mA from a 12V input.
C1
1µF
80
60
C1: TAIYO YUDEN EMK107BJ105MA
C2: TAIYO YUDEN EMK316BJ106ML
C3: TAIYO YUDEN EMK212BJ105MG
L1: TAIYO YUDEN NR4018T220M
RSENSE: 1Ω||1.2Ω
VIN
3V TO
5V
EFFICIENCY (%)
The LT3591 can drive up to ten white
LEDs, as long as the maximum switch
current is below 500mA. Figure 1
shows a typical application circuit
driving ten white LEDs from a single
Li-Ion battery. The LEDs are connected
in series, which results in accurate
LED current matching regardless of
variations in their forward voltages
— no additional circuitry required.
The power switch, Schottky diode,
compensation components and openLED protection are all integrated into
the LT3591’s tiny 3mm × 2mm DFN
package. Only four external components are used in Figure 1. This
minimizes the design effort, solution
cost and board space. The fixed 1MHz
switching frequency allows the use of
tiny inductors and capacitors, while
still keeping efficiency high. All these
make the LT3591 ideal for portable
applications.
PWM
FREQ
Q1
Si2308DS
100k
C1: TAIYO YUDEN EMK107BJ105MA
C2: MURATA GRM31CR71H225KA88
L1: TAIYO YUDEN NR4018T220M
Figure 4. Li-Ion to ten LEDs with direct PWM dimming
31
L DESIGN IDEAS
Shrink MP3 Players and
Digital Cameras with Two New
Dual-Input USB/Wall-Adapter Linear
Li-Ion/Li-Polymer Battery Chargers
by Alfonso Centuori
Introduction
USB or wall adapter? Many of today’s
digital cameras, PDAs, mobile phones
and MP3 players can charge their
batteries from either source, requiring a flexible charging circuit. The
LTC4096 and LTC4097 are specifically
designed to charge single cell lithiumion/lithium-polymer batteries from a
wall adapter or another input such as
a USB port, all in just 48mm² of board
space (LTC4097).
Using a constant current/constant
voltage algorithm, these chargers can
deliver up to 1.2A of charge current
(programmable) from the wall adapter
and up to 1A of charge current (programmable) from the other power input
(usually USB), with a final float volt-
1.2A (WALL)
476mA (USB)
LTC4096
WALL
ADAPTER
USB
PORT
1µF
age accuracy of ±0.6%. The LTC4096
and LTC4097 each include two
internal P-channel power MOSFETs
and thermal regulation circuitry with
no blocking diode or external sense
resistor required—a basic charger
circuit requires only three external
components.
Both parts simplify status reporting,
start-up, charging and shutdown with
simple hookup. The CHRG open-drain
status pin indicates the battery charge
state. The PWR open-drain status
pin of the LTC4096 (VNTC pin of the
LTC4097) reports when at least one
of the inputs has sufficient voltage to
charge a battery. The PWR pin can
source up to 120mA of current to
1µF
ON OFF
DCIN
BAT
USBIN
PWR
IDC
RIDC
845Ω
1%
4.2V
Li-Ion
CELL
CHRG
IUSB
RIUSB
2.1k
1%
+
1k
SUSP
ITERM
GND
RITERM
2k
1%
Figure 1. Just a few components are needed to create feature-laden, LTC4096-based,
dual input USB/wall-adapter Li-Ion or Li-Polymer battery charger.
WALL
ADAPTER
USB
PORT
1.2A (WALL)
95mA/476mA (USB)
LTC4097
DCIN
1µF
VNTC
HPWR
RNTCBIAS
100k
1k
NTC
ON OFF
RIUSB
2.1k
1%
RIDC
845Ω
1%
IDC
CHRG
ITERM
GND
+
RITERM
2k
1%
RNTC
100k
4.2V
Li-Ion
CELL
Figure 2. This LTC4097 USB/wall-adapter battery charger is similar to the charger in Figure 1,
but adds NTC battery-temperature monitoring safety and the ability to change USB power modes
on the fly.
32
The LTC4096 and LTC4097 provide a
great deal of design flexibility, including programmable charge current and
programmable current termination.
The charge currents are programmed
using a resistor from the IDC and IUSB
pins to ground as follows:
ICHG =
1000 V
(Wall Adapter Present)
RIDC
ICHG =
1000 V
RIUSB
(USB, HPWR = High for LTC4097)
SUSP
IUSB
Programmability
BAT
USBIN
1µF
power up a microprocessor or other
general circuitry, solving potential
start-up problems. The LTC4097 has
an additional safety feature in that it
can qualify charging based on battery
temperature via the NTC input. The
ITERM pin provides a means to implement programmable current based
termination schemes.
Internal thermal feedback regulates the charge current to maintain
a constant die temperature during
high power operation or high ambient
temperature conditions.
Both devices can be shut down to
reduce the drain on any of the input
sources. In shutdown, the DCIN (wall
adapter) supply current reduces to
20µA, the USBIN supply reduces to
10µA and the battery drain to less
than 2µA.
Both the LTC4096 and LTC4097
terminate the charge cycle based
on the battery current. The current
detection threshold, IITERM, is set by
connecting a resistor, RITERM, from
ITERM to ground. The following
Linear Technology Magazine • March 2007
DESIGN IDEAS L
Table 1. Power source selection
VUSBIN > 4.2V and
VUSBIN > BAT + 30mV
VUSBIN < 4.2V or
VUSBIN < BAT + 30mV
VDCIN > 4.2V and
VDCIN > BAT + 30mV
Charger powered from wall adapter source
VPWR = VDCIN – RDC-PWR • IPWR
USBIN current < 25µA
Charger powered from wall adapter source
VPWR = VDCIN – RDC-PWR • IPWR
VDCIN < 4.2V or
VDCIN < BAT + 30mV
Charger powered from USB source
VPWR = VUSBIN – RUSB-PWR • IPWR
No charging
PWR: Hi-Z
formula programs the termination
current:
ITERM =
100 V
RITERM
The condition of the CHRG pin
indicates the charge state. A strong
pull-down on the CHRG pin indicates
that the battery is charging. When
the current termination threshold is
reached, the CHRG pin assumes a
high impedance state and the charge
cycle is terminated.
USB Compatibility
Both chargers are USB compatible.
Figures 1 and 2 show USB compatible setups for the LTC4096 and
LTC4097, respectively. In both cases,
wall adapter input takes priority over
USB input, with the maximum charge
current set to 1.2A by the 845Ω IDC
resistor. When a wall adapter is not
present and USB power is available,
the devices draw current from USBIN.
The 2.1k resistor at the IUSB pin sets
the USB charge current to 476mA, well
within the limits of the high power USB
specification.
The LTC4097 has an additional pin,
HPWR, that allows mode selection of
high power (≤500mA) or low power
(≤100mA) USB charging on the fly. A
logic high on the HPWR pin sets the
charge current to 100% (476mA) of the
current programmed by the IUSB pin
resistor, while a logic low on the HPWR
DCIN
USBIN
1
2
+
4.2V
+
–
–
DCIN UVLO
ΔV
+–
DCON
USBON
4V
USBIN UVLO
+
+
–
–
BAT
10
–
BAT
10
DCON
–+
ΔV
–
CURR-LIM
CURR-LIM
+
+
3 120mA MAX
PWR (LTC4096) OR VNTC (LTC4097)
Figure 3. Simplified schematic shows the output of the power present output (PWR)
pin (VNTC on LTC4097) as it relates to the DCIN and USBIN inputs.
Linear Technology Magazine • March 2007
pin sets the charge current to 20%
(95mA) of the current programmed
by the IUSB pin resistor. If the HPWR
pin is not driven externally, a weak
pull down on the HPWR pin defaults
to the low power state. The HPWR pin
provides a simple control for managing
charge current as shown in Figure 2
with HPWR in its high state and 95mA
with HPWR in its low state for LTC4097
(or just 476mA for LTC4096).
Avoiding Unnecessary
Charge Cycles
LTC4096 and LTC4097 are designed
to avoid unnecessary charge cycles to
extend the life of Li-Ion or Li-Polymers
batteries. When power is first applied or
when exiting shutdown, the LTC4096
and LTC4097 check the voltage on
the BAT pin to determine its initial
state. If the BAT pin voltage is below
the recharge threshold of 4.1V (4.15V
for LTC4096), which corresponds to
approximately 80%–90% battery capacity, LTC4096 and LTC4097 enter
charge mode and begin a charge cycle.
If the BAT pin is above 4.1V (4.15V for
LTC4096), the battery is nearly full and
the charger does not initiate a charge
cycle and enters standby mode. When
in standby mode, the chargers continuously monitor the BAT pin voltage.
When the BAT pin voltage drops below
the recharge threshold, the charge
cycle is automatically restarted. This
feature eliminates the need for periodic
charge cycle initiations, ensures that
the battery is always fully charged and
reduces the number of unnecessary
charge cycles, thereby prolonging
battery life.
PWR/VNTC Functionality
Both parts provide a power supply
status output pin (PWR in the LTC4096
continued on page 35
33
L DESIGN IDEAS
Four Rails from One
Small Footprint Regulator
by Kevin Soch
Introduction
Easy Board Layout
The quad output LTC3544B is a
monolithic buck regulator capable of
simultaneously providing four independent voltage supply rails at over
90% efficiency. The four outputs are
rated at maximum output currents of
300mA, 200mA, 200mA, and 100mA.
At light load currents, pulse skipping
operation maintains both high efficiency and low output voltage ripple.
The input voltage can range from 2.25V
to 5.5V and the output voltage levels
are independently programmable from
0.8V to VIN. Space saving features of
the LTC3544B include a 3mm × 3mm,
16-pin QFN package and a fixed,
2.25MHz switching frequency, which
allows the use of a minimum num-
Particular attention was paid to the
placement of the package pins to
ensure a logical and compact board
layout, particularly with respect to
the power paths. Figure 1 is a photo
of the LTC3544B demo board with
the power components primarily on
the top. The feedback elements (not
shown) reside on the bottom of the
board. Total circuit footprint for this
board is approximately 225mm2.
Figure 1. The LTC3544B is designed to
facilitate simple and compact board layout
ber of small, surface mount external
components.
High Level of Integration
The LTC3544B provides a simple,
extremely compact solution for applications requiring multiple voltage
supply rails. Many of the components
typically required to operate switching regulators have been integrated
into the LTC3544B. Internal loop
compensation eliminates the need
for external compensation resistors
and capacitors. Integrated synchronous switches eliminate the need for
external Schottky diodes. An integrated soft-start function eliminates
the need for external capacitors or
control ramps.
VOUT100
10mV/DIV
VOUT200A
10mV/DIV
VOUT200B
10mV/DIV
VOUT300
100mV/DIV
VIN = 3.6V
40µs/DIV
TA = 25 C
300mA LOAD STEP ON VOUT300
OTHER CHANNELS LOADED 50% OF MAXIMUM
Figure 2. Channel to channel load transient
crosstalk is negligible.
VSUPPLY
3.6V
C9
4.7µF
L2
4.7µH
VOUT2
1.5V
C2
4.7µF
R3
93.1k
C6
20pF
4
1
R4
107k
VOUT3
0.8V
3
5
2
R6
100k
RUN200B
A potential problem with multiple
output regulators is the interaction
between channels when one of the
channels undergoes a load transient.
Figure 2 shows the response on the
VOUT100
VOUT200A
VOUT200B
VOUT300
RUNx
VIN = 3.6V
200µs/DIV
TA = 25°C
ALL CHANNELS UNLOADED
Figure 3. Integrated soft-start limits inrush
current and prevents voltage overshoot.
C10
4.7µF
15
VCC
7
PVIN
RUN100
SW200B
SW100
VFB200B
VFB100
12
13
L1
10µH
C5
20pF
11
RUN200A
RUN300
SW200A
SW300
VFB200A
GNDA
14
PGND
6
VFB300
R1
59k
R2
118k
LTC3544B
L3
4.7µH
C3
4.7µF
16
Minimal Channel Cross-Talk
9
8
10
L4
4.7µH
C8
20pF
R7
162k
R8
76.8k
VOUT1
1.2V
C1
4.7µF
VOUT2
2.5V
C4
4.7µF
Figure 4. Minimal external components are required to create four separate voltage rails.
34
Linear Technology Magazine • March 2007
DESIGN IDEAS L
100
VOUT = 2.5V
90 TA = 25°C
VOUT300
100mV/DIV
AC COUPLED
EFFICIENCY (%)
80
70
IL
250mA/DIV
60
50
40
ILOAD
250mA/DIV
30
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
Figure 5. Pulse skipping assures both
low ripple and high peak efficiency.
100mA and both 200mA channels to a
0mA to 300mA load step on the 300mA
channel. Here, the first three channels
are each loaded at 50% of their rated
load. In each case, the crosstalk is on
the order of 1mV to 2mV.
Soft-Start Prevents
Inrush Currents
VIN = 3.6V
20µs/DIV
VOUT = 2.5V
TA = 25°C
LOAD STEP = 300µA TO 300mA
Figure 6. Current mode architecture assures
fast recovery from load transients.
the inrush currents associated with
rapid charging of the output filter caps
during start-up. Figure 3 shows the
output of each of the four channels
during start-up. In addition to limiting
the inrush currents, soft-start also prevents output voltage overshoot when
starting up under light loads.
High Overall Efficiency
The LTC3544B includes an integrated
soft-start function. By ramping up
the output voltage over a period of
approximately 1ms, soft-start reduces
Figure 4 shows a typical use of the
LTC3544B to generate 100mA at
1.2V, 200mA at both 0.8V and 1.5V,
and 300mA at 2.5V. Efficiency versus
LTC3773, continued from page 29
allows inductor current to reverse, regardless of load. Discontinuous mode
(PLLIN/FC floating) does not allow
inductor current to reverse, and does
not set a minimum peak inductor current, resulting in constant frequency
operation at light loads.
operation at light loads for high efficiency across a wide range of loads
(Figure 12). Burst Mode operation
clamps minimum inductor current
peaks to 20% of the maximum programmed current limit, and does not
allow inductor current to reverse.
The pin can also be tied to VCC for
continuous conduction mode, which
operates at a constant frequency and
LTC4096/97, continued from page 33
and VNTC in the LTC4097), which
indicates when sufficient input power
is available to the charger. Given its
120mA drive capability, it can be
used for many applications where
immediate input power is necessary
to start-up a microprocessor or some
other circuitry. Its output voltage is
equal to DCIN or USBIN if either input
source is present (above undervoltage
lockout thresholds). If both DCIN and
USBIN are present, the voltage at PWR
Linear Technology Magazine • March 2007
Conclusion
Rather than providing a single output
like other 3-phase buck controllers,
and VNTC equal the DCIN voltage
(even if USBIN voltage is higher). If
neither DCIN nor USBIN is valid (below
undervoltage lockout thresholds), then
PWR and VNTC assume a high impedance state. Table 1 shows how these
output pins work and Figure 3 shows
their basic electrical schematics.
Conclusion
The LTC4096 and LTC4097 are complete linear Li-Ion or Li-Polymer battery
chargers compatible with portable
load current and supply, and transient
response for the 300mA channel are
shown in Figures 5 and 6. In addition
to the obvious board real-estate savings, a 4-output monolithic regulator
has another important advantage over
four individual regulators. Overall
efficiency is improved because internal overhead circuitry such as the
oscillator and voltage reference are
shared between the four regulators,
thereby minimizing the power loss
per regulator.
Conclusion
The LTC3544 is a unique part with
tremendous flexibility. It greatly
simplifies system and board design
where multiple voltage supply rails
are needed without sacrificing the
features and performance found in
individual regulators. The LTC3544B
is ideally suited for battery powered
applications where multiple or isolated voltage rails are required, board
space is at a premium and a minimum
number of external components are
desired. L
the LTC3773 regulates one, two or
three outputs ranging from 0.6V to
5V. They may be tracked, sequenced,
or allowed to start up and shut down
independently. With an array of protection features, adjustable frequency,
and three modes of switching operation, the LTC3773 is the most versatile
3-phase DC/DC controller on the
market. L
USB applications. They are designed
to accommodate charging from a wall
adapter or a USB input. Their versatility, low quiescent current, simplicity,
high level of integration and small
size make them an easy fit in many
portable USB applications. LTC4096 is
available in a small, 10-lead low profile
3mm × 3mm DFN package. A version
without trickle charge is also available (the LTC4096X). The LTC4097 is
available in a small 12-lead low profile
3mm × 2mm DFN package. L
35
L DESIGN IDEAS
Synchronous Buck Controller
Regulates from Input Voltages
as Low as 2.2V
Introduction
Low voltage power supplies require
special design considerations, especially if a low voltage bus must support
more than a few amps. The difficulty
usually arises in balancing electronic
efficiency and volumetric efficiency.
Suppose, for example, that a system
requires several amps of 1.8V power,
and that both 3.3V and 12V are locally
available as input buses. Assuming
that sufficient current is available from
both, there are compelling reasons
to choose the lower voltage bus for
the input power source: it allows a
small valued inductor, yields reduced
magnetic core loss and less switching
voltage stress, benefiting both the
design footprint and efficiency.
Unfortunately, many control ICs
available for this type of application
derive their MOSFET drive power from
the input voltage. This means that, if
3.3V is chosen for the input bus, those
ICs are only able to provide about 2.5V
to the MOSFET gates. The designer
is either required to add circuitry
to generate a higher voltage for the
MOSFET driver or select a MOSFET
that operates reliably and efficiently
with a gate voltage of only 2.5V.
The LT3740 is a valley mode, No
RSENSE™ synchronous buck controller
that can accept input voltages as low
93
92
EFFICIENCY (%)
91
90
89
88
87
86
VIN = 3.3V
0
2
4
6
LOAD CURRENT (A)
8
10
Figure 2. Efficiency for the circuit in Figure 1
36
47µH
VIN
SHDN
10µF
BIAS
M1
HAT2172H
TGATE
0.9µH
SW
15k
XREF
SN+
1Ω
BGATE
0.22µF
15k
VIN
3V to 12V
1µF
LT3740
SWB
BGDP
2.2nF
M2
HAT2165H
VOUT
1.8V
10A
D1
B320A
1k
100µF
×3
39pF
VC
15k
by David Ng
SN–
105k
PGND
RANGE GND FB
80.6k
Figure 1. Synchronous buck converter produces 1.8V
at 10A from a 3V–12V input. All capacitors are ceramic.
as 2.2V, and as high as 25V, but still
provides at least 7V to the MOSFET
gate drives through an internal boost
regulator. It features No RSENSE operation to maximize efficiency, three user
selectable current limit ranges and a
flexible soft-start system capable of
tracking an external command voltage.
The LT3740 is available in a space
saving 5mm × 3mm DFN package.
1.8V Buck Converter in 1.5in2
Figure 1 shows an example of a LT3740
synchronous buck converter. The
design produces 1.8V at 10A from an
input range of 3V to 12V. All of the
capacitors are ceramic. The internal
boost converter, along with L1 and
C1, regulates the voltage at the BIAS
pin to 7V above the input voltage.
The voltage at the BIAS pin is used
internally to power the top MOSFET
driver. The bottom gate drive power
pin, BGDP, is connected to BIAS, so
it is also 7V above the input voltage.
The IC internally limits the amplitude
of both the top and bottom gate driver
output voltages to about 6V, which
is high enough to ensure that the
MOSFET is fully on, but low enough
to keep switching losses down.
The LT3740’s switching frequency
optimizes both electrical and volumetric efficiency. The IC’s 300kHz
operation is high enough to allow
the user to choose physically small
power inductors and capacitors, but
is also low enough to keep switching
losses to a minimum. The regulator
in Figure 1 takes up less than 1.5in2
of board space.
The design takes advantage of the
LT3740’s ability to read the switch
current through the bottom MOSFET
on-resistance. The chip features three
user selectable current limit thresholds to optimize efficiency: 50mV,
80mV and 110mV. For an output voltage as low as 1.8V, the voltage given
up to resistively sense the current
can have an appreciable impact on
the converter efficiency, so the lowest
current threshold is used by simply
grounding the RANGE pin. For higher
input and output voltages, the higher
current limit settings may be used for
improved signal to noise ratio. The efficiency of the LT3740 design, plotted
against output load in Figure 2, peaks
at 92% at half load and is still above
90% at full load.
continued on page 39
Linear Technology Magazine • March 2007
DESIGN IDEAS L
Dual High Speed Amplifier Doubles
as Differential 100Mbps Line Receiver
by Cheng Wei Pei and Mitchell Lee
Introduction
Increasing the data rate on unshielded
twisted-pair cable such as Category 5
(Cat 5) cable increases the demands
on line drivers and receivers, including
the need for short propagation delays,
clean transient response and blazing
fast rise/fall times. For line receivers,
the challenge is reproducing the original signals with good fidelity despite
signal losses over hundreds of feet of
cable. The receivers must be able to
resolve the attenuated digital signals
and convert them into full-scale logic
levels. When nothing else will do the
job, high speed dual amplifiers can
act as digital data receivers for high
data rate applications. Translated into
traditional amplifier terminology, high
speed data receivers require amplifiers
with high slew rate, large bandwidth,
fast output overdrive recovery and a
clean transient step response. The
LT6411 dual amplifier meets all of
these requirements while maintaining
low power consumption (80mW with a
single 5V supply) and small size (3mm
× 3mm × 1mm 16-pin QFN package).
In addition, the LT6411 has flexibility
of selecting different gains for different
applications with a minimal number
of additional components.
Figure 1 shows the internal block
diagram of the LT6411. It is a high
speed dual amplifier with built-in gain
5V
VCC
4
LT6411
+
–
370Ω
370Ω
370Ω
2
OUTPUT (V)
370Ω
–
+
DGND
0
–1
–3
EN
–4
Figure 1. Block diagram of the LT6411. Gain
resistors are included internally, which means
minimal external components are necessary
for operation.
0
2
4
6
8 10 12 14 16 18 20
TIME (ns)
Figure 2. Large-signal transient response
of the LT6411. Slew rate is shown to be
approximately 3000V/µs, and overshoot/
ringing is minimal. Settling time is only 4ns.
5V
LT6411 VCC
+
–
5V
50 FEET CAT-5 CABLE
INPUT
370Ω
+
370Ω
137Ω
100Ω
LTC1688
OUT
– 370Ω
370Ω
–
+
EN
DGND
VEE
Figure 3. The LTC1688 100Mbps differential driver drives 50 feet of Category 5E (Enhanced)
twisted pair cable, and the LT6411 receives and buffers the signal. The LT6411’s gain of 3
configuration restores the signal amplitude after cable attenuation.
4.5
4
4
3.5
3.5
3
3
2.5
2.5
VOLTS
VOLTS
1
–2
VEE
4.5
2
1.5
2
1.5
1
1
0.5
0.5
0
–30
VIN = 2.5VP-P
AV = 2
VS = ±5V
RL = 150Ω
TA = 25°C
3
–20
–10
0
NANOSECONDS
(a)
10
20
30
0
–30
–20
–10
0
NANOSECONDS
10
20
30
(b)
Figure 4. Figure 4a shows the attenuated 100Mbps input to the LT6411 (after 50 feet of cable), and figure 4b shows the LT6411 output. Input-tooutput propagation delay is less than 4ns, which includes the LT6411’s output overdrive recovery time.
Linear Technology Magazine • March 2007
37
L DESIGN IDEAS
VS
5V
LT6411 VCC
5V
300 FEET CAT-5 CABLE 49.9Ω
INPUT
499Ω
370Ω
+
–
+
370Ω
100Ω
LTC1688
+
–
OUT
49.9Ω
499Ω
370Ω
EN
–
+
DGND
370Ω
–
AV = 1
VEE
Figure 5. The LT6411’s simple architecture allows higher gain settings when necessary. This
circuit shows a gain of 10 configuration to recover the signal after the of 300 feet of Category 5
cable (non-Enhanced).
GND
(INPUT)
GND
(OUTPUT)
5ns/DIV
Figure 6. The top trace shows the LT6411’s input, attenuated by the 300 feet of cable. The
bottom trace shows the output of the LT6411. Propagation delay from input to output with
a gain of 10 is approximately 4.5ns.
resistors and a power-saving enable
feature. The high 600MHz bandwidth
and >3000V/µs slew rate allows the
part to track and buffer signals with
high fidelity, and the built-in resistors
reduce component count, simplifying
high speed board layout. Figure 2
shows the transient step response of
the LT6411 to a 2.5V, fast-rise-time
pulse with a minimal amount of overshoot and ringing. In data receiver
terms, that means minimal eye closure
and inter-symbol interference at high
data rates.
Data Receiver
When receiving data from cables, gain
is often useful to compensate for cable
losses. The higher the data rate (i.e.
transmission frequency), the more
loss from the transmission medium.
Figure 3 shows an LTC1688 100Mbps
differential data driver followed by 50
feet of Category 5E (Enhanced) cable.
The LT6411 is shown with a gain
38
–
+
–
+
–
+
–
+
INPUT
1V/DIV
OUTPUT
1V/DIV
of 3V/V.The shunt 137Ω resistor is
selected to match the input impedance of the LT6411 stage to the 100Ω
impedance of the twisted pair cable.
Figure 4 (top) shows the differential
signal at the input of the LT6411,
after attenuation from the cable. The
bottom trace in Figure 4 is the LT6411
output. The propagation delay of the
LT6411 is under 4ns and the output
eye diagram shows a well-behaved
transient response with fast edges.
The fast, clean output saturation
recovery of the LT6411 enables high
fidelity digital signal recovery, giving it
a logic-level output without the minimum input signal level limitations of
logic devices.
Receiving Longer Cables
When the cable length gets longer,
more gain may be necessary to recreate the original input data signal.
Figure 5 shows the LT6411 with a
gain of 10V/V. Because the gain and
+
AV = 2 (AC)
AV = 1 (DC)
–
RTERM
Figure 7. In lower-gain configurations
(1–2V/V), the LT6411 can be set up with a
high impedance input, for multi-drop (multiple
receivers along the line) applications. For a
gain of 2, the gain-setting pins should be tied
to an AC ground (as shown). If these pins are
tied to DC ground, there will be a DC gain of 2,
and the amplifier’s output may saturate at the
top rail.
feedback resistors of the LT6411 are
internal, a cross-coupled configuration is used for higher gains. One
benefit of this configuration is that
the value of the gain resistors (in this
case, 49.9Ω) can be selected to match
the impedance of the cable. Figure 6
shows the input and output of the
LT6411 in this configuration.
Multiple Receiver Applications
In cases where the transmission cable
is short and high gain is unnecessary,
the LT6411 can also be used as a high
input impedance receiver. One such
application is where multiple receivers and/or transmitters are spread
along a transmission line and the line
is terminated at each end. Figure 7
shows two methods of configuring the
LT6411 in a high input impedance
configuration, useful for monitoring
data on the twisted-pair without improperly loading it.
Linear Technology Magazine • March 2007
DESIGN IDEAS L
Single-Ended Output
The LT6411 produces a differential
output, but if a single-ended logic
output is needed, there are multiple
options for data conversion. One such
way is shown in Figure 8, in which
the MC10H350 PECL-TTL translator
performs the conversion. To translate
OPTION
5V
+
OVDD
LT1715
–
5V
700mVPP
(DIFFERENTIAL)
MINIMUM
5V
(INPUT
CIRCUITRY
OMITTED)
Conclusion
OUT
OGND
R3
200Ω
R4
200Ω
R1
200Ω
5V
LT6411
TTL OUTPUT
R2
200Ω
1/4 MC10H350
PECL-TTL TRANSLATOR
PECL LEVELS
Figure 8. If a single-ended output is needed, there are many options available for translators.
One example is ON Semiconductor’s MC10H350 PECL-TTL translator. The 200Ω resistors shift
the output of the LT6411 up to PECL voltage levels. Alternatively, a level-translating comparator
such as the LT1715 could be used to give a variety of logic output levels.
LT3740, continued from page 36
The LT3740 uses a valley mode current control system that boasts a fast
response to load changes. As shown
in Figure 3, this design responds to
0A–10A step load change in 10µs,
yielding a voltage transient of less
than 50mV.
the voltage levels from the LT6411 to
PECL input voltage levels, two resistive dividers level-shift and attenuate
the output signal of the LT6411. Alternatively, a high speed comparator
such as Linear Technology’s LT1715
can also perform this task without the
level-shifting resistors.
The LT6411 is a dual high speed amplifier with flexible features and superb
AC characteristics, making it suitable
for use as a high data rate receiver.
The ability to select different gain
configurations with minimal external
components makes the LT6411 easy
to use. Its small footprint and low
power consumption allow it to fit into
almost any application without painful
compromises, especially for portable
or peripheral applications where space
and power are at a premium. L
Conclusion
The LT3740 is a synchronous buck
controller that boasts a rich feature set
which allows the designer to optimize
power and volumetric efficiency by exploiting the advantages of a low input
voltage. Through a combination of its
onboard boost regulator, user programmable current limit thresholds,
fast transient response and flexible
soft-start system, the designer can
produce a small, efficient, full featured
converter. L
Soft-Start
The LT3740 is also equipped with a
flexible soft-start design that allows for
either ramped current or tracking. If
the XREF pin is held above 1V, and an
RC timer is applied to the SHDN pin,
the converter soft-starts by ramping
the current available to the load. If the
SHDN pin is high, enabling the chip,
and a 0V to 0.8V tracking signal is
applied to the XREF pin, the internal
reference of the LT3740 follows the
tracking signal.
LTC3706/26, continued from page 13
used to provide a gradual increase
in peak current during the soft-start
interval. The circuit of Figure 7 also
includes an optional falling-edge delay
circuit on the gate of synchronous
switch Q4. This delay has been used
to optimize the dead time for this
specific application, thereby improving
Linear Technology Magazine • March 2007
VOUT
50mV/DIV
INDUCTOR
CURRENT
5A/DIV
20µs/DIV
Figure 3. Output voltage and inductor current response to a
0A–10A step load transient applied to the circuit in Figure 1
the efficiency by about 1%. Figure 8
shows the transient response that is
achieved using the circuit of Figure 7,
and Figure 9 shows the efficiency at
VIN = 12V and VIN = 24V.
Conclusion
The new LTC3706/26 controller and
LTC3705/25 driver bring an un-
precedented level of simplicity and
performance to the design of isolated
power supplies. Each controller-driver
pair works in concert to offer high
efficiency, low cost solutions using
off-the-shelf components. The devices
are versatile and easy to use, covering
a broad range of forward converter
applications. L
39
L DESIGN IDEAS
1A Synchronous Boost Converters for
Portable Applications up to 7.5V
by Eddy Wells
Introduction
40
high efficiency over many decades.
The LTC3458 is optimized for higher
voltage applications with a reduced
maximum load current.
95
90
EFFICIENCY (%)
Today’s battery powered devices
require efficient and compact power
conversion solutions with minimal
design effort. The LTC3458 and
LTC3458L are full-featured, step-up
DC/DC converters intended for applications with load currents up to
1A. Their 1.5V–6V input voltage range
is well suited for multi-cell alkaline,
Li-Ion, or USB power. Despite their
small size, both parts are extremely
versatile, with programmable output
voltage, current limit, switching frequency, burst threshold and soft-start
period. Other features include short
circuit protection, controlled inrush
current and true output disconnect
in shutdown.
Both regulators use current mode
control, which provides fast transient
response to both line and load steps.
In fixed frequency operation, the
oscillator can be synchronized to an
external source or set between 400kHz
and 1.5MHz, allowing the designer
to optimize component size and efficiency. Burst Mode operation at light
loads extends battery life, with typical
quiescent currents of just 12µA. The
threshold between fixed frequency and
Burst Mode operation is adjustable
with an external resistor. Both parts
are offered in a thermally enhanced
12-pin DFN (4mm × 3mm) package.
The LTC3458 and LTC3458L have
identical functions and pin-out, but
differ in how they trade-off switch
RDS(ON) with maximum VOUT rating.
The LTC3458L has a 0.2Ω (typical)
N-Channel MOSFET and a 0.3Ω PChannel MOSFET with a maximum
VOUT rating of 6.0V. The LTC3458
has higher switch resistance (0.3Ω
N-Channel, 0.4Ω P-Channel), but VOUT
can be programmed as high as 7.5V. An
efficiency comparison of the two parts
at 3.6VIN, 5.0VOUT at 1MHz is shown in
Figure 1. As shown in the figure, the
LTC3458L is more efficient (especially
at load currents above 250mA) and
5V from a Li-Ion/USB Input
85
Devices requiring a regulated 5V output from a Li-Ion battery or powered
USB port must contend with an input
voltage range between 3.1V and 5.25V.
Although the LTC3458 and LTC3458L
are primarily step-up converters,
their output-disconnect architecture
provides regulation when the output
is below the input voltage. Since VIN
is near the regulated 5V output when
powered from the USB, converter efficiency is still acceptable even when
stepping down.
80
LTC3458L
LTC3458
75
0.1
1
10
100
LOAD CURRENT (mA)
1000
Figure 1. LTC3458 and LTC3458L
5V efficiency vs load current
able to deliver more current to the load.
Between 0.1mA and 100mA (threshold programmable), both parts enter
into Burst Mode operation, providing
10µH
USB = 5V
Li-Ion = 3.5V
LTC4066
USB
4.35V TO
5.25V
BAT
SW
VIN
IN
LTC3458L
VOUT
VOUT
100µF
GND/PGND
Li-Ion
3.1V TO 4.2V
1.0M
SHDN
+
SYNC
FB
COMP
0.22µF
RT
SS
0.01µF
ILIM
287k
124k
VOUT
5V
500mA
BURST
GND/PGND
133k
324k
33k
10pF
22µF
560pF
Figure 2. Regulated 5V supply from USB or Li-Ion input
VOUT
AC COUPLED
100mV/DIV
VIN
1V/DIV
IL
200mA/DIV
500µs/DIV
Figure 3. Transient response for USB cable insertion and 22µF COUT
Linear Technology Magazine • March 2007
DESIGN IDEAS L
L1
Li-Ion
2.7V to 4.2V
CIN
2.2µF
SW
VIN
fOSC = 850kHz
LTC3458
GND/PGND
VOUT
6.4V TO 6.8V
VOUT
Z1
ON OFF
SHDN
FB
SYNC
COMP
D1
0.01µF
SS
RT
33k
COUT
2.2µF
0.01µF
ILIM
243k
D2
BURST
124k
CIN, COUT: TAIYO YUDEN JMK107BJ225MA
D1, D2: LUXEON EMITTER LUMILED WHITE
LXHLMW1D (2.9V AT 350mA)
L1: Wurth 12µH 774775112
0.01µF
RBURST
ILED (mA) =
RBURST: 35.7k FOR 350mA,
47.5k FOR 250mA,
82.5k FOR 150mA
Z1: CENTRAL SEMI 6.8V ZENER DIODE SOT-23 CMPZ5235B
Figure 4. Dual Lumiled LED application using the BURST pin to regulate current
LTC3458 Driving Two High
Current White LEDs
The BURST pin is normally used to
set the load current threshold where
the part transitions between fixed
frequency and Burst Mode operation.
A fraction of the internal P-channel
synchronous rectifier current (approx
LT3591, continued from page 31
The LT3591 is also applicable for
buck and buck-boost circuits because
of its high side sense. Figure 2 gives
an example of a buck circuit using the
LT3591. The LED current can go up
to 350mA. As shown in Figure 3, its
efficiency reaches 87%.
Dimming Control
Three different dimming control methods are available for the LT3591: filtered
PWM, DC voltage level and direct PWM
Linear Technology Magazine • March 2007
1⁄10,000)
is sourced from BURST and
internally compared to a 1V threshold.
When BURST is >1V the part operates
in fixed frequency mode and when
<1V in Burst Mode operation. To
maximize efficiency through the fixed
100
90
EFFICIENCY (%)
Another challenge in this application is responding to a step change on
input voltage when the USB cable is
inserted or removed. The LTC3458L’s
current mode architecture and programmable compensation allow the
designer to minimize the resulting
transient on the output. Figure 3
shows the LTC3458L’s transient response with a 3.6V to 5V input step
with the configuration of Figure 2. The
resulting ripple on VOUT is less than
35mV with a 22µF output capacitor.
150mA, 6.4V
350mA, 6.8V
70
60
50
NOTE: LUMILED CURRENT REGULATION
~10% OVER VIN RANGE
2.0
2.5
3.0 3.5
4.0 4.5
INPUT VOLTAGE (V)
5.0
10, 000 • VREF
RBURST
where VREF = VFB = 1.23V. Since FB
regulates above the 1V Burst Mode
threshold, the part operates in fixed
frequency mode. A zener diode Z1
(VF = 6.8V) is added to limit the output voltage in case the LED string
is opened. The resulting electrical
efficiency with respect to input voltage and LED current is impressive
and shown in Figure 5. LED current
accuracy is near 10% over the entire
Li-Ion operating range.
Conclusion
250mA, 6.6V
80
frequency/Burst Mode transition,
the threshold is typically set between
50mA to 100mA.
An application where the LTC3458
is used to regulate current (as opposed
to output voltage) in a string of high
current LEDs is shown in Figure 4.
Current regulation in the LED string is
achieved by connecting FB to BURST,
which mirrors a portion of the output
or LED current. Approximate output
current is programmed by the value of
RBURST using the following formula:
5.5
Figure 5. Efficiency vs VIN for the circuit in
Figure 4
dimming control. The direct PWM dimming circuit has the highest dimming
range and maintains a constant chromaticity for the LEDs. A typical dimming
Want to know more? Visit:
www.linear.com
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1-800-4-LINEAR
The LTC3458 and LTC3458L provide
compact solutions for battery powered
devices requiring step-up conversion.
Thermally enhanced packaging and
low switch resistance at heavy loads
along with Burst Mode operation at
light loads result in excellent efficiency
over many decades. The programmable
function pins allow both parts to be
customized for use in a wide variety
of applications. L
application is shown in Figure 4. The
PWM signal controls the turn-on and
turn-off of the chip and the LED string.
With a 100Hz PWM signal, the dimming
range can reach 80:1.
Conclusion
LT3591 is a white LED driver designed
to drive up to ten LEDs in series. It is
ideal for portable applications because
of its small solution size, high level of
integration and high efficiency. L
41
L NEW DEVICE CAMEOS
New Device Cameos
DC/DC Converter with
Selectable Frequency Modes
in a 2mm × 3mm DFN
The LTC3543 is a high efficiency
600mA monolithic step-down switching regulator intended for low power
applications such as Lithium-Ion battery powered devices. It operates within
a 2.5V to 5.5V input voltage range and
has three different frequency modes
of operation.
Efficiency is extremely important in
battery powered applications, and the
LTC3543 keeps efficiency high with an
automatic, power saving Burst Mode
operation, which reduces gate charge
losses at low load currents. With no
load, the converter draws only 45µA,
and in shutdown, the device draws
less than 1µA, making it ideal for low
current applications.
Burst Mode operation is an efficient
solution for low current applications,
but sometimes noise suppression
is a higher priority. To reduce noise
problems, a pulse-skipping mode is
available, which decreases the ripple
noise at low currents. Although not
as efficient as Burst Mode operation
at low currents, pulse-skipping mode
still provides high efficiency for moderate loads. In dropout, the internal
P-channel MOSFET switch is turned
on continuously, thereby maximizing
the usable battery life.
Three different frequency modes
are possible on the LTC3543: Fixed
Frequency, Spread Spectrum, or Synchronous. In Fixed Frequency mode,
the regulator operates at a constant
2.25MHz making it possible to use
capacitors and inductors that are less
than 1.2mm in height. In Spread Spectrum mode, the switching frequency is
randomly varied from 2MHz to 3MHz.
By spreading the switcher’s operating
frequency, a significant reduction in
peak radiated and conducting noise
can be realized. In Synchronous mode,
the LTC3543’s switching frequency
can be synchronized to a 1MHz to
3MHz external clock.
The small size, efficiency, low ex­
ternal component count, and design
42
flexibility of the LTC3543 make it an
ideal DC/DC converter for portable
devices using a Lithium-Ion battery.
25Msps 14-Bit ADC in
5mm × 5mm TQFP Package
for Industrial and Automotive
Applications
The LTC2246H is a 25Msps sampling
14-bit A/D converter designed for digitizing high frequency, wide dynamic
range signals. The LTC2246H is offered
in a leaded 5mm × 5mm TQFP package
for use in industrial and automotive
applications.
The TQFP leaded package makes
the LTC 2246H ideal for applications
in high temperature and high vibration
environments. The leaded package
provides superior strain relief compared to the QFN leadless package.
The LTC2246H provides data sheet
performance over a temperature range
of –40°C to 125°C.
Like the original LTC2246, the
LTC2246H provides a 73dB SNR and
80dBFS of spurious free dynamic
range for input frequencies up to
140MHz. The DC specs include ±1LSB
INL (typ), ±0.5LSB DNL (typ) and no
missing codes over temperature. The
transition noise is a low 1LSBRMS.
A single 3V supply allows low power
operation, and a separate digital output supply allows the outputs to drive
0.5V to 3.6V logic. A single-ended clock
input controls converter operation. An
optional clock duty cycle stabilizer allows full performance for a wide range
of clock duty cycles.
Octal Supply Supervisor
Allows Monitoring Positive
and Negative Voltages
The LTC2910 is an octal voltage
monitor intended for monitoring
multiple voltages in a variety of
applications. Each input has a low
0.5V threshold, featuring 1.5% tight
threshold accuracy over the entire
operating temperature range. When
any input falls below the 0.5V threshold, common RST and RST outputs
assert. The low fixed threshold volt-
age allows monitoring any voltage
level with the selection of just two
resistances.
Each input also features glitch
rejection. The LTC2910 integrates the
output of the first stage comparator
allowing each input to filter glitches
without adding hysteresis, which
would cause additional accuracy errors. With this type of glitch filtering,
the glitch duration that triggers the
output is dependent on the glitch
magnitude.
A three-state polarity-select pin
(SEL) selects one of three possible
polarity combinations for the input
thresholds. With this, up to two inputs will trigger when a voltage is
above the 0.5V threshold. This allows
overvoltage monitoring or negative
supply monitoring when used with
the available 1V reference (REF). The
SEL pin is connected to GND, VCC,
or left unconnected during normal
operation. This allows the different
polarities to be selected without the
need for external components.
The RST and RST outputs each
have a weak internal pull-up to VCC
and a strong pull-down to ground.
This arrangement allows each pin
to have open drain behavior while
possessing other beneficial characteristics. The weak pull-up eliminates the
need for an external pull-up resistor
when the rise time on the pin is not
critical. The open drain configuration allows for wired-OR connections
when more than one signal needs
to pull down on the pin. The output
also has an externally adjustable
timeout function that holds the pin
asserted for a set period of time after
all faults have cleared. This assures a
minimum reset pulse width allowing
a settling time delay for a monitored
voltage after it has entered the valid
region of operation. By connecting
a capacitor between the TMR pin a
ground, virtually any timeout value
can be chosen.
The LTC2910 is available in spacesaving 16-lead SSOP and 16-lead
(5mm × 3mm) DFN packages. L
Linear Technology Magazine • March 2007
DESIGN TOOLS L
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to speed up and simplify the simulation of switching
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• Macromodels for most of Linear Technology’s switching regulators as well as models for many of LTC’s high
performance linear regulators, op amps, comparators,
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FilterCAD — FilterCAD 3.0 is a computer-aided design
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SPICE Macromodel Library — A library includes LTC
op amp SPICE macromodels for use with any SPICE
simulation package.
Linear Technology Magazine • March 2007
43
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www.linear.com
Linear Technology Magazine • March 2007