V17N3 - OCTOBER

LINEAR TECHNOLOGY
OCTOBER 2007
IN THIS ISSUE…
COVER ARTICLE
Low Dropout Regulator Can Be Directly
Paralleled to Spread The Heat ............1
Robert Dobkin
Linear in the News… ...........................2
DESIGN FEATURES
16-Channel LED Driver Drives up to 160
White LEDs with 5000:1 PWM Dimming
...........................................................6
Keith Szolusha
2-Phase Synchronous Buck Controller
Delivers Maximum Features in
Minimum Footprint ...........................10
Eric Gu and Theo Phillips
Measure Microamps to Amps or Reduce
Power Dissipation by 99%, You Decide!
.........................................................13
Brendan J. Whelan
Pushbutton On/Off Controller Provides
µProcessor Reset Monitor and Input
Supply Monitoring.............................17
Victor Fleury
LED Driver Yields 3000:1 True Color
PWM Dimming with Any Buck, Boost
or Buck-Boost Topology from a Wide
3V–40V Input Range ..........................20
Xin Qi
White LED Driver and OLED Driver
with Integrated Schottkys and Output
Disconnect in 3mm × 2mm DFN .........23
Alan Wei
Light Up 12 LEDs from a Single-Cell
Li-Ion Battery via Highly Integrated
3mm × 2mm Dual-LED-String Driver
.........................................................25
Ben Chan
Low Offset 2-Wire Bus Buffer Provides
Capacitance Buffering, Stuck Bus
Recovery, and Tolerates High VOL .....28
John Ziegler
DESIGN IDEAS
....................................................32–40
(complete list on page 32)
New Device Cameos ...........................41
Design Tools ......................................43
VOLUME XVII NUMBER 3
Low Dropout
Regulator Can Be
Directly Paralleled
to Spread The Heat
Introduction
by Robert Dobkin
The 3-terminal adjustable linear regu- output current, all-surface-mount aplator has been around since 1976, but plications where only a limited amount
since then, little has changed in its of heat can be dissipated in any single
essential architecture. A 1.2V refer- spot on a board—applications that
ence is boosted to generate a regulated previously demanded a switching
output somewhere above a minimum regulator.
1.2V. What if, however, you throw
When regulators are surface mountaway the voltage
ed on a system
reference and reboard, conducplace it with a
tive dissipation
The LT3080 is the first
precision current
and air -cooling
adjustable linear regulator
source? The result
limits the amount
that can be directly
is a giant leap
of power that can
paralleled to spread the
forward in linear
be dissipated in
current load and thus
regulator capabileach chip. With
ity, performance
a typical board,
spread dissipated heat.
and versatility.
allowing a max
This makes it possible to
The LT3080 is the
use linear regulators in high operating temfirst adjustable
perature of 60°C
output current, all-surfacelinear regulator to
to 70°C, a linmount applications where
do just that. This
ear regulator can
deceptively simsafely dissipate
only a limited amount of
ple architectural
approximately 1W
heat can be dissipated in
change allows this
any single spot on a board— to 2W. This numnew regulator to
ber depends on
applications that previously
be directly paralthe ability of the
demanded a switching
leled to spread the
board to spread
regulator.
current load and
the heat and airthus spread dissiflow across the
pated heat among
board. If high powthe ICs. Spreading the heat makes it er requirements cause the regulator
possible to use linear regulators in high to generate more heat than the board
continued on page Sales Offices .....................................44
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM,
UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
Video Stars
EDN magazine launched their new online video design ideas
with Linear Technology Staff Scientist Jim Williams’ video
clip, “Measure Nanoamps to Ensure Accurate Computer
Clocks.” EDN also ran a video design idea from Applications Design Manager Mark Thoren, “Build Your Own
Laboratory Precision Voltage Reference.” Check them out
at www.edn.com
New 3-Terminal Regulator Introduced
Linear recently introduced the
LT3080, a 1.1A 3-terminal LDO
that can be easily paralleled for
spreading dissipated heat, and
requires only a single resistor to
adjust the output. The device,
developed by Linear Technology
cofounder and Chief Technical
Oficer Bob Dobkin, is a follow-on
to his earlier contribution in this
key product area.
According to online publication
AnalogZone, “Reducing the set resistor down to a single component
is an amazing jump in three-terminal adjustable regulators, but
even bigger is the ability to take
the output of the regulator down
to 0 V. This opens up a wealth of
applications.”
The 3-terminal adjustable
regulator was irst introduced in
1976 with the LM317. It used two
external resistors that would boost
the output voltage anywhere from
1.2V up to 40V. It was speciied with good line and load
regulation, 3V dropout, 1.5A output current and had a well
controlled current limit that was constant with temperature.
This device was a big advance over the earlier regulators
which were ixed and had poor current limit.
One shortcoming of the LM317, and even its higher
current offspring, is the inability to adjust the output
below 1.2V. New high performance digital circuits require
voltages below 1.2V and there is reason to believe that
these voltages will drop further.
Another problem with older regulators is the inability
to parallel the devices. Paralleling allows higher output
current and enables spreading the dissipated power over
a larger area. This is a boon in all-surface-mount or high
density systems, where localized peak temperatures are
limited and their compact design precludes the use of heat
sinks and extra wires.
High performance switching regulators have moved
in to ill this gap, providing lower output voltages and
2
minimizing the heat buildup. The downside of switcher
solutions is cost and complexity.
The new LT3080 solves all of these problems. The output
is adjustable with a single resistor down to zero and devices
can be paralleled for higher output current or to spread
the heat. The input to output dropout is 1.3V when used
as a 3-terminal regulator, or the collector of the power
device can be connected separate
from the control circuitry to allow
dropout voltages of 300mV. This
allows high eficiency if auxiliary
supplies are available for powering
the control circuitry. The LT3080
can eliminate the need to have a
switching regulator for power levels
that are easily handled by a linear
regulator.
The LT3080 its well with modern circuit design. Lower operating
voltages, higher currents, higher
density and surface mounting all
preclude standard IC regulators.
The new architecture allows for a
regulator that its well with high
performance circuits. The device
is speciied up to 40V, increasing
its versatility and providing good
margin for transients in automotive applications.
Surge Protector
Another innovative product recently
launched by Linear is the LT4356 surge
protector, an overvoltage protection regulator, with
overcurrent protection and inrush current limiting for
high availability systems. In applications where electronic
systems must cope with high voltage surges of short duration, such as load dump in automobiles, the LT4356
provides solid front-end protection for valuable, safety
critical downstream components. The wide input operating range of 4V to 80V enables continuous operation
during cold crank conditions where the battery voltage
can be as low as 4V. With its high input voltage rating,
the LT4356 can handle transient voltages of 100V and
higher, and provides reverse input protection to –30V
without damage to itself or the load. The LT4356 lends
itself well to automotive, industrial and avionics applications, as well as positive high voltage distributed power
Hot Swap™ systems.
The device is a featured new product in the most recent
issue of Auto Electronics magazine, which highlighted the
product’s ability to protect electronic systems from high
voltage surges in automotive applications. L
Linear Technology Magazine • October 2007
DESIGN FEATURES L
LT080, continued from page Internal Precision Current
Source Makes it Possible to
Parallel the LT3080
A precision “zero” TC 10µA internal
current source is connected to the noninverting input of a power operational
ampliier. The power op amp provides
a low impedance buffered output from
the voltage on the non-inverting input.
A single resistor from the non-inverting
input to ground sets the output voltage
and if this resistor is set to zero, zero
output results. As can be seen, any
output voltage can be obtained from
zero up to the maximum deined by
the input power supply.
What is not so obvious from this
architecture is the beneits of using a true internal current source
as the reference as opposed to the
bootstrapped voltage reference of older
regulators. A true current source alLinear Technology Magazine • October 2007
2.5
10.20
10.15
SET PIN CURRENT (µA)
2
VIN – VOUT (V)
can dissipate, the regulator must be
mounted separately on a heat sink. In
all-surface-mount systems, this is not
an option, so the limitation of power
dissipation (1W for example) limits the
output current.
Figure 1 shows the maximum output current at different input-output
differentials that can be obtained for
a regulator with both 1W and 2W
dissipation. 2W dissipation is a reasonable limitation on a single regulator.
Paralleling LT3080s increases the
maximum total output current by
spreading the heat, helping to maintain low peak temperatures.
The LT3080 is also especially well
suited to applications needing multiple
rails. The new architecture adjusts
down to zero with a single resistor, handling modern low voltage digital ICs.
Adjusting to zero output makes it possible to shut off the powered circuitry
when the input is preregulated—such
as a 5V or 3.3V input supply. External
resistors in series with IN can help
spread the heat, keeping the system
all surface mount.
Finally, the new regulator is made
in a 40V bipolar process. This allows
high input voltage as well low operating voltage, since bipolar transistors
turn on at 0.6V.
POWER DISSIPATION = 2W
1.5
POWER DISSIPATION = 1W
POWER DISSIPATION = 0.5W
1
0.5
10.10
10.05
10.00
9.95
9.90
9.85
0
0
1
2
3 4 5 6 7
LOAD CURRENT (A)
8
9
9.80
–50 –25
10
Figure 1. The available output current as
a function of input-output differential and
allowable power dissipation. At 2W, 1A
output currents are possible even with 1V
to 2V input-to-output differential.
lows the regulator to have gain and
frequency response independent of
the output voltage since the loop
gain does not change. Traditional
adjustable regulators, such as the
LT1086, have a change in loop gain
and bandwidth with output voltage
as well as bandwidth changes when
the set/adjustment pin is bypassed
0
25 50 75 100 125 150
TEMPERATURE (°C)
Figure 2. Temperature performance of
the LT3080’s precision current source.
to ground. With the LT3080, however,
the loop gain remains unchanged with
changing output voltage or bypassing.
Output regulation is no longer ixed
at a percentage of the output voltage
but rather a ixed number of millivolts.
With a true current source, all the gain
in the buffer ampliier provides regulation; none of it is needed to amplify the
reference to a higher output voltage.
This, and the LT3080’s precise DC
Table 1. Comparison of the LT3080 to traditional 1A regulators
LT317
LT1086
LT3080
Dropout (V)
3V
1.5V
1.3V or 300mV
Min Load (mA)
10
10
0.3
Min Output (V)
1.2
1.2
0
IOUT (A)
1.5
1.5
1.1
Parallel Operation
—
—
L
External Resistors
2
2
1
Table 2. Some key specifications for the LT3080
Parameter
Value
Load Regulation, IOUT = 10mA to 1.1A
<1mV
Line Regulation, IN = 2V to 40V
<1mV
SET Pin Current
10µA ±1%
Min Load Current
0.3mA
SET to OUTPUT Offset
1mV
Operating Temp Range
–55°C to 125°C
Dropout (3-Terminal) 1.1A
1.3V
Dropout (4-Terminal) 1.1A
0.3V
Ripple Rejection (120Hz)
75 dB
3
L DESIGN FEATURES
characteristics, makes it possible to
easily parallel regulators (see below:
“It is Easy to Parallel the LT3080”).
VCONTROL
10µA
High Performance
No sacriices were made in regulator
performance for the LT3080. Line
and load regulation are excellent over
temperature. Its low dropout and a
new architecture make it extremely
versatile. On chip trimming keeps the
accuracy of the reference current below
one percent, and the offset voltage
between the SET pin and the output
to under 2mV.
Line regulation is virtually immeasurable, a few nanoamps, since the
internal circuitry double-regulates the
current source section. The temperature performance of the reference is
shown in Figure 2 and is nearly lat
from –55°C to 150°C. Thermal limiting is set at about 160°C. Quiescent
current is only about 300µA, allowing
this device to be used in light load
and battery-powered applications.
High frequency ripple rejection is also
excellent, making the LT3080 a good
it as a post regulator to switching
regulators when low output ripple is
needed.
+
–
SET
4
VCONTROL Pin Offers Additional
Ways to Spread the Heat
Clearly, one of the driving design objectives for this new regulator was to
enable the thermal design for surface
mount boards—notably eliminating
the need for heat sinks. Paralleling
LT3080s makes a signiicant difference, but another feature also helps.
The collector of the output transistor
is available at the VCONTROL pin (see
Figure 3). This can decrease peak
temperatures in two ways.
First, the dropout on the collector
is 400mV (IN pin) so it can take a
lower voltage supply than is used for
the LT3080’s control circuitry (1.3V
OUT
Figure 3. Block diagram of the LT3080.
Four terminals are available from the
package to allow the device to be used
in a low dropout mode with only 300mV
input-to-output dropout.
The SET pin is very high impedance
and the output voltage is set by the
10µA current times an external resistor. Even a 0.1µF capacitor is large
enough to bypass the SET pin at 60Hz,
allowing for reduction of output noise
and pickup into the SET pin.
With a capacitor on the SET pin,
output noise is 40µVRMS—about the
VIN
3.3V
≥ VOUT +1.3V
Operation of the LT3080
Figure 3 shows a block diagram of the
LT3080. The simplest application, as
a 3-terminal adjustable regulator, is
shown in Figure 4. The VCONTROL and IN
pins are tied together. (These two pins
can connect to different supplies for
additional thermal beneits, described
below.) The only added components
are input and output capacitors and
a resistor to set the output voltage. In
this case, the output is set to 1.8V,
which at 1.3V dropout works with a
3.3V input. Input and output capacitors are required for stability—they can
be ceramic, tantalum, or electrolytic
capacitors. Unlike older 3-terminal
regulators, the minimum load current
is guaranteed at only 1mA for this
device. By making the adjustment
resistor zero or tying the SET pin to
the ground with a switch, the output
goes to zero, turning off connected
circuitry. Typically, the quiescent
current is under 300µA.
same as many low noise regulators. In
other applications, the SET pin can be
driven with an ampliier or a reference
voltage to be used as a power buffer.
With multiple regulators, the SET pins
and outputs can be tied together for
paralleling the regulator (described
below). Grounding the SET pin brings
the output to zero.
LT3080
IN
LT3080
IN
VCONTROL
+
–
1µF
OUT
VOUT
1.8V
SET
1µF
1µF
180k
Figure 4. Basic hookup for the LT3080 regulator. The IN and VCONTROL pins are tied
together and a single resistor sets the output voltage. A 1µF output capacitor ensures
stability. If the adjustment resistor is adjusted to zero, the output is zero.
RD
2.9Ω
VIN
5V
(4.7V MIN)
LT3080
IN
VCONTROL
+
–
1µF
OUT
VOUT
1.8V
SET
1µF
180k
RD =
VIN(MIN) − ( VOUT + 0.4V )
IOUT(MAX)
Figure 5. Adding a resistor in series with the collector of the output device to
remove some of the power dissipation from the regulator. This disperses heat
around the surface mount board rather concentrating it at the regulator.
Linear Technology Magazine • October 2007
DESIGN FEATURES L
VIN
LT3080
Table 3. Trace resistance for
ballast resistors in mΩ/in
VCONTROL
+
–
10mil Width 20mil Width
OUT 10mΩ
SET
VIN
4.5V TO 30V
VIN
LT3080
VCONTROL
+
–
1µF
SET
OUT 10mΩ
VOUT
3.3V
2A
100µF
165k
Figure 6. Paralleling of two regulators. Need more current? Add more regulators.
Current sharing is assured by the 10mΩ PC board traces, which act as ballast resistors.
dropout). Lowering the input-to-output voltage on the power transistor
increases eficiency and thus reduces
dissipation.
Second, a resistor can be inserted
in series with the collector. Adding
this resistor splits power dissipation
between the internal power transistor and an external resistor so that
some of the heat from the IC can be
moved to elsewhere on the PC board.
Figure 5 shows such a design using a
2.9Ω resistor. The dropout voltage for
the output transistor is only 400mV,
so several volts can be dropped across
the external resistor, minimizing the
heating of the IC. At full load, the
external resistor drops approximately
3V and dissipates 3W. To minimize
peak temperatures on a PC board,
this resistor can be split into several
1Ω resistors and thus further spread
dissipated heat. The power dissipation
in the LT3080 peaks at about 750mW
when the power dissipation in the
resistor and the power dissipation in
the transistor are equal. The copper
planes in the PC board can easily
handle this power.
Of course the LT3080 can be operated in 3-terminal mode by simply
connecting the VCONTROL pin to the
power input pin, but this limits the
input to the 1.3V dropout of the
regulator. Alternately, by tying the IN
Linear Technology Magazine • October 2007
pin to a lower voltage than VCONTROL,
it is possible to produce a 1.1A, 2.5V
to 1.8V or 1.8V to 1.2V regulator with
low dissipation—likewise for other low
IN – OUT differentials. To achieve the
same peak operating temperatures,
the dissipation constrained design
current must be lower for higher IN
– OUT differentials, such as 5V to 3.3V
or 3.3V to 1.5V.
1oz Weight
54.3
27.1
2oz Weight
27.1
13.6
It is Easy to Parallel the LT3080
The architecture of the LT3080 allows
direct paralleling unlike any other type
of regulator. Parallel linear regulators
distribute the current load and distribute power dissipation around the
system board. Need more power but
can’t afford more spot heating? Add
more regulators. Even paralleling 5–10
devices is reasonable.
Practical current sharing by parallel
LT3080s is made possible by internal
trimming, which keeps the offset voltage between the adjustment pin and
the output under 2mV. Figure 6 shows
how easy it is to parallel LT3080s.
Simply tie the SET pins of the LT3080s
together, and do the same for the IN
pins. This is the same whether it’s in
3-terminal mode or has a separate IN
supply. The outputs are also connected
in common but with a small piece of
PC trace in series with each OUTPUT
continued on page 27
Figure 7. Thermograph shows two regulators, each dissipating 0.7W
from a 0.7V input-to-output differential at 2A total load. The result
is a 28°C rise over ambient at each IC on a two sided PC board.
5
L DESIGN FEATURES
16-Channel LED Driver
Drives up to 160 White LEDs
with 5000:1 PWM Dimming
by Keith Szolusha
Introduction
0.47µF
SW4
L4
L5
SW5
L6
SW6
SW7
L7
0.47µF
SW13
L13
L12
SW12 L11 SW11 VIN SW10 L10
0.47µF
0.47µF
0.47µF
0.47µF
SW8
L8
PWM9
PWM10
PWM11
PWM12
PWM13
PWM14
PWM15
PWM16
GND
RSET
SW9 L9
100µH
L15 SW15 VIN SW14 L14
0.47µF
100µH
100µH
0.47µF
0.47µF
LED
BRIGHTNESS
CONTROL
75.0k*
L3
100µH
100µH
100µH
SW3
0.47µF
LT3595
100µH
10µF
SW2
The PWM dimming capability of the
LT3595 is as high as 5000:1. Figure 2
shows the 5000:1 PWM dimming waveform and a very square looking LED
current waveform. Even at a mere 2µs
on-time, a 20mA LED current snaps up
100µH
VCC
3V TO
5.5V
L2
0.47µF
5000:1 PWM Dimming
100µH
LED
BRIGHTNESS
CONTROL
L1 SW1
OPENLED
PWM1
PWM2
PWM3
PWM4
PWM5
PWM6
PWM7
PWM8
SHDN
VCC
L16 SW16
0.47µF
100µH
100k
100µH
VCC
0.47µF
100µH
0.47µF
100µH
0.47µF
100µH
10µF
100µH
VIN
15V TO
45V
Each channel requires only a tiny chip
inductor and an even tinier ceramic
output capacitor. The only other required components are a single input
capacitor and current-determining set
resistor (Figure 1). All sixteen channels of catch diodes, power switches,
and control logic with compensation
are squeezed inside the LT3595’s
relatively small 56-pin, 5mm × 9mm
QFN package.
The LT3595 boasts 92% peak
eficiency at a 2MHz switching frequency.
The LT3595 buck mode LED
driver has 16 individual
channels—each driving up
to 50mA from inputs up to
45V. One advantage of its
56-pin QFN package is the
availability of individual
PWM pins for each of the
16 channels. This allows
independent control over the
brightness of different areas
of a monitor or display. For
instance, the secondary
picture of a picture-inpicture display can have
a different brightness
than the main picture.
100µH
The light behind the large displays
comes increasingly from the smallest
lights: LEDs. A lot of LEDs. They light
large screen LCD televisions, giant
LED billboards and even stadium
advertisements. In such big displays,
driving hundreds of LEDs requires a
large quantity of high voltage drivers
that can accurately control a number of
long strings of LEDs, each string with
its own high PWM dimming ratios. A
simple, low-component-count solution is a must, especially in consumer
electronics.
The LT3595 buck mode LED driver
has 16 individual channels—each driving up to 50mA from inputs up to 45V.
It is possible with the LT3595 to drive
160 bright, white LEDs driven from a
single converter. Each channel has a
separate PWM input that is capable
of up to 5000:1 PWM dimming ratio.
* SETS PER
CHANNEL
CURRENT
TO 20mA
0.47µF
Figure 1. A 16-channel LED driver. The 15V–45V input is used to drive three white LEDs per channel with 5000:1 PWM dimming.
6
Linear Technology Magazine • October 2007
DESIGN FEATURES L
and turns off in sync with the 100Hz
PWM signal. Higher PWM dimming
ratios are achievable with lower PWM
frequencies, but 100Hz guarantees
that there is no visible licker.
One advantage of a 56-pin QFN
package is the availability of individual
PWM pins for each of the 16 channels.
In some applications, the brightness
of the entire screen is uniform and all
of the PWM pins can be tied together
and driven from a single PWM waveform. Ideally, every point on the screen
or display has the same brightness
VIN
45V
0.47µF
0.47µF
determined by a single PWM setting.
However, it may be a feature for some
billboards or television screens with
picture-in-picture to show small sections or regions of the display in higher
brightness than others for forefront
and background effects. In this case,
it is an advantage to be able to provide
some higher dimming PWM waveforms
to several channels and run different
brightness on other channels. PWM
can also be used to completely turn
off some channels or sections of a
display while leaving others on. This is
VPWM
5V/DIV
ISW
20mA/DIV
ILED
10mA/DIV
400ns/DIV
VIN = 15V
3 LEDS AT 20mA
T = 10ms
tON = 2 s
Figure 2. 5000:1 PWM dimming
waveforms for the circuit in Figure 1.
0.47µF
0.47µF
0.47µF
0.47µF
0.47µF
0.47µF
100µH
100µH
100µH
100µH
100µH
100µH
100µH
100µH
10µF
VCC
L1 SW1
OPENLED
PWM1
PWM2
PWM3
PWM4
PWM5
PWM6
PWM7
PWM8
SHDN
VCC
L16 SW16
LED
BRIGHTNESS
CONTROL
3V TO
5.5V
L2
SW2
SW3
L3
SW4
L4
L5
SW5
L6
SW6
SW7
L7
LT3595
L15
SW15 VIN SW14
L14
SW13
L13
L12
SW12 L11 SW11 VIN SW10 L10
SW8
L8
PWM9
PWM10
PWM11
PWM12
PWM13
PWM14
PWM15
PWM16
GND
RSET
SW9 L9
LED
BRIGHTNESS
CONTROL
30.1k*
100k
10µF
0.47µF
0.47µF
0.47µF
0.47µF
0.47µF
0.47µF
100µH
100µH
100µH
100µH
100µH
100µH
100µH
100µH
* SETS PER
CHANNEL
CURRENT
TO 50mA
0.47µF
0.47µF
Figure 3. A 16-channel LED driver for 160 white LEDs from a 45V input. PWM dimming ratio is 5000:1.
Linear Technology Magazine • October 2007
7
L DESIGN FEATURES
particularly useful for individual pixel
control of giant billboards and gives
the designer control of an amazing 16
pixels per IC.
EFFICIENCY (%)
100
Adjustable 50mA
LED Current per Channel
LED brightness is normally set by
static current. The LT3595 can drive
as high as 50mA per channel directly
through a string of LEDs. A single
external set resistor is all that is
needed to set the LED current for all
16 channels. Each channel has the
same programmed LED current—set
between 10mA and 50mA. LED current
accuracy is within 8% from channelto-channel.
The ixed frequency, current mode
control scheme provides stable operation over a wide range of input
and output voltages and currents.
Direct control of the LED current
through internal sense resistors for
each channel and internal switches
and control circuitry for each channel provide excellent constant current
source regulation for LED driving. The
internal 100mA power switches and
95
90
85
20
10
40
30
ILED (mA)
50
Figure 4. Efficiency of the 160-LED
driver shown in Figure 3 is over 92%.
exposed thermal pad of the 56-pin QFN
provide enough power and thermal
management to handle the power and
heat of 16 channels at 50mA.
45V Input to Any
Number of LEDs
The LT3595 has a 45V maximum input
voltage on its two VIN pins. With an 80%
maximum duty cycle at 2MHz switching frequency, this allows a fairly low
dropout and up to 35V LED output
per string. On the lip side, the low
minimum on-time of the IC (around
70ns) allows down to a single white
LED to be run at 10mA–50mA from
a 42V input.
Each channel of the LT3595 can
support any number of LEDs as long
as the total string voltage is between
3V to 35V. The only other requirement
is that the duty cycle is below 80% and
on-time is above the minimum rating.
One channel can have the maximum
number of LEDs and another channel can have the minimum number
of LEDs. This is typical in RGB applications where each color requires
a different number of LEDs, such as
8 red, 8 green, and 4 blue.
16 Fully Integrated and
Independent Channels
The block diagram in Figure 5 shows
the fully integrated design of the
LT3595. Each channel includes a
100mA, 48V NPN power switch,
Schottky diode, sense resistor, error
ampliier, compensation components
and other bias and control circuitry.
CIN
10µF
PWM1-16
52
VCC
3.3V
VCC
C1
10µF
VREG
L1-16
VREG
DFC
CONTROL
1 CHANNEL
16X
24
51
SHDN
RSET
VIN
VIN
COUT1-16
0.47µF
SW1-16
REF
V/I
–
+
PWM
RSET
Σ
Q
R
S
L1-16
100µH
+
–
ISNS
GND
57
RAMP
GENERATOR
OPENLED
2MHz
OSCILLATOR
23
CONTROL
Figure 5. Block diagram for the LT3595
8
Linear Technology Magazine • October 2007
DESIGN FEATURES L
The 16 channels run independently,
but regulate to the same LED current
at the same, internally ixed switching
frequency of 2MHz.
Each channel has its own PWM
pin and separate dimming logic. Nevertheless, all the channels must be
synchronized to the rising edge of the
PWM signal, where dimming is created
by varying duty cycle. Of course, the
falling edges can be asynchronous.
The maximum junction temperature is rated at 125ºC. The 31ºC/W
thermal capabilities of the 56-pin QFN
can be accomplished with proper layout of the IC for excellent grounding
and thermal management. Without a
decent ground plane or correct connection of the thermal pad, the thermal
impedance of the IC can creep up to
unacceptably high levels.
a. Top layer
b. Layer 2
c. Layer 3
d. Bottom layer
Low Shutdown Current
When the shutdown pin is pulled
low, all 16 channels turn off and the
part consumes a quiescent current
of just 15µA. Low shutdown current
saves battery energy and extends
its lifetime. In shutdown, the open
LED comparator is disabled and not
valid. If the shutdown pin is left high
and the PWM pins are pulled low,
the LEDs turn off, but the quiescent
current remains around 280µA. The
open LED pin function is still valid in
this case.
Recommended Layout
The LT3595 comes in a thermally
enhanced 56-pin 5mm × 9mm QFN
package. This fully integrated part
minimizes layout, complexity, and cost
of otherwise high component count
multichannel LED driver solutions.
With 31ºC/W thermal resistance, it
is possible to run at full 50mA LED
current and high number of LEDs
without violating the 125ºC junction
temperature rating.
Layout is important for the LT3595.
The ground connection is only tied to
the thermal pad (pin 57). Therefore,
the input capacitors, set resistors, and
control logic such as PWM signals,
shutdown signal and overtemperature
monitor must all be tied to the comLinear Technology Magazine • October 2007
Figure 6. Suggested board layout for the LT3595
mon ground at the thermal pad. To
minimize circuit noise and ripple, it
is best if the input capacitors and set
resistor are attached to ground on the
backside of the board with the shortest
connection possible between ground
and their respective pins. Figure 6
shows the recommended layout. For
a 5mm × 9mm 56-pin QFN package, it
may be best if the traces and vias are
small. The layout is optimized if vias
have a drill size of 6mil (.006 inches)
or less with pad of 12mil or less. Clearance between metal traces and pads
should be set at 5mil or below.
Conclusion
The LT3595 is a 16-channel buck
mode LED driver with 5000:1 PWM
dimming. The high 45V input voltage, 50mA LED current, and 2MHz
switching frequency make this a very
powerful multichannel LED driver for
big screen televisions, billboards and
stadium displays. The fully integrated
solution in the compact 5mm × 9mm
QFN package makes the designs small
and simple. 5000:1 PWM dimming
is one of the highest PWM dimming
capabilities available in an integrated
DC/DC converter LED driver IC. The
inductors, the input and output capacitors, the set resistor and the LEDs
are the only required external circuitry.
The 16 independently controlled channels maximize the lexibility of the
LT3595. L
Want to know more?
visit:
www.linear.com
or call
1-800-4-LINEAR
9
L DESIGN FEATURES
2-Phase Synchronous Buck Controller
Delivers Maximum Features in
by Eric Gu and Theo Phillips
Minimum Footprint
Introduction
The LTC3850 is a feature-rich dual
channel synchronous step-down
switching regulator available in a 4mm
× 4mm QFN package. It is designed to
meet today’s high performance power
application needs. With constant
frequency peak current mode control for clean operation over a broad
range of duty cycles, the LTC3850 is
a response to customer requests for
a cost-effective solution that balances
ease of use, eficiency, precision and
performance.
Familiar Features,
and Some New Ones
CCM
BURST
10mV/DIV
DCM
1µs/DIV
Figure 1. Three modes of operation.
Continuous mode features predictable,
constant frequency operation. Burst Mode®
operation has the best light-load efficiency,
with somewhat higher output ripple. Pulse skip
mode is a compromise between the other two.
D3
M1
VIN PGOOD EXTVCC INTVCC
TG1
0.1µF
BG1
0.1µF
S
COUT1
100µF
X2
S
1800pF
20k
1%
4.75k
1%
100pF
0.1µF
TK/SS1
M2
0.1µF
S
2.2µH
BG2
SENSE2 +
4.12k
1%
S
S
SENSE2 –
S
1.5k
1%
0.1µF
RUN2
VFB1
ITH1
S
63.4k
1%
22µF
50V
D4
PGND
SENSE1–
33pF
VIN
7V TO
20V
FREQ/PLLFLTR
RUN1
S
S
Figure 2. The LTC3850 is a peak current mode
controller. As such, it uses a compensating
ramp on the inductor upslope to ensure
stability at duty cycles greater than 50%.
Alone, the ramp would cause current limit to
drop at high duty cycles, but the LTC3850
uses a patent-pending scheme to prevent
this behavior. Here, the LTC3850 is operating
in current limit, and peak current is wellcontrolled when duty cycle swings from 66%
to 22%.
10k, 1%
SENSE1+
S
1.33k
1%
S
LTC3850
MODE/PLLIN
S
4ms/DIV
VIN = 5V TO 15V
VOUT1 = 3.3V IN CURRENT LIMIT
RSENSE = 8mΩ
BOOST2
SW2
S
ILIM
VOUT1
3.3V
5A
TG2
BOOST1
SW1
3.3µH
6.19k
1%
VIN
5V TO 15V
1.5V
= 107ns
(20 V) • (700kHz)
4.7µF
S
IL
1A/DIV
The LTC3850’s two channels run
out of phase, which reduces the
input RMS current ripple and thus
the input capacitance requirement.
Switching frequency can be adjusted
from 250kHz to 780kHz, either set
with a voltage on the FREQ/PLLFLTR
pin, or synchronized to a signal into
the MODE/PLLIN pin using a phaselocked loop. During high frequency
operation, the LTC3850 can operate
normally at low duty cycles due to its
short top switch minimum on-time.
For example, a 20V to 1.5V converter
operating at 700kHz requires a minimum on-time of less than
IL1
VIN = 12V
VOUT1 = 3.3V
LOAD = 100mA
The LTC3850 can cycle its strong
top gate drivers in just 90ns, making this low duty cycle application a
reality.
S
S
33pF
VFB2
ITH2
SGND
S
3.16k
1%
5.49k
1%
S
25.5k
1%
2200pF
TK/SS2
0.1µF
S
100pF
20k
1%
VOUT2
1.8V
5A
COUT2
100µF
X2
L1, L2: COILTRONICS HCP0703
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
Figure 3. Schematic for a 2-channel, 5A/500kHz regulator with DCR sensing.
10
Linear Technology Magazine • October 2007
DESIGN FEATURES L
EFFICIENCY (%)
85
1
80
75
70
0.1
65
POWER LOSS
60
55
50
0.01
Figure 4. Efficiency and power
loss for Figure 3, Channel 1.
light load ripple compared to Burst
Mode operation and improves light
C1
1000pF
R12
7.5k
R2
20k
R5
10k
RVIN
2.2Ω
M1
SENSE1– SENSE1+ RUN1 FREQ/ MODE/ SW1 TG1
PLLFLTR PLLIN
TK/SS1
CSS2
0.1µF
R4
25.5k
VOUT1
2.5V/15A
D3
CMDSH-3
M2
+
D1
B34OLA
330µF
4V
2X
VIN
LTC3850
BG2
PGND
SENSE2–
CVIN
1µF
INTVCC
TK/SS2
C15
47pF
RSENSE1
0.003Ω
BOOST1
ITH2
C12
100pF
L1
0.68µH
BG1
VFB2
R3
20k
CIN
180µF
CB1
0.1µF
VFB1
R18
4.99k
VIN*
7V TO 14V
PLLIN
400kHz
ITH1
C11
1000pF
+
C2
0.01µF
R1
43.2k
C10
33pF
0.01
10
0.1
1
LOAD CURRENT (A)
load eficiency compared to forced
continuous mode.
Tracking provides a predictable way
of slewing the output voltages up or
down. Tracking generally holds the
feedback voltage to the lesser of the
internal reference voltage or the voltage
on a TRACK pin. The LTC3850 goes
farther by combining track and soft
start functions in a single pin for each
channel and by tailoring the mode of
switching operation to the state of the
TK/SS pins.
When TK/SS is ramping up from
ground to 0.8V, either from its 1.3µA
internal current source or by tracking
another supply, the channel remains
CVCC
4.7µF
PGND
GND
D4
CMDSH-3
BOOST2
M3
SENSE2+ SGND RUN2 ILIM EXTVCC PGOOD SW2 TG2
L2
0.68µH
CB2
0.1µF
C5
1000pF
M4
+
C6
100pF
EFFICIENCY
90
C4
1000pF
CSS1
0.1µF
C7
1000pF
VIN = 12V
95 VOUT = 3.3V
R10
10Ω
R9
10Ω
10
100
POWER LOSS (W)
At heavy loads, the LTC3850 operates in constant frequency PWM mode.
At light loads, it can switch in any of
three modes (Figure 1). Burst Mode
operation switches in pulse trains of
one to several cycles, with the output
capacitors supplying energy during
intervening sleep periods. This provides the highest possible light load
eficiency. Forced continuous mode
offers PWM operation from no load to
full load, providing the lowest possible
output voltage ripple. Pulse skipping
mode operates at a constant frequency,
but always turns off the synchronous
switch before inductor current is allowed to reverse. This method reduces
RSENSE2
0.003Ω
330µF
4V
2X
VOUT2
1.8V/15A
D2
B340LA
PGOOD
R22
10Ω
RPG
100k
R20
10Ω
L1, L2: SUMIDA CEP125-OR6MC
COUT1, COUT2: SANYO 4TPD330M
M1, M3: RJK0305DPB
M2, M4: RJK0301DPB
* FOR VIN = 5V ± 0.5V, TIE VIN AND INTVCC PINS TOGETHER.
Figure 5. Schematic for a 2-channel, 15A/400kHz regulator.
Linear Technology Magazine • October 2007
11
L DESIGN FEATURES
in pulse skipping mode until the TK/SS
voltage reaches 0.64V. This prevents
the regulator from sinking current
from the output while it is at 80% or
less of the target voltage. When TK/SS
ramps up from 0.64V to 0.74V, the
channel operates in forced continuous
mode to ensure that the power good
indicator (PGOOD) makes just one
transition from low to high when the
output comes into regulation (within
±7.5% of the precision 0.8V reference). Once 0.74V (the undervoltage
threshold) is reached, the regulator
transitions to the mode of switching
operation programmed on the MODE/
PLLIN pin. When TK/SS is descending
from 0.8V, forced continuous mode
takes over when VTK/SS and VFB ramp
from 0.74V down to 0.1V, allowing the
LTC3850 to pull down the output at
the programmed slew rate. Once TK/
SS ramps down to 0.1V, the channel
begins operating in pulse skipping
mode. Switching stops when TK/SS
is less than the feedback voltage.
Each channel also features a
separate RUN pin with a precision
1.2V turn-on threshold. When the
LTC3850’s own current source is
used to charge the soft-start capacitor, bringing a channel’s RUN pin high
causes its soft-start capacitor to begin charging within about 80µs. As
an alternative, either RUN pin can
remain high while TK/SS is held low,
which keeps the internal 5V regulator enabled as a standby supply. This
feature can be used to power a wakeup circuit which controls the state of
both TK/SS pins.
The LTC3850’s two channels
run out of phase, which
reduces the input RMS
current ripple and thus
the input capacitance
requirement.
error of 5%-10%. The LTC3850 can use
either method, with a choice of three
pin-selectable current limits.
When an output sees a short circuit, the LTC3850 protects the input
supply and power components by
limiting peak current cycle by cycle.
The main MOSFET turns off when the
inductor’s peak current sense threshold (VSENSE(MAX)/RSENSE) is reached.
VSENSE(MAX) can be set to 30mV, 50mV,
or 75mV, for a wide range of output
current levels. Duty cycle has little
effect on this current limit (Figure 2).
For load currents greater than the
programmed maximum but less than
a hard short, the LTC3850 gracefully
folds back the top MOSFET’s on-time,
reducing the output voltage.
The LTC3850 also protects against
undervoltage input and overvoltage
output voltages. The RUN pins can
be referenced to a voltage divider from
VIN, so that their precision thresholds
control the state of the outputs. If
the output voltage is more than 7.5%
above its target, the bottom MOSFET
can remain on until regulation is
recovered. If the LTC3850 is allowed
continued on page 6
a. Top view
Two Ways to Sense Current
The LTC3850 features a fully differential comparator to sense current
through the inductor. The SENSE+
and SENSE– pins can be connected
to a sense resistor in series with the
inductor, or to an RC network in
parallel with the inductor for energy
eficient sensing across the inductor’s
DC resistance (DCR sensing). Using
1% tolerance sense resistors offers an
accurate current limit, but incurs I2R
losses in the resistors. DCR sensing
prevents this power loss, but uses a
sense element with a typical built in
12
b. Bottom view
Figure 6. The circuit of Figure 5 can be laid out inside a square inch on a two-layer board.
Linear Technology Magazine • October 2007
DESIGN FEATURES L
Measure Microamps to Amps or
Reduce Power Dissipation by 99%,
by Brendan J. Whelan
You Decide!
Introduction
Why Use High Side Sense?
In many applications, the sense load
is ground referred. The simplest way
to measure current in these systems is
low side sensing, which involves adding a small sense resistor between the
load and the system ground (Figure 1).
Almost any ampliier can be used to
amplify the sense voltage, and no level
translation is required.
Low side sensing, for all its simplicity, has several inherent problems.
First, the sense resistor affects the
return voltage on the load—where
the return is now the sum of the
system’s ground potential and the
voltage across the sense resistor. The
load now loats above system ground
by the sense voltage, which can be
signiicant—a traditional low side
current sense circuit of some precision requires a voltage around and
above 100mV.
Of course, as the load current
changes, the sense voltage reacts as
it should, thus affecting the ground
potential seen by the load. A moving
ground reference is no reference at all,
resulting in load errors and signiicant
noise. Transient load currents can
present the load with tremendous
Linear Technology Magazine • October 2007
3
4
5
+
2
10Ω
TO
MEASURED
CIRCUIT
1
LTC2050HV
–
OUT
3V/AMP
LOAD CURRENT
IN MEASURED
CIRCUIT
10k
3mΩ
0.1µF
LOAD CURRENT
GND
Figure 1. Classic high precision low side current sense
ground noise, reducing the performance of the monitoring system and
injecting this noise directly into the
load.
Second, there may be coupling
between the load and ground, due to
shielding. This coupling can alter the
effective resistance of the sense resistor, especially over frequency, thereby
reducing system performance.
Finally, safety may be compromised.
In the event that the sense resistor
fails, or becomes disconnected, the
ground node of the load is energized to
the full supply voltage. This is a safety
hazard, as the node that is normally
connected to ground is now held at
dangerous potentials. It may not be
obvious that such a fault has occurred,
so it may be assumed that the ground
terminal of the load is held at a safe
voltage. Low voltage circuitry tied to
the grounded side of the load may also
be damaged, thus requiring additional
work and expense in repair.
High side sensing eliminates these
problems by allowing the system load
to be safely and securely referred to
ground. The high side of the load
can be measured relative to ground
without the sense resistor noise. The
sense resistance can be more carefully
controlled. Most importantly, a fault
in the sense resistor disconnects the
load from the supply, not ground, so
safety is assured.
So, why isn’t high side sensing used
more often? The problem is that these
advantages are tempered by a lack of
simplicity. First, high supply voltages
with high voltage transients demand
a robust monitoring circuit. Second,
the sense voltage must be accurately
translated to ground. The LTC6102
addresses both of these problems with
ease, while adding additional features
to maximize accuracy and lexibility.
Solve the Dynamic
Range Problem
It is no great technical feat to measure
high load currents, but accurately
monitoring high currents and low
110
100
RSENSE = 100mΩ RSENSE = 10mΩ
RSENSE = 1Ω
90
DYNAMIC RANGE (dB)
A required, but often overlooked, element of any industrial or automotive
monitoring/control system is a current sensing circuit that can maintain
accuracy over the entire load range.
Many applications use circuits that
can provide only moderate precision
and dynamic range. In many cases
the current sense solution is woefully
inadequate, with poor resolution and
signiicant power dissipation in the
sense resistor. The LTC6102 addresses
both of these problems while boosting
performance via a comprehensive set
of current sense features.
10V
80
100dB: MAX
VSENSE = 1V
70
40dB: MAX
VSENSE = 1mV
60
50
RSENSE = 10µΩ
40
30
RSENSE = 100µΩ
RSENSE = 1mΩ
20
0.001
0.01
0.1
1
10
100
MAXIMUM POWER DISSIPATION (W)
DYNAMIC RANGE RELATIVE
TO 10µV, MINIMUM VSENSE
MAX ISENSE = 1A
MAX ISENSE = 10A
MAX ISENSE = 100A
Figure 2. Dynamic range vs maximum
power dissipation in RSENSE
13
L DESIGN FEATURES
V+
VIN
RIN
+IN
–INS
+
VCM
V
–
–
–INF
V+
0.1µF
VREG
LTC6102
OUT
VOUT
ROUT
VOUT = VIN •
ROUT
RIN
Figure 3. Level translation
currents on the same line, or resolving very small variations on large
load currents, requires a monitoring
circuit with a wide dynamic range. For
example, a system that typically runs
at 1A, but has dynamic loads up to
100A, would require at least 40dB of
dynamic range for accurate measurement. If the typical load current must
be measured with 1% accuracy, then
80dB of dynamic range is required. A
battery system that calculates total
battery charge over a range of load
currents from 1mA to 100A would
require 100dB or more!
For many current monitoring circuits, the dynamic range is limited at
the high end by the maximum input
voltage of the current sense ampliier,
usually speciied between 100mV and
500mV for integrated high side current
sensing ampliiers. At the low end, input offset voltage limits the resolution.
VOS can be >1mV for many available
integrated circuits, resulting in a
dynamic range of 40dB–50dB, which
is inadequate for many systems. The
resolution is further degraded over
temperature as the input offset can
drift signiicantly.
The LTC6102 solves this problem
by providing a maximum offset voltage of 10µV with a drift of less than
50nV/°C. The maximum input voltage
of the part is 2V, giving a dynamic range
of 106dB and a minimum resolution
of 10µV. In simple terms, this allows
a system to measure currents from
1mA to 200A without changing gain
14
or over-ranging. Current sense circuits
that use the LTC6102 can easily be designed to provide high precision while
accommodating temporary current
surges or dropouts. This allows more
accurate end-of-charge calculation
and improved overall reliability.
Don’t Need Dynamic Range?
Trade in Dynamic Range for
Reduced Power Loss
If you don’t need to measure a large
range of currents, the built-in dynamic range of the LTC6102 allows
the use of very low value resistors.
Reducing value of the sense resistor
translates directly to improved power
dissipation.
For instance, only 40dB of dynamic
range is required for a system that
must measure currents from 1A to
100A. Nevertheless, if a sense ampliier
with 1mV input offset is used, then
the maximum sense voltage must be
no less than 100mV. At 100A, this
dissipates 10W in the sense resistor.
For accurate resistance at this high
dissipation, a large, expensive custom
sense resistor may be required, as well
as a heat sink. The system must also
be designed to provide the additional
10W, plus it must dissipate the resulting heat effectively.
If, however, the LTC6102 is used
for this current measurement, then
the maximum sense voltage can be
reduced to 1mV without degrading
performance. In fact, the low drift of
the LTC6102 can provide improved
precision over temperature when
compared to other solutions. At the
same time, dissipation is reduced to
100mW, a 99% reduction in power
dissipation in the sense resistor, signiicantly simplifying or eliminating
thermal design requirements.
Figure 2 shows the dynamic range
vs power dissipation for 1A, 10A and
100A loads. Each line represents a
ixed current. Dynamic range and
power dissipation are optimized by adjusting the value of the sense resistor
(RSENSE). The sense resistor extremes
are illustrated in the igure. It is easy
DANGER! Lethal Potentials Present — Use Caution
ISENSE
VSENSE
–
500V
+
RSENSE
+IN
L
O
A
D
V–
–INS
RIN
100Ω
–INF
+ –
DANGER!!
HIGH VOLTAGE!!
V+
VREG
0.1µF
OUT
LTC6102
51V
BZX884-C51
M1
BAT46
VOUT
M1 AND M2 ARE FQD3P50 TM
ROUT
VOUT =
• VSENSE = 49.9 VSENSE
RIN
M2
ROUT
4.99k
2M
Figure 4. Simple 500V current monitor
Linear Technology Magazine • October 2007
DESIGN FEATURES L
to adjust the circuit performance using available sense resistor values.
Dynamic range is the ratio between
the maximum voltage across the
sense resistor and the input offset of
the LTC6102, while power dissipation
is the power dissipated in the sense
resistor at the listed current.
ILOAD
VSENSE
–
V+
RSENSE
L
O
A
D
+
ROUT
4.99k
VOUT =
ROUT
• VSENSE = 249.5VSENSE
RIN
the maximum output current of the
LTC6102. If RIN = 10kΩ, then the input
voltage can be as large as 10V. The
gain is still set by ROUT/RIN, so either
gain or attenuation may be chosen to
allow the input signal to be translated
to a useful output signal.
Simple and Flexible Design
The high precision and wide dynamic
range of the LTC6102 are just the tip
of the iceberg. A collection of features
make the part easy to use, robust and
lexible for many applications.
Wide Supply Range
The LTC6102 is speciied for operation from 4V to 60V, and survives 70V
supplies. The LTC6102HV is speciied
for operation up to 100V, with a maximum of 105V. In addition, just a few
fC =
1
2 • π • ROUT • COUT
–INF
OUT
0.1µF
LONG WIRE
ADC
ROUT
COUT
REMOTE ADC
Figure 6. Remote current sense with simple noise filter
Linear Technology Magazine • October 2007
VOUT
LTC2433-1
TO µP
*PROPER SHUNT SELECTION COULD ALLOW
MONITORING OF CURRENTS IN EXCESS OF 1000A
Figure 5. 10A current sense with 10mA resolution and 100mW maximum dissipation
V+
LTC6102
OUT
1µF
5V
VREG
LOAD
VREG
0.1µF
V+
LTC6102HV
–INS
V–
–INF
+ –
V–
RIN–
–
5V TO 105V
–INS
TIE AS CLOSE TO RIN AS POSSIBLE
+IN
RIN
20Ω
+IN
Precision Level Translation
Unlike many application-specific
current sensing ampliiers, the architecture of the LTC6102 is similar
to standard operational ampliiers.
The design includes high impedance
inputs and external feedback as well
as low input offset. This allows the
LTC6102 to be used in a variety of
voltage ampliication circuits as well
as current sensing applications. Because of its inherent level translation
capability, the LTC6102 can amplify
a wide variety of signals while simultaneously rejecting the common-mode
component.
Figure 3 shows a level translation
circuit that ampliies a voltage signal.
The LTC6102 mirrors the input voltage onto RIN, which is then translated
to ROUT. It is important to note that
in this circuit the supply pin of the
LTC6102 is tied to the negative terminal of the input signal. Both input
pins are within a few microvolts of the
supply pin, so the input voltage may
exceed the full scale input range of
the LTC6102 without introducing an
error in the output. This circuit works
as long as the current through RIN,
deined as VIN/RIN, does not exceed
+
1mΩ*
external components can increase the
operating voltage to several hundred
volts or more without a loss of precision (Figure 4).
High Impedance Inputs
Unlike current-steering type sense
amps that have input bias currents
of several microamperes, the LTC6102
has <100pA input bias, allowing measurement of very small currents.
Simple, Flexible Gain Control
The gain of the LTC6102 can be set
with two external resistors. Gain error
is limited only by these external components, not poorly speciied internal
resistors or saturation voltages. The
external input resistor allows a wide
choice of gains, as well as control of
input and output impedances. For example, choosing a small input resistor
allows large gain with relatively small
output impedance, reducing noise
and making it easier to drive an ADC
without additional buffering.
Open-Drain Output
Additional lexibility and performance
are provided by the open drain output.
With no internal pull down device, the
minimum output voltage is not limited
by a saturation voltage, so the output
can drive all the way to ground. The
output can also be referred to a voltage
above ground simply by connecting
the output resistor to that voltage.
The sensing circuit can be physically
15
L DESIGN FEATURES
located far from the ADC without losing accuracy due to the resistance of
the long output wire. The output can
also be cascoded for additional levelshifting capability.
High Speed
The LTC6102 can support signals up
to 200kHz, allowing the monitoring
of fast-changing load currents. High
speed also allows the LTC6102 to settle
quickly after load transients, providing
uninterrupted precision.
Fast Response Time
Protection circuitry must often react
within microseconds to avoid system
or load damage during fault conditions. The LTC6102 can respond to
an input transient in 1µs.The output
signal may then be used to turn off
a MOSFET pass device or turn on a
load protection circuit before system
damage occurs.
Kelvin Input
The copper traces on the PC Board add
to RIN, creating a gain error that drifts
0.4%/°C. By connecting –INS very near
to RIN, this effect is minimized, so very
small (1Ω or less) input resistors may
be used. Small input resistors allow
large gains with relatively small output impedance. Reducing the effect of
parasitic series resistance also helps
maintain large dynamic range, even
with relatively large input resistors.
LTC850, continued from page 2
to operate with a main input voltage
approaching the programmed output
voltage, its duty cycle can be as high
as 97%.
Dual Output, 5A Regulator
with DCR Sensing
Figure 3 shows the schematic for a
500kHz, 2-output regulator requiring no sense resistors. By using the
inductor’s DC resistance as the current
sense element, the application dissipates as little power as possible—at full
load current, eficiency is well above
90%, as Figure 4 shows.
16
All That and Small Size, Too
Today’s applications don’t just require
precision; they also need it in the smallest package possible. In order to meet
that demand, the LTC6102 is available
in a 3mm × 3mm DD package, which
requires no more board than a SOT-23.
Where space is not such a premium,
or where a leaded package is desired,
the LTC6102 is also available in an
8-lead MSOP package.
Applications
Figure 5 depicts a simple current
sensing circuit. RSENSE converts the
load current to a sense voltage. The
LTC6102 applies a gain of 249.5 and
shifts the level of the signal from the
positive supply to ground. The sense
resistor value may be chosen to maximize the dynamic range by setting a
large maximum sense voltage (VSENSE),
or to limit power dissipation by choosing a smaller value. The high gain is
made possible by both the Kelvin input,
which allows the use of a small input
resistor with little gain error, and the
very low input offset, which produces
less than 2.5mV error at the output.
The small input resistor allows ROUT to
be set to 4.99k, which is small enough
to be compatible with high resolution
converter inputs. The addition of an
LTC2433-1 is a simple way to convert
the result.
For systems that are subjected to
electrical interference, or for remote
sensors, a capacitor may be placed
Dual Output, 15A Regulator
with Sense Resistors
Figure 5 shows the schematic for an
eficient 400kHz, 2-output regulator.
Figure 6 shows that this circuit’s core
occupies less than a square inch on a
2-layer board. Peak inductor current
is limited to 25A by the maximum current sense threshold looking across
the sense resistor (50mV / 2mΩ).
Taking inductor ripple current into
account, the output current limit is
around 20A for each channel. Higher
load current will cause the LTC3850
to protect the power stage using current foldback.
across ROUT to filter the output,
reducing noise and high frequency
interference (Figure 6). This adds a
simple pole to the output without
affecting the DC result. In remote sensing, the LTC6102 should be placed in
the sensor location, and the output can
be run long distances to a converter.
Since the output is current, not voltage, there is no loss in the wire. The
output resistor and capacitor should
be placed at the processor end of the
wire to reduce noise and ensure accuracy.
Conclusion
Many current sensing applications
can beneit from a high side sense
method. High side current sensing
circuits must be able to work at high
voltages determined by the supply
range, even under fault conditions, and
must usually level-shift the signal to
ground or another reference level. They
must accomplish these tasks while
preserving the precision and accuracy
of the signal. The LTC6102 zero-drift
current sensing ampliier offers the
highest precision DC speciications.
Wide supply range, low input offset
and drift, accurate gain, fast response,
and simple conigurability make the
LTC6102 and LTC6102HV ideal for
many current sensing applications.
For a complete guide to current
sense applications, visit www.linear.
com/currentsense. L
Conclusion
The LTC3850 delivers copious features
in small packages. Available in 4mm ×
4mm 28-pin QFN (0.4mm lead pitch),
4mm × 5mm 28-pin QFN (0.5mm lead
pitch), or 28-pin narrow SSOP, it
can run at high eficiency using DCR
sensing and Burst Mode operation.
Tracking, strong on-chip drivers,
multiphase operation, and external
sync capability ill out its menu of
features. Ideal for notebook computers, PDAs, handheld terminals and
automotive electronics, the LTC3850
delivers multiphase power to mission
critical applications. L
Linear Technology Magazine • October 2007
DESIGN FEATURES L
Pushbutton On/Off Controller Provides
µProcessor Reset Monitor and Input
by Victor Fleury
Supply Monitoring
Introduction
System designers often grapple with
ways to debounce and control the
on/off pushbutton of portable devices.
Traditional debounce designs use
discrete logic, lip-lops, resistors and
capacitors. Some designs require an
onboard microprocessor to monitor the
pushbutton, but this puts a burden on
the microprocessor—if it hangs up, all
device on/off control is lost. Also, in
high voltage, multicell battery applications the low voltage circuits require
an LDO power supply. In the end,
what should be a simple monitoring
circuit consumes an oversized share
of the space and complexity of the
system. Plus, its draw on the power
budget is high even when the system
is off, since the microprocessor must
keep awake, constantly watching the
pushbutton.
The LTC2953 pushbutton on/off
controller with voltage monitoring
alleviates the headaches of discrete
implementations and provides a
self-contained alternative to microprocessor based pushbutton monitoring.
The LTC2953 integrates all the lexible
timing circuits needed to debounce the
on/off pushbutton of portable systems
and provides a simple yet powerful
interface that allows for controlled
power up and power down.
The part also includes input and
output supply monitors. A power fail
comparator issues an early warning
when it detects a low battery condition,
while a UVLO comparator prevents
a user from applying system power
from a dead battery (or low supply).
Additionally, an adjustable single
supply supervisor provides a 200ms
reset output delay after the monitored
supply rises above the programmed
voltage.
The LTC2953’s wide input voltage
range (2.7V to 27V) is designed to
operate from single-cell to multicell
Linear Technology Magazine • October 2007
VIN
+
8.4V
100k
2150k
UVLO
23.2k
VIN
DC/DC
SHDN
EN
499k
LTC2953-1
PFI
VOUT
100k
VM
100k
100k
196k
100k
ON/OFF
PB
RST
PFO
INT
KILL
GND
PDT
RST
GPIO
INT SYSTEM
LOGIC
KILL
1µF
tPDT = 6.4 SECONDS
Figure 1. A complete pushbutton and voltage monitoring
system is easy to set up with the LTC2953-1.
PB, UVLO AND KILL
IGNORED
PB
tDB, ON
tKILL, ON BLANK
EN
(LTC2953-1)
KILL
DO NOT CARE
SYSTEM SETS KILL HIGH
Figure 2. Timing diagram shows a pushbutton-controlled system power on.
PB OR UVLO
SHORT PULSE
tDB, OFF
INT
tINT, Min
KILL
EN
(LTC2953-1)
SYSTEM SETS
KILL LOW
SYSTEM
POWER OFF
Figure 3. Timing diagram for normal power off sequence
17
L DESIGN FEATURES
battery stacks, thus eliminating the
need for a high voltage LDO. The part’s
features allow the system designer to
turn off power to all circuits except
the LTC2953, whose very low quiescent current (14µA typical) extends
battery life. The device is available in
a space saving 12 lead 3mm × 3mm
DFN package.
PB OR UVLO
LONG PULSE
16 CYCLES
PDT
tPD, Min
tPDT
EN
(LTC2953-1)
Orderly Power On
The pushbutton input of the LTC2953
controls the logic state of the open
drain enable output. Figure 1 shows
the EN output of the LTC2953-1 driving a DC/DC converter. To turn on
system power, the pushbutton input
must be debounced (held low continuously for at least 32ms). See the timing
diagram shown in Figure 2. Note that
once power has been enabled, the
system must set the KILL input high
within 512ms.
Figure 4. Timing diagram for forced power off, in the
case where the user must bypass system logic control.
+
8.4V
VIN
LTC2953-1
DEBOUNCE
UVLO
COMPARATOR
R14
2150k
EN
UVLO
–
VTH = 5.4V
DEBOUNCE
AND DELAY
50mV
Orderly Power Off
The LTC2953 provides two ways to
manually turn off system power: issuing an orderly power off request,
and forcing an immediate power off.
An orderly power off involves a simple
push and release of the on/off button.
For instance, for the circuit in Figure 1, if an end user is using an MP3
player, he presses and releases the
on/off button, which subsequently
drives the INT output low for a minimum of 32ms. The system logic that
monitors the LTC2953’s INT output
then initiates various pre-powerdown and housekeeping tasks, and
asserts KILL low when all is well.
The LTC2953 then shuts down the
DC/DC converter—turning off system
power. See the timing diagram shown
in Figure 3.
The other type of shutdown is a
manual reset. This allows the user to
force power off if the system logic or
µP fails to respond to the interrupt
signal. To do so, the end user presses
and holds the pushbutton down. The
length of time required to force a power
down is given by a ixed internal 64ms
delay plus an adjustable power down
timer delay. The adjustable delay is set
by placing an optional external capacitor on the PDT pin. See Figure 4.
18
INT
R13
23.2k
0.5V
+
PFI
R12
196k
PFO
–
VTH = 6.04V
4mV
0.5V
+
POWER FAIL
COMPARATOR
Figure 5. De-glitched UVLO comparator monitors battery stack
UVLO
0.5V
SUPPLY
GLITCH
0.55V
tDB, OFF
INT
UVLO
tINT, Min
LOW SUPPLY CONDITION
0.5V
tDB, OFF
INT
tPD, Min + tPDT
EN
LOW SUPPLY
LOCKS OUT
ENABLE
Figure 6. Low supply initiates system power down and locks out enable
Linear Technology Magazine • October 2007
DESIGN FEATURES L
Table 1. Pushbutton product family
Part Number
Supply
Voltage (V)
Supply
Current (µA)
ON
Timer
OFF
Timer
Kill Timer
Comments
Package
LTC2950
2.7 to 26
6
Adj
Adj
1024ms
Active high enable output (LTC2950-1)
Active low enable output (LTC2950-2)
TSOT-8
DFN-8
LTC2951
2.7 to 26
6
128ms
Adj
Adj
Active high enable output (LTC2951-1)
Active low enable output (LTC2951-2)
TSOT-8
DFN-8
LTC2952
2.7 to 28
25
Adj
Adj
Extendable
Pushbutton PowerPath controller
with system monitoring
TSSOP-20
QFN-20
LTC2953
2.7 to 27
14
32ms
Adj
Pushbutton controller with supply
monitor, UVLO and power fail
comparators
DFN-12
Adj
Interrupt logic for menu driven
applications.
Active high enable output (LTC2954-1)
Active low enable output (LTC2954-2)
TSOT-8
DFN-8
LTC2954
2.7 to 26
6
Power Fail Comparator Issues
Low Supply Warning
The LTC2953 provides an uncommitted power fail comparator that can
serve as the irst warning of a decaying battery or a low supply. The PFO
output is driven low when the PFI
input voltage drops below 0.5V. This
comparator provides real time supply information and does not affect
the functionality of the enable and
interrupt outputs. A system designer
can use the power fail comparator to
identify the source of a power down interrupt request: the pushbutton or the
UVLO. If the PFO output is low when
the interrupt output is asserted, then
the UVLO input initiated the power
down request (see Figure 5).
Adj
5.4V for an indeinite length of time,
the LTC2953 automatically shuts off
system power. See the Figure 6 timing
diagram.
supply drops below a predetermined
adjustable level, the LTC2953 does not
allow system power on (see Figure 6
timing diagram).
UVLO Locks Out
Pushbutton Input
Pushbutton Controlled
Supply Sequencing
The LTC2953 prevents a user from
turning on system power with a dead
battery or low supply. If system power
is off and the voltage on the UVLO input
is below 0.5V, the pushbutton input is
ignored. This means that if a battery or
The circuit in Figure 7 uses the
LTC2953-2 to sequence three supply
rails. Power on supply sequencing
begins by pressing the pushbutton
for 32ms. This asserts the EN output
low, which turns on the V1 supply.
continued on page 42
V1
3.3V
3.3V
R5
100k
VIN
UVLO Comparator Rejects
Short Supply Glitches
The application shown in Figure 5
monitors a 2-cell Li-Ion battery stack.
The UVLO comparator has glitch immunity to prevent short spikes on
the supply line from issuing a power
down request. All glitches shorter than
32ms are ignored. If the battery voltage drops below 5.4V for longer than
32ms, however, the LTC2953 asserts
the interrupt output for a minimum
of 32ms. When both INT and PFO
are driven low, this alerts the system
logic that a signiicant battery glitch
has occurred. For cases where the
battery voltage falls and stays below
Linear Technology Magazine • October 2007
R3
866k
EN
VM
PB
R9
100k
LTC2953-2
VTH = 2.66V
ON/OFF
R2
200k
VIN
DC/DC
#1
SHDN VOUT
RST
VTH = 2.01V
R15
604k
V2
2.5V
PFI
R16
200k
VIN
DC/DC
#2
PFO
SHDN VOUT
V3
1.8V
Figure 7. Pushbutton controlled supply sequencing
19
L DESIGN FEATURES
LED Driver Yields 3000:1 True Color
PWM Dimming with Any Buck, Boost
or Buck-Boost Topology from a Wide
3V–40V Input Range
by Xin Qi
Introduction
High power LEDs are quickly expanding their reach as a light source for
TV projection, scanners, and various
automotive and avionic products.
All require a constant LED current,
whether in buck, boost, buck-boost
or SEPIC conigurations. Pulse Width
Modulation (PWM) is the preferred
dimming method for these LED systems to preserve LED color over a wide
dynamic range of light intensities.
The LT3518 is a highly integrated
2.3A full-featured LED driver capable of providing 3000:1 True Color
PWM™ dimming ratio in a variety of
topologies for high power LED driver
applications.
The LT3518 features a 45V power
switch, 100mV high side current
PWM
PWM
OSC
OSC
SW
SW
TD
TD = 200ns
Figure 1. Regular LED driver timing diagram
Figure 2. LT3518 timing diagram
sense and accurate open LED protection. It combines a traditional voltage
feedback loop and a current feedback
loop to operate as a constant current
and/or constant voltage source. The
programmable soft-start limits inrush
current during startup, preventing
input current spikes. The LT3518’s
wide operating input range of 3V to 40V
makes it ideally suitable for automotive
applications. The 10:1 analog dimming range further extends the total
dimming range to 30,000:1. A PMOS
disconnect switch driver is integrated
to improve the transient response to
the PWM control signal. The programmable operating frequency of 250kHz
to 2.5MHz allows optimization of the
external components for eficiency or
component size. To reduce switching
noise interference, the LT3518 is synchronizable to an external clock.
LED ARRAY
RSENSE(EXT)
CFILT(EXT)
PMOS
PVIN
R1
MAIN SWITCH
DRIVER
R2
Q1
MAIN
SWITCH
NMOS
M1
PWM
LED
DRIVER
PWM
Figure 3. External PMOS disconnect switch driver for a conventional LED driver
LED ARRAY
CFILT(EXT)
RSENSE(EXT) PMOS
PVIN (VISP)
VISP
PMOS DRIVER
MAIN SWITCH
DRIVER
Q1
MAIN
SWITCH
VISP – 7V
LT3518
PWM
Figure 4. LT3518 internal PMOS driver
20
Highly Effective PWM
Dimming Control
Alignment of Internal Clock and
External PWM signal
Most LED drivers operate with an
independent, free-running internal
oscillator. Each switching cycle begins
when the internal oscillator transitions
from high to low. When PWM dimming,
the switch is turned off when the PWM
signal is low. After the PWM signal is
driven high, the switch has to wait
for the next oscillator high-low transition to turn on, as shown in Figure 1.
The turn on delay varies from 0 to
one full oscillator cycle, which limits
Linear Technology Magazine • October 2007
DESIGN FEATURES L
L1
8.2µH
VIN
8V TO 16V
SHDN VIN
PWM
PWM
SYNC
SYNC
TGEN
C1
2.2µF
CTRL
D1
SW
ISP
LT3518
ISN
VREF
TG
CTRL
FB
VC
ONE WIRE CONNECTION FOR
LED STRING. THE OTHER SIDE
OF LED STRING CAN BE
RETURNED TO GROUND ANYWHERE.
RSENSE
330mΩ
RT SS
M1
R1
1M
GND
LED2
C4
0.1µF
300mA
RT
16.9k
R2
30k
C3
0.1µF
C2
6.8µF
LED1
LED8
C1: KEMET C1206C225K2RAC
C2: TDK C5750X7R1H685M
C3, C4: MURATA GRM21BR71H104KA01B
D1: ZETEX ZLLS2000TA
L1: TOKO B992AS-8R2N
LEDS: LUXEON I (WHITE)
M1: ZETEX ZXMP6A13GTA
Figure 5. 1-wire boost 300mA LED driver with LED open protection
the achievable PWM dimming ratio.
This extra cycle becomes an obstacle
when high PWM dimming ratios are
required.
The LT3518 adopts a new timing
scheme, illustrated in Figure 2, to run
the converter. Instead of using a freerunning oscillator, the LT3518 aligns
the internal oscillator to the external
PWM signal. When the PWM signal
is low, the internal clock is disabled.
The PWM rising edge wakes up the
internal oscillator with a ixed 200ns
delay. In this manner, the LT3518
has a fast response to the PWM input
signal, thus improving the achievable
PWM dimming ratio.
PMOS Disconnect Switch Driver
Recent LED driver designs disable all
internal loads to the VC pin during
the PWM low period, which preserves
the charge state of the VC pin on the
external compensation capacitor. This
feature reduces the transient recovery
time, further increasing the achievable PWM dimming ratio. However, to
achieve the best PWM dimming ratio
for a buck/buck-boost mode LED
driver, other ICs still rely on several
additional external components to
drive a PMOS disconnect switch. As
The LT3518’s wide operating
input range of 3V to 40V
makes it ideally suitable for
automotive applications.
shown in Figure 3, a typical PMOS
disconnect switch driver consists of
an NMOS transistor and a level shift
resistor network formed by R1 and R2.
This kind of PMOS driver must juggle
L1
4.3µH
VIN
8V TO 16V
SHDN VIN
PWM
TGEN
Figure 6. PWM dimming waveform for Figure 5
at 120Hz PWM frequency and VIN = 10V
Linear Technology Magazine • October 2007
C5
0.22µF
300mA
LT3518
ISP
VREF
RSENSE
330mΩ
CTRL
C1
2.2µF
ISN
SYNC
C4
0.1µF
2µs/DIV
R1
3.92M
SW
R2
124k
PWM
PWM
5V/DIV
IL
1A/DIV
D1
FB
VC
ILED
200mA/
DIV
the tradeoffs between fast transient
response and high power consumption. The diverse input voltage and
LED voltage combinations also make
the level shifter design dificult.
In contrast, the LT3518 incorporates a PMOS driver inside, which
can transition a 1nF gate capacitance
PMOS switch in 200ns with a small
holding current, typically 600µA. In
this way, the LT3518 simpliies board
layout, reduces the bill of material,
and avoids the dilemma of trading
off the power consumption for a fast
transient response. Additionally, the
LT3518 includes an internal level
shifter to ensure the that the TG pin
TG
RT SS
GND
M1
C2
4.7µF
RT
6.04k
2MHz
C3
0.1µF
C1: KEMET C0806C225K4RAC
C2: KEMET C1206C475K3RAC
C3, C4: MURATA GRM21BR71H104KA01B
C5: MURATA GRM21BR71H224KA01B
D1: ZETEX ZLLS2000TA
L1: TOKO B992AS-4R3N
LEDS: LUXEON I (WHITE)
M1: ZETEX ZXMP6A13GTA
Figure 7. Buck-boost LED driver for automotive applications
21
L DESIGN FEATURES
PWM
5V/DIV
ILED
200mA/
DIV
VIN
3.3V
IL
1A/DIV
500ns/DIV
Figure 8. 3000:1 PWM dimming waveform of
application circuit of Figure 7 at 120Hz PWM
frequency and VIN = 12V.
Applications
1-Wire High PWM Dimming Boost
LED Driver
Many LED drivers feature high side
current sensing that enables the parts
to function as a 1-wire current source.
To improve PWM dimming ratio in
boost coniguration, those LED drivers typically rely on a low side NMOS
disconnect switch, unfortunately
limiting the 1-wire operation. On the
contrary, the unique internal PMOS
driver of the LT3518 makes 1-wire operation feasible in boost coniguration
while keeping a high PWM dimming
ratio. Figure 5 shows the LT3518
driving eight 300mA LEDs in boost
coniguration. This setup only needs
to provide 1-wire for the top side of the
LED string, while the other side of the
LED string can be returned to ground
anywhere. Figure 6 shows a 1000:1
PWM dimming waveform captured by
using this setup.
Buck-Boost PWM LED Driver
For an application in which the VIN
and VOUT ranges overlap, a buckboost topology is preferred. To make
the LT3518 with a low side switch
function as a buck-boost converter,
the LED current should be returned
to VIN. Thus, the LEDs see a voltage
of VOUT -VIN. Figure 7 depicts a buckboost PWM LED driver for automotive
applications. In this setup, the single
C3
10µF
M1
C2
2.2µF
L1
15µH
1.5A
D1
ISN TG
ISP
VIN
SW
C1
2.2µF
SHDN
VREF
CTRL
FB
PWM
PWM
SS
LT3518
SYNC
RT
TGEN VREF VC GND
RT
16.9k
1MHz
C4
0.1µF
is 7V or less below ISP pin. The internal PMOS driver can also be used to
implement fault protection. When a
fault is detected (e.g., an input surge),
the LED array will be disconnected
and protected by pulling down the
PWM input.
22
RSENSE
68mΩ
PVIN
24V
C5
0.1µF
C1: KEMET C0805C225K4RAC
C2: MURATA GRM31MR71E225KA93
C3: MURATA GRM32DR71E106KA12B
C4, C5: MURATA GRM21BR71H104KA01B
D1: ZETEX ZLLS2000TA
L1: TOKO B992AS-150M
LEDS: LUXEON K2 (WHITE)
M1: ZETEX ZXMP6A13GTA
Figure 9. Buck mode 1.5A LED driver
battery input voltage is able to vary
from 8V to 16V. The 6.04kΩ R T resistor
sets the system up for 2MHz switching,
which permits a higher PWM dimming ratio than the standard 1MHz
switching frequency. The 3000:1 PWM
dimming ratio shown in Figure 8 is
achieved at 120Hz PWM frequency.
High Current Buck
PWM LED Driver
The LT3518 features a 2.3A switch,
which makes it capable of driving
1.5A LEDs in buck coniguration.
Special attention should be paid to
the internal power consumption when
driving high current LEDs. Both high
switching frequency and high power
input voltage (PVIN) tend to cause
high power consumption and heat
up the silicon. With 1MHz switching
frequency and 24V PVIN, the circuit
shown in Figure 9 can provide 1000:1
PWM dimming ratio as shown in the
waveforms in Figure 10.
When a high power input voltage
drives a few LEDs in buck coniguration, open LED protection should be
considered. Unlike the boost coniguration, the output voltage needs to be
level-shifted to a signal with respect
to ground as illustrated in Figure 11.
In this manner, the unique constant
voltage loop of the LT3518 can regulate the output voltage of the buck
coniguration at the predeined value,
thus protect LEDs.
Conclusion
The LT3518 is a high current, high
voltage and high accuracy LED driver
offering high PWM dimming ratios a
variety of topologies. Its versatility,
simplicity and reliability make it very
attractive in most LED applications.
The LT3518 is available in the tiny footprint QFN UF16 package and leaded
FE16 package. It provides a complete
solution for both constant-voltage and
constant-current applications. L
+
PWM
5V/DIV
R1
RSEN(EXT)
VOUT
ILED
1A/DIV
LT3518
–
Q1
LED
ARRAY
FB
LI
1A/DIV
R2
2µs/DIV
Figure 10. 1000:1 PWM dimming waveform of
the application circuit of Figure 9 at 120Hz
PWM frequency.
Figure 11. Open LED protection
setup for buck configuration
Linear Technology Magazine • October 2007
DESIGN FEATURES L
White LED Driver and OLED Driver
with Integrated Schottkys and Output
Disconnect in 3mm × 2mm DFN
by Alan Wei
Introduction
Linear Technology Magazine • October 2007
VIN
3V TO 5V
4.7µF
15µH
0.47µF
15µH
16V
24mA
1µF
CAP1 SW1
10Ω
SW2 CAP2 VOUT2
VIN
10µF
LT3498
LED1
20mA
CTRL1
GND1
GND2
CTRL2
FB2
2.21MΩ
OFF ON
SHUTDOWN
AND
CONTROL
OFF ON
SHUTDOWN
AND
DIMMING
CONTROL
Figure 1. Li-Ion to six white LEDs and an OLED display
ing battery life in application modes
where the LED driver is temporarily
disabled.
Figure 1 shows a typical application
driving 6 LEDs and an OLED. Figures
2 and 3 show the eficiency of the LED
driver and OLED driver respectively.
Features
LED Driver High Side Sense
The LED driver of the LT3498 features
a unique high side LED current sense
that enables the part to function as
a 1-wire current source. This allows
the cathode side of the bottom LED
in the string to be returned to ground
anywhere, resulting in a simple 1wire LED connection. Traditional
LED drivers use a grounded resistor
to sense LED current, requiring a
2-wire connection to the LED string
since the ground must return to the
part ground. In addition, high side
sense allows the LT3498 LED driver to
operate in unique applications (buck
mode or buck boost mode, where the
LED string is returned to the input)
where traditional LED drivers cannot
be used.
80
80
75
75
EFFICIENCY (%)
EFFICIENCY (%)
70
70
65
60
400
VIN = 3.6V
VOUT2 = 16V
350
EFFICIENCY
FOR VOUT2
65
250
60
200
55
150
50
55
50
45
0
5
10
20
40
0.1
100
POWER LOSS
FROM VOUT2
LED CURRENT (mA)
1
10
OLED CURRENT (mA)
Figure 2. Efficiency of the
LED driver in Figure 1
Figure 3. Efficiency of the
OLED Driver in Figure 1
15
300
POWER LOSS (mW)
The LT3498 is a dual boost converter
featuring both an LED driver and
OLED driver in a single 3mm × 2mm
DFN package. It provides an internal
power switch and Schottky diode for
each converter as well as an output
disconnect PMOS for the OLED driver.
Both converters can be independently
shutdown and dimmed. This highly
integrated power solution is ideal for
dual display portable electronics with
tight space constraints.
The LED driver is designed to drive
up to six white LEDs in series from a LiIon cell. It is capable of regulating the
LED current in a series coniguration,
providing equal brightness throughout
an LED string regardless of variations
in forward voltage drop. The 2.3MHz
switching frequency allows the use
of small external components and
keeps switching noise out of critical
wireless and audio bands. It features
a high side LED current sense, which
allows the converter to be used in a
wide variety of application conigurations. The LED driver also contains
internal compensation, open-LED
protection, analog or PWM controlled
dimming, a 32V power switch and a
32V Schottky diode.
The OLED driver of the LT3498
features a novel control technique resulting in low output voltage ripple as
well as high eficiency over a wide load
range. During operation, the converter
controls power delivery by varying both
the peak inductor current and switch
off time. The off time is not allowed to
exceed a ixed level, guaranteeing that
the switching frequency stays above
the audio band. This unique control
scheme makes it ideal for noise sensitive applications such as MP3 players
and mobile phones. When operated by
itself, the OLED driver consumes a
low 230µA quiescent current, extend-
50
0
100
23
L DESIGN FEATURES
240
16
14
VOUT2 VOLTAGE (V)
200
SENSE VOLTAGE (mV)
18
T = 25°C
T = –50°C
T = 125°C
160
120
80
12
10
8
6
4
40
0
2
0
500
1000
VCTRL1 (mV)
1500
0
2000
Figure 4. LED sense voltage vs CTRL1 pin voltage
0
500
1500
1000
CTRL2 VOLTAGE (V)
Figure 5. VOUT2 voltage vs CTRL2 pin voltage
VIN
3V TO 5V
RSENSE1
10Ω
PWM
10kHz TYP
CIN
1µH
L1
15µH
LT3498
R1
100kΩ
2000
CAP1 SW1
VIN SW2 CAP2 VOUT2
CTRL1
LT3498
COUT1
1µF
C1
0.1µF
LED1
CTRL1 GND1
GND2
CTRL2
FB2
Q1
Si2304BDS
Figure 6. Filtered PWM dimming
5V
100k
PWM
FREQ
0V
Figure 7. Li-Ion to four white LEDs with direct PWM dimming
24
sets the LED current (see Figure 4).
The CTRL2 pin regulates the VOUT2
voltage in a similar fashion as shown
in Figure 5.
Filtered PWM dimming works similarly to DC voltage dimming, except
that the DC voltage input to the CTRL
pins comes from an RC-iltered PWM
signal. The corner frequency of the R1
and C1 should be much lower than
the frequency of the PWM signal for
proper iltering. Filtered PWM dimming
is shown in Figure 6.
10000
PWM DIMMING RANGE
Dimming & Shutdown Control
The LT3498 features a single pin
shutdown and dimming control for
each converter. To shutdown the
LT3498, simply pull both control pins
below 75mV. To enable each individual
converter, increase the control pin
(CTRL1 for the LED Driver and CTRL2
for the OLED Driver) voltage to 125mV
or higher. On the LED side, the LED
current can be set by modulating the
CTRL1 pin. On the OLED side, the
VOUT2 voltage can be set by modulating
the CTRL2 pin. There are three types
of dimming methods available in the
LT3498: DC voltage dimming, iltered
PWM signal dimming and direct PWM
dimming.
The LED current and VOUT2 voltage
are proportional to the DC voltages at
the CTRL1 and CTRL2 pins, respectively. To dim the LEDs or lower the
VOUT2 voltage, reduce the voltage on the
CTRL1 and CTRL2 pins. The dimming
range of the LED driver extends from
1.5V at the CTRL1 pin for full LED current down to 125mV. The CTRL1 pin
directly controls the regulated sense
voltage across the sense resistor that
PULSING MAY BE VISIBLE
1000
100
10
1
10
100
1000
PWM FREQUENCY (Hz)
Figure 8. LED dimming range
vs PWM dimming frequency
10000
Direct PWM dimming is typically
used because it achieves a much wider
dimming range compared to using a
iltered PWM or a DC voltage. Direct
PWM dimming uses a MOSFET in
series with the LED string to quickly
connect and disconnect the LED
string. Figure 7 displays direct PWM
dimming of the LEDs in a Li-Ion to 4
white LED application. A PWM signal is
applied to the CTRL pin and MOSFET
where the PWM signal controls both
the turn-on and turn-off of the part.
Figure 8 shows the linearity of PWM
dimming across a range of frequencies.
The available dimming range depends
on the settling time of the application
and the PWM frequency used. The
application in Figure 7 achieves a
dimming range of 250:1 using a 100Hz
PWM frequency.
OLED Driver
PMOS Output Disconnect
The low-noise boost converter of the
LT3498 features a PMOS output disconnect switch. This PMOS switch is
continued on page 8
Linear Technology Magazine • October 2007
DESIGN FEATURES L
Light Up 12 LEDs from a Single-Cell
Li-Ion Battery via Highly Integrated
3mm × 2mm Dual-LED-String Driver
by Ben Chan
Introduction
The LT3497 is a dual step-up converter capable of driving up to 12
white LEDs from a single-cell Li-Ion
input. The device is capable of driving
asymmetric LED strings with independent dimming and shutdown control,
perfect for driving backlight circuits
in battery-powered portable devices,
such as cellular phones, MP3 players,
PDAs, digital cameras, and portable
GPS devices.
The LT3497 directly regulates
LED current, providing consistent
brightness for all LEDs regardless
of variations in their forward voltage
drop. Important features including
internal compensation, open-LED protection, DC/PWM dimming control, a
35V power switch and a 35V Schottky
diode are all integrated into the part,
making the LT3497 LED driver an
ideal solution for space-constrained
portable devices. In addition, the
2.3MHz switching frequency allows the
use of tiny inductors and capacitors.
Figure 1 shows a typical 12-whiteLED application. Figure 2 shows the
eficiency of the circuit.
The LT3497 features a
unique high side LED
current sense that enables
the part to function as a
1-wire current source—the
cathode side of the bottom
LED in the string can
be returned to ground
anywhere, allowing a simple
1-wire LED connection.
Linear Technology Magazine • October 2007
Dimming & Shutdown Control
The LT3497 features single pin shutdown and dimming control for each
converter. The LED current in the
two drivers can be set independently
by modulating the CTRL1 and CTRL2
pins. There are three different types
of dimming methods: DC voltage dimming, iltered PWM signal dimming
and direct PWM dimming.
VIN
3V TO 5V
L2
15µH
L1
15µH
SW1
VIN
C1
1µF
RSENSE1
10Ω
C3
1µF
SW2
CAP1
CAP2
RSENSE2
10Ω
LT3497
C2
1µF
LED1
LED2
CTRL1 GND CTRL2
OFF ON
OFF ON
SHUTDOWN
AND DIMMING
CONTROL 1
SHUTDOWN
AND DIMMING
CONTROL 2
C1, C2: TAIYO YUDEN GMK212BJ105KG
C3: TAIYO YUDEN LMK212BJ105MG
L1, L2: MURATA LQH32CN150K53
Features
Figure 1. Li-Ion powered driver for twelve white LEDs
80
240
VIN = 3.6V
6/6LEDs
200
SENSE VOLTAGE (mV)
75
EFFICIENCY (%)
High Side Sense
The LT3497 features a unique high
side LED current sense that enables
the part to function as a 1-wire current
source—the cathode side of the bottom
LED in the string can be returned to
ground anywhere, allowing a simple
1-wire LED connection. Traditional
LED drivers use a grounded resistor to
sense LED current requiring a 2-wire
connection to the LED string. High side
sense moves the sense resistor to the
top of the LED string. In addition, high
side sense allows the LT3497 to operate in unique applications (Buck-Mode
or Buck-Boost Mode) where traditional
LED drivers cannot be used.
70
65
60
55
50
160
120
80
40
0
5
10
15
LED CURRENT (mA)
20
Figure 2. Efficiency of the circuit in Figure 1
0
0
500
1000
VCTRL (mV)
1500
2000
Figure 3. LED sense voltage
versus CTRL pin voltage
25
L DESIGN FEATURES
3V TO 5V
PWM
10kHz TYP
LT3497
R1
100k
C1
0.1µF
1µF
L1
15µH
CTRL1,2
SW1
L2
15µH
VIN
SW2
CAP1
Figure 4. Filtered PWM Dimming
RSENSE1
10Ω
1µF
The LED currents are proportional
to the DC voltages at the CTRL1 and
CTRL2 pins, so DC voltage dimming
is achieved by reducing the voltage
on the CTRL pin. The dimming range
of the part extends from 1.5V at the
CTRL pin for full LED current down to
100mV. The CTRL pin directly controls
the regulated sense voltage across the
sense resistor that sets the LED current (see Figure 3).
Filtered PWM dimming works
similarly to DC voltage dimming except
that the DC voltage input to the CTRL
pins comes from an RC-iltered PWM
signal. The corner frequency of the R1
and C1 should be much lower than
the frequency of the PWM signal for
proper iltering. Filtered PWM dimming
is shown in Figure 4.
Direct PWM dimming is typically
used because it achieves a much wider
dimming range compared to using a
iltered PWM or a DC voltage. Direct
PWM dimming uses a MOSFET in
series with the LED string to quickly
connect and disconnect the LED
string. Figure 5 displays direct PWM
dimming in a Li-Ion to a 4-and-4 white
LED application. A PWM signal is applied to the CTRL pin and MOSFET
where the PWM signal controls both
VIN
CAP1
RSENSE1
10Ω
100k
5V
0V
0V
PWM
FREQ
PWM
FREQ
100k
10000
PWM DIMMING RANGE
NORMALIZED SENSE VOLTAGE (%)
100
10
1
PULSING MAY BE VISIBLE
1000
100
10
1
0.1
0.1
10
1
PWM DUTY CYCLE (%)
10
100
Figure 6. Linearity of PWM Dimming of
Figure 5 at 100Hz
the turn-on and turn-off of the part.
Figure 6 shows the linearity of PWM
dimming. The available dimming range
depends on the settling time of the application and the PWM frequency used.
The application in Figure 5 achieves a
dimming range of 250:1 using a 100Hz
PWM frequency. Figure 7 shows the
C3
1µF
Applications
Li-Ion Powered Driver
for 12 White LEDs
Figure 1 highlights the LT3497’s
impressive input and output voltage
range. This circuit is capable of driving two strings of six LEDs each with
20mA of constant current. As shown
VIN = 3.6V
2/6LEDs
75
RSENSE2
10Ω
OFF ON
SHUTDOWN
AND DIMMING
CONTROL 2
C1, C2: TAIYO YUDEN GMK212BJ105KG
C3: TAIYO YUDEN LMK212BJ105MG
L1: MURATA LQH32CN100K53
L2: MURATA LQH32CN150K53
Figure 8. Li-Ion to a 2-LED and 6-LED Display
10000
available dimming ranges for different
PWM frequencies.
80
CAP2
100
1000
PWM FREQUENCY (Hz)
Figure 7. Dimming Ratio Range vs Frequency
SW2
LED1
LED2
CTRL1 GND CTRL2
26
5V
Figure 5. Li-Ion to eight white LEDs with direct PWM dimming.
LT3497
OFF ON
SHUTDOWN
AND DIMMING
CONTROL 1
Q2
Si2318DS
C2
1µF
70
EFFICIENCY (%)
SW1
1µF
Q1
Si2318DS
L2
15µH
L1
10µH
RSENSE2
10Ω
LED1
LED2
CTRL1 GND CTRL2
VIN
3V TO 5V
C1
1µF
CAP2
LT3497
65
60
55
50
45
0
5
10
15
20
LED CURRENT (mA)
Figure 9. Efficiency of the circuit in Figure 8
Linear Technology Magazine • October 2007
DESIGN FEATURES L
in Figure 1, the circuit works from a
single Li-Ion (3V) battery or 5V wall
adapter. Figure 2 shows eficiency with
a 3.6V input.
Li-Ion to a 2-LED and
6-LED Display
Figure 8 (Buck-Boost/Boost coniguration) shows a white LED driver used
to backlight two displays: a 6-LED
main and a 2-LED sub display. This
design generates a constant 20mA in
each white LED string from a Li-Ion
(3V~4.2V) or 5V adapter input. Two
independent dimming and shutdown
controls (CTRL1 and CTRL2) simplify
power management and extend battery life. Figure 9 shows the eficiency
of the circuit.
Conclusion
The LT3497 is a dual channel white
LED driver capable of driving up to 12
white LEDs from a single cell Li-Ion
input. The device features 35V internal power switches, internal Schottky
diodes, DC or PWM dimming control,
open LED protection and optimized
internal compensation. The LT3497
is an ideal solution for a wide range of
applications including multipanel LCD
backlighting, camera lash or space
constrained portable applications
such as cellular phones, MP3 players,
PDAs and digital cameras. L
LT080, continued from page 5
pin serving as ballast to equalize the
currents. PC trace resistance in milliohms/inch is shown in Table 3. Only
a tiny area is needed for ballasting.
Figure 6 shows two devices with a
small 10mΩ ballast resistor, which at
full output current gives better than
80% equalized sharing of the current.
The external resistance of 10mΩ (5mΩ
for the two devices in parallel) only
adds about 10mV of output regulation
drop at an output of 2A. Even with the
1V output, this only adds 1% to the
regulation.
Thermal Performance
Two LT3080 3mm × 3mm QFN devices
are mounted on a double sided PC
board. They are placed approximately
1.5 inches apart and the board is
mounted vertically for convection cooling. Two tests were set up to measure
the cooling performance and current
sharing of these devices.
The irst test was done with approximately 0.7V input-to-output
differential and a 1A load per device.
This setup produced 700mW dissipation in each device and a 2A output
current. The temperature rise above
ambient is approximately 28°C and
both devices were within ±1°C of each
other. Both the thermal and electrical
sharing of these devices is excellent.
The thermograph in Figure 7 shows
the temperature distribution between
these devices, where the PC board
reaches ambient within about 0.5in
from the devices.
Figure 8 shows what happens when
the power is increased to 1.7V across
each device. This produces 1.7W disLinear Technology Magazine • October 2007
Figure 8. Thermograph shows a 65°C rise for two regulators, each
dissipating 1.7W from a 1.7V input-to-output differential at 2A load.
sipation in each device and a device
temperature of about 90°C, about 65°C
above ambient. Again, the temperature matching between the devices is
within 2°C, showing excellent tracking
between the devices. The board temperature drops to about 40°C within
0.75 inches of each device.
While 95°C is an acceptable operating temperature for these devices, this
rise is in a 25°C ambient environment.
For higher ambient temperatures, the
temperature rise must be controlled
to prevent the device temperature
from exceeding 125°C. A 3-meterper-second airlow across the devices
decreases the device temperature
by about 20°C, providing a margin
for higher operating ambient temperatures. Also, this example is for a
2-layer board. A 4-layer board would
provide better power dissipation.
Conclusion
The LT3080’s breakthrough design and
high performance DC characteristics
allows it to be paralleled for high current all-surface-mount applications.
It is also adjustable to zero output,
an impossible feat with a traditional
3-terminal adjustable linear regulator.
It is optimized for new circuit applications and all-surface-mount system
assembly techniques—especially high
performance, high density circuit
boards. L
27
L DESIGN FEATURES
Low Offset 2-Wire Bus Buffer Provides
Capacitance Buffering, Stuck Bus
Recovery, and Tolerates High VOL
by John Ziegler
Introduction
High availability computing, networking and data storage systems employ
system management buses such as I2C
or SMBus to monitor system health.
These simple serial buses allow system controllers to monitor parameters
such as temperature and voltage,
read vital product information from
individual cards, and make changes
to the system, such as controlling fan
speed. As these systems increase in
complexity, several implementation
issues arise with the system management bus.
First, each additional device on the
bus adds a capacitive load. The bus
capacitance of a large system makes
meeting rise time speciications very
dificult. While a strong pull-up resistor
can reduce the rise time, the penalty
is increased VOL and decreased noise
margin. Second, some devices that can
communicate via I2C or SMBus have a
VOL that is near or above the maximum
allowed by the standards. Third, power
cannot be cycled whenever a new card
is added to the system. Finally, since
any device can hold the bus low, each
additional device increases the chance
of a stuck bus caused by a single
confused device.
The LTC4309 solves all of these
problems by acting as a buffer between two physically separate 2-wire
buses. The input side of the LTC4309,
SDAIN and SCLIN, connects to one
2-wire bus (backplane), while the
output side, SDAOUT and SCLOUT,
connects to the other bus (I/O card).
The LTC4309 provides bidirectional
buffering, keeping the backplane
and card capacitances isolated from
each other.
The LTC4309’s low, pull-up independent offset voltage allows multiple
devices to be put in series while meeting VOL and maintaining noise margin.
28
The LTC4309 solves many
I2C and SMBus problems by
acting as a buffer between
two physically separate
2-wire buses.
The input side of the
LTC4309, SDAIN and SCLIN,
connects to one 2-wire bus
(backplane) while the output
side, SDAOUT and SCLOUT,
connects to the other bus
(I/O card). The LTC4309
provides bidirectional
buffering, keeping the
backplane and card
capacitances isolated
from each other.
A large system can be broken into many
smaller buses by inserting LTC4309s
throughout the system, reducing the
capacitance of each electrically isolated bus. The LTC4309’s rise time
accelerators help to further reduce
the rise time.
The LTC4309 has connection circuitry that connects and passes a logic
low even if the input voltage is above the
bus speciication VOL. The low, pull-up
independent offset reduces the impact
to the VOL of buffering the bus.
Since the LTC4309’s SDA and SCL
pins are high impedance when inactive
or powered down, the LTC4309 can
be inserted into a live bus without
corrupting the bus. The LTC4309’s
capacitance buffering feature also
isolates the capacitance of the card
from the live bus during, and after,
insertion.
Finally, the LTC4309’s stuck bus
detection circuitry can detect when
a bus is stuck in a low condition and
disconnect the stuck portion of the
system while attempting to recover
the stuck bus.
Circuit Operation
Start Up
A block diagram of the LTC4309 is
shown in Figure 1. When the LTC4309
irst receives power on its VCC pin, either during power up or live insertion,
it starts in an undervoltage lockout
(UVLO) state, ignoring any activity on
the SDA or SCL pins until VCC rises
above 2.0V (typ). This is to ensure that
the LTC4309 does not try to function
until it has enough voltage to do so.
During this time, the 1V precharge circuitry is active and forces 1V through
100k nominal resistors to minimize the
worst-case voltage differential these
pins see at the moment of connection, thus minimizing the disturbance
caused by the I/O card.
Once the LTC4309 comes out of
UVLO and the ENABLE input is high,
it monitors both the input and output
sides for either a stop bit or bus idle
condition to indicate the completion
of data transactions. When transactions on both sides of the LTC4309
are complete, the back-to-back buffers shown in Figure 1 (referred to
below as “connection circuitry”) are
activated, joining the SDA and SCL
buses on the input side with those
on the output. Once the connection
is made, the READY pin is released,
allowing it to pull up and signal that
the connection is complete. READY
remains high as long as the connection circuitry is active. If the ENABLE
pin is grounded, the LTC4309 does
not connect and I/Os remain in a
high impedance state until ENABLE
is pulled high.
Linear Technology Magazine • October 2007
DESIGN FEATURES L
Connection Circuitry
When the connection circuitry is activated, the functionality of the SDAIN
and SDAOUT pins, as well as SCLIN
and SCLOUT, are identical. When an
external device pulls any SDA or SCL
pin below a threshold of 1.65V (for
VCC > 2.9V) or 1.35V (for VCC < 2.9V)
the LTC4309 detects a low and pulls
the other side down to a voltage that
is 60mV above the forced voltage.
This low offset is practically independent of bus voltage level and pull-up
resistance. The LTC4309 remains
connected until the input and output
are above 0.6V and it senses a rising
edge on both the input and output or
until one side is above the 0.45 • VCC
connection threshold. The LTC4309’s
connection circuitry ensures that the
input and output enter a logic high
state only when all devices on both
sides of the LTC4309 have released
the bus and the pull-ups have pulled
the bus high. This important feature
ensures that clock stretching, clock
arbitration and the acknowledge protocol always work, regardless of how the
devices in the system are connected
to the LTC4309.
Another key feature of the connection circuitry is that, while it joins the
two buses together, it still maintains
electrical isolation between them, thus
providing capacitance buffering for
both sides. With the LTC4309’s low
offset and tolerance to devices having high VOL, multiple devices can be
cascaded on a single bus. This allows
larger systems to be divided into many
smaller, less capacitive and therefore
faster buses. The LTC4309 is capable
of driving capacitive loads ranging from
0pF to more than 1000pF on all of its
data and clock pins.
Stuck Bus Detection and Recovery
Slave devices on a bus use the clock
signal to sample the data. Occasionally, devices become confused and get
stuck in a low state, causing a “stuck”
VCC2
VCC
VCC2
8mA
VCC
8mA
CONNECT
IBOOSTSDA
IBOOSTSDA
SDAIN
SDAOUT
100k
SLEW RATE
DETECTOR
SLEW RATE
DETECTOR
100k
PRECHARGE
VCC2
VCC
PC
CONNECT
8mA
100k
IBOOSTSCL
PC
CONNECT
8mA
100k
CONNECT
IBOOSTSCL
SCLIN
SCLOUT
SLEW RATE
DETECTOR
SLEW RATE
DETECTOR
ACC
30ms
TIMER
+
–
1.65V/1.4V
1.35V/1.1V
+
–
1.65V/1.4V
1.35V/1.1V
FAULT
DISCEN
ENABLE
1.65V/1.4V
1.35V/1.1V
+
–
1.65V/1.4V
1.35V/1.1V
+
–
1.4V/1.3V
+
–
LOGIC
IBOOSTSCL
IBOOSTSDA
CONNECT
READY
PC CONNECT
CONNECT
GND
UVLO
95µs
DELAY
Figure 1. Block diagram of LTC4309. The input side of the LTC4309, SDAIN and SCLIN, connects to one 2-wire bus (backplane), while the output
side, SDAOUT and SCLOUT, connects to the other bus (I/O card). The LTC4309 provides bidirectional buffering, keeping the backplane and card
capacitances isolated from each other.
Linear Technology Magazine • October 2007
29
L DESIGN FEATURES
bus. The LTC4309 monitors both the
data and clock buses independently
for a stuck bus condition. If either
data or clock is in a low state for more
than 30ms, the LTC4309 determines
that the bus is “stuck.” The LTC4309
signals a fault condition by pulling
the FAULT and READY pins low and
disables the connection circuitry, disconnecting the stuck bus and freeing
the portion of the bus that is not stuck.
At this time, the LTC4309 attempts
to free the stuck bus by generating
up to 16 clock pulses on SCLOUT.
Once the 16 pulses are completed, or
the clock pulses terminate due to the
bus becoming unstuck, a stop bit is
generated to clear the bus for further
communication. If a master wants to
force reconnection of the bus after the
LTC4309 has disconnected the bus
due to a fault condition, the master
can pull the ENABLE pin low and immediately high again. This resets the
30ms timer and forces the LTC4309
to reconnect.
The LTC4309’s stuck bus recovery
feature is illustrated in Figure 2. After
SDAOUT has been held low for 30ms,
SDAIN IS RELEASED
WHEN FAULT IS DETECTED
SDAIN
FAULT
BACKPLANE
FAULT GOES HIGH
WHEN FAULT IS CLEARED
LTC4309 RELEASES SDAOUT
TO GENERATE STOP BIT
SDAOUT
SCLOUT
SCLOUT IS TOGGLED
TO UNSTICK BUS
Figure 2. The stuck bus recovery feature of the LTC4309 disconnects
stuck buses and uses auto clocking to recover the stuck bus.
the LTC4309 detects the stuck bus.
The LTC4309 pulls the FAULT pin
low, and releases the SDAIN bus. The
SCLOUT pin is then toggled at 8.5kHz
in an attempt to free the bus. In this
example, after 11 clock edges the bus
becomes unstuck and the FAULT
pin is released. Note that SDAOUT
temporarily goes high at the same
time that FAULT goes high, but this
is not visible in the igure due to the
time scale and due to the LTC4309
quickly pulling SDAOUT back low so
that it can generate a Stop Bit on the
BACKPLANE
CONNECTOR
VCC
FAULT GOES LOW
TO SIGNAL FAULT
CARD
CONNECTORS
I/O PERIPHERAL CARD 1
C1
0.01µF
C2
0.01µF
VCC2
R1
10k
R2
10k
R3
10k
bus. The LTC4309 holds SDAOUT low
for 125µs, then releases SDAOUT to
generate the Stop Bit.
If automatic disconnection is not
desired, this feature can be disabled
by connecting the DISCEN pin to GND.
The LTC4309 still monitors both sides
for a stuck bus condition and pulls
FAULT low if a fault occurs, but does
not disconnect the bus or attempt
to free the stuck bus. A master can
disconnect the stuck bus manually
by pulling the LTC4309’s ENABLE
pin low. This forces the connection
R4
10k
VCC2
SDA
SDAIN
SCL
SCLIN
FAULT
FAULT
READY
READY
ENABLE
ENA1
R7
10k
VCC
LTC4309
R5
10k
R6
10k
DISCEN
SDAOUT
CARD 1_SDA
SCLOUT
CARD 1_SCL
ACC
GND
I/O PERIPHERAL CARD N
C3
0.01µF
C4
0.01µF
VCC2
SDAIN
SCLIN
VCC
LTC4309
R8
10k
R9
10k
DISCEN
SDAOUT
CARD N_SDA
SCLOUT
CARD N_SCL
ACC
FAULT
READY
ENABLE
ENAN
R10
10k
GND
Figure 3. The LTC4309 in a live insertion and capacitance buffering application
30
Linear Technology Magazine • October 2007
DESIGN FEATURES L
VCC
VCC2
2.7k
2.7k
VCC
VCC4
VCC3
VCC2
2.7k
2.7k
VCC2
LTC4309
VCC
2.7k
2.7k
LTC4309
VCC
VCC2
2.7k
2.7k
LTC4309
SDA1
SDAOUT
SDAIN
SDAIN
SDAOUT
SDAOUT
SDAIN
SDA4
SCL1
SCLOUT
SCLIN
SCLIN
SCLOUT
SCLOUT
SCLIN
SCL4
Figure 4. The LTC4309 provides level translating, and allows cascading of multiple buffers while meeting system VOL requirements.
circuitry to disconnect the inputs
from the outputs, and put the I/O
pins in a high impedance state. Once
the master has cleared the stuck bus,
the LTC4309 ENABLE pin can be
pulled high. When the bus is idle, the
LTC4309 reconnects the input to the
output as described previously.
Rise Time Accelerators
The ACC pin controls the state of the
rise time accelerators. If the ACC pin
is tied to GND, all four accelerators
are activated. To disable the input
side accelerators only, tie the ACC
and VCC2 pins to GND. Connect the
ACC pin to VCC to disable all four rise
time accelerators. When activated,
the rise time accelerators switch in
8mA of slew limited pull-up current
at VCC = 3.3V during bus rising edges
to quickly slew the SDA and SCL lines
once their DC voltages exceed 0.8V and
the initial rise rate on the pin exceeds
0.8V/µs. The slew limiting is achieved
by monitoring the rising edge; if the
edge is rising faster than 1V/10ns,
the pull-up current is reduced. This
helps prevent signal integrity issues in
lightly loaded systems where a strong
pull-up could make the rising edge
fast enough to create transmission
line relections on the bus.
Live Insertion and Removal,
and Capacitance Buffering
Application
The application shown in Figure 3
highlights the live insertion/removal
and the capacitance buffering features
of the LTC4309. Assuming that a staggered connector is available, make
ground, VCC and VCC2 the longest pins
to guarantee that SDAIN and SCLIN
receive the 1V pre-charge voltage
before they connect. Make SDAIN
and SCLIN medium length pins to
ensure that they are irmly connected
Linear Technology Magazine • October 2007
while ENABLE is low. Make ENABLE
the shortest pin and connect a weak
resistor from ENABLE to ground on
the I/O card. This ensures that the
LTC4309 remains in a high impedance state while SDAIN and SCLIN
are making connection during live
insertion. During live removal, having
ENABLE disconnect irst ensures that
the LTC4309 enters a high impedance
state in a controlled manner before
SDAIN and SCLIN disconnect.
Note that if an I/O card were plugged
directly into the backplane, the card
capacitance would add directly to the
backplane capacitance, making rise
and fall time requirements dificult
to meet. Inserting a LTC4309 on the
edge of the card, however, isolates the
card capacitance from the backplane.
The LTC4309 drives the capacitance
of everything on the card, and the
backplane must drive only the capacitance of the LTC4309. As more
I/O cards are added and the system
grows, placing a LTC4309 on the
edge of each card breaks what would
be one large, unmanageable bus into
several manageable segments, while
still allowing all segments to be active
at the same time. If breaking the bus
up further is desired, the LTC4309’s
low offset and high VOL tolerance allows cascading of multiple devices.
Moreover, the LTC4309’s rise time
accelerators provide strong pull-up
currents during bus rising edges, so
that even heavily loaded bus lines
meet system rise time requirements
with ease.
Level Translator and
Cascading Applications
The LTC4309’s very low offset,
typically 60mV, allows cascading of
multiple devices while still meeting
VOL speciications. Figure 4 illustrates
an application where three LTC4309s
have been used to break a bus into
four isolated buses. The total offset
of the cascaded devices is approximately 180mV. This feature can be
used in conjunction with the level
translating feature of the LTC4309 and
each isolated section of the bus can
operate off a different supply voltage.
The LTC4309 functions for voltages
ranging from 2.3V to 5.5V on VCC and
1.8V to 5.5V on VCC2.
Simplified 8-Pin Option
in the LTC4307
The LTC4307 is a simpliied 8-pin version of the LTC4309. For the LTC4307,
the DISCEN, ACC, FAULT and VCC2
pins are removed. The rise time accelerators and stuck bus recovery
are always enabled. Since there is no
FAULT pin, the READY pin should
be monitored to determine if a fault
condition occurs.
Conclusion
The LTC4309 low offset buffer allows
I/O cards to be hot-plugged into live
systems and breaks one large capacitive bus into several smaller ones,
while still passing the SDA and SCL
signals to every device in the system.
The low, pull-up independent offset
allows cascading of multiple devices,
breaking the bus into smaller, less
capacitive sections. Slew limited rise
time accelerators further decrease the
rise time and allow the bus to operate
at higher frequencies, or with better
data integrity. Stuck bus recovery
helps maintain system integrity by
detecting and clearing stuck buses.
The LTC4309’s tolerance to high
VOL allows capacitance buffering on
buses with other devices that may not
meet VOL speciications. With these
features, the LTC4309 simpliies the
design process of complex 2-wire bus
systems. L
31
L DESIGN IDEAS
Compact and Versatile
Monolithic Synchronous Buck
Regulators Deliver 1.25A in Tiny
TSOT23, DFN and MS10 Packages
by Jaime Tseng
Introduction
Adding More Options
To meet industry demands to squeeze
more power from smaller packages
the LTC3564 monolithic synchronous
buck regulator provides 1.25A from a
tiny TSOT23-5 package. Its siblings,
the LTC3565 and LTC3411A, also
1.25A monolithic synchronous bucks,
come in 10-lead 3mm × 3mm DFN
and MS10 packages. The LTC3564’s
internal switching frequency is set at a
ixed 2.25Mhz to allow the use of tiny
inductors and ceramic output capacitors. Switching at this high frequency
does not compromise eficiency. In
Burst Mode operation, the LTC3564
only needs 20µA of quiescent current
and <1µA in shutdown. The internal
150mΩ power MOSFETs keep the
power dissipation low and eficiencies as high as 94% at maximum load
current.
The additional pins of the LTC3565
and LTC3411A give them a versatility
edge over the LTC3564. Both parts
can program their internal frequency,
synchronize to an external clock,
select the mode of operation among
Burst Mode operation, pulse-skipping, or forced continuous mode, and
provide a PGOOD indicator output.
For noise-sensitive applications,
pulse-skipping mode decreases the
output ripple noise at low currents.
Although not as eficient as Burst Mode
operation at light load, pulse-skipping
mode still provides high eficiency for
moderate loads. In forced continuous
DESIGN IDEAS
Compact and Versatile Monolithic
Synchronous Buck Regulators
Deliver 1.25A in Tiny TSOT23,
DFN and MS10 Packages ...................32
mode a steady operating frequency
is maintained at all load conditions,
making it easier to reduce noise and
RF interference—important for some
applications. In order to squeeze into
a TSOT23-5 package, the LTC3564
forgoes a few features such as PGOOD,
the ability to adjust the switching
frequency and the mode select. The
frequency and mode of operation are
internally set at 2.25MHz and Burst
Mode operation respectively.
All three devices employ a constant
frequency, current mode architecture
that operates from an input voltage
range of 2.5V to 5.5V and provides an
VIN
2.5V TO 5.5V
C1
22µF
VIN
SYNC/MODE
PGOOD
PVIN
SVIN
LTC3411A
C2
22µF
887k
SHDN/RT
SGND
VFB
PGND
324k
1000pF
VOUT
2.5V/1.25A
SW
ITH
13k
L1
2.2µH
412k
Jaime Tseng
Single-IC Converter Operates Buck
and Boost to Provide an Output that
is Within the Input Voltage Range .....34
Figure 1. Battery to 2.5V at 1.2A application of the LTC3411A
David Burgoon
VIN
2.5V TO 5.5V
Feature-Rich Monolithic Triple Buck
Regulator Supplies Up To 2.4A from
a 3mm × 3mm Package .....................35
C1
22µF
Kevin Soch
Single-Wire Camera LED Charge Pump
Allows Multiple Output Current Levels
With Single-Resistor Programmability
.........................................................37
VIN
SYNC/MODE
PGOOD
RUN
LTC3565
ITH
Mohammed H. Jafri
RT
Compact Controller is a
Basic Building Block for Wide Array
of DC/DC Conversion Solutions .........39
13k
1000pF
SVIN
L1
2.2µH
SW
1.3M
VFB
SGND
324k
PVIN
VOUT
2.5V/1.25A
C2
22µF
PGND
412k
Victor Khasiev and Hong Ren
Figure 2. Battery to 1.2V at 1.2A application of the LTC3565
32
Linear Technology Magazine • October 2007
DESIGN IDEAS L
3
CIN
22µF
CER
VIN
SW
4
100
1µH
22pF
COUT
22µF
LTC3564
5
VFB
RUN
GND
2
VOUT
1.8V
1
634k
316k
Figure 3. Battery to 1.2V at 1.2A application of the LTC3564
adjustable regulated output voltage
down to 0.6V (0.8V for LTC3411A),
which make them ideal for single-cell
Li-Ion or 3-cell NiCd and NiMH applications. The 100% duty cycle capability
for low dropout allows maximum energy to be extracted from the battery.
In dropout, the output voltage is determined by the input voltage minus
the voltage drop across the internal
P-channel MOSFET and the inductor
resistance.
The switching frequency of the
LTC3565 and LTC3411A can be set
between 400kHz and 4MHz with an
external resistor or synchronized to
an external clock. The LTC3411A is a
drop-in replacement for the popular
LTC3411, but with improved eficiency
at higher VIN and improved response
to fault conditions.
Adaptive Current Reversal
Comparator
In each of the parts, a patent pending
adaptive current reversal comparator
monitors the current reversal across
the synchronous switch. In discontinuous mode, to emulate the behavior of
an ideal diode, the synchronous switch
turns on when the inductor current is
positive and turns off when the inductor current is negative. Because the
comparator has a inite propagation
VOUT1
100mV/DIV
AC COUPLED
IL
500mA/DIV
VIN = 3.3V
40µs/DIV
VOUT1 = 2.5V
LOAD STEP = 250mA TO 1.2A
Figure 5. Load step response
Linear Technology Magazine • October 2007
delay, the inductor current trip point is
offset before zero. This offset depends
on the output voltage of the regulator
and the inductor value used on the
board. In the LTC3564, LTC3565 and
The LTC3564, LTC3565 and
LTC3411A employ a constant
frequency, current mode
architecture that operates
from an input voltage range
of 2.5V to 5.5V and provides
an adjustable regulated
output voltage down to 0.6V
(0.8V for LTC3411A), which
make them ideal for single
cell Li-Ion or 3-cell NiCd and
NiMH applications.
LTC3411A, the offset of the current
reversal comparator is automatically
adjusted for any output voltage and
inductor value to ensure the synchronous switch is always turned off at the
right inductor current value.
Fault Protection
All three parts are protected against
output short-circuit and output overdissipation conditions. The output can
90
80
EFFICIENCY (%)
VIN
2.5V
TO 5.5V
70
60
50
40
30
20
0
0.0001
0.1
1
10
be shorted to ground or VIN in any
mode without fear of damage. When a
VOUT short to VIN is removed the output
returns immediately to its regulated
output voltage if forced continuous
mode is selected. This allows the use
in a pre-biased application where
the output is held at higher than
the regulated output when the part
is shutdown. When there is a power
over-dissipation condition and the
junction temperature reaches 160°C,
the thermal protection circuit turns off
the power switches. Normal operation
does not resume until the part cools
off and the junction temperature drops
back to 150°C.
Conclusion
Three monolithic synchronous stepdown voltage regulators provide up
to 1.25A of output current in a tiny
footprint. The LTC3564, LTC3565
and LTC3411A also offer high switching frequency, high eficiency and a
number of versatile features that make
them an excellent choice for portable
applications. L
IL
500mA/
DIV
IL
500mA/
DIV
Figure 6. Operating waveforms
0.01
Figure 4. Efficiency vs load current for the
circuit of Figure 1 in various operating modes
SW
2V/DIV
400ns/DIV
0.001
LOAD CURRENT (A)
SW
2V/DIV
VIN = 3.3V
VOUT = 2.5V
L = 1µH
BURST MODE
PULSE SKIP
FORCE CONTINUOUS
10
VIN = 3.3V
VOUT = 2.5V
L = 2.2µH
1µs/DIV
Figure 7. Operating waveforms
33
L DESIGN IDEAS
Single-IC Converter Operates
Buck and Boost to Provide an Output
that is Within the Input Voltage Range
by David Burgoon
Introduction
Generating an output voltage that
is always above or below the input
voltage range can easily be handled
by conventional boost or buck regulators, respectively. However, when the
output voltage is within the input voltage range, as in many Li-Ion battery
powered applications requiring a 3V
or 3.3V output, conventional designs
fall short, suffering variously from low
eficiency, complex magnetics, polarity inversion and circuit complexity.
The LTC3785 buck-boost controller
facilitates a simple, eficient, low partscount, single-converter solution that is
easy to implement and does not have
any of the drawbacks associated with
conventional circuits.
tion (OVP) and a 2.7V–10V output
range.
The circuit produces seamless operation throughout the input voltage
range, operating as a synchronous
buck converter, synchronous boost
1
215k
127k
3
31.6k 2
215k
127k 470pF
4
42.2k 5
6
49.9k 7
8
9
15
14
Figure 1 shows a synchronous,
4-switch, buck-boost design that
provides a 3.3V, 3A output from a
2.7V–10V input—perfect for a Li-Ion
and/or loosely regulated wall adapter
input. The controller provides shortcircuit protection, offering a choice of
burp-mode or latch-off operation for
severe overload faults. Other features
include soft-start, overvoltage protec-
13
RUN/SS
VIN
FB
VCC
LT3785EMS
VC
ISVIN
VSENSE
VBST1
ILSET
TG1
CCM
SW1
RT
ISSW1
MODE
BG1
NC
VDRV
BG2
ISVOUT
ISSW2
SW2
VBST2
GND
Figure 2. Input-side and output-side switch
waveforms along with inductor current for
buck mode (10VIN)
4.7µF
VIN
2.7V TO 10V
22
CMDSH-3
21
20
0.22µF Q1A
FDS6894A
19
18
L1
2.2µH
TDK
RLF7030T
17
16
Q1B
VOUT
3.3V
3A
10
CMDSH-3
11
47µF
6.3V
12
0.22µF
Q2A
FDS6894A
Figure 1. Schematic of buck-boost converter using LTC3785
to provide 3.3V at 3A out from a 2.7V–10V source
VSW1
5V/DIV
IL1
2A/DIV
VSW2
5V/DIV
VSW2
5V/DIV
1µs/DIV
23
22µF
16V
Q2B
IL1
2A/DIV
VSW2
5V/DIV
TG2
24
25
VSW1
5V/DIV
IL1
2A/DIV
34
continued on page 6
2.2nF
VOUT
3.3V, 3A Converter Operates
from 2.7V–10V Source
VSW1
5V/DIV
converter, or a combination of the
two through the transition region. At
input voltages well above the output,
the converter operates in buck mode.
Switches Q1A and Q1B commutate
the input voltage, and Q2A stays
1µs/DIV
Figure 3. Input-side and output-side switch
waveforms along with inductor current for
boost mode (2.7VIN)
1µs/DIV
Figure 4. Input-side and output-side switch
waveforms along with inductor current for
buck-boost mode (3.8VIN)
Linear Technology Magazine • October 2007
DESIGN IDEAS L
Feature-Rich Monolithic Triple Buck
Regulator Supplies up to 2.4A from
a 3mm × 3mm Package
by Kevin Soch
Introduction
The triple output LTC3545 is a monolithic synchronous buck regulator
capable of supplying three independent voltage supply rails, each with
maximum output current of 800mA
and peak eficiency over 90%. The
3mm × 3mm QFN package and default
internal 2.25MHz switching frequency
allow for a simple and compact multiple power supply solution. The input
voltage range of 2.25V to 5.5V is perfect
for batteries and the output voltage is
resistor programmable down to 0.6V.
Features include selectable high eficiency Burst Mode operation or low
ripple pulse-skipping mode, soft-start,
power sequencing, and the option for
externally driven 1.0MHz to 3.0MHz
switching frequency.
High Level of Integration
Many of the external components required to operate a typical switching
regulator have been integrated into the
LTC3545. Internal loop compensation
The placement of the package pins
ensures the isolation of the sensitive
feedback pins and a logical and compact board layout, particularly with
respect to the power paths. Figure 1
is a photo of the LTC3545 demoboard
with the power components primarily
on the top. The feedback components
(not shown) reside on the bottom of the
board. Total circuit footprint for this
board is approximately 300mm2.
Power Sequencing Example
Figure 1. The LTC3545 is a compact solution
to the problem of multiple voltage supplies.
eliminates the need for external compensation resistors and capacitors,
integrated synchronous switches eliminate the need for external Schottky
diodes, and an integrated soft-start
function eliminates the need for external capacitors or control ramps.
Figure 2 shows the schematic of an
application providing three voltage
supply rails with power sequencing.
The outputs are externally programmed to 1.8V, 1.2V, and 1.5V. In
this application, PGOOD1 is connected
to the RUN2 pin and PGOOD2 is connected to the RUN3 pin. The power
on sequence is shown in Figure 3.
The soft-start feature prevents large
RUN1
5V/DIV
VOUT1
1V/DIV
VIN
2.25V TO 5.5V
GNDA
R8
500k
VOUT2
1V/DIV
C4
10µF
C5
10µF
R7
500k
VIN
RUN1
PVIN1
PGND1
VOUT3
1V/DIV
L2
1.5µH
SW2
C7
20pF
PGOOD1
RUN2
VFB2
R4
226k
PGOOD2
RUN3
VOUT1
1.8V
C1
10µF
L1
1.5µH
C6
20pF
R1
511k
SYNC/MODE
R3
226k
C2
10µF
L3
1.5µH
SW3
VFB1
VFB3
R2
255k
C8
20pF
VOUT3
1.5V
VOUT3
2mV/DIV
VOUT1
100mV/DIV
R6
200k
GNDA
Figure 3. PGOOD pins allow simple power
sequencing. Soft-start reduces inrush currents
and prevents output voltage overshoot.
VOUT2
2mV/DIV
LTC3545
SW1
1ms/DIV
VOUT2
1.2V
R5
301k
C3
4.7µF
PGND1
LOADSTEP1
500mA/DIV
200µs/DIV
Figure 2. A high level of integration minimizes the number of necessary external components.
Linear Technology Magazine • October 2007
Figure 4. Channel-to-channel transient
crosstalk is negligible.
35
L DESIGN IDEAS
inrush currents while charging the
output caps during startup, as well
as minimizing voltage overshoot when
starting into light loads.
For those applications requiring a
power good output on the third channel, the LTC3545-1 version of the part
substitutes a PGOOD3 output in place
of the MODE/SYNC pin. The option
of an external clock is not available
on this version, and the part enters
Burst Mode operation at light load
currents.
Minimal Channel Crosstalk
High Efficiency
A potential problem with multiple with Low Ripple
output regulators is the interaction
between channels when one of the
channels undergoes a load transient.
Figure 4 shows the response on channels 2 and 3 to a 0mA to 500mA load
step on channel 1. Channels 2 and
3 are each loaded at 400mA. In each
case, the crosstalk is on the order of
1mV to 2mV.
1
100
90
EFFICIENCY
80
0.1
EFFICIENCY (%)
SW
2V/DIV
VOUT
20mV
/DIV
VIN = 2.5V
VIN = 3.6V
VIN = 4.2V
60
50
0.01
40
POWER LOSS
30
IL
100mA
/DIV
20
10
0
0.0001
1µs/DIV
Figure 5. At low load currents, Burst Mode
operation improves efficiency without
degrading output voltage ripple.
TA = 25°C
VOUT = 2V
fOSC = 2.25MHz
SINGLE CHANNEL
Burst Mode OPERATION
POWER LOSS (W)
70
0.001
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
Figure 6. Burst Mode operation maintains
high efficiency at low load currents.
At low load currents, the LTC3545
operates in either pulse-skipping mode
or Burst Mode operation depending
on the state of the MODE/SYNC pin.
Though pulse-skipping mode exhibits
lower output ripple, the ripple in Burst
Mode operation is still quite low while
maintaining the added advantage of
better eficiency at the lightest loads.
The Burst Mode operation and Burst
Mode eficiency are shown in Figures
5 and 6.
Conclusion
The LTC3545 is a unique part with
tremendous flexibility. It greatly
simpliies system and board design
where multiple voltage supply rails
are needed without sacriicing the
features and performance found in
individual regulators. The LTC3545
is ideally suited for battery powered
applications where multiple or isolated
voltage rails are required and board
space is at a premium. L
LTC785, continued from page 4
36
100
10
90
EFFICIENCY
80
1
70
POWER
LOSS
60
50
40
BURST MODE
OPERATION
30
20
10
0
0.001
0.1
VIN = 2.7V
VIN = 3V
VIN = 4.2V
VIN = 10V
VIN = 2.7V
VIN = 3V
VIN = 4.2V
VIN = 10V
0.01
0.1
1
LOAD CURRENT (A)
POWER LOSS (W)
EFFICIENCY (%)
on, connecting L1 to the output. As
the input voltage is reduced and approaches the output, the converter
approaches maximum duty cycle on
the input (buck) side of the bridge, and
the output (boost) side of the bridge
starts to switch, thus entering the
buck-boost or 4-switch region of operation. As the input is reduced further,
the converter enters the boost region
at the minimum boost duty cycle.
Switch Q1A stays on, connecting the
inductor to the input, while switches
Q2A and Q2B commutate the output
side of the inductor between the output
capacitor and ground.
In boost mode, this converter has
the ability to limit input current and to
shut down and disconnect the source
from the output—two very desirable
features that a conventional boost
converter cannot provide. Figures 2,
3, and 4 show input-side and outputside switch waveforms along with
inductor current for buck (10VIN),
boost (2.7VIN), and buck-boost (3.8VIN)
modes of operation.
0.01
0.001
10
Figure 5. Efficiency in normal mode
and Burst Mode operation
limit. Even higher eficiencies are
possible by using a larger inductor
and better MOSFETs as they become
available. Eficiency at 10V in would
beneit from an inductor with a lowloss ferrite core, especially at light
loads. This circuit easily its in 0.6in2
with components on both sides of the
board. The curves show how Burst
Mode operation improves eficiency
at extremely light loads, dramatically
enhancing battery life in applications
such as memory that must maintain
housekeeping functions even when
the system is turned off.
95% Efficiency
Conclusion
Figure 5 shows eficiency in both
normal (not forced continuous conduction) and Burst Mode operation. Very
high eficiency of 95% is achieved at
typical loads. This level of performance
results in part from sophisticated
controller features including high side
drivers for N-channel MOSFETs and
RDS(ON) current sensing for current
The LTC3785 buck-boost controller
overcomes the deiciencies of traditional designs with a smooth-transition,
4-switch, single-IC solution. It is elegant in its simplicity, high in eficiency
and requires only a small number of
inexpensive external components.
The LTC3785 is available in a small
4mm × 4mm QFN package as well as
a 28-lead SSOP. L
Linear Technology Magazine • October 2007
DESIGN IDEAS L
Single-Wire Camera LED Charge
Pump Allows Multiple Output
Current Levels With Single-Resistor
Programmability
by Mohammed H. Jafri
Introduction
ENT
ILED
LOW
LOW
SHUTDOWN
LOW
HIGH
1029/RSETT
HIGH
LOW
2965/RSETF
HIGH
HIGH
3993/RSETF
Linear Technology Magazine • October 2007
90
Multiple Current Ratios
LED drivers often use external resistors to program LED current. The LED
current is related to the programming
resistor current through a ixed ratio.
By employing multiple current ratios,
the LTC3218 can be programmed for
80
70
60
50mA
150mA
300mA
50
40
2.9
3.1
3.3
3.5
3.7
3.9
4.1
4.3
4.5
VIN (V)
Figure 1. Efficiency vs VIN
for various LED currents
2.2µF
CP
2.2µF
DISABLED
DISABLED
CPO
ENF
ENABLED
ENT
ISETF
ENT
ILED
0
0
0 (SHUTDOWN)
0
1
100mA (TORCH)
1
0
290mA
1
1
390mA (FLASH)
4.7µF
LTC3218
ENABLED
ENF
CM
VIN
2.9V TO 4.5V
ILED
LED
AOT2015
GND
ISETT
10.2k
1%
Figure 2. Typical application, using a single resistor to program LED currents
2.2µF
CP
2.9V TO 4.5V
DISABLED
DISABLED
CM
VIN
2.2µF
Table 1. Output current modes
for all ENT and ENF settings
ENF
100
time, the LTC3218 features a built-in
timer. This timer shuts down the part
if it has been enabled in lash-mode
(ENF = HIGH) for more than 2 seconds.
The timer is reset by bringing the part
into shutdown and re-enabling it.
EFFICIENCY (%)
The number of features in cell phones
continues to grow, even as the phones
themselves physically shrink, driving a
need for space saving circuits to control
these features. The LTC3218 is such a
device. It can drive a white LED with
multiple current levels, requiring only
three 0603 ceramic capacitors and one
0402 resistor. Its low proile, 3mm ×
2mm, DFN package allows for an application circuit footprint of less than
30mm2, making it an ideal driver for a
cell phone camera lash. Additionally,
due to its single-wire, high side current
sensing design, only one high current
trace is required to run to the anode of
the LED. The cathode of the LED can
be grounded locally, eliminating the
need for a separate return trace. The
LTC3218 can operate from a single-cell
Li-Ion battery, with an input voltage
range of 2.9V to 4.5V.
The LTC3218 generates the regulated output voltage needed to maintain
the desired LED current. By remaining
in the current regulated, 1x mode for
as much of the battery voltage range
as possible, eficiency is maximized.
The LTC3218 steps up to 2x mode
only when needed. Figure 1 shows the
eficiency of the LTC3218 for various
current levels.
To protect the LED from experiencing high currents for long periods of
CPO
ENABLED
ENF
ENABLED
ENT
ILED
GND
ISETF
ENF
ENT
ILED
0
0
0 (SHUTDOWN)
0
1
VARIABLE BY PWM TO 100mA (TORCH)
1
0
290mA
1
1
390mA (FLASH)
4.7µF
LTC3218
RSETF
10.2k
LED
ISETT
R1
2k
1µF
R2
8.16k
R1 = ≥ 1kΩ
PWM
Figure 3. LED driver uses pulse-width modulation to implement dimming and brightness control
37
L DESIGN IDEAS
three different current levels using
a single programming resistor. The
current ratios are selected using the
ENT and ENF pins. Table 1 shows the
three different current ratios, and the
ENT/ENF settings required to select
them. RSETT refers to the resistor
connected between the ISETT pin and
GND, and RSETF refers to the resistor
connected between the ISETF pin and
GND. In the case where single-resistor programming is desired, the ISETT
and ISETF pins can be shorted together
and connected to a resistor to GND.
Figure 2 shows an example of this
coniguration, along with the resulting
output current levels.
LT498, continued from page 24
turned on when the part is enabled.
When the part is in shutdown, the
PMOS switch turns off, allowing the
VOUT2 node to go to ground. This type
of disconnect function is often required
for OLED applications.
Li-Ion Powered Driver
for Four White LEDs
and OLED display
Figure 9 highlights the LT3498’s simplicity and versatility. From a single
3mm × 2mm DFN, this circuit is caVIN
3V TO 5V
CAP1 SW1
Figure 3 shows how the LTC3218
can be conigured to control LED
brightness with just a few external
components. By pulse-width modulating the gate of M1, the reference
current in resistor R1 can be varied.
The maximum LED current is determined by:
ILED(MAX ) =
850 • 1.21V
R SETT
where RSETT = R1 + R2 and the onresistance of M1 is small compared to
RSETT. Resistor R1 should be greater
than 1kΩ to provide adequate isola-
pable of driving four LEDs in series,
with 20mA of constant current as well
as an OLED display. The eficiency for
the LED driver in Figure 9 is shown in
Figure 10. As shown above in Figure 1,
the circuit can operate from a single
Li-Ion battery (down to 3V) or 5V wall
adapter and drive up to six LEDs in
series at 20mA and an OLED display
at 16V, 24mA out.
Conclusion
The LT3498 is a dual output boost
converter that is capable of driving
CIN
4.7µF
L1
15µH
C1
1µF
Dimming and Brightness
Control
Conclusion
Due to its small size and low external parts count, the LTC3218 is
ideally suited for compact, camera
LED applications. Features such as
its single resistor programmability,
multiple current ratios and 2-second
lash timeout make the part simple to
use, without the need for complicated
control algorithms. Its low shutdown
current and high eficiency make it
perfect for situations where battery
power is at a premium. L
up to 6 white LEDs and an OLED
display from a single-cell Li-Ion input. The device features 32V internal
power switches, 32V internal Schottky
diodes, independent DC or PWM dimming control, open LED protection,
OLED output disconnect and internal
compensation. The LT3498 offers a
highly integrated, space-saving solution for a wide range of applications
including space-constrained and
noise-sensitive portable applications
such as cellular phones, MP3 players
and digital cameras. L
C2
0.47µF
L2
10µH
16V
24mA
SW2
VIN
tion between the 1µF capacitor and
the internal servo-ampliier.
CAP2 VOUT2
80
C3
10µF
75
RSENSE1
10Ω
20mA
LED1
CTRL1
GND1
GND2
OFF ON
SHUTDOWN
AND
DIMMING
CONTROL
CTRL2
FB2
RFB2
2.21MΩ
OFF ON
SHUTDOWN
AND
CONTROL
EFFICIENCY (%)
LT3498
70
65
60
55
50
0
5
10
15
20
LED CURRENT (mA)
CIN, C2: X5R OR X7R WITH SUFFICIENT VOLTAGE RATING
C1: TAIYO YUDEN GMK212BJ105KG
C3: TAIYO YUDEN TMK316BJ106ML
L1: MURATA LQH32CN150K53
L2: MURATA LQH32CN100K53
Figure 10. Efficiency of
the LED driver in Figure 9
Figure 9. Li-Ion to four white LEDs and an OLED display
38
Linear Technology Magazine • October 2007
DESIGN IDEAS L
Compact Controller is a Basic
Building Block for Wide Array of
DC/DC Conversion Solutions
by Victor Khasiev and Hong Ren
Introduction
CIN
2.2µF
x2
EFFICIENCY
80
48V Input,
3.3V, 3A Output Flyback
Figure 1 shows a nonisolated stepdown converter for telecom and
industrial applications with a 36V to
72V input range and a 3.3V, 3A output, impressive for such a compact
converter. Eficiency is over 85%,
resulting in low power loss.
4
1•
221k
Q2
D3
D1
70
2000
60
50
POWER LOSS
40
5•
2
51Ω
20
10
0
100
LOAD CURRENT (mA)
Figure 2. Efficiency of the converter
in Figure 1 peaks at 86%.
•
VOUT
3.3V
3A
7
8
COUT
100µF
6.3V
x3
9
10
D2
150pF
200V
VCC
ITH
Q1
NGATE
RUN/SS
220Ω
SW
FB
GND
15k
VIN = 72V
VIN = 60V 500
VIN = 48V
VIN = 36V
0
10000
SECONDARY
RTN
LTC3873
2.2nF
1500
1000
30
4.7µF
IPRG
2500
T1
PA1861NL
221k
PRIMARY
RTN
90
Applications
OPTIONAL
36V TO 72V
VIN
3000
100
POWER LOSS (mW)
One of interesting features of
this IC is its programmable
current limit. The current
sense voltage can be
set to 290mV, 110mV or
185mV. This feature allows
flexibility in MOSFET
selection. If a higher sensing
threshold is selected, the
circuit is less sensitive
to noise and PCB layout.
48V Input,
3.3V, 3A Output Isolated Flyback
Figure 3 shows an isolated application. In this case, feedback is provided
by the LT4430 optocoupler driver,
which controls the PWM via ITH pin
of LTC3873.
q No RSENSE™ eliminates the need
for current-sensing resistor.
q Programmable soft-start
q Adjustable current limit enables a
wide range of power MOSFETs
q Pulse-skipping mode maintains
constant frequency operation at
light loads.
q Extremely small packages:
2.8mm × 2.9mm 8-lead SOT-23
or 3mm × 2mm QFN.
EFFICIENCY (%)
The LTC3873 is a compact PWM
controller that can be used in boost,
lyback and SEPIC DC/DC converters.
Other features include:
q Wide input range, suitable
for telecom and industrial
applications
21.5k
0.068Ω
D1: PDZ6.8B
D2: UPS840
D3: BAS516
Q1: FDC2512
Q2: MMBTA42
0.1µF
12.4k
100pF
Figure 1. A nonisolated flyback converter
Linear Technology Magazine • October 2007
39
L DESIGN IDEAS
T1
PA1861NL
ISOLATION BARRIER
VIN
36V TO 72V
221k
OPT
4.7 F
100V
MMBTA42
OPT
PDZ6.8B
OPT
221k
4
1•
BAS516
5•
2
51
•
VOUT
3.3V
3A
7
8
100 F
6.3V
3
9
10
UPS840
FDC2512
2.2
LTC3873
1
2
3
4
IPRG
SW
ITH RUN/SS
FB
VCC
GND
GATE
2
8
7
1
BAT54CWT1G
0.068
6
3
5
1 F
OPT
4.7 F
1210
AND
0805
0.1 F
274
6.8k
NEC
PS2801-1
1
BAT760
4
3
BAS516
LT4430
1
2
3
2
VIN
OPTO
GND COMP
OC 0.6V FB
6
22nF
5
4
100k
330pF
3.01k
22.1k
0.47 F
2200pF
250V AC
Figure 3. Isolated converter can be controlled by the LT4430 optoisolator driver, which also provides soft-start and overshoot control.
9V–15V Input,
12V, 2A Output SEPIC
Figure 4 shows a SEPIC that converts
input voltages that can be higher or
lower than the output. The advantage
of a SEPIC over a lyback converter is
in the higher eficiency and lower EMI.
A SEPIC converter does not provide
isolation.
Adjustable Current Limit
One of interesting features of this IC
is programmable current limit. The
current sense voltage can be set to
T1
4.56µH
BH510-1009
BH ELECTRONICS
1
4
VIN
9V TO 15V
10µF
×3
+
100µF
20V 2
•
Conclusion
•
3
10µF
25V
UPS840
301Ω
+
100k
LTC3873
1
2
3
10nF
4
33.2k
11k
IPRG
SW
ITH
RUN/SS
FB
VCC
GND
NGATE
Si4840
8
47µF
16V
×3
10µF
16V
VOUT
12V
2A
The LTC3873 is a constant frequency,
current mode controller. It requires no
sense resistor and can be used in a
wide variety of applications as a boost,
lyback and SEPIC converter. L
7
6
5
4.7µF
0.1µF
Figure 4. A SEPIC converter for applications with higher power levels
and input voltages that can be higher or lower than the output voltage
40
290mV, 110mV or 185mV by tying
the IPRG pin to VIN, tying the IPRG pin
to GND or leaving it loating, respectively. This feature allows lexibility in
MOSFET selection. If a higher sensing
threshold is selected, the circuit is less
sensitive to noise and PCB layout. Keep
in mind that a higher sense voltage
results in higher power dissipation in
the MOSFET.
Want to know more?
Visit:
www.linear.com
or call
1-800-4-LINEAR
Linear Technology Magazine • October 2007
NEW DEVICE CAMEOS L
New Device Cameos
DC/DC Converter with
Selectable Frequency Modes
in a 2mm × 3mm DFN
The LTC3543 is a high eficiency
600mA monolithic step-down switching regulator intended for low power
applications such as Lithium-Ion battery powered devices. It operates within
a 2.5V to 5.5V input voltage range and
has three different frequency modes
of operation.
Eficiency is extremely important in
battery powered applications, and the
LTC3543 keeps eficiency high with an
automatic, power saving Burst Mode
operation, which reduces gate charge
losses at low load currents. With no
load, the converter draws only 45µA,
and in shutdown, the device draws
less than 1µA, making it ideal for low
current applications.
Burst Mode operation is an eficient
solution for low current applications,
but sometimes noise suppression
is a higher priority. To reduce noise
problems, a pulse skipping mode is
available, which decreases the ripple
noise at low currents. Although not
as eficient as Burst Mode operation
at low currents, pulse skipping mode
still provides high eficiency for moderate loads. In dropout, the internal
P-channel MOSFET switch is turned
on continuously, thereby maximizing
the usable battery life.
The LTC3543 offers three different
frequency modes: ixed frequency,
spread spectrum, or synchronous. In
ixed frequency mode, the regulator
operates at a constant 2.25MHz, making it possible to use capacitors and
inductors that are less than 1.2mm
in height. In spread spectrum mode,
the switching frequency is randomly
varied from 2MHz to 3MHz. By spreading the switcher’s operating frequency,
a signiicant reduction in peak radiated and conducting noise can be
realized. In synchronous mode, the
LTC3543’s switching frequency can
be synchronized to a 1MHz to 3MHz
external clock.
The small size, eficiency, low external component count, and design
Linear Technology Magazine • October 2007
lexibility of the LTC3543 make it an
ideal DC/DC converter for portable
devices using a Lithium-Ion battery.
Easy-to-Use, Ultra-Tiny
16-Bit ΔΣ ADC
The LTC2450 is an ultra-tiny 16-bit
analog-to-digital converter. It uses a
single 2.7V to 5.5V supply, accepts
a single-ended analog input voltage,
and communicates through an SPI
interface. It also includes an integrated
oscillator that does not require any
external components.
A delta-sigma modulator serves as
a converter core and provides singlecycle settling time for multiplexed
applications.
The converter is available in a 6pin, 2mm × 2mm DFN package. The
LTC2450 includes a proprietary input
sampling scheme that reduces the
average input sampling current by several orders of magnitude. The LTC2450
is capable of up to 30 conversions per
second and, due to the very large oversampling ratio, has extremely relaxed
anti-aliasing requirements.
The LTC2450 includes continuous
internal offset and full-scale calibration algorithms, which are transparent
to the user, ensuring accuracy over
time and over the operating temperature range. The converter uses its
power supply voltage as the reference
voltage and the single-ended, rail-torail input voltage range extends from
GND to VCC. Following a conversion,
the LTC2450 can automatically enter
sleep mode and reduce its power to
less than 200nA. If the user samples
the ADC once a second, the LTC2450
consumes an average of less than
50µW from a 2.7V supply.
1.1A Low Noise LDO Offers
High Power Density
The LT1965 is a low noise, low voltage
1.1A LDO with high power density. The
LT1965 features a low dropout voltage
of only 300mV at full load, with wide
VIN capability of 1.8V to 20V and low
adjustable output from 1.2V to 19.5V.
Ultra-low output noise of only 40µVRMS
reduces noise in instrumentation,
RF, DSP and logic supply systems
and is beneicial for post-regulating
switching power supplies. Output
tolerance is tightly regulated to within
±3% over line, load and temperature.
The device’s low quiescent current
of 500µA (operating) and less than
1µA (shutdown) make it an excellent
choice for applications requiring high
output drive capability with low current consumption.
The LT1965 regulator optimizes
stability and transient response with
low ESR, ceramic output capacitors
as small as 10µF. These tiny external capacitors can be used without
any necessary series resistance as is
common with many other regulators.
Internal protection circuitry includes
reverse-battery protection, no reverse current, current limiting with
foldback, and thermal limiting.
For applications requiring large
input-to-output differentials, the
LT1965 offers a very compact and
thermally effective solution. The IC
features a wide breadth of packaging options, ranging from modern
high power density, small footprint,
thermally eficient DFN and MSOPE
packages to more traditional DD-Pak
and TO-220 power packaging.
Powerful Family of
Synchronous N-Channel
MOSFET Drivers Boosts the
Efficiency and Voltage Range
of DC/DC Converters
The LTC4442, LTC4443, LTC4444,
LTC4445, and LTC4447 family of
synchronous N-channel MOSFET
drivers maximizes DC/DC converter
eficiency with peak output currents
as high as 5A, propagation delays as
low as 14ns, and high voltage operation up to 100V. From buck to boost to
buck-boost, these drivers can improve
the eficiency and extend the operating voltage range of a wide variety of
converter topologies.
The LTC4442 features powerful 5A
drivers capable of producing 5ns–12ns
transition times on 3nF loads. These
41
L NEW DEVICE CAMEOS
rapid transition times substantially
reduce the power loss in a DC/DC
converter by minimizing the switching losses in MOSFETs with high gate
capacitance. Adaptive shoot-through
protection circuitry is also integrated
to prevent power loss due to MOSFET
cross-conduction current. In addition,
the LTC4442 includes undervoltage
lockout detectors that monitor the gate
drive supply and disable operation if
the voltage is too low. The LTC4442
operates with a 6V to 9.5V gate drive
supply, and its loating high side driver
is capable of handling 38V supply
voltages.
The LTC4443 includes all of the
features of the LTC4442, but also
integrates the Schottky diode required
for the high side bootstrapped supply
to simplify layout and reduce parts
count. The LTC4445 is a dual version
of the LTC4443, with two independent
channels that are ideal for two-phase
or 2-channel applications.
For lower gate drive supply applications, the LTC4447’s rail-to-rail
outputs are optimized to source 4A
and sink 5A of current while operating
from a 4V to 6.5V supply. With 14ns
propagation delays and 5ns transition
times driving 3nF loads, this high
speed driver minimizes power loss due
to switching losses and synchronous
MOSFET body diode conduction. The
low forward drop Schottky diode required for the high side bootstrapped
supply is also integrated to simplify
converter design and reduce board
area. Like the LTC4442, the LTC4447’s
high side driver handles voltages up
to 38V.
The LTC4444 is a powerful synchronous N-channel MOSFET driver that
has been optimized for higher voltage
applications. With its two CMOS-
compatible inputs connected to the
Top Gate and Bottom Gate pins of a
controller IC, the LTC4444 instantly
extends the voltage range of a DC/DC
converter to 100V. Its powerful 3A pullup and 0.8Ω pull-down output drivers
generate 10ns rise times and 5ns fall
times on 1nF capacitive loads from a 7V
to 14V driver supply. Adaptive shootthrough prevention and undervoltage
lockout detectors are integrated to
guarantee that the system is eficient
and well-controlled.
The LTC4442 and LTC4444 gate
drivers are available in the thermallyenhanced MSOP package, and the
LTC4443, LTC4445, and LTC4447 are
available in DFN packages. This family
of rugged and powerful gate drivers is
available in the 40°C to 85°C industrial
temperature range. L
to RST during power down supply
sequencing). When V2 decays to 2V,
V3 is immediately disabled (see the
timing diagram in Figure 8).
Pushbutton Product Family
LTC295, continued from page 9
Using the reset comparator and 200ms
after V1 reaches 80% of its inal value
(2.66V), the V2 supply is enabled.
When the V2 DC voltage reaches 80%
of its inal value (2V), the V3 supply
in enabled.
A user initiates a power down
supply sequence by again pressing
the pushbutton for 32ms. When EN
is released and pulls up to VIN, V1
disconnects irst. When the V1 supply
decays to 2.66V, V2 is immediately
disabled (there is no delay from VM
PB
LTC2953-1 and
LTC2953-2 Versions
The LTC2953-1(EN) and LTC29532(EN) differ only by the polarity of
the EN/EN pin. The LTC2953-1 is
intended to drive a DC/DC converter
while the LTC2953-2 drives an external power PFET.
POWER ON
POWER OFF
32ms
EN
80%
80%
V1
200ms
V2
80%
80%
V3
Figure 8. Timing diagrams for sequencing three supplies
42
Table 1 summarizes Linear Technology’s family of pushbutton products.
The LTC2950, LTC2951 and LTC2954
provide a complete standalone solution for interfacing a manual on/off
pushbutton to system power and
system logic. The LTC2953 adds voltage monitoring functions to allow for
failsafe operation. The LTC2952 offers
selectable dual power path ideal diode
controllers.
Conclusion
The LTC2953 is a low power, wide
input voltage range (2.7V to 27V)
pushbutton on/off controller with
input and output voltage monitoring.
The LTC2953 provides a simple and
complete solution to manually toggling
power to many types of systems. It
includes a power fail comparator that
issues an early warning of a decaying
supply, along with a UVLO comparator
that prevents a user from turning on
a system with a low supply or dead
battery. The LTC2953 furthers system
reliability by integrating an adjustable
single supply supervisor. The device
is available in a space saving 3mm ×
3mm DFN package. L
Linear Technology Magazine • October 2007
DESIGN TOOLS L
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Product Press Releases — New products are announced
constantly
Solutions Brochures — Complete solutions for automotive electronics, high speed ADCs, LED drivers, wireless
infrastructure, industrial signal chain, handheld, battery
charging, and communications and industrial DC/DC
conversion applications.
Product Selection
Purchase products directly from Linear Technology either
through the methods below or contact your local LTC
sales representative or licensed distributor.
The focus of Linear Technology’s website is simple—to
get you the information you need quickly and easily. With
that goal in mind, we offer several methods of inding the
product and applications information you need.
Linear Express — Purchase online with credit terms.
Linear Express is your new choice for purchasing any
quantity of Linear Technology parts. Credit terms are
available for qualifying accounts. Minimum order is
only $250.00. Call 1-866-546-3271 or email us at
[email protected].
Packaging (www.linear.com/packaging) — Visit our
packaging page to view complete information for all of
Linear Technology’s package types. Resources include
package dimensions and footprints, package cross
reference, top markings, material declarations, assembly
procedures and more.
Quality and Reliability (www.linear.com/quality)
— The cornerstone of Linear Technology’s Quality,
Reliability & Service (QRS) Program is to achieve 100%
customer satisfaction by producing the most technically
advanced product with the best quality, on-time delivery
and service. Visit our quality and reliability page to view
complete reliability data for all of LTC’s products and
processes. Also available is complete documentation on
assembly and manufacturing lows, quality and environmental certiications, test standards and documentation
and failure analysis policies and procedures.
Lead Free (www.linear.com/leadfree) — A complete
resource for Linear Technology’s Lead (Pb) Free Program
and RoHS compliance information.
Simulation and Software
(www.linear.com/purchase)
Credit Card Purchase — Your Linear Technology parts
can be shipped almost anywhere in the world with
your credit card purchase. Orders up to 500 pieces
per item are accepted. You can call (408) 433-5723
or email [email protected] with questions regarding
your order.
Design Support
Part Number and Keyword Search — Search Linear
Technology’s entire library of data sheets, Application
Notes and Design Notes for a speciic part number or
keyword.
Sortable Parametric Tables — Any of Linear Technology’s product families can be viewed in table form,
allowing the parts to be sorted and iltered by one or
many functional parameters.
Applications Solutions — View block diagrams for a
wide variety of automotive, communcations, industrial
and military applications. Click on a functional block to
generate a complete list of Linear Technology’s product
offerings for that function.
Linear Technology offers several powerful simulation
tools to aid engineers in designing, testing and troubleshooting their high performance analog designs.
LTspice/SwitcherCAD™ III (www.linear.com/swcad)
— LTspice / SwitcherCAD III is a powerful SPICE simulator and schematic capture tool speciically designed
to speed up and simplify the simulation of switching
regulators. LTspice / SwitcherCAD III includes:
• Powerful SPICE simulator speciically designed for
switching regulator simulation
• Complete and easy to use schematic capture and
waveform viewer
• Macromodels for most of Linear Technology’s switching regulators as well as models for many of LTC’s high
performance linear regulators, op amps, comparators,
ilters and more.
• Ready to use demonstration circuits for over one hundred of Linear Technology’s most popular products.
FilterCAD — FilterCAD 3.0 is a computer-aided design
program for creating ilters with Linear Technology’s
ilter ICs.
Noise Program — This program allows the user to
calculate circuit noise using LTC op amps and determine
the best LTC op amp for a low noise application.
SPICE Macromodel Library — A library includes LTC
op amp SPICE macromodels for use with any SPICE
simulation package.
Linear Technology Magazine • October 2007
43
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Linear Technology Corp. Ltd.
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Linear Technology Corp. Ltd.
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Excellence Times Square Building
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GERMANY
Linear Technology GmbH
Osterfeldstrasse 84, Haus C
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ITALY
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Via Colleoni, 17
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KOREA
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Phone: +82 (2) 792-1617
FAX: +82 (2) 792-1619
SINGAPORE
Linear Technology Pte. Ltd.
507 Yishun Industrial Park A
Singapore 768734
Phone: +65 6753-2692
FAX: +65 6752-0108
FRANCE
Linear Technology S.A.R.L.
Parc Tertiaire Silic
2 Rue de la Couture, BP10217
94518 Rungis Cedex
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Phone: +33 (1) 56 70 19 90
FAX: +33 (1) 56 70 19 94
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SE-164 40 Kista
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UNITED KINGDOM
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Marlow, Buckinghamshire SL7 1FD
United Kingdom
Phone: +44 (1628) 477066
FAX: +44 (1628) 478153
TAIWAN
Linear Technology Corporation
8F-1, 77, Nanking E. Rd., Sec. 3
Taipei, Taiwan
Phone: +886 (2) 2505-2622
FAX: +886 (2) 2516-0702
Linear Technology Corporation
1630 McCarthy Blvd.
Milpitas, CA 95035-7417
TEL: (408) 432-1900
FAX: (408) 434-0507
www.linear.com
© 2007 Linear Technology Corporation/Printed in U.S.A./37.5K
Linear Technology Magazine • October 2007