V18N3 - SEPTEMBER

LINEAR TECHNOLOGY
SEPTEMBER 2008
IN THIS ISSUE…
COVER ARTICLE
Replace Batteries in Power RideThrough Applications with Robust
Supercaps and 3mm × 3mm
Capacitor Charger ..............................1
Jim Drew
Linear in the News… ...........................2
DESIGN FEATURES
New Family of Integrated Power
Controllers Combine Fast Battery
Charging, PowerPath™ Control
and Efficient DC/DC Converters
in Less Than 20mm2 ..........................4
Sam Nork
Charging and Discharging Methods
That Extend Li-Ion Battery Life ...........7
Fran Hoffart
Serial Interface for High Speed Data
Converters Simplifies Layout over
Traditional Parallel Devices .............13
Clarence Mayott
Synchronous Buck Controller in
3mm × 3mm QFN Fits Automotive and
Industrial Applications with 4V–38V
Input Capability ................................16
Mark Mercer
Feature-Packed Charger Handles All
Battery Chemistries and Produces
3A/50W for Fast Charging from a
4mm × 4mm QFN ...............................18
James A. McKenzie
Dual Hot Swap™ Controller in
3mm × 2mm DFN is Perfect for
Backplane or Card Resident
1V–6V Applications ..........................21
CY Lai
0V to 18V Ideal Diode Controller Saves
Watts and Space over Schottky .........24
Pinkesh Sachdev
Low Voltage, High Current Step-Down
µModule™ Regulators Put a (Nearly)
Complete Power Supply in a
15mm × 9mm × 2.8mm Package ........28
Judy Sun, Sam Young and Henry Zhang
DESIGN IDEAS
....................................................30–36
(complete list on page 30)
New Device Cameos ...........................37
Design Tools ......................................39
Sales Offices .....................................40
VOLUME XVIII NUMBER 3
Replace Batteries in
Power Ride-Through
Applications with
Robust Supercaps
and 3mm × 3mm
Capacitor Charger
Introduction
Supercapacitors (or ultracapacitors)
are inding their way into an increasing
number of applications for shortterm energy storage and applications
that require intermittent high energy
pulses. One such application is a
power ride-through circuit, in which
a backup energy source cuts in and
powers the load if the main power
supply fails for a short time. This type
of application has been dominated
by batteries in the past, but electric
double layer capacitors (EDLCs) are
fast making inroads as their priceper-farad, size and effective series
resistance per capacitance (ESR/C)
continue to fall.
In a power ride-through application, series-stacked capacitors must
be charged and cell-voltage balanced.
Supercaps are switched into the power
path when needed and the power to
the load is controlled by a DC/DC
converter. The LTC3225 supercapacitor charger has a number of features
that make it a good choice for power
ride-through applications. It comes
in a small, 10-lead 3mm × 3mm DFN
package and features programmable
by Jim Drew
One advantage
supercapacitors have over
batteries is their long life.
A capacitor’s cycle life is
quoted as greater than
500,000 cycles; batteries
are specified for only a few
hundred cycles. This makes
the supercapacitor an ideal
“set and forget” device,
requiring little or
no maintenance.
charging current, automatic cell voltage balancing, low drain current on the
supercapacitors and a patent pending,
low noise, constant current charger.
Supercapacitor
Characteristics
Supercapacitors come in a variety of
sizes, for example a 10F/2.7V supercap is available in a 10mm × 30mm
2-terminal radial can with an ESR of
continued on page L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, TimerBlox, True
Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks
of the companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
Linear Technology Analog Channel
Starting this month, you can tune in to the Linear Technology Analog Channel on the web. The channel kicks off
with a series of video design ideas covering a broad range
of topics from some of the industry’s premier analog gurus. To see the videos, visit the Linear Technology website
at www.linear.com/LTchannel or the EDN website at
www.edn.com/videocast/video_tech_clips.html. New
videos will be added periodically, so check in often, or sign
up to receive the Linear Insider email to be notiied when
new videos are available at www.linear.com/mylinear.
Check out the following videos, now available online.
“Direct Paralleling, High Power Density LDO”
with Robert Dobkin, Vice President, Engineering
and Chief Technical Officer
The LT3080 is a new architecture for linear regulators. It
provides better regulation, a simple output adjustment with
a single resistor where the output can be adjusted down
to zero. Also, this architecture allows easy paralleling of
regulators for “no heat sink” operation in all-surface-mount
applications. The LT3080 video shows circuit operation and
applications for paralleling, spreading the heat, general
purpose power supplies and current sources.
“High Voltage, Low Noise, DC/DC Converters:
A Kilovolt with 100µV of Noise”
with Jim Williams, Staff Scientist
Photomultipliers (PMT), avalanche photodiodes (APD),
ultrasonic transducers, capacitance microphones, radiation detectors and similar devices require high voltage,
low current bias. Additionally, the high voltage must be
pristinely free of noise; well under a millivolt is a common
requirement with a few hundred microvolts sometimes
necessary. The video details circuits featuring outputs
from 200V to 1000V with output noise below 100µV
measured in a 100MHz bandwidth. Special techniques
enable this performance, most notably power stages optimized to minimize high frequency harmonic content. An
additional aid to achieving low noise is that load currents
rarely exceed 5mA. This freedom permits output iltering methods that are usually impractical. A lab-based
circuit noise measurement demonstration concludes the
presentation.
“A Thermocouple Meter Reference Design
Using the LTC2492 Delta Sigma ADC”
with Mark Thoren, Applications Engineering Manager,
Mixed Signal Products
Thermocouples are perhaps the most common temperature sensor in use. And while they are extremely simple
and rugged, the output is very small—tens of microvolts
per degree Celsius. Traditionally, thermocouple measurement circuits use a cold junction compensation circuit to
2
drive the thermocouple negative terminal and a low offset
ampliier with enough gain to use the entire input span
of a 12- or 16-bit ADC.
Linear Technology’s LTC2492 greatly simpliies thermocouple instrument design. A simple ilter and protection
circuit is all that is required to build a rugged, ready-to-use
meter. Some software tricks take care of cold junction compensation and the thermocouple’s non-linear output.
Investment in Environment Award
Linear Technology’s LTC4151 High Voltage I2C Current
and Voltage Monitor has been selected as a inalist for an
E-Legacy Investment in Environment Award by Electronic
Product Design in the UK. This award highlights electronics
companies that are devveloping environmentally responsible products.
In a world where power conservation is increasingly
important, designers need to give greater consideration
to the overall impact and cost of operation of their end
product. Accurately monitoring power consumption provides information to understand and manage the power
requirements in a high voltage system to avoid wasted
resources.
The LTC4151 breaks the mold of traditional current
sense solutions by combining the high voltage capability
of a Hot SwapTM controller with the accuracy of a 12-bit
ADC. The LTC4151 provides a true power measurement
between 7V and 80V, rather than just a current reading
or voltage reading on its own. This is extremely valuable
data for high voltage applications, as it alerts designers to
overloading and power loss in their system. The LTC4151
is ideal for a wide range of applications. By measuring
up to 80V, the LTC4151 can accurately measure industrial, telecom and automotive signals at 48V, and survive
transient surges up to 80V. This power monitor can also
measure 72V systems with ±10% tolerances (79.2V). By
measuring as low as 7V, the LTC4151 can accurately
monitor 10V and 12V industrial systems. L
Linear Technology Magazine • September 2008
DESIGN FEATURES L
LTC225, continued from page 25mΩ while a 350F/2.5V supercapacitor with an ESR of 1.6mΩ is available
in a D-cell battery form factor. One
advantage supercapacitors offer over
batteries is their long life. A capacitor’s
cycle life is quoted as greater than
500,000 cycles; batteries are speciied
for only a few hundred cycles. This
makes the supercapacitor an ideal
“set and forget” device, requiring little
or no maintenance.
Two parameters of the supercapacitor that are critical to an application are
cell voltage and initial leakage current.
Initial leakage current is a misnomer
in that the initial leakage current is
really dielectric absorption current
which disappears after some time. The
manufacturers of supercapacitors rate
their leakage current after 100 hours of
applied voltage while the initial leakage
current in those irst 100 hours may
be as much as 50 times the speciied
leakage current.
The voltage across the capacitor
has a signiicant effect on its operating life. When used in series, the
supercapacitors must have balanced
cell voltages to prevent over-charging
of one of the series capacitors. Passive cell balancing, where a resistor
is placed across the capacitor, is a
popular and simple technique. The
disadvantage of this technique is that
the capacitor discharges through the
balancing resistor when the charging
circuit is disabled. The rule of thumb
for this scheme is to set the balancing
resistor to 50 times the worst case
leakage current, estimated at 2µA/
Farad. Given these parameters, a 10F,
2.5V supercapacitor would require a
2.5k balancing resistor. This resistor
would drain 1mA of current from the
supercapacitor when the charging
circuit is disabled.
An alternative is to use a non-dissipative active cell balancing circuit,
such as the LTC3225, to maintain cell
voltage. The LTC3225 presents less
than 4µA of load to the supercapacitor
when in shutdown mode and less than
1µA when input power is removed. The
LTC3225 features a programmable
charging current of up to 150mA,
charging two series supercapacitors
to either 4.8V or 5.3V while balancing
the voltage on the capacitors.
Power Ride-Through
Applications
To provide a constant voltage to the
load, a DC/DC converter is required
between the load and the supercapacitor. As the voltage across the
supercapacitor decreases, the current
drawn by the DC/DC converter increases to maintain constant power to
the load. The DC/DC converter drops
out of regulation when its input voltage reaches the minimum operating
voltage (VUV).
To estimate the requirements for
the supercapacitor, the effective circuit
resistance (R T) needs to be determined.
R T is the sum of the capacitors’ ESRs
and the circuit distribution resistances.
R T = ESR + RDIST
Assuming 10% of the input power is
lost in the effective circuit resistance
when the DC/DC converter is at the
minimum operating voltage, the worst
case R T is
R T(MAX ) =
0.1 • VUV 2
PIN
The voltage required across the
supercapacitor at the undervoltage
lockout threshold of the DC/DC converter is;
VC(UV ) =
VUV 2 + PIN • R T
VUV
The required effective capacitance
can then be calculated based on the
required ride-through time (TRT), and
the initial voltage on the capacitor
(VC(0)) and VC(UV).
CEFF =
2 • PIN • TRT
VC(0)2 − VC(UV )2
The effective capacitance of a series
connected bank of capacitors is the effective capacitance of a single capacitor
divided by the number of capacitors
while the total ESR is the sum of all
the series ESRs.
The ESR of a supercapacitor
decreases with higher frequency.
Manufacturers usually specify the ESR
continued on page 2
Si4421DY
1.8V
5.0V
D
S
G
VIN1
VOUT1
VIN2
FB1
LTM4616
22µF
Si4421DY
GND
100µF
4.87k
ITHM1
1.2V
VOUT2
D
VIN
COUT
FB2
+
C+
VIN
SENSE
LTC4412
CX
C–
+
10F
SHDN
GND
GATE
CTL
STAT
100µF
100µF
10k
G
10F
LTC3225
1µF
S
ITHM2
470k
GND
2.2µF
VSEL
PROG
GND
12K
+
= NESSCAP ESHR-0010C-002R7 OR ILLINOIS CAPACITOR 106DCN2R7Q
Figure 1. A 5V power ride-through application
Linear Technology Magazine • September 2008
3
L DESIGN FEATURES
New Family of Integrated Power
Controllers Combine Fast Battery
Charging, PowerPath Control and
Efficient DC/DC Converters in
by Sam Nork
Less Than 20mm2
Introduction
The quickest way to build an eficient
power system for a battery-powered
portable application is to use an
IC that combines all power control
functions into a single chip, namely a
Power Management Integrated Circuit
(PMIC). PMICs seamlessly manage
power low from various power sources
(wall adapters, USB and batteries) to
power loads (device systems and the
charging battery), while maintaining
current limits where required (such
as that speciied for USB). To this
end, PMICs typically feature built-in
PowerPath™ control, DC/DC conver-
USB/WALL
4.35V TO 5.5V
TO OTHER
LOADS
USB COMPLIANT
STEP-DOWN
REGULATOR
CC/CV
BATTERY
CHARGER
CURRENT
CONTROL
OPTIONAL
0V
CHARGE
+
Li-Ion
T
LTC3555/LTC3555-X
3.3V/25mA
ALWAYS ON LDO
ENABLE
CONTROLS
5
0.8V TO 3.6V/400mA
1
TRIPLE
HIGH EFFICIENCY
STEP-DOWN
SWITCHING
REGULATORS
RTC/LOW
POWER LOGIC
0.8V TO 3.6V/400mA
2
0.8V TO 3.6V/1A
3
RST
2
I2C PORT
MEMORY
I/O
CORE
µPROCESSOR
I2C
Figure 1. High efficiency PowerPath manager and triple step-down regulator
Table 1. Power management ICs with Li-ion/polymer battery chargers
Integrated Converters and Load Current Capabilities
Part Number
PowerPath
Topology
Interface
Buck
LTC3555/-1/-3
Switching
I2C
1A, 400mA × 2
LTC3556
Switching
I2C
400mA × 2
LTC3566
Switching
LTC3567
Switching
LTC3586*
Switching
400mA × 2
LTC3557/-1
Linear
LTC3455
Linear
LDO
Package
25mA
4mm × 5mm
QFN-28
1A
25mA
4mm × 5mm
QFN-28
1A
25mA
4mm × 4mm
QFN-24
1A
25mA
4mm × 4mm
QFN-24
20mA
4mm × 6mm
QFN-38
600mA, 400mA × 2
25mA
4mm × 4mm
QFN-28
600mA, 400mA
Controller
4mm × 4mm
QFN-24
I2C
LTC3558
400mA
LTC3559/-1
400mA × 2
Buck-Boost
1A
400mA
Boost
0.8A
3mm × 3mm
QFN-20
3mm × 3mm
QFN-16
*For an application of the LTC3586 see “Complete Power Solution for Digital Cameras and Other Complex Compact Portable Applications” in this issue
4
Linear Technology Magazine • September 2008
DESIGN FEATURES L
Switching PowerPath Control
Efficiently Harnesses
Available External Power
To speed up charging, some of Linear’s
new PMICs employ a unique current
limited synchronous buck switching charger architecture that uses
more power from the USB or adapter
than other topologies. This is a big
improvement over battery fed and
linear PowerPath control schemes.
(For a more detailed description of
the switching PowerPath architecture,
BATTERY CHARGE CURRENT
600
EXTRA CURRENT
FOR FASTER CHARGING
500
500mA USB CURRENT LIMIT
400
300
200
100
0
VBUS = 5V
5X MODE
BATTERY CHARGER PROGRAMMED FOR 1A
2.8
3
3.2 3.4 3.6
3.8
BATTERY VOLTAGE (V)
4
4.2
Figure 2. Switching power manager charge
current vs battery voltage with a 500mA input
current limit. Peak charge current = 700mA.
see the cover article in the June 2008
issue of Linear Technology magazine
titled “Speed Up Li-ion Battery Charging and Reduce Heat with a Switching
PowerPath Manager.”)
For instance, portable products
with large capacity batteries (1Ahr
plus) face a direct tradeoff between
charge time and charger power dissipation—especially when a linear
charging method is used. At relatively
low charge currents, a linear charger
dissipates a modest amount of power,
but at currents required to quickly
charge high capacity batteries, a linear
charger can dissipate 2W or more.
A switching PowerPath topology is
an improvement over the commonly
used linear PowerPath topology, and
both are an improvement over battery
fed applications. A linear PowerPath
powers the application directly from
an external source rather than from
the battery itself and provides “instant
TO OTHER
LOADS
USB COMPLIANT
STEP-DOWN
REGULATOR
100
80
OPTIONAL
0V
+
CHARGE
T
Li-Ion
LTC3556
ALWAYS ON LDO
DUAL HIGH EFFICIENCY
BUCKS
HIGH EFFICIENCY
BUCK-BOOST
SEQ
I2C
3
IOUT3 = 50mA
90
CC/CV
BATTERY
CHARGER
ENALL
on” capability if the battery is dead or
missing (as long as the load current
is less than the input current limit).
However, neither a linear charger nor
linear power manager is well-suited
for high current charging due to poor
eficiency under certain conditions.
USB is now a common source of
power, but charging/powering from
the USB host is complicated by the
host’s 2.5W limit. To take advantage of
the limited USB power, all components
in the power path must be as eficient
as possible.
A key attribute in these new PMICs
is a battery-tracking (Bat-Track™)
synchronous buck design with logic
programmable input current limit to
ensure USB compatibility. When USB
or adapter power is available, the
LTC35xx power manager generates a
VOUT supply equal to VBAT + 300mV.
The 300mV difference voltage is suficient to keep the battery charger
just out of dropout and deliver the
programmed charge current at high
eficiency. As with linear power managers, the load current is provided irst,
and current that is left over is directed
to the battery. Input current limit is
controlled via an external resistor to
set absolute current and two logic
pins to control the ratio (e.g. 100mA,
500mA, 1A and Suspend).
Charging eficiency of over 80%
with a completely discharged battery
is achievable vs 60% or so for a linear
charger. Or said another way, the
switching power path dissipates only
50% of the power dissipated by a linear
I2C PORT
1
2
3
3.3V/25mA
0.8V TO 3.6V/400mA
0.8V TO 3.6V/400mA
2.5V to 3.3V/1A
PGOODALL
RTC/LOW
POWER LOGIC
EFFICIENCY (%)
USB/WALL
4.5V TO 5.5V
700
CHARGE CURRENT (mA)
sion and battery charging functions.
PMICs can be applied in everything
from consumer electronics such as
MP3 players and Bluetooth headsets
to specialized portable medical and
industrial equipment.
Table 1 shows the wide variety of
integrated charger and DC/DC combinations now available from Linear
Technology. The latest additions to
the family, the LTC3555, LTC3556,
LTC3566, LTC3567 and LTC3586, are
primarily targeted toward relatively
high power Li-Ion applications and
contain blocks capable of high eficiency at high current levels. (To see
an application of the LTC3586, see
“Complete Power Solution for Digital
Cameras and Other Complex Compact
Portable Applications” in the Design
Ideas section of this issue.)
The most noteworthy feature of the
new parts is the use of a proprietary
switching PowerPath design, which
improves eficiency over linear power
path or battery fed solutions.
IOUT3 = 200mA
70
IOUT3 = 1000mA
60
50
40
30
20
MEMORY
CORE
µP
10
0
VOUT3 = 3.3V
TA = 27°C
2.7
HDD/IO
3.1
3.5
3.9
4.3
VIN3 (V)
4.7
3556 TA01
Figure 3. 1A buck-boost efficiency vs VIN (LTC3556, LTC3566/7, LTC3586)
Linear Technology Magazine • September 2008
5
L DESIGN FEATURES
charger under worst case conditions.
The LTC35xx switching power managers can charge at up to 1.2A max
and provide seamless switchover to
battery power when the external power
is removed. In USB applications, the
constant power (vs constant current)
nature of the switching PowerPath
controller makes it possible to charge
with more than 500mA from a ixed
500mA USB input source, as shown
in Figure 2.
USB/WALL
4.5V TO 5.5V
CURRENT
CONTROL
Need a Buck-Boost?
Not a Problem…
Most high end portable products need
a minimum of three key power supplies: one for the µP core (~1.0V–1.5V),
one for memory (~1.8V), and one for
the I/O and main system supply
(~3.3V). The LTC3555 covers all three
with its built-in three synchronous
bucks. However, some applications,
particularly the more feature-rich variety, face occasional high peak power
transients during wireless transmissions or when a hard drive spins up.
The effective voltage of the battery
drops during these transient currents
due to the battery series resistance
CC/CV
BATTERY
CHARGER
OPTIONAL
0V
+
CHARGE
T
Li-Ion
LTC3586
3.3V/20mA
ALWAYS ON LDO
4
EN
3
HIGH EFFICIENCY
BOOST
4
MEMORY/
CORE µP
0.8V TO 3.6V/400mA
2
HIGH EFFICIENCY
BUCK-BOOST
RTC/LOW
POWER LOGIC
0.8V TO 3.6V/400mA
1
DUAL HIGH EFFICIENCY
BUCKS
MODE
Higher Current Chargers Go
Hand-In-Hand with Higher
Current Regulators
An obvious companion to a high
performance battery charger is a corresponding set of DC/DC regulators
with similar peak current handling
and high eficiency. As shown in Table
1, the latest PMICs offer between one
and four DC/DCs of varied topologies
with peak currents reaching 1A. The
new parts provide a variety of speciic
options to meet the high performance
needs of speciic applications.
TO OTHER
LOADS
USB COMPLIANT
STEP-DOWN
REGULATOR
2.5V to 3.3V/1A
I/O
SYSTEM
5V/800mA
AUDIO/
MOTOR
2
ILIM
FAULT
Figure 4. The LTC3586 is a high efficiency PowerPath controller, alwayson LDO, dual buck, buck-boost, plus boost—all in a 4mm × 6mm package
(BSR), trace impedance or power path
losses. This poses a problem for the
3.3V supply, which can drop out of
regulation even if the battery is still
signiicantly charged. In such cases,
a buck-boost regulator can save the
day by riding through such battery
transients—maintaining regulation
as if nothing happened. Several new
PMICs contain buck-boost DC/DCs
speciically for this purpose. As shown
in Figure 3, the PMIC buck-boosts can
provide a high eficiency 3.3V output
with an input that ranges from 2.7V
to 5.5V.
The LTC3566 and LTC3567 products include a 1A buck-boost supply
in addition to a high performance
switching PowerPath controller as cornerstone high performance building
blocks. The LTC3556 ups the integration further by including two 400mA
buck regulators to accompany the
charger and buck-boost supply. The
LTC3586 contains all of the blocks of
the LTC3556, but ups the integration
one step further…
Need an Additional 5V Boost?
The LTC3586 Has It Covered
While the buck-boost regulators are
capable of regulating a 5V supply,
some applications require both. To
meet this need, the LTC3586 includes
not only a full complement of low voltage regulators, it also includes a high
continued on page 5
CHRGEN
VIN1
ENABLE
SWAB1
MODE
ILIM
DECODE
LOGIC
D/A
VOUT1
L
CHRGEN
VIN4
4
1A, 2.25MHz
BUCK-BOOST
REGULATOR
EN1
SW4
SWCD1
DVCC
LTC3586
SDA
VOUT4
CPL
R1
COUT
FB4
R2
FB1
I2C PORT
VC1
SCL
GND
6, 12, 17, 25
Figure 5. Boost converter application circuit
6
Figure 6. The LTC3567 I/O and DC/DC output voltage control interface
Linear Technology Magazine • September 2008
DESIGN FEATURES L
Charging and Discharging Methods
That Extend Li-Ion Battery Life
by Fran Hoffart
Introduction
Much emphasis has been put on increasing lithium-ion battery capacity
to provide the longest product run
time in the smallest physical size, but
there are instances where a longer
battery life, an increased number of
charge cycles or a safer battery is more
important than battery capacity. This
article presents methods relating to
charging and discharging Li-ion batteries that can considerably increase
battery life.
Rechargeable lithium-ion, including lithium-ion polymer batteries
can be found in practically every
high performance portable product
and the reason for this is well justiied. Compared to other rechargeable
batteries, lithium-ion batteries have
a higher energy density, higher cell
voltage, low self-discharge, very good
cycle life, are more environmentally
friendly and are simple to charge
and maintain. Also, because of their
relatively high voltage (2.5V to 4.2V)
many portable products can operate
from a single cell, thereby simplifying
an overall product design.
Lithium-Ion Battery Basics
Before covering the battery charger’s
role in extending battery life, a quick
review of the lithium-ion battery is
necessary. Lithium is one of the lightest metals, one of the most reactive
and has the highest electrochemical
The Letter “C”
The letter “C” is a battery term used to indicate the battery manufacturers
stated battery discharge capacity, which is measured in mAh. For example,
a 2000mAh rated battery can supply a 2000mA load for one hour before
the cell voltage drops to it’s zero capacity voltage. In the same example,
charging the battery at a C/2 rate would mean charging at 1000mA (1A).
The letter “C” becomes important in battery chargers because it determines
the correct charge current required and the length of time needed to fully
charge a battery. When discussing minimum charge current termination
methods, a 2000mAh battery using C/10 termination ends the charge cycle
when the charge current drops below 200mA. L
potential making it the ideal material
for a battery. A Li-ion battery contains
no lithium in a metallic state, but instead uses lithium ions that shuttle
back and forth between the positive
electrode (cathode) and the negative
electrode (anode) of the battery during
charge and discharge.
Types of Lithium-Ion Batteries
Although there are many different
types of Li-ion batteries, the most
popular chemistries presently in
production can be narrowed down to
three, all relating to the cathode materials used in the battery. The lithium
cobalt chemistry has become more
popular in laptops, cameras and cell
phones mainly because of its greater
charge capacity. Other chemistries are
used where high discharge currents
are required, where safety is a concern
or where cost is an issue. Also, new
hybrid Li-ion batteries under development use a combination of electrode
materials to take advantage of beneits
of each chemistry.
Unlike a few other battery chemistries, Li-ion battery technology is not
yet mature. Research is ongoing with
new types of batteries that have even
higher capacities, longer life and im-
Table 1. Most common lithium-ion batteries
Cathode Materials
Lithium Cobalt Oxide
(Most Common)
Advantages
❏ High Capacity
❏ Lower ESR
Lithium Manganese Oxide
❏ Higher Charge and Discharge Rates
❏ Higher Temperature Operation
❏ Inherently Safer
❏ Very Low ESR
Lithium Phosphate
(Newest, A123 and Saphion)
❏ Very High Charge and Discharge Rates
❏ High Temperature Operation
❏ Inherently Safer
Linear Technology Magazine • September 2008
Disadvantages
❏ Lower Charge and Discharge Rates
❏ Higher Cost
❏ Lower Capacity
❏ Lower Life Cycle
❏ Shorter Lifetime
❏ Lower Discharge Voltage
❏ Lower Float Voltage
❏ Lower Capacity
7
L DESIGN FEATURES
proved performance than present day
batteries. The table shown in Table 1
includes some important characteristics of each battery type.
Perhaps one of the worst
locations for a Li-ion battery
is in a laptop computer when
used daily on a desktop
with the charger connected.
Laptops typically run warm
or even hot, raising the
battery temperature, and the
charger is maintaining the
battery near 100% charge.
Both of these conditions
shorten battery life.
Lithium-Ion Polymer Batteries
A lithium-ion polymer battery is
charged, discharged and has similar
characteristics as a standard Li-ion
battery. The main difference between
the two is that a solid ion conductive
polymer replaces the liquid electrolyte
used in a standard Li-ion battery,
although most polymer batteries also
contain an electrolyte paste to lower
the internal cell resistance. Eliminating the liquid electrolyte allows
the polymer battery to be housed in
a foil pouch rather than the heavy
metal case required for standard Liion batteries. The ability to fabricate
the battery in many different shapes,
including very thin form factors, and
lower production costs are making the
Li-ion polymer battery very popular.
discharging eventually reduces the
battery’s active material and causes
other chemistry changes that result
in increased internal resistance and
permanent capacity loss. Batteries
can even lose permanent capacity
when not used, sitting on the shelf.
The permanent capacity loss is greatest at elevated temperatures with the
battery voltage maintained close to
4.2V (fully charged).
For maximum storage life, batteries
should be stored with a 40% charge
(3.6V) at 40°F (refrigerator). Perhaps
one of the worst locations for a Li-ion
battery is in a laptop computer when
used daily on a desktop with the
charger connected. Laptops typically
run warm or even hot, raising the
battery temperature, and the charger
is maintaining the battery near 100%
charge. Both of these conditions
shorten the battery life, which could
be as short as 6 months to a year.
Battery Lifetime
All rechargeable batteries wear out,
and Li-ion cells are no exception. Battery manufacturers usually consider
end-of-life for a battery to be when the
battery capacity drops to 80% of the
rated capacity, although the battery
can still deliver usable power below
80% charge capacity, the run time is
shortened.
The number of charge/discharge
cycles is commonly used when referring to battery life, but cycle life and
battery life (or service life) can be different lengths of time. Charging and
CONSTANT CURRENT
CONSTANT VOLTAGE
CHARGE CURRENT (A)
90
80
4.0
CHARGER FLOAT VOLTAGE
3.5
100%
2.5
80%
2.0
60%
1.5
40%
1.0
20%
0.5
0
0
70
CHARGE CAPACITY
3.0
60
50
40
CHARGE CURRENT
30
CHARGE CAPACITY (%)
CELL VOLTAGE
100
4.5
20
CHARGE RATE = 1C
10
0
0
0.5
1
1.5
2
2.5
3
CHARGE TIME (HOURS)
If possible, remove the battery and
use the AC adapter for powering the
laptop when the computer is used on
a desktop. A properly cared for laptop
battery can have a service life of 2 to
4 years, or more.
Lithium-Ion Battery Capacity Loss
There are two types of battery capacity losses, recoverable loss and
permanent loss. After a full charge,
a Li-ion battery typically loses about
5% capacity in the irst 24 hours, then
approximately 3% per month because
of self-discharge and an additional
3% per month if the battery pack
has pack protection circuitry. These
self-discharge losses occur when the
battery remains around 20°C, but
increases considerably with higher
temperature and also as the battery
ages. This capacity loss can be recovered by recharging the battery.
The permanent capacity loss is like
the name implies, permanent, not
recoverable by charging. This loss is
linked to battery life because when
the permanent capacity loss drops
to approximately 80%, the battery is
considered at the end of its life. Permanent capacity loss is mainly due to
the number of full charge/discharge
cycles, the battery voltage and battery temperature. The more time the
battery remains near 4.2V or 100%
charge level (lower voltage for Li-ion
Phosphate) the faster the capacity loss
occurs. This is true whether the battery
is being charged or just remaining in a
fully charged condition with the voltage
near 4.2V. Always maintaining a Li-ion
battery in a fully charged condition
shortens its lifetime. The chemical
changes that shorten the battery lifetime, begin when it is manufactured,
and these changes are accelerated
by high loat voltage and high temperature. Permanent capacity loss is
unavoidable, but it can be held to a
minimum by observing good battery
practices when charging, discharging
or simply storing the battery. Using
partial discharge cycles can greatly
increase cycle life and charging to
less than 100% capacity can increase
battery life even further.
Figure 1. Typical charge profile showing charge current, voltage and capacity
8
Linear Technology Magazine • September 2008
DESIGN FEATURES L
Factors That Determine Li-ion
Battery Cycle Life or Service Life
Battery life is affected by a combination of several factors. To increase a
battery’s life, use some of the following
techniques.
q Use partial discharge cycles.
Using only 20% or 30% of the
battery capacity before recharging
extends cycle life considerably.
A general rule is from 5 to 10
shallow discharge cycles are
equal to one full discharge cycle.
Although partial discharge cycles
can number in the thousands,
at the same time, keeping the
battery in a fully charged state
also has an effect on shortening
battery life. Full discharge cycles
(down to 2.5V or 3V depending on
chemistry) should be avoided if
possible.
q Avoid charging to 100% capacity.
Selecting a lower loat voltage can
do this. Reducing the loat voltage
increases cycle life and service
life at the expense of reduced
battery capacity. A 100mV to
300mV drop in loat voltage can
increase cycle life more than 5x.
Li-ion cobalt chemistries are more
sensitive to a higher loat voltage
than other chemistries. Li-ion
phosphate cells have a lower loat
voltage than the more common
Li-ion batteries.
q Select the correct charge
termination method. Selecting
a charger that uses minimum
charge current termination
(C/10 or C/x) can also extend
battery life by not charging to
100% capacity. For example,
ending a charge cycle when the
current drops to C/5 is similar to
reducing the loat voltage to 4.1V.
In both instances, the battery is
charged to approximately 85% of
capacity, which can signiicantly
improve overall battery life.
q Limit battery temperature. High
temperatures accelerate chemical
changes within the battery,
which shorten battery life, while
charging below 0°C promotes
metal plating at the battery
anode, which can develop into an
Linear Technology Magazine • September 2008
internal short, producing heat
and making the battery unstable
and unsafe. Many battery
chargers have provisions for
measuring battery temperature to
assure charging does not occur at
temperature extremes.
q Avoid high charge and discharge
currents as they reduce cycle life.
High currents place excessive
stress on the battery. Some
chemistries are more suited for
higher currents such as Li-ion
manganese and Li-ion phosphate.
q Avoid very deep discharges below
2V or 2.5V, as this quickly and
permanently damages a Li-ion
battery. Internal metal plating
can occur causing a short circuit
making the battery unusable
and unsafe. Most Li-ion batteries
have electronic circuitry within
the battery pack that opens the
battery connection if the battery
voltage is less than 2.5V, exceeds
4.3V or if the battery current
when charging or discharging
exceeds a predeined threshold.
Li-Ion Charging Methods
The recommended way to charge a Liion battery is to provide a ±1% voltage
limited constant current to the battery
until it becomes fully charged, and
then stop. Methods used to determine
when the battery is fully charged
include timing the total charge time,
monitoring the charge current or a
combination of the two.
The irst method applies a voltage
limited constant current ranging from
C/2 to 1C for 2.5 to 3 hours thus bring-
Table 2. Battery chargers that provide a lower float voltage for increased battery life
Product
Description
Float Voltage
LTC1730-4.1
Pulse Charger
4.1V
LTC1731-4.1
Linear Charger Controller
4.1V
LTC1731-8.2
2-Cell Linear Charger Controller
8.2V
LTC1732-4.1
Linear Charger Controller
4.1V
LTC1733-4.1
Linear Charger
4.1V
LTC1734-4.1
Linear Charger
4.1V
LTC3455-1
Linear Charger/DC-DC/USB Manager
4.1V
LTC3555-3
Linear Charger/DC-DC/USB Manager
4.1V
LTC3557-1
Pre-Reg Charger & USB Manager
4.1V
LTC3559-1
Linear Charger/Dual DC-DC
4.1V
LTC4001-1
Switching Charger
4.1V
LTC4007, LTC4007-1
Switching Charger Controller
12.3V & 16.4V
LTC4050-4.1
Linear Charger
4.1V
LTC4064-4.0
Linear Charger
4.0V
LTC4066-1
Linear Charger and USB Manager
4.1V
LTC4085-1
Linear Charger and USB Manager
4.1V
LTC4008
Switching Charger Controller
Adjustable
LTC1980
Switching Charger Controller
Adjustable
LTC4089-1
HV/High Eficiency Charger
4.1V
LTC4098-1
Charger/USB Manager
4.1V
LTC1760, LTC1960
Dual Smart Battery Charger Controller
Set by battery
LTC4100
Smart Battery Charger Controller
Set by battery
9
L DESIGN FEATURES
4.2
2000
100
1000
80
UNSAFE REGION
60
4.2V FLOAT VOLTAGE
CELL VOLTAGE (V)
CAPACITY
500
GRAPHITE ANODE
3.8
BATTERY CAPACITY (%)
1500
4
100
BATTERY CAPACITY (%)
CHARGE/DISCHARGE CYCLES
120
# OF CYCLES
90
80
4.1V FLOAT VOLTAGE
70
3.6
CARBON ANODE
3.4
3.2
3V
CUT-OFF
VOLTAGE
3
2.8
60
2.5V
CUT-OFF
VOLTAGE
2.6
0
4
4.1
4.2
4.3
4.4
4.5
0
200
400
600
800
1000
1200
CHARGE TERMINATION (FLOAT) VOLTAGE (V)
NUMBER OF CHARGE CYCLES
Figure 2. Charger float voltage vs
battery capacity and cycle life
Figure 3. Cycle life and capacity
vs 4.1V and 4.2V float voltages
ing the battery up to 100% charge. A
lower charge current can also be used
but requires more time.
The second method is similar but
it requires monitoring the charge current. As the battery charges, the voltage
rises, exactly as in the irst method.
When it reaches the programmed
voltage limit, which is also called the
loat voltage, the charge current begins to drop. When it irst begins to
drop, the battery is about 50% to 60%
charged. The loat voltage continues
to be applied until the charge current
drops to a suficiently low level (C/10
to C/20) at which time the battery is
approximately 92% to 99% charged
and the charge cycle ends. Presently,
there is no safe method for fast (less
than one hour) charging a standard
Li-ion battery to 100% capacity.
Applying a continuous voltage to
a battery after it is fully charged is
not recommended as this accelerates
permanent capacity loss, can cause
the battery to swell and may result
in internal lithium metal plating. This
plating can develop into an internal
short circuit resulting in overheating making the battery thermally
unstable. The length of time required
is months.
Some Li-ion battery chargers allow
a thermistor to be used to monitor battery temperature. The main
purpose is to prevent charging if the
battery temperature is outside the
recommended window of 0°C to 40°C.
Unlike NiCd or NiMH batteries, Li-ion
cell temperature rises very little when
charging. See Figure 1 for a typical
Li-ion charge proile showing charge
current, battery voltage, and battery
capacity vs time.
10
What Determines
Battery Float Voltage?
The main determining factor of a
battery’s loat voltage is the electrochemical potential of the active
materials used in the battery’s cathode, which for lithium is approximately
4V. The addition of other compounds
raises or lowers this voltage. The second factor is a tradeoff between cell
capacity, cycle life, battery life and
safety. The curve shown in Figure 2
shows the relationship between cell
capacity and cycle life.
Most manufacturers of standard Liion cells have set a 4.2V loat voltage
as the best compromise of capacity and
cycle life. Using 4.2V as the constant
voltage limit (loat voltage), a battery
can typically deliver about 500 charge/
discharge cycles before the battery
capacity drops to 80%. A lower loat
voltage for Li-ion phosphate batteries
allows the number of charge/discharge
cycles to be much higher. One charge
cycle consists of a full charge to a full
discharge. Multiple shallow discharges
add up to one full charge cycle.
Although charging to a capacity less
than 100% using either a reduced loat
voltage or minimum charge current
termination results in initial reduced
battery capacity, as the number of
cycles increases beyond 500, the battery capacity of the lower loat voltage
can exceed that of the higher loat
voltage. Figure 3 illustrates how the
0
20
40
60
80
100
DISCHARGE CAPACTY (%)
Figure 4. Li-ion discharge voltage profile
for different anode materials
recommended loat voltage compares
with a reduced loat voltage with regard to capacity and the number of
charge cycles.
Because of the different Li-ion battery chemistries and other conditions
that can affect battery life, the curves
shown here are only estimates of the
number of charge cycles and battery
capacity levels. Even similar battery
chemistries from different manufacturers can have dramatically different
results due to minor differences in
battery materials and construction
methods.
Battery manufacturers specify a
charge method and a loat voltage
the end user must use to meet the
battery speciications for capacity,
cycle life and safety. Charging above
the recommended loat voltage is not
recommended. Many batteries include
a battery pack protection circuit, which
temporarily opens the battery connection if the maximum battery voltage
is exceeded. Once opened, connecting the battery pack to the charger
normally resets the pack protection.
Battery packs often have a voltage
printed on the battery, such as 3.6V
for a single cell battery. This voltage
is not the loat voltage, but rather the
average battery voltage when the battery is discharging.
Selecting a Battery Charger
for Extending Battery Life
Although a battery charger has no
control over a battery’s depth-of-discharge, discharge current and battery
temperature, all of which have an effect
Linear Technology Magazine • September 2008
DESIGN FEATURES L
on battery life, many chargers have
features that can increase battery life,
sometimes dramatically.
A battery charger’s role in extending
battery lifetime is mainly determined
by the charger’s loat voltage and
charge termination method. Many
Linear Technology Li-ion chargers
feature a ±1% (or lower) ixed loat
voltage of 4.2V, but there are some
offerings in 4.1V and 4.0V, as well as
adjustable loat voltages. Table 2 lists
battery chargers that feature a reduced
loat voltage that can increase battery
life when used to charge a 4.2V Li-ion
battery.
Battery chargers not offering lower
loat voltage options are also capable of
increasing battery life. Chargers that
provide minimum charge current termination methods (C/10 or C/x) can
provide a longer battery life by selecting the correct charge current level at
which to end the charge cycle.
Although C/10 termination brings
the battery to only ~92% capacity,
there is a siginiicant increase in
cycle life over charging the battery to
full capacity. A C/5 termination level
can double the cycle life, though the
battery charge capacity drops to approximately 85%. Table 3 lists Linear
Technology chargers that provide
either C/10 (10% current threshold)
or C/x (adjustable current threshold)
charge termination mode.
Longer Run Time or
Longer Battery Life,
Can You Have Both?
With present battery technology and
without increasing battery size, the
answer is no. For maximum run time,
the charger must charge the battery
to 100% capacity. This places the battery voltage near the manufacturers
recommended loat voltage, which is
typically 4.2V ±1%. Unfortunately,
charging and maintaining the battery
near these levels shortens battery life.
One solution is to select a lower loat
voltage, which prohibits the battery
from achieving 100% charge, although
this means selecting a higher capacity
battery to provide the same run time.
Of course, in many portable products,
a larger sized battery may not be an
option.
Also, using a C/10 or C/x minimum
charge current termination method
can have the same effect on battery
life as using a lower loat voltage.
Reducing the loat voltage by 100mV
reduces capacity by approximately
15% but can double the cycle life. At
the same time terminating the charge
cycle when the charge current has
dropped to 20% (C/5) also reduces
the capacity by 15% and achieves the
same doubling of cycle life.
Typical Li-Ion Battery Voltage
when Discharging
As expected, during discharge, the
battery voltage slowly drops. The
discharge voltage profile vs time
depends on a number of items including discharge current, battery
temperature, battery age and the type
of anode material used in the battery.
Presently, most Li-ion batteries use
Table 3. Battery chargers that feature minimum charge
current termination method for increased battery life
Product
Description
Termination Method
LTC3550, LTC3550-1
Linear Charger & DC/DC Converter
C/x
LTC3552, LTC3552-1
Linear Charger & DC/DC Converter
C/x
LTC4001
Switching Charger
C/x
LTC4054, LTC4054X, LTC4054L
Linear Charger
C/10
LTC4058, LTC4058X
Linear Charger
C/10
LTC4061
Linear Charger
C/x or Adj. Timer
LTC4062
Linear Charger
C/x or Adj. Timer
LTC4063
Linear Charger
C/x or Adj. Timer
LTC4068, LTC4068X
Linear Charger
C/x
LTC4075
Dual Input Linear Charger
C/x
LTC4075HVX
Dual Input Linear Charger
C/x
LTC4076
Dual Input Linear Charger
C/x
LTC4077
Dual Input Linear Charger
C/10
LTC4078
Dual Input Linear Charger
C/x
LTC4088-1, LTC4088-2
Linear Charger/USB Manager
C/x
LTC4096, LTC4096X
Dual Input Linear Charger
C/x
LTC4097
Dual Input Linear Charger
C/x
Linear Technology Magazine • September 2008
11
L DESIGN FEATURES
either a petroleum based coke material or graphite. The voltage proiles
for each are shown in Figure 4. The
more widely used graphite material
produces a latter discharge voltage
between 20% and 80% capacity, then
drops quickly near the end, whereas
the coke anode has a steeper voltage
slope and a lower 2.5V cutoff voltage.
The approximate remaining battery
capacity is easier to determine with
a coke material by simply measuring
the battery voltage.
Parallel or Series
Connected Cells
For increased capacity, Li-ion cells
are often connected in parallel. There
are no special requirements other
than they should be the same chemistry, manufacturer and size. Series
connected cells require more care
because cell capacity matching and cell
balancing circuitry is often required
to assure that each cell reaches the
same loat voltage and the same level
of charge. Connecting two cells (that
have individual pack protection circuitry) in series is not recommended
because a mismatch in capacity can
result in one battery reaching the
overvoltage limit, thus opening the
battery connection. Multicell battery
packs should be purchased assembled
with the appropriate circuitry from a
battery manufacturer.
Conclusion
LTC3225 is used to charge the supercapacitors at 150mA and maintain cell
balancing, while the LTC4412 provides
an automatic switchover function. The
LTM4616 dual output switch mode
µModule DC/DC converter generates
the 1.8V and 1.2V outputs.
Figure 2 shows a 12V power system
that uses six 10F, 2.7V supercapacitors
in series charged by three LTC3225’s
set to 4.8V and a charging current
of 150mA. The three LTC3225’s are
powered by three loating 5V outputs
generated by the LT1737 lyback controller. The output of the stack of six
supercapacitors is set up in a diode
OR arrangement via the LTC4355 dual
ideal diode controller. The LTM4601A
µModule DC/DC regulator produces
1.8V at 11A from the OR’d outputs.
The LTC4355’s MON1 in this application is set for 10.8V.
The lifetime of a Li-ion battery is determined by many factors of which the
most important are battery chemistry,
depth of discharge, battery temperature and battery capacity termination
level. The number of available charge/
discharge cycles can be increased
by selecting a charger that allows
charging to less than 100% capacity,
such as one that features a lower loat
voltage or one that terminates earlier
in the charge cycle. L
Authors can be contacted
at (408) 432-1900
LTC225, continued from page at 1kHz, while some manufactures
publish both the value at DC and at
1kHz. The capacitance of supercapacitors also decreases as frequency
increases and is usually speciied at
DC. The capacitance at 1kHz is about
10% of the value at DC. When using
a supercapacitor in a ride-through
application where the power is being
sourced for seconds to minutes, use
the effective capacitance and ESR
measurements at a low frequency,
such as 0.3Hz.
Applications
Figure 1 shows two series connected
10F, 2.7V supercapacitors charged
to 4.8V that can hold up 20W. The
Conclusion
Supercapacitors are meeting the needs
of power ride-through applications
where the time requirements are in the
seconds to minutes range. Capacitors
offer long life, low maintenance, light
weight and environmentally friendly
solutions when compared to batteries.
To this end, the LTC3225 provides a
compact, low noise solution to charging and cell balancing series connected
supercapacitors. L
IDEAL DIODE
12V
DC/DC
VIN
LT1737 FLYBACK
M1
IRF7427
VOUT
1.8V
LTM4601A
+
DC-A
1µF
10F
LTC3225
LTC4355
+
GND
GND
GND
10F
UV DETECTOR
DC-B
+
1µF
LT1737
10F
LTC3225
+
DC-C
10F
GND
+
1µF
10F
LTC3225
+
10F
+
= NESSCAP ESHR-0010C-002R7 OR ILLINOIS CAPACITOR 106DCN2R7Q
Figure 2. A 12V power ride-through application
12
Linear Technology Magazine • September 2008
DESIGN FEATURES L
Serial Interface for High Speed Data
Converters Simplifies Layout over
Traditional Parallel Devices
by Clarence Mayott
Introduction
Current Mode Logic and
8B/10B Encoding Allows
High Speed Serial Data
Transfer
AMPLITUDE (dBFS)
The LTC2274 achieves excellent signal
to noise ratio (SNR) performance of
77.6dBFS and spurious free dynamic
range (SFDR) of 100dB at baseband,
as shown in Figure 1. The input topology of the LTC2274 family is based on
its predecessor, the LTC2207 family,
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
–130
The LTC2274 ADC replaces
the usual parallel interface
with a novel high speed
serial interface, thus
reducing the typical number
of required data input/
output lines from 16 CMOS
or 32 LVDS to a single, selfclocking, differential pair
communicating at 2.1Gbps.
and achieves similar AC performance.
However, the LTC2274 differs from the
LTC2207 in its output structure. The
LTC2274 uses an 8B/10B encoder to
encode and serialize the data before it
is transmitted. 8B/10B encoding is a
process that takes 8 bits of data and
encodes them into 10 bits to ensure
zero DC offset and a limited run length.
To encode a 16-bit word, the LTC2274
must transmit 20 bits of serial data.
This requires that the serial data must
be transmitted at 20 times the clock
frequency of the ADC. Sampling at
105Msps requires the LTC2274 to
transmit serial data at 2.1GHz. This
is beyond the usable range of LVDS
signaling, and therefore requires a
faster, more robust differential signaling scheme. The LTC2274’s differential
signaling uses current mode logic
(CML), which is capable of transmitting
data in excess of 10GHz.
Current mode logic uses a differential output transistor pair (usually
N-type) to steer current into resistive
loads. The output swing and offset
depends on the bias current and
termination resistance. The output
driver bias current is typically 16mA,
generating a signal swing potential
of 400mVP–P (800mVP–P differential)
across the combined internal and
external termination resistance of 25Ω
on each output. LVDS typically uses
3.5mA to develop its signal swing, and
the capacitance of the ESD protection
diodes becomes a limiting factor for
transmission speed. CML uses more
current, and therefore this capacitance
becomes less of a limiting factor to
data throughput.
CML is typically faster than LVDS.
A typical LVDS output stage requires
four transistors to steer current into
the load, usually using both P-channel
and N-channel devices. A mixture of
N- and P-channel makes it dificult to
produce devices that have the same
characteristics. P-channel devices are
often slower—that is, if an N-channel
1.0E+00
1.0E–02
BIT ERROR RATE (BER)
The LTC2274 is a 105Msps, 16-bit
ADC that simpliies the digital connection between the ADC and FPGA by
replacing the usual parallel interface
with a novel high speed serial interface,
thus reducing the typical number of
required data input/output (I/O) lines
from 16 CMOS or 32 LVDS parallel
data lines to a single, self-clocking,
differential pair communicating at
2.1Gbps. This frees up valuable FPGA
pins and board space. It also allows
lexibility to route across analog and
digital boundaries—in noise sensitive applications, the serial interface
provides an effective isolation barrier
between digital and analog circuitry
and serves to eliminate coupling between the digital outputs and analog
inputs to reduce digital feedback.
100mV/DIV
1.0E–04
1.0E–06
1.0E–08
1.0E–10
1.0E–12
1.0E–14
79.4ps/DIV
0
10
20
30
40
FREQUENCY (MHz)
0
0.2
50
Figure 1. Typical LTC2274 performance
at 105Msps fIN = 4.93MHz
Linear Technology Magazine • September 2008
a. CMLOUT eye diagram 2.1GBps
0.4
0.6
0.8
UNIT INTERVAL (UI)
1.0
b. CMLOUT Dual-Dirac BER
bathtub curve, 2.1GBps
Figure 2. Signal integrity of CMLOUT
13
L DESIGN FEATURES
and a P-channel device are cascaded,
the P-channel cannot pull up the signal
as fast as the N-channel can pull down.
This causes the output waveform to be
distorted, which can lead to bit errors,
and limits the speed at which LVDS
can transfer data.
The LTC2274 CML driver is implemented with only N-channel devices,
which allows faster throughput rates.
Since CML only sinks current, it has
true differential signal, which improves
signal integrity. The eye diagram and
bathtub curves of the LTC2274 are
shown in Figure 2. The eye diagram
shows very little variation cycle to
cycle of the CML logic output, and the
bathtub curve shows that total jitter
in the signal is less than 0.35UI (unit
interval). This equates into a very clean
uniform signal that can easily received
by a properly terminated receiver.
SERIAL CML DRIVER
OVDD
50Ω
14
50Ω
CMLOUT+
50Ω
TRANSMISSION LINE
50Ω
50Ω
CMLOUT–
DATA+
50Ω
TRANSMISSION LINE
DATA–
16mA
GND
a. Recommended CML termination, directly-coupled mode
SERIAL CML DRIVER
SERIAL CML RECEIVER
OVDD
50Ω
Termination of CML
CML must be terminated for proper
operation. Figure 3a shows a recommended design in which an FPGA
receiver uses internal 50Ω pull up resistors for termination. These resistors
pull up to the OVDD of the LTC2274.
OVDD must be between 1.2V and 3.3V
to ensure proper operation. The signal
has a common mode voltage of OVDD
– 0.2V. The directly-coupled differential termination of Figure 3b may be
used in the absence of a receiver termination voltage within the required
range. In this case, the common mode
voltage is shifted down to approximately 400mV below OVDD, requiring
an OVDD in the range of 1.4V to 3.3V.
If the serial receiver’s common mode
input requirements are not compatible
with the directly-coupled termination
modes, the DC balanced 8B/10B
encoded data permits the addition of
DC blocking capacitors as shown in
Figure 3c. In this AC-coupled mode,
the termination voltage is determined
by the receiver’s requirements. The
coupling capacitors should be selected appropriately for the intended
operating bit-rate, usually between
1nF and 10nF. In AC coupled mode,
the output common mode voltage is
approximately 400mV below OVDD, so
the OVDD supply voltage should be in
SERIAL CML RECEIVER
1.2V TO 3.3V
50Ω
CMLOUT+
1.4V TO 3.3V
50Ω
TRANSMISSION LINE
100Ω
CMLOUT–
DATA+
50Ω
TRANSMISSION LINE
DATA–
16mA
GND
b. CML termination, directly-coupled differential mode
SERIAL CML DRIVER
SERIAL CML RECEIVER
1.4V TO 3.3V
VTERM
OVDD
50Ω
50Ω
CMLOUT+
50Ω
TRANSMISSION LINE
0.01µF
50Ω
50Ω
0.01µF
CMLOUT–
DATA+
50Ω
TRANSMISSION LINE
DATA–
16mA
GND
c. CML termination, AC-coupled mode
Figure 3. CML termination schemes
Linear Technology Magazine • September 2008
DESIGN FEATURES L
the range of 1.4V to 3.3V. If possible,
using a lower OVDD can reduce power
consumption. The termination scheme
is largely based on the receiver. When
choosing the OVDD voltage, refer to the
receiver’s data sheet to terminate the
CML lines properly.
CML uses true double termination.
Generally, LVDS is only terminated at
the receiver, which means that any
signal relection back to the source
relects back to the receiver with little
attenuation. This limits the data rate
and trace length that LVDS can drive.
The truly differential nature of CML
radiates less energy than LVDS and
CMOS signals, allowing devices to be
in closer proximity to antennas, mixers or other sensitive analog front end
systems. CML also has common mode
termination. This gives CML a better
common mode behavior than LVDS.
LVDS is only terminated differentially,
which does not reject any common
mode signal that may appear on the
transmission line—another limiting
factor in LVDS signaling.
CML Power Consumption
With a constant 16mA of bias current
and a voltage swing of 800mV differential, CML logic consumes a moderate
amount of power. For an equal data
rate, CML logic consumes less total
power than PECL and LVPECL. A
single CML driver uses more power
than a single LVDS driver, but only
marginally more that the three pairs
of LVDS drivers required for a typical
LVDS serial bus.
8B/10B Encoding Makes for
Simple Connection
ADC requires three or more pairs,
and a typical parallel ADC can require
more than 16 pairs.
The complexity of decoding 8B/10B
lies in the receiver. Fortunately Xilinx,
Altera and Lattice have solutions to
receive data from the LTC2274 and
decode the 8B/10B data, simplifying
the collection of 8B/10B data. Other
8B/10B decoding solutions may be
available. The FPGA required to receive
data from the LTC2274 must be able
to receive high speed serial transmissions of 2GHz or more.
The 8B/10B encoding process results
in an average DC offset of zero, allowing
the data to be routed through transformers or iber channel transceivers
that can provide isolation between
the digital and analog realm. 8B/10B
encoding also does not require a framing signal or a data clock, whereas
both are required in traditional serial
communication. 8B/10B encoding
transmits data over a single pair of
data lines, whereas a typical serial
Conclusion
on the complexity and limitations of
the power supply circuits.
The LTC35xx family provides very
simple and lexible control of all essential power supply functions. The
LTC3566 and LTC3586 employ dedicated I/O control pins for enabling,
disabling and changing DC/DC
operating modes. Voltages on these
parts are ixed and set with external resistor dividers. The LTC3555,
LTC3556 and LTC3567 accommodate
either I2C control or simple I/O pins
to control the supplies. The LTC3556
provides a three-state SEQ pin to allow the power up sequence of its three
DC/DC converters to be programmed
via pin-strapping. Those parts with
I2C VOUT control power-up at their
maximum VOUT (as determined by the
FB servo point and external dividers)
when enabled via simple I/O, and
can independently reduce VOUT by as
much as 50% in equal 16-step increments via I2C.
All DC/DC converters in all the
PMICs discussed here can survive
an indeinite output fault. The parts
all provide a RST output and all converters are actively pulled down in
shutdown to ensure proper power-up
sequencing. The LTC3586 contains an
additional fault handing feature that
automatically powers down all DC/DC
converters whenever a valid fault is
detected. In short, the entire family
is designed for simple, lexible and
trouble-free control and operation.
Without sacrificing resolution or
sample rate, the LTC2274 delivers full
16-bit performance at 105Msps over
a single pair of transmission lines,
greatly simplifying layout and saving
valuable board space. This mitigates
interaction with other circuitry in
software deined radio, base station or
industrial applications which involve
many channels of an ADC routed to
one FPGA. L
LTC5xx, continued from page power synchronous boost converter
(Figure 5).
The fully integrated boost in the
LTC3586 can regulate up to a 5V output with up to 800mA from a battery
voltage as low as 3V. The regulator has
built in output disconnect making it
well-suited for USB OTG supplies or
for powering motors in printer and
camera applications. The current
mode synchronous boost is internally
compensated and operates at a ixed
2.25MHz switching frequency. Pulseskipping at low loads achieves low
noise output for driving high power
audio circuits.
I2C, Programmable
Sequencing and Easy I/O
Despite the progress in new cutting
edge features and design, one old
problem does not go away: power
supply control. Power supplies require
startup and power down sequencing,
fault detection/reporting/handling
and voltage and operating mode adjustments. Getting it all right can be
a system control nightmare depending
Linear Technology Magazine • September 2008
Conclusion
Linear Technology’s latest PMIC
products improve the performance
and simplify the design of a wide variety of portable power management
applications. Instead of kitchen sink
alternatives with large packages,
Linear Technology offers a number of
devices with various feature mixes in
small packages. These new PMICs are
simple to use, highly integrated and
high performance, allowing for shorter
design times, greater PCB lexibility,
and better power/thermal management than traditional solutions. L
15
L DESIGN FEATURES
Synchronous Buck Controller in
3mm × 3mm QFN Fits Automotive
and Industrial Applications with
4V–38V Input Capability
by Mark Mercer
Introduction
The LTC3851 has ±1% output voltage
tolerance over temperature. The part’s
low minimum on-time (90ns, typical)
allows for low duty cycle operation
even with switching frequencies as
high as 750kHz.
Two Current Sensing Options
The LTC3851 features a high input
impedance current sense comparator.
This allows the use of the inductor’s
DC resistance (DCR) as the current
sense element in conjunction with an
RC ilter. DCR current sensing allows
the designer to eliminate the need for a
discrete sense resistor, thereby maximizing eficiency and lowering solution
cost. Alternately, higher current sense
accuracy may be achieved by connecting the SENSE+ and SENSE– pins to a
precision sense resistor in series with
the inductor. The LTC3851 offers the
choice of three pin-selectable maximum current sense thresholds (30mV,
1000
750
OSCILLATOR FREQUENCY (kHz)
The LTC3851 is a versatile synchronous step-down switching regulator
controller that is available in a space
saving 16-lead 3mm × 3mm QFN or
convenient narrow SSOP packages. Its
wide input range of 4V to 38V makes it
well-suited for regulating power from a
variety of sources, including automotive batteries, 24V industrial supplies
and unregulated wall transformers.
The strong onboard drivers allow the
use of high power external MOSFETs
to produce output currents up to 20A
with output voltages ranging from
0.8V to 5.5V.
The constant frequency peak current mode control architecture provides
excellent line and load regulation along
with load current sharing capability
and dependable cycle-by-cycle current
limiting. OPTI-LOOP® compensation
simpliies loop stability design and
provides well-behaved regulation over
a broad range of operating conditions.
500
250
100
10
36
60
160
RFREQ (k)
1000
Figure 1. Relationship between oscillator
frequency and resistor connected between
FREQ/PLLFLTR and GND
50mV and 75mV) to accommodate a
wide range of DCR values and output
current levels.
As with all constant frequency,
peak current mode control switching
regulators, the LTC3851 utilizes slope
compensation to prevent sub-harmonic oscillations at high duty cycles. This
VIN
4.5V TO 32V
MODE/PLLIN
VIN
+
RFREQ
82.5k
FREQ/PLLFLTR
CIN
22µF
HAT2170H
TG
0.1µF
RUN
0.1µF
BOOST
LTC3851
0.1µF
TK/SS
15k
2200pF
L1
0.68µH
SW
VOUT
3.3V
15A
CMDSH05-4
330pF
ITH
3.01k
INTVCC
47pF
154k
1%
4.7µF
VFB
BG
SENSE–
GND
SENSE+
ILIM
HAT2170H
48.7k
1%
+
COUT
330µF
s2
0.047µF
30.1k
COUT: SANYO 6TPE330MIL
CIN: SANYO 63HVH22M
L1: VISHAY IHLP5050-EZERR68M01
Figure 2. High efficiency 3.3V/15A power supply with DCR sensing
16
Linear Technology Magazine • September 2008
DESIGN FEATURES L
is accomplished internally by adding
a compensating ramp to the inductor
current signal. Normally, this results
in a >40% reduction of maximum
inductor peak current at high duty
cycles. However, the LTC3851 uses
a novel scheme that allows the maximum peak inductor current to remain
stable throughout all duty cycles.
100
Burst Mode
OPERATION
90
EFFICIENCY (%)
80
70
60
50
PULSESKIPPING
MODE
CONTINUOUS
CONDUCTION MODE
40
30
20
VIN = 12V
VOUT = 3.3V
10
Versatility
0
0.01
During heavy load operation, the
LTC3851 operates in constant frequency, continuous conduction mode.
At light loads, it can be conigured
to operate in high eficiency Burst
Mode® operation, constant frequency
pulse-skipping mode or forced continuous conduction mode. Burst Mode
operation offers the highest eficiency
because energy is transferred from the
input to the output in pulse trains
of one to several cycles. During the
intervening period between pulse
trains, the top and bottom MOSFETs
are turned off and only the output
capacitor provides current to the load.
Forced continuous conduction mode
results in the lowest output voltage
ripple, but is the least eficient at light
loads. Pulse-skipping mode offers
a compromise—lower output ripple
than Burst Mode operation and more
eficiency than continuous conduction mode.
0.1
1
10
LOAD CURRENT (A)
100
Figure 3. Efficiency vs load current with three
modes of operation for the circuit of Figure 2
The switching frequency of the
LTC3851 may be programmed from
250kHz to 750kHz by the resistor,
RFREQ, connected to the FREQ/PLLFLTR pin. This provides the lexibility
needed to optimize eficiency. Figure 1
shows a plot of the switching frequency
vs RFREQ. Additionally, the switching
frequency may be synchronized to
an external clock. While doing so,
the LTC3851 will operate in forced
continuous conduction mode.
The output voltage can be ramped
during start-up by means of an adjustable soft-start function, or it can
track an external ramp signal. Track
and soft-start control are combined in
a single pin, TK/SS. Whenever TK/SS
is less than 0.64V, the LTC3851 operates in pulse-skipping mode. This
feature allows for starting up into a
pre-biased load. When TK/SS is between 0.64V and 0.74V, the regulator
operates in forced continuous mode
to ensure a smooth transition from
start-up to steady state. Once TK/SS
exceeds 0.74V, the mode of operation is determined by the state of the
MODE/PLLIN pin.
The RUN pin enables or disables
the LTC3851. This pin has a precision
1.22V turn-on threshold which is useful for power supply sequencing. The
turn-off threshold is 1.10V. There is
an internal 2µA pull-up current source
on the RUN pin.
The LTC3851’s fault protection
features include foldback current
limiting, output overvoltage detection
and input undervoltage detection. If
an overload event causes the output
to fall to less than 40% of the target
regulation value, then the LTC3851
folds back the maximum current sense
threshold. This reduces the on-time in
order to minimize power dissipation in
the top MOSFET. If the output voltage
is more than 10% above the target
regulation value, the bottom MOSFET
turns on until the output falls back
into regulation. If the input voltage is
allowed to fall low enough such than
the output of the internal linear regulator falls below 3.2V, then switching
operation is disabled. This feature
continued on page VIN
6V TO 14V
PLLIN
350kHz
0.01µF
MODE/PLLIN
VIN
FREQ/PLLFLTR
TG
+
10k
RJK0305DPB
CIN
180µF
0.1µF
RUN
1000pF
BOOST
LTC3851
0.1µF
TK/SS
7.5k
1000pF
L1
0.68µH
SW
RSENSE
0.002Ω
VOUT
1.5V
15A
CMDSH-3
100pF
ITH
33pF
INTVCC
42.2k
1%
4.7µF
VFB
BG
SENSE–
GND
SENSE+
ILIM
RJK0301DPB
48.7k
1%
+
COUT
330µF
s2
1000pF
10Ω
COUT: SANYO 2R5TPE330M9
L1: SUMIDA CEP125-OR6MC
10Ω
Figure 4. High efficiency 1.5V/15A power supply synchronized to 350kHz
Linear Technology Magazine • September 2008
17
L DESIGN FEATURES
Feature-Packed Charger Handles
All Battery Chemistries and Produces
3A/50W for Fast Charging from a
4mm × 4mm QFN
by James A. McKenzie
Introduction
The LTC4009, LTC4009-1 and
LTC4009-2 are a family of high power
battery charger ICs that achieve a small
circuit size and high performance without compromising functionality. The
family operates with high eficiency
while packing the most desirable
charging and protection features into
a space-eficient 20-lead 4mm × 4mm
QFN package. When combined with
just a few external components and
termination control, the LTC4009
family facilitates construction of
chargers capable of delivering up to
3A to batteries with output power
levels approaching 50W. These ICs are
especially well-suited to implementing
microprocessor-controlled chargers
for all chemistry types, including
smart batteries.
High Performance
The LTC4009 family builds upon the
proven quasi-constant frequency,
constant off-time PWM control architecture found in other Linear
Technology battery chargers such as
the LTC4006, LTC4007, LTC4008,
and LTC4011. This buck topology
provides continuous switching with
synchronous rectiication, even with
no load current.
Normally the charger operates over
a wide duty cycle range in a manner
similar to a ixed-frequency PWM controller running at 550kHz. However,
if the input or output voltage drives
the duty cycle to extremes, below 20%
or above 80%, the LTC4009 smoothly
adjusts the operating frequency
downward to avoid pulse-skipping
that might otherwise begin to occur
at 550kHz. Under very low dropout
conditions requiring high duty cycle
operation, the internal watchdog
timer on the LTC4009 prevents the
charger from switching below 25kHz.
This allows the charger to achieve a
Table 1. LTC4009 family features
Feature
LTC4009
LTC4009-1
LTC4009-2
LTC4008
Output Voltage Selection
External Resistor
Divider
Pin Programmable
at 4.1V/cell
Pin Programmable
at 4.2V/cell
External Resistor
Divider
±0.5% + Divider Error
±0.6%
±0.6%
±0.8% + Divider Error
Maximum Charge Current
3A
3A
3A
4A
Charge Current Accuracy
±5%
±5%
±5%
±5%
Input Current Limit Accuracy
±4%
±4%
±4%
±7%
Input Current Limit/Indicator
L
L
L
L
External PWM Switching MOSFETs
All NFET
All NFET
All NFET
PFET/NFET
Nominal PWM Frequency
550kHz
550kHz
550kHz
300kHz
Shutdown Pin
L
L
L
Merged with ACP
C/10 Indicator
L
L
L
L
Charge Current Monitor
L
L
L
L
Termination Method
External
External
External
External
Output Voltage Accuracy
(Room Temperature)
18
Fault Indicator
L
Thermistor Interface
L
INFET Control
L
Linear Technology Magazine • September 2008
DESIGN FEATURES L
maximum duty cycle of 98% or higher
without producing frequencies that
could extend down into the audible
range.
With a synchronous rectiier, not
only are high current applications
supported at eficiency levels greater
than 90%, but switching activity is
continuous and independent of the
load current. Maintaining full continuous conduction mode in the inductor
at inal output voltage, under no-load
conditions, avoids pulse-skipping
which can generate audible noise and
provide poor load regulation.
The input current limit accuracy
is typically ±3% and a maximum of
±4% over the full operating temperature range. Output voltage accuracy
is typically ±0.5% and a maximum of
±0.8% over temperature.
Small PCB Footprint
Besides its small surface mount package size, the LTC4009 family offers
other features that drive down the
total solution size.
For instance, as shown in Figure 1,
the family supports direct drive of both
an N-channel MOSFET power switch
and N-channel MOSFET synchronous
FROM
ADAPTER
15V AT 2A
capabilities tailored to synchronous
buck PWM switching topologies.
Increasing the switching frequency
to 550kHz and adjusting internal bias
circuits to allow higher charge current
ripple minimize both the inductor size
and output capacitance requirements.
This is particularly important because
these components tend to dominate
the overall solution size due to continual improvements in IC and passive
SMD packaging technology.
The physical layout of a typical
3A application is shown in Figure 3,
requiring only 240mm2 of board
space.
When combined with just
a few external components
and termination control, the
LTC4009 family facilitates
construction of chargers
capable of delivering
up to 3A to batteries
with output power levels
approaching 50W. These
ICs are especially well
suited to implementing
microprocessor-controlled
chargers for all chemistry
types, including smart
batteries.
A Rich Tradition
rectiier. N-channel MOSFETs are
desirable in high current applications because of their lower RDS(ON),
and the LTC4009 family uses a novel
adaptive gate drive that is insensitive
to MOSFET inertial delays to avoid
overlap conduction losses. Many
suppliers now source dual N-channel
MOSFETs in a single space-eficient
package, often with individual drive
D2
The LTC4009 family builds upon the
general purpose features offered by
the LTC4008 and the output voltage
programming convenience afforded
by the LTC4006. Each member of the
LTC4009 family contains the same
charge current and input current limit
programming features, along with a
full complement of charge monitoring, safety and fault management
functions. The LTC4009 has a fully
adjustable output voltage, which is set
with a simple resistor divider. Charge
25mΩ
POWER TO SYSTEM
0.1µF
3k
D1
RED
0.1µF
CHRG
D5
5.1k
CLP
DCIN
22.1k
CLN
DCDIV
20µF
BOOST
0.1µF
2.43k
TGATE
LTC4009
SW
TO/FROM
MCU
ACP
INTVDD
ICL
BGATE
Q2
D3
D4
Q3
L1
6.8µH
2µF
SHDN
GND
3.01k
ITH
CSP
0.1µF
33mΩ
3.01k
6.04k
CSN
BAT
FBDIV
PROG
4.7nF
26.7k
BULK
CHARGE
Q1
53.6k
20µF
294k
+
10pF
VFB
31.2k
3-CELL
Li-Ion
BATTERY
STACK
D2, D4, D5: MBR230LSFT1
D3:CMDSH-3
Q1: 2N7002
Q2, Q3: Si7212DN
L1: IHLP-2525CZ-11
Figure 1. A 12.6V, 3A lithium-ion charger
Linear Technology Magazine • September 2008
19
L DESIGN FEATURES
current accuracy is maintained at
output voltages below 6V, making the
LTC4009 ideal for charging nickelbased chemistries or supercaps.
The LTC4009-1 and LTC4009-2
are easy to use in lithium-based
battery products containing one to
four series cells. Each has a range
of output voltages that can be selected simply by strapping two pins
to either ground or the output of the
onboard 5V regulator, as shown for the
LTC4009-1 in Figure 2. No other external components are required to set
this precision voltage. The LTC4009-1
provides 4.1V/cell settings in support
of evolving consumer product safety
standards or coke-anode cells, while
the LTC4009-2 utilizes 4.2V/cell for
conventional full-capacity charging
of graphite-anode lithium-ion packs.
The ICs contain a dedicated PMOS
switch that during shutdown removes
the additional current drained from
the battery by the resistive feedback
divider, whether external or internal.
Table 1 compares the features of the
LTC4009 family to the LTC4008.
Battery Charge Management
The LTC4009 family contains all of the
features required for complete external
charge control and state monitoring
with a logic-level shutdown control
input and three open-drain status
outputs. All charging is unconditionally suspended and battery drain
is reduced to its lowest levels if the
SHDN input is asserted by driving the
pin to ground. DC input supply voltage is sensed by feeding an external
resistor voltage divider to the DCDIV
sense input. The AC present status
output indicates whether or not this
input voltage is within a valid range
for charging under all modes of operation, whether charging is in progress
or suspended. There is a charge status
output that indicates when the battery
is being charged. The drive level of
this pin changes from low impedance
(about 2k) to a 25µA pull-down current
source to indicate that the charge current has dropped to one-tenth of the
programmed full-scale bulk value.
These control inputs and status
outputs of the LTC4009, along with
20
D1
18V AT 3A FROM
ADAPTER
33mΩ
0.1µF
20µF
0.1µF
DCIN
22.1k
DCDIV
CLP
5.1k
CLN
BOOST
0.1µF
LTC4009-1
1.74k
Q2
TGATE
D2
CHRG
TO/FROM
MCU
POWER TO
SYSTEM
ACP
SW
INTVDD
ICL
BGATE
SHDN
GND
ITH
CSP
D4
2µF
Q3
D3
L1
6.8µH
3.01k
6.04k
50mΩ
3.01k
0.1µF
CHARGE
CURRENT
MONITOR
CSN
PROG
4.7nF
BAT
FVS1
20µF
26.7k
FVS0
INTVDD
+
4-CELL
Li-Ion
BATTERY
STACK
D1, D3, D4: MBR230LSFT1
D2: CMDSH-3
Q1, Q2: Si7212DN
L1: IHLP-2525CZ-11
Figure 2. A 16.4V, 2A lithium-ion charger
the analog current monitor output,
can be used by the host system to
perform necessary preconditioning,
charge termination and safety timing
functions.
Charge Fault Management:
Safety First
The LTC4009 family has a built-in fault
management system that suspends
charging immediately for various internal and external fault conditions.
First, battery overvoltage protection is
provided with a comparator that looks
for sudden loss of battery load during
charge. This comparator instantly sus-
Figure 3. A typical LTC4009 application layout
pends PWM activity when the battery
voltage rises above the programmed
output voltage by 6%. This protects the
charger, the battery, and downstream
components in charger-fed topologies
under the condition where the battery
is suddenly removed or if internal
battery pack electronics momentarily
disconnect the load in order to perform
functions such as calibration or pulse
charging.
Next, the DC sensing input has both
under and overvoltage threshold limits
to ensure the system is protected from
starting or running during unsafe
conditions, such as transient input
overvoltage or an overloaded DC input
adapter.
These parts have several means of
detecting or avoiding reverse charge
current. For instance, reverse current
can occur in a synchronous system
if a slightly overcharged battery is
inserted, with the resulting reverse
current discharging the battery and/or
damaging other system components.
To prevent reverse currents, the parts
in this family irst monitor the PROG
pin for reverse current. PROG outputs
a voltage analog of the average charge
current lowing in the system. Next, the
continued on page Linear Technology Magazine • September 2008
DESIGN FEATURES L
Dual Hot Swap Controller in
3mm × 2mm DFN is Perfect for
Backplane or Card Resident
1V–6V Applications
by CY Lai
Introduction
High availability electronics, such
as those used in telecom, real-time
transaction processing, hospitals and
air trafic control systems, cannot afford any down time. These systems
must continue to operate even as
components are added or removed
(hot swapped) for expansion, upgrades
or maintenance. The Hot Swap™
systems must be carefully designed
to avoid burned PCB traces and
power brownouts, which can result
in system resets and data loss. The
LTC4224 ensures dependable Hot
Swap design while simplifying and
shrinking total solution size. It does
this by combining two feature-rich and
independent Hot Swap controllers for
1V–6V applications in a 3mm × 2mm
DFN package.
Figure 1 shows a functional block
diagram of the LTC4224. The ON pin
is used to turn an external N-channel
MOSFET on or off via the GATE pin.
When commanded on, an internal
charge pump pulls the GATE above
the supply rail to fully enhance the
MOSFET, reducing its series resistance to several mΩ. The LTC4224’s
ability to derive power from the higher
SENSE1
5.5V
CHARGE
PUMP
–
25mV
10µA
ACL1
GATE1
VCC
+
+
–
VCC1
VCC
GATE
PULLDOWN
+
ON1
ECB1
0.8V
–
0.8V
FAULT
+
2.4V
VCC
LOGIC
CONTROL
–
10µA
VCC2 UV
ON2
ECB2
+
–
CHARGE
PUMP
+
–
GATE
PULLDOWN
10µA
25mV
+
ACL2
5.5V
VCC
–
SENSE2
GND
Figure 1. Functional block diagram of the LTC4224
Linear Technology Magazine • September 2008
ON2
+
–
0.8V
VCC2
ON1
+
–
VCC1 UV
VCC
10µA
0.8V
GATE2
of its two supplies allows it to control
load voltages as low as 1V. Active current limiting (ACL) acts on the GATE
when the load current causes more
than 25mV of voltage drop across the
sense resistor. An electronic circuit
breaker (ECB) performs the role of a
timekeeper, latching off the MOSFET
in the event of a prolonged current
overload.
Operation
Figure 2 shows the LTC4224 together
with two N-channel MOSFETs and two
sense resistors in a 5V backplane resident Hot Swap application. Initially,
the ON pin is pulled high in the absence of an add-in card and the GATE
is held low, shutting off the MOSFET.
When an add-in card is fully inserted
into the backplane connector, the ON
pin is pulled low through the ground
connections on the card connector.
Spurious ON transitions can occur
as the connectors mate. To prevent
the MOSFET from turning on prematurely, the LTC4224 waits out these
short term transitions with an internal
10ms debounce delay that restarts
every time ON transitions high.
To turn on the MOSFET, an internal charge pump sources 10µA to
soft-start the GATE with a slew rate
of 10µA/CISS, where CISS is the external MOSFET’s gate capacitance. The
start-up inrush current lowing into
the load capacitor COUT is limited to
(COUT/CISS) • 10µA. However, if the
sense resistor voltage drop becomes
too large, the inrush current is limited
at 25mV/RSENSE by the ACL. The ECB
monitors the ACL, and if it detects that
the current limit is still active 5ms after
the GATE began ramping, the MOSFET
latches off and FAULT pulls low. If
21
L DESIGN FEATURES
R1
0.004Ω
Q1
Si7336ADP
5V
5A
5V
PRSNT1
R3
10k
VCC1
SENSE1
GATE1
FAULT
ON1
LTC4224
CARD
CONNECTOR1
BACKPLANE
CONNECTOR2
CARD
CONNECTOR2
ON2
GND
VCC2
BACKPLANE
CONNECTOR1
SENSE2
GATE2
PRSNT2
R2
0.004Ω
5V
5A
Q2
Si7336ADP
Figure 2. Hot Swap application for two add-in cards
COUT cannot be suficiently charged
within this period, connect a capacitor from GATE to ground to lower the
inrush current, as shown in Figure 3.
With CGATE, the inrush current is reduced to (COUT/(CGATE + CISS)) • 10µA.
Adjusting CGATE so that the inrush
current stays below the ECB threshold
prevents ECB faults with large load
capacitors.
Overcurrent Protection
An important feature of the LTC4224
is its 25mV electronic circuit breaker
(ECB) threshold with a 10% tolerance.
This low ECB threshold allows the use
of sense resistors with lower power ratings and hence smaller packages. In
addition, the ECB threshold must not
cut excessively into the supply voltage
tolerance of downstream circuits. For
instance, if the downstream circuits
can tolerate at most a 5% variation on
the 1V supply, the ECB threshold of an
upstream Hot Swap controller must be
signiicantly lower than 50mV.
To guard against damage to the
external MOSFET from excessive
power dissipation, active current
limiting (ACL) regulates the gate to
limit the sense resistor’s voltage drop
to about 25mV. To minimize external
components, the current limit loop
is compensated by the parasitic gate
capacitance CISS of the MOSFET and
remains stable for CISS values as low
as 600pF. During ACL, the ECB activates and initiates an internal time-out
period of 5ms. The waveform in Figure
4 shows the LTC4224 limiting the
GATE1
5V/DIV
R1
0.015Ω
current and subsequently latching
off the MOSFET due to a mild current
overload at the output lasting longer
than 5ms. FAULT is pulled low; this
could either instruct the microprocessor to take actions or light an LED to
attract operator’s attention.
In the event of a severe short-circuit,
the current typically overshoots the
current limit level signiicantly as the
gate overdrive of the external MOSFET
is large initially. The LTC4224 responds in less than 0.1µs to swiftly
discharge the gate with a 100mA current sink. Figure 5 shows the LTC4224
bringing the current under control in
less than 0.5µs when a 3.3V rail is
shorted into a 10mΩ load without any
load capacitance. Also due to the fast
ACL is the absence of gate undershoot,
despite the speed at which the gate is
discharged. The potential peak current
is dictated by DC resistances along the
power path (trace resistance + RDSON of
the MOSFET + RSENSE + 10mΩ), while
the path’s parasitic inductance limits
the current slew rate.
After the MOSFET latches off, the
ON pin must be pulled above 0.8V
to reset the internal fault latch. Alternatively, recycle the supply below
its UV level. The LTC4224-1 latches
off after a fault, while the LTC4224-2
automatically tries to apply power four
seconds after latching off.
Optical Transceiver
Hot Swap Application
Optical transceivers such as those
speciied for the popular XENPAK/X2
GATE2
5V/DIV
Q1
5V
CLOAD
VCC1 SENSE1
VOUT2
5V/DIV
VOUT1
5V/DIV
RG
10Ω
IGATE
CGATE
IOUT1
1A/DIV
IOUT2
8A/DIV
GATE1
LTC4224
2ms/DIV
Figure 3. A method to adjust inrush current
by gate capacitor. RG prevents parasitic selfoscillation in Q1
22
Figure 4. Active current limiting latches
off the external MOSFET following a mild
overcurrent
0.5µs/DIV
Figure 5. Fast current limit isolates severe
short circuit fault in less than 0.5µs
Linear Technology Magazine • September 2008
DESIGN FEATURES L
3mm × 2mm DFN package. Fast
current limiting ensures that system
disturbances are minimized during
a severe overload and that faults are
R1
0.015Ω
R2
0.010Ω
Q2
R3
390
VCC1
SENSE1
VCC2
SENSE2
GATE1 GATE2
MOD DETECT
ON1
D1
LTC4224
FAULT
ON2
RMOD_DET
1k
GND
GND
CONNECTOR
PLUG-IN CARD
Figure 6. XENPAK/X2 optical module Hot Swap application
3.3V
5V
R2
R1
2
8
11
5
1
9
6
10
FDS6911
3
5
4
4
6
3
7
2
Linear Technology Magazine • September 2008
3.3V
2A
3.3V
BULK SUPPLY
BYPASS CAPACITOR
7
The LTC4224 simpliies the design of
low voltage Hot Swap applications by
integrating two Hot Swap controller
and timing delay circuits in a tiny
5V
1A
Q1
BULK SUPPLY
BYPASS CAPACITOR
1
Conclusion
FDS6911
5V
5V/5A, 3.3V/5A
Hot Swap Application
The LTC4224 can also reside on an
add-in card as shown in Figure 8.
There are no bulk capacitors on the
inputs as these could draw large inrush current. In their place are the
Transient Voltage Suppressors (Z1
and Z2) and RC Snubber networks.
During current transients, inductive
kickback can cause the input supply to
swing beyond the absolute maximum
(ABSMAX) rating of LTC4224’s input
pins without the TVS. By clamping the
voltage, the TVS protects the LTC4224
from damage and an ABSMAX rating
of 9V provides margin for the selection of the TVS. Snubbers damp the
parasitic LC tanks to eliminate ringing
on the input supplies. The Si7336ADP
has been chosen for its SOA, 20V
Gate-Source breakdown voltage and
low RDSON.
quickly isolated. The LTC4224 offers
a complete and robust Hot Swap solution for XENPAK/X2 optical modules
that can be implemented in an SO8
footprint. L
8
Multi- Source Agreements (MSA) are
employed in high speed networking
routers as an interface between optical and electrical signals. The MSA
mandates hot plug capability for
transceiver modules, which are supplied with 5V, 3.3V and 1.xV.
A Hot Swap application based on
the LTC4224 for the 5V and 3.3V rails
is shown in Figure 6. Typically, a dedicated DC/DC converter controls the
1.xV rail and limits the inrush current
for each module. As the optical module
consumes relatively little power, a dual
FET such as the FDS6911 is a good
candidate for the power switches,
saving cost and minimizing area. For
the tiniest solution, sense resistors in
a 0603 case size are selected. Figure 7
shows the full solution, which its in
the footprint of an SO8 package. In an
application where all the three supply
rails need to be hot swapped, three
LTC4224s can be used to control the
power to two modules, all in a solution
no larger than the footprint of three
SO8 packages.
LTC4224
BOTTOM SIDE
TOP SIDE
Figure 7. A compact PCB layout of the sense resistors, MOSFET and the LTC4224
R1
0.004Ω
Q1
Si7336ADP
5V
Z1
RSNUB1
10Ω
CSNUB1
100nF
R2
0.004Ω
Q2
Si7336ADP
3.3V
Z2
RSNUB2
10Ω
CSNUB2
100nF
R3
10k
VCC1
PWRFLT
FAULT
PWREN
ON1
SENSE1
VCC2
SENSE2
5V
5A
GATE1
3.3V
5A
GATE2
LTC4224
ON2
GND
BACKPLANE
CONNECTOR
GND
CARD
CONNECTOR
Z1: SMAJ6.5A
Z2: SMAJ5.0A
Figure 8. A 5V and 3.3V card resident Hot Swap application
23
L DESIGN FEATURES
0V to 18V Ideal Diode Controller
Saves Watts and Space over Schottky
by Pinkesh Sachdev
Introduction
Schottky diodes are used in a variety
of ways to implement multisource
power systems. For instance, high
availability electronic systems, such
as network and storage servers, use
power Schottky diode-OR circuits to
realize a redundant power system.
Diode-ORing is also used in systems
that have alternate power sources,
such as an AC wall adapter and a
backup battery feed. Power diodes
can be combined with capacitors to
hold up a load voltage during an input brownout. In this case, the power
diodes are placed in series with the
input voltage, with the capacitors on
the load side of the diode. While the
capacitors provide power, the reversebiased diode isolates the load from the
sagging input.
Schottky diodes sufice for these
applications when currents are below
a few amperes, but for higher currents, the excess power dissipated in
the diode due to its forward voltage
drop demands a better solution. For
instance, 5A lowing through a diode
with a 0.5V drop wastes 2.5W within
the diode. This heat must be dissipated
with dedicated copper area on the
PCB or heat sinks bolted to the diode,
both of which take signiicant space.
The diode’s forward drop also makes
Si4438DY
TO LOAD
2.9V TO 18V
OPTIONAL
0.1µF
CPO SOURCE VIN
GATE
0.1µF
OUT
MOSFET ON
STATUS
STATUS
VCC
UV
LTC4352
FAULT
OV
REV
FAULT
GND
Figure 1. The LTC4352 controlling an N-Channel MOSFET replaces a power diode and associated
heat sink to save power, PCB area, and voltage drop. Also shown: the small PCB footprint of the
ideal diode circuit using a 3mm × 3mm DFN-12 packaged LTC4352 and SO-8 size MOSFET.
it impractical for low voltage applications. This problem calls out for an
ideal diode with a zero forward voltage
drop to save power and space.
The LTC4352 ideal diode controller
in tandem with an N-channel MOSFET
creates a near-ideal diode for use with
0V to 18V input supplies. Figure 1 illustrates the simplicity of this solution.
This ideal diode circuit can replace a
power Schottky diode to create a highly
eficient power ORing or supply holdup
application. Figure 2 shows the power
savings of the ideal diode circuit over
a Schottky diode. 3.5W is saved at
10A, and the saving increases with
load current. With its fast dynamic
response, the controller excels in low
voltage diode-OR applications which
are more sensitive to voltage droop.
Si7336ADP
SUPPLY
INPUT
OUTPUT
TO LOAD
0.1µF
VIN
SOURCE
GATE
CPO
VIN VCC
VCC
100µA
LDO
0.1µF
25mV
+
CHARGE
PUMP
AMP
+
–
4.0
–
+
POWER DISSIPATION (W)
3.0
OV
POWER
SAVED
1.0
REV
0.5
0
MOSFET (Si7336ADP)
1V
0
2
4
6
LOAD CURRENT (A)
8
+
–
MOSFET
ON
DETECT
VCC
10µA
LOGIC
OPEN
MOSFET
DETECT
FAULT
10
Figure 2. As load current increases, so do the
power savings gained from using an ideal diode
(LTC4352 + Si7336ADP) instead of a power
Schottky diode (SBG1025L).
24
–
+
2.0
1.5
STATUS
0.5V
2.5
VCC
10µA
UV
3.5
DIODE (SBG1025L)
OUT
GND
Figure 3. Simplified internals of the LTC4352
Linear Technology Magazine • September 2008
DESIGN FEATURES L
What Makes It Ideal?
CONSTANT
RDS(ON)
25mV
RDS(ON)
CURRENT (A)
The LTC4352 monitors the differential
voltage across the MOSFET source
(the “anode”) and drain (the “cathode”)
terminals. The MOSFET has an intrinsic source-to-drain body diode which
conducts the load current at initial
power-up. When the input voltage is
higher than the output, the MOSFET is
turned on, resulting in a forward voltage drop of ILOAD • RDS(ON). The RDS(ON)
can be suitably chosen to provide an
easy 10x reduction over a Schottky
diode’s voltage drop. When the input
drops below the output, the MOSFET
is turned off, thus emulating the behavior of a reverse biased diode.
An inferior ideal diode control technique monitors the voltage across the
MOSFET with a hysteretic comparator.
For example, the MOSFET could be
turned on whenever the input to output voltage exceeds 25mV. However,
choosing the lower turn-off threshold
can be tricky. Setting it to a positive
forward voltage drop, say 5mV, causes
the MOSFET to be turned off and
on repeatedly at light load currents.
Setting it to a negative value, such as
–5mV, allows DC reverse current.
LTC4352
CONSTANT
VOLTAGE
0
SCHOTTKY
DIODE
0.025
0.5
FORWARD VOLTAGE (V)
Figure 4. The forward I-V characteristic of the
LTC4352 ideal diode vs a Schottky diode.
The LTC4352 implements a linear
control method to avoid the problems
of the comparator-based technique.
It servos the gate of the MOSFET to
maintain the forward voltage drop
across the MOSFET at 25mV (AMP
of Figure 3). At light load currents,
the gate of the MOSFET is slightly
above its threshold voltage to create a resistance of 25mV/ILOAD. As
the load current increases, the gate
voltage rises to reduce the MOSFET
resistance. Ultimately, at large load
currents, the MOSFET gate is driven
fully on, and the forward voltage drop
Fast Switch Control
Most ideal diode circuits suffer slower
transient response compared to conventional diodes. The LTC4352, on
the other hand, responds quickly to
changes in the input to output voltage. A powerful driver turns off the
MOSFET to protect the input supply
and board traces from large reverse
currents. Similarly, the driver turns
on the switch rapidly to limit voltage
droop during supply switchover in
diode-OR applications.
Figure 5 shows a fast switchover
event occurring in a 3.3V ideal diode-
VIN1
Q1
Si4438DY
VIN1
3.5V
rises linearly with load current as ILOAD
• RDS(ON). Figure 4 shows the resulting
ideal diode I-V characteristic.
In a reverse voltage condition, the
gate is servoed low to completely turn
off the MOSFET, thus avoiding DC
reverse current. The linear method
also provides a smooth switchover
of currents for slowly crossing input
supplies in diode-OR applications. In
fact, depending on MOSFET and trace
impedances, the input supplies share
the load current when their voltages
are nearly equal.
VIN2
VIN2
VIN1
0.1µF
VOLTAGE
(2V/DIV)
CPO SOURCE VIN
GATE
VCC
0.1µF
VLOAD
OUT
STATUS
UV
LTC4352
OV
FAULT
REV
GND
TIME (5µs/DIV)
Q3
Si4438DY
VIN2
3.3V
CL
100µF
0.1µF
CPO SOURCE VIN
VCC
0.1µF
UV
GATE OUT
VOLTAGE
(2V/DIV)
STATUS
LTC4352
OV
REV
$VGATE = VGATE – VSOURCE
$VGATE1
IL
8A
$VGATE2
FAULT
GND
TIME (5µs/DIV)
a. Ideal diode-OR of 3.5V and 3.3V input supply.
b. Supply switchover from VIN1 to VIN2 due to short-circuit
on VIN1 shows minimal disturbance on load voltage.
Figure 5. Ideal diode-OR fast switchover
Linear Technology Magazine • September 2008
25
L DESIGN FEATURES
OR circuit. Initially VIN1 supplies the
entire load current since it is higher
than VIN2. In this state, MOSFET Q1
is on and Q3 is off. A short circuit
causes VIN1 to collapse below VIN2.
The LTC4352’s fast response shuts
off Q1 and turns on Q3 so that the
load current can now be supplied by
VIN2. This fast switchover minimizes
disturbance on the load voltage so that
downstream circuits can continue to
operate smoothly.
To achieve fast switch turn-on,
the LTC4352 uses an internal charge
pump with an external reservoir capacitor. This capacitor is connected
between the CPO and SOURCE pins.
CPO is the output of a charge pump
that can deliver up to 100µA of pullup current. The reservoir capacitor
accumulates and stores charge, which
can be called upon to produce 1.5A of
transient GATE pull-up current during a fast turn-on event. The reservoir
capacitor voltage drops after the fast
turn-on since it charge-shares with
the input gate capacitance (CISS) of
the MOSFET. For an acceptable drop,
the reservoir capacitor value should
be around 10 times the CISS of the
MOSFET.
It is easy to disable fast turn-on.
Omitting the reservoir capacitor slows
down the gate rise time as determined
by the CPO pull-up current charging
CISS. Slow gate turn-on may cause the
load to droop roughly a volt below the
input as current lows through the
MOSFET body diode until the channel
is enhanced. This may be acceptable
Q2
Si7336ADP
Table 1. Operating state of the LTC4352 ideal diode as indicated by the STATUS and FAULT lights
STATUS Green LED
FAULT Red LED
at higher input voltage applications,
such as 12V.
Do What No Diode
Has Done Before
The LTC4352 goes above and beyond
the functionality of a diode by incorporating input undervoltage and
overvoltage protection, outputs to
report status and fault information,
open MOSFET detection, and the ability to allow reverse current.
Figure 6 shows the LTC4352 in a 5V
ideal diode circuit with undervoltage
and overvoltage protection. The UV
and OV pins have comparators with a
0.5V trip threshold and 5mV hysteresis
(Figure 3). The resistive dividers from
the input supply to these pins set up an
input voltage window, typically 4.36V
to 5.78V, where the ideal diode function operates. The STATUS pin pulls
low to light up a green LED whenever
the gate is high and power is lowing
through the external MOSFET. For VIN
Q1
Si7336ADP
TO LOAD
5V
0.15µF
5V
5V
1k
31.6k
1%
VIN CPO
SOURCE
1k
1%
GATE
OUT
FAULT
UV
D2
1k
D1
FAULT
MOSFET ON
STATUS
OV
LTC4352
VCC
0.1 µF
3.09k
1%
GND
REV
D1: GREEN LED LN1351C
D2: RED LED LN1261CAL
Figure 6. A 5V ideal diode circuit with input undervoltage and overvoltage protection.
Ideal diode function operates for 4.36V < VIN < 5.78V, else GATE is low.
26
Ideal Diode
Operating State
LED State
MOSFET
UV/OV
OFF
NO
ON
NO
OFF
YES
OPEN
NO
outside the input voltage window, the
gate is held off and the FAULT pin pulls
low to signal a fault condition. A red
LED, D2, provides visual indication.
Back-to-back MOSFETs are needed to
block conduction through their intrinsic source-to-drain body diodes in the
gate low condition. A single MOSFET,
Q1, could be used in the case where
only a VIN out-of-range indication is
suficient. But care should be taken
that the load current lowing through
Q1’s body diode, when its gate is low,
does not cause excessive heat dissipation in the MOSFET.
The MOSFET switch could fail open
circuit or its RDS(ON) may degrade over
years of operation, increasing the voltage drop across the switch. A large drop
also results when excessive current
lows through the MOSFET, possibly
due to an output short circuit. The
LTC4352 detects such failures and
lags it through its FAULT pin. The
open MOSFET detection circuit trips
whenever it senses more than 250mV
of forward voltage drop across the
MOSFET—even with the gate turned
on. Note that this condition only causes
the FAULT pin to pull low, but no action is taken to turn off the switch.
Table 1 translates STATUS and FAULT
LED status to the operating state of
the LTC4352.
The input at the REV pin conigures
the LTC4352’s behavior for reverse
current. It is tied low for normal diode
operation, which blocks reverse current from lowing through the external
MOSFET. Driving REV above 1V turns
the gate completely on to its limit, even
during reverse current conditions.
Linear Technology Magazine • September 2008
DESIGN FEATURES L
Q2
Si7336ADP
Q1
Si7336ADP
TO LOAD
12V
CLOAD = 10mF
Z1
UV
(0.5V/DIV)
RG
10Ω
10k
LOAD
CG
0.1µF
105k
VOLTAGE
(5V/DIV)
SOURCE GATE OUT
VIN
UV
LTC4352
5.11k
GATE
SUPPLY
CURRENT
(5A/DIV)
CPO
OV
GND
GND
BACKPLANE
TIME (5ms/DIV)
Z1: DIODES INC. SMAJ12A
CONNECTORS
PLUG-IN CARD
b. After short pin makes contact and UV is above 0.5V,
GATE starts ramping up. Once it crosses the MOSFET
threshold voltage, LOAD follows with the same dV/dt.
Here, inrush is limited to 8.3A peak for a 10mF CLOAD.
a. Omitting the CPO capacitor and adding an RC network
on the gate allows inrush current control on a Hot Swap board.
Figure 7. Controlling inrush current
Only undervoltage, overvoltage, and
VCC undervoltage lockout can override
this to turn-off the gate. This feature
is handy either in power path control
applications which allow reverse
current low to occur, or for testing
purposes.
Inrush Control on a
Hot Swap Board
When the diode power input lows
across a connector on a hot swap
board, the LTC4352 can do doubleduty to control the inrush current.
Again, back-to-back MOSFETs are
required for this application to block
conduction through the MOSFET body
diodes. The inrush current is limited
by slowing the rise rate of the load
voltage. This is done by limiting dV/dt
on the MOSFET gate and operating it
in a source-follower coniguration.
Figure 7 illustrates an application where the LTC4352 is used for
inrush control. Since the goal is
to limit dV/dt on the gate, the fast
turn-on characteristic of the ideal
diode is disabled by omitting the CPO
reservoir capacitor. The gate current
is now limited to the CPO pull-up
current of 100µA. To further reduce
dV/dt, an RC network is added on
the gate. The resistor decouples the
capacitor during fast turn-off due to
reverse current or overvoltage faults.
Linear Technology Magazine • September 2008
Resistor RG prevents high frequency
oscillations in Q2.
When the board is hot-plugged, the
long power pins make contact irst.
The LTC4352 powers up, but holds
the gate off since UV is low. After a few
milliseconds of board insertion delay,
the short UV pin makes contact. If
VIN is above 10.8V, the MOSFET gate
starts ramping up. The MOSFET turns
on as the gate reaches the threshold
voltage, and current starts charging
the output. Q2 operates in the source
follower mode and suffers the most
power dissipation. Its VDS starts off
at VIN and decreases to 25mV/2.
Care should be taken that the power
dissipated during inrush falls within
the safe operating area (SOA) of the
MOSFET.
0V TO 18V
5V
C1
0.1µF
Down to Earth Operation
The VIN operating range extends all
the way down to 0V. However, when
operating with inputs below 2.9V,
an external supply is needed on the
VCC pin. This supply should be in the
range 2.9V to 6V. For a 2.9V to 4.7V
subset of this range, VIN should always
be lower than VCC. A 0.1µF bypass
capacitor is also needed between the
VCC and GND pins. Figure 8 shows an
ideal diode circuit, where a 5V supply
powers up the VCC pin. In this case,
VIN can operate all the way down to
0V and up to 18V.
For input supplies from 2.9V to 18V,
the external supply at the VCC pin is
not needed. Instead, an internal low
dropout regulator (LDO in Figure 3)
continued on page Q1
Si7336ADP
C2
0.1µF
CPO SOURCE VIN
VCC
UV
OUT
STATUS
LTC4352
OV
REV
GATE
TO LOAD
FAULT
GND
Figure 8. A 0V to 18V ideal diode circuit. By powering the VCC pin with an external
supply in the 4.7V to 6V range (here 5V), VIN can operate down to 0V and up to 18V.
27
L DESIGN FEATURES
Low Voltage, High Current Step-Down
µModule Regulators Put a (Nearly)
Complete Power Supply in a
15mm × 9mm × 2.8mm Package
by Judy Sun, Sam Young and Henry Zhang
Introduction
Endlessly increasing power density
requirements are a major driving force
behind the continuous need to ind
new power supply solutions. Switching
regulators are the top choice for high
current applications because of their
high eficiency and high performance,
but high power density doesn’t come
for free with a switcher. Components
must be carefully chosen and laid
out to maximize eficiency, transient
response and thermal performance.
Making a high density switching power
supply requires signiicant design and
test time, or does it?
The LTM4604 and LTM4608 LTC
µModule switching regulators make
it possible to create high density
designs with minimal effort. Both
are high density power supplies for
≤5.5V input voltage, high output current, step-down applications. Each
µModule regulator comes in a 15mm
× 9mm LGA surface mount package
and is nearly self-contained—only a
few passive components are required
to complete a power supply design. The
switching controller, MOSFETs, inductor and all support components are
Easy Design with
Few Components
VIN
3.3V
Figure 1 shows a typical 2.5V/4A
design with LTM4604 and Figure 2
shows the resulting eficiency. Ceramic
input capacitors are integrated into
the µModule package—additional
input capacitors are only required if
a load step is expected up to the full
4A level. Additional required output
capacitance is typically in the range
of 22µF to 100µF. A single resistor on
the FB pin sets the output voltage.
For applications needing more
output current, the LTM4608 its the
bill. Figure 3 shows a 1.8V/8A design
with LTM4608 and Figure 4 shows
its eficiency. As with the LTM4604,
the number of necessary external
components has been reduced to a
minimum, signiicantly simplifying
the design effort. Nevertheless, a very
fast transient response to the line and
load changes is guaranteed by the optimized design of the µModule’s high
switching frequency and current mode
control architecture. Furthermore, a
number of features can be enabled on
the LTM4604 and LTM460408 to suit
the needs of various applications.
10MF
6.3V
VIN
PGOOD
VOUT
2.5V
4A
VOUT
LTM4604
COMP
22MF
6.3V
r2
FB
RUN/SS TRACK
GND
VIN
2.37k
Figure 1. Only a few components are required
for a 2.5V/4A design with LTM4604.
already carefully chosen and laid out
in the package. Low proile packages
(2.3mm and 2.8mm, respectively) allow
them to be easily mounted in unused
space on the bottom of PC boards and
simplify thermal management.
The LTM4604 features a 2.375V
to 5.5V input range and a 0.8V to
5V output range, while the LTM4608
takes a 2.7V to 5.5V input to a 0.6V to
5V output. The LTM4604 can deliver
up to 4A continuous current with up
to 95% eficiency. The slightly higher
proile of the LTM4608 allows it to
deliver up to 8A continuous current
thanks to its high eficiency design and
low thermal impedance package.
100
95
EFFICIENCY (%)
VIN = 3.3V
VIN
3V TO 5.5V
10μF
90
85
VIN
75
70
VOUT
SVIN
FB
SW
ITH
RUN
80
CLKIN
LTM4608
100μF
VOUT
1.8V
8A
4.87k
ITHM
PLLLPF
PGOOD
TRACK
MGN
CLKOUT GND SGND
VOUT = 2.5V
0
1.0
2.0
3.0
LOAD CURRENT (A)
4.0
Figure 3. Only a few components are required for a 1.8V/8A design with the LTM4608.
Figure 2. High efficiency is achieved with the
LTM4604 in the application of Figure 1
28
Linear Technology Magazine • September 2008
DESIGN FEATURES L
100
VIN
3V TO 5.5V
95
EFFICIENCY (%)
VIN = 3.3V
10µF
90
VOUT
LTM4608
RUN
80
75
VOUT = 1.8V
0
2
4
6
LOAD CURRENT (A)
100µF
6.3V
X5R
100pF
SVIN
SW
VIN = 5V
85
70
CLKIN
VIN
FB
ITH
PLLLPF
ITHM
TRACK
PGOOD
MODE
BSEL
PHMODE
MGN
VOUT
1.5V
16A
3.32k
CLKOUT GND SGND
10
8
Figure 4. High efficiency is achieved with the
LTM4608 in the application of Figure 3.
Wealth of Features
10µF
Both LTM4604 and LTM4608 feature RUN pin control, output voltage
tracking selections and power good
indicators. For systems requiring
voltage sequencing between different
power supplies, the sequencing function can be implemented by controlling
the RUN pins and the PGOOD signals
with a few additional components.
Fault protection features include
overvoltage protection, over current
protection and thermal shutdown.
The LTM4608 offers some additional features. Burst Mode® operation,
pulse-skipping mode or continuous
current mode can be selected to improve light load eficiency. Burst Mode
operation provides the highest eficiency at very light load, while forced
continuous current mode leads to the
lowest output ripple. Pulse-skipping
mode offers a compromise between
Burst Mode operation and continuous
mode, offering good light load eficiency
while keeping output voltage ripple
CLKIN
VIN
VOUT
SVIN
SW
LTM4608
RUN
FB
100µF
6.3V
X5R
ITH
PLLLPF
ITHM
TRACK
PGOOD
MODE
BSEL
PHMODE
MGN
CLKOUT GND SGND
Figure 5. Two LTM4608s are easily paralleled to provide
1.5V/16A output with interleaved switching operation.
down. Programmable output voltage
margining is supported for ±5%, ±10%
and ±15% levels. The LTM4608 also
allows frequency synchronization and
spread spectrum operation to further
reduce switching noise harmonics.
Parallel for More Power
With cycle-by-cycle current mode
control, the LTM4604 and LTM4608
can be easily paralleled to provide
more output power with excellent current sharing. The LTM4608 includes
CLKIN and CLKOUT pins to make it
possible to operate paralleled devices
out of phase of one another to reduce
input and output ripple. A total of 12
phases can be cascaded to run simultaneously with respect to each other
by programming the PHMODE pin of
each LTM4608 to different levels.
Figure 5 shows an example of two
LTM4608s in parallel to provide 16A
output current. Figure 6 shows the
measured current sharing performance of the circuit, illustrating that
the DC current sharing error is less
continued on page OUTPUT CURRENT OF EACH LTM4608 (A)
9
8
IOUT2
7
6
IOUT1
5
4
3
2
1
0
0
2
4
6
8 10 12 14
TOTAL LOAD CURRENT (A)
16
18
Figure 6. Bench test shows excellent current
sharing between two paralleled LTM4608s over
the entire load range.
Linear Technology Magazine • September 2008
Figure 7. Good thermal balance is maintained between two
paralleled LTM4608 boards supplying 16A output current.
29
L DESIGN IDEAS
5-Output High Current Power
Supply for TFT-LCDs in a Low Profile
QFN Features Space-Saving 2MHz
Switching Regulators
by Kevin Huang
Introduction
The LT3513 is a highly integrated,
5-output regulator designed to provide all the supply voltages typically
required by large TFT liquid crystal
displays (LCDs) with a single IC. The
part features a step-down switching
regulator to produce a 3.3V or 5V
logic voltage from a wide voltage range
input, such as automotive battery.
A lower voltage logic supply can be
generated from the irst supply by
adding an external NPN driven by an
internal linear regulator. The other
three on-chip regulators provide the
three bias voltages required by LCDs: a
high power boost regulator to generate
AVDD, a low power boost regulator to
generate VON and an inverting regulator to provide VOFF.
Switching regulators are chosen
over linear regulators to accommodate
a wide input voltage range (providing
both step up and step down functions)
and to minimize power dissipation.
The LT3513’s wide input range, 4.5V
to 30V, allows it to accept a variety of
power sources, including automotive
batteries, distributed supplies and
wall transformers. The low proile
38-pin QFN package has an exposed
metal pad on the backside to maximize
thermal performance.
5-Output High Current Power Supply
for TFT-LCDs in a Low Profile QFN
Features Space-Saving 2MHz
Switching Regulators ........................30
Kevin Huang
32VIN Synchronous Buck Regulators
with Integrated FETs Deliver up to 12A
from Sub-1mm Height Packages ........32
Stephanie Dai and Theo Phillips
Complete Power Solution for Digital
Cameras and Other Complex Compact
Portable Applications ........................34
Brian Shaffer
VIN
8V TO 30V
VOFF
–10V
20mA
2.2µF
L1
10µH
VLOGIC
5V
10µF 178k
DFLS240L
UVLO LDOPWR VIN
L2
10µH
10k
10µF
SW2
SW4
69.8k
PGOOD
D4
RUN-SS1
BIAS
10k
L3
4.7µH
RUN-SS3/4
0.22µF
CT
SW1
SENSE+
SENSE–
FB1
22µF
10k
RUN-SS2
BOOST
30.1k
AVDD
8V
80mA
FB2
NFB4
47nF
LT3513
VON_CLK
E3
BD
FB5
VC1
42.2k
L4
6.8µH
10µF
7.5k
10k
SW3
FB3
VC2
4.7k
VC3
30k
4.7nF 1.5nF
2.7nF
L1: SUMIDA CDR6D28MNNP-100NC
L3: SUMIDA CDR6D28MNNP-4R5NC
GND
15nF
VC4
15nF
VON
22V
20mA
232k
VONSINK
DFLS240L
15nF
VON_CLK
VON
ZXTAM322
VLDO
3.3V
0.5A
100k
Si2333DS
60.4k
VLOGIC
5V
0.5A
53.6k
0.47µF
ZHCS400
165k
VLOGIC
5V
10k
0.47µF
2.2µF
13k
2.2nF
L2: COILCRAFT LPO3310-103MLC
L4: COILCRAFT LPO3310-682MLC
Figure 1. A complete 5-output 2MHz TFT-LCD power supply
Operation
All of the regulators are synchronized to a 2MHz internal clock,
allowing the use of small, low cost
inductors and ceramic capacitors.
Since different types of panels may require different bias voltages, all output
voltages are adjustable for maximum
lexibility. Programmable soft-start
capability is included in each of the
regulators to limit inrush current.
Figure 1 shows a 5-output TFT LCD
power supply that can accommodate
an 8V to 30V input voltage. The irst
switching converter produces a 5V
logic supply using a buck regulator.
The internal linear regulator with an
external NPN produces a 3.3V logic
supply using the 5V supply as input.
The second switcher is used to boost
the 5V supply to an 8V, 80mA AVDD
RUN/SS 2V/DIV
VLOGIC 5V/DIV
AVDD 10V/DIV
VOFF 10V/DIV
VE3 20V/DIV
VON 20V/DIV
IIN(AVG) 1A/DIV
5ms/DIV
Figure 2. Startup waveforms of the power supply in Figure 1
30
Linear Technology Magazine • September 2008
DESIGN IDEAS L
bias supply. Another boost converter
and an inverter generate VON and
VOFF, which also use the 5V supply
as input.
When power is irst applied to the
input, the RUN-SS1 capacitor starts
charging. When its voltage reaches
0.8V, Switcher 1 is enabled. The capacitor at the RUN-SS1 pin controls
the ramp rate for the Switcher 1 output, VLOGIC and inrush current in L1.
Switchers 2, 3 and 4 are controlled
by the BIAS pin, which is usually
connected to VLOGIC. When the BIAS
pin is higher than 2.8V, the capacitors
at the RUNSS-2 and RUN-SS3/4 pin
begin charging to enable Switchers 2,
3 and 4. When AVDD reaches 90% of
its programmed voltage, the PGOOD
pin is pulled low. When AVDD, VOFF and
E3 all reach 90% or their programmed
voltages, the CT timer is enabled and a
20µA current source begins to charge
CT. When the CT pin reaches 1.1V, the
output PNP turns on, connecting E3
to VON. Figure 2 shows the start up
sequence of the circuit in Figure 1.
If one of the regulated voltages,
VLOGIC, AVDD, VOFF or E3 dips more
than 10%, the internal PNP turns off
to shut down VON. This action protects
the panels, as VON must be present to
turn on the TFT display. The PGOOD
LTM404, LTM408, continued from page 29
than 5% at full load. Excellent current sharing results in well balanced
thermal stresses on the paralleled
LTM4608s, which in turn makes
for a more reliable system. Figure 7
demonstrates the small temperature
difference between these two paralLTC452, continued from page 27
generates a 4.1V supply at the VCC
pin. For VIN below 4.1V, VCC follows
approximately 50mV below VIN. The
0.1µF VCC capacitor is still needed for
bypassing and LDO stability.
Conclusion
An ever-present theme in electronic
system design has been to pack more
computation in smaller form factors
and tighter power budgets. Another
Linear Technology Magazine • September 2008
pin can drive an optional PMOS device
at the output of the boost regulator to
disconnect the load at AVDD from the
input during shutdown. The converter
uses all ceramic capacitors. X5R and
X7R types are recommended, as these
materials maintain capacitance over
a wide temperature range.
All four switchers employ a constant frequency, current mode control
scheme. Switching regulator 1 uses a
feedback scheme that senses inductor current, while the other switching
regulators monitor switch current.
The inductor current sensing method
avoids minimum on-time issues and
maintains the switch current limit at
any input-to-output voltage ratio. The
other three regulators have frequency
foldback scheme, which reduces the
switching frequency when its FB pin
is below 0.75V. This feature reduces
the average inductor current during
start up and overload conditions,
minimizing the power dissipation
in the power switches and external
components.
Layout Considerations
Proper PC board layout is important
to achieve the best operating performance. Paths that carry high switching
current should be short and wide to
leled LTM4608 boards supplying 16A
output current.
minimize parasitic inductance. In a
buck regulator, this loop includes
the input capacitor, internal power
switch and Schottky diode. In a boost
regulator, this loop includes the output capacitor, internal power switch
and Schottky diode. Keep all the loop
compensation components and feedback resistors away from the high
switching current paths. The LT3513
pin out was designed to facilitate PCB
layout. Keep the traces from the center
of the feedback resistors to the corresponding FB pins as short as possible.
LT3513 has an exposed ground pad
on the backside of the IC to reduce
thermal resistance. A ground plane
with multiple vias into ground layers
should be placed underneath the part
to conduct heat away from the IC.
Conclusion
The LT3513 is a comprehensive, but
compact, power supply solution for
TFT-LCD panels. Its wide input range
and low power dissipation allow it
to be used in a wide variety of applications. All four of the integrated
switching regulators have a 2MHz
switching frequency and allow the
exclusive use of the ceramic capacitors to minimize circuit size, cost and
output ripple. L
The LTM4604 and LTM4608 15mm
× 9mm µModule regulators are complete power supply solutions for low
input voltage and high output cur-
rent applications. They signiicantly
simplify circuit and layout designs
by effortlessly itting into the tightest
spaces, including the bottom of the
PCB. Despite their compact form,
these µModules are rich in features,
and they can be easily paralleled when
more output current is needed. L
trend has been to lower the voltage of
distributed power, which increases the
current to maintain power levels. Given
these constraints, board designers
must scrutinize each diode in a high
current power path for its power and
area consumption.
The LTC4352 MOSFET controller
provides the same functionality as a
diode but at higher eficiencies and
cooler temperatures, especially as
currents increase. It also incorporates
useful features such as fast switch
control, 0V operation, undervoltage
and overvoltage protection, open
MOSFET detection, ability to allow
reverse current, Hot Swap capability, and fault and status outputs. All
of this functionality comes wrapped
in space-saving 12-pin DFN (3mm ×
3mm) and MSOP packages, making
it possible to produce an ideal diode
solution in a smaller footprint than
conventional diodes. L
Conclusion
31
L DESIGN IDEAS
32VIN Synchronous Buck Regulators
with Integrated FETs Deliver up to 12A
from Sub-1mm Height Packages
by Stephanie Dai and Theo Phillips
Introduction
Monolithic buck regulators are easy
to hook up and they make it possible
to squeeze an entire DC/DC converter
into very tight spaces. Although monolithics are an easy it, they aren’t the
perfect it for every application. For
instance, they typically lack the capability to eficiently convert high input
voltages (>12V) to low voltages at high
output currents (>4A), thus leaving the
job to a traditional controller IC and
external MOSFETs.
A new family of devices, though,
offers the advantages of monolithics
with the low duty cycle and high eficiency of discrete components. The
LTC3608, LTC3609, LTC3610 and
LTC3611 are synchronous buck converters that bring high power density
and simpliied design to point-of-load
applications. With a maximum input of
A new family of devices
offers the advantages of
monolithic DC/DC converters
with the low duty cycle and
high efficiency of discrete
components. The LTC3608,
LTC3609, LTC3610 and
LTC3611 are synchronous
buck converters that bring
high power density and
simplified design to point-ofload applications.
32V they utilize current-mode control
up to a 2MHz switching frequency,
deliver up to 12A of load current, and
L1
0.5µH
LTC3608
VIN
5V TO 18V
PVIN
RON
187k
1%
CIN
10µF
25V
3s
SW
+
C1
0.22µF
ION
COUT
100µF
s2
BOOST
VOUT
1.2V
8A
CMDSH-3
RUN/SS
CSS
0.1µF
INTVCC
4.7µF
6.3V
100k
PGOOD
30.1k
1%
EXTVCC
0.01µF
VFB
30.1k
1%
1Ω
SVIN
VIN
0.47µF
25V
VON
FCB
VRNG
ITH
100pF
SGND
C1: TAIYO YUDEN JMK316BJ226ML-T
CIN: TAIYO YUDEN TMK432BJ106MM
COUT: TDK C4532X5R107M
L1: SUMIDA CDEP85NP-R50MC-125
0.1µF
7.68k
1%
1500pF
PGND
KEEP POWER GROUND AND SIGNAL GROUND SEPARATE.
CONNECT AT ONE POINT.
= PGND
= SGND
Figure 1. Typical application of the LTC3608
32
are packaged in thermally enhanced
packages less than 1mm in height. A
typical application of the LTC3608 is
shown in Figure 1.
Features
The LTC3608, LTC3609, LTC3610 and
LTC3611 integrate high performance
synchronous buck controllers with
super-low RDS(ON) DMOS MOSFETs
to produce compact high eficiency
converters (Figure 2). Two package
sizes are available, each having a
high voltage or high current option
(Table 1). Each device features a sub100ns on-time, allowing very low duty
cycle operation and high switching
frequency. The current-mode control
architecture of these parts simpliies
tuning of loop stability and allows
excellent transient response with a
variety of output capacitor types, including all-ceramic output capacitor
applications.
The LTC3610 can operate in forced
continuous mode, which provides the
lowest possible output ripple and EMI,
or discontinuous mode, which has
better light load eficiency because
inductor current is not allowed to
reverse.
Current into the ION pin sets the
on-time—a resistor RON from VIN to the
ION pin reduces on-time as VIN rises,
thus limiting changes in switching
frequency. Furthermore, response to a
load step can be very fast since the loop
does not have to wait for an oscillator
pulse before the top switch is turned
on and current begins increasing.
The current limit, which is inferred
from the maximum allowable sense
voltage across the on-resistance of
the bottom FET, can be adjusted by
applying a voltage to the VRNG pin.
Maximum load current limits for each
Linear Technology Magazine • September 2008
DESIGN IDEAS L
90
VIN = 12V
85 FREQ = 550kHz
VOUT
200mV/DIV
75
70
DCM
CCM
65
60
200mV
55
LOAD STEP 1A-8A
VIN = 12V
VOUT = 1.2V
FCB = 0V
Conclusion
The LTC3608, LTC3609, LTC3610
and LTC3611 buck regulators offer the eficiency and power output
capability of separate (controller +
discrete) MOSFET solutions with
the ease-of-use and space-saving
advantages of traditional MOSFETon-the-die monolithics. These parts
also yield higher eficiencies than
80
IL
5A/DIV
EFFICIENCY (%)
part are shown in Table 1. Soft-start
and latch off functions are controlled
by the RUN/SS pin, preventing inrush
current and current overshoot during
startup, and providing the option of
latch-off if an under voltage or short
circuit is presented. An open drain
power-good pin monitors the output
and pulls low if the output voltage is
±10% from the regulation point.
50
0.01
Figure 3. Transient response for the typical
LTC3608 application represented in Figure 1
with a load step of 1A to 8A
0.1
1
LOAD CURRENT (A)
10
Figure 2. Efficiencies for a typical
LTC3608 application in discontinuous
conduction mode (DCM) and continuous
conduction mode (CCM)
traditional monolithic solutions.
They conserve power, save space, and
simplify power designs. They reduce
discrete components over controller-based solutions, making them a
good it in everything from low power
portable device applications such as
notebook and palmtop computers
to high-power industrial distributed
power systems. L
Table 1. Integrated MOSFET buck regulators
LTC3610
LTC3611
LTC3608
LTC3609
PVIN Max
24V
32V
18V
32V
ILOAD Max
12A
10A
8A
6A
Package
9mm × 9mm × 0.9mm
64-pin
9mm × 9mm × 0.9mm
64-pin
7mm × 8mm × 0.9mm
52-pin
7mm × 8mm × 0.9mm
52-pin
RDS(ON)
Top FET
12mΩ
15mΩ
14mΩ
19mΩ
RDS(ON)
Bottom FET
6.5mΩ
9mΩ
8mΩ
12mΩ
LTC4009, continued from page 20
LTC4009 family monitors the voltage
across the input blocking diode for
unexpected voltage reversal. Initial
startup, restarts from fault conditions,
and charge current reduction during
input current limit are also carefully
controlled to avoid producing reverse
current.
All members of the family provide
an input current limit lag to tell the
system when the adapter is running
at over 95% of its current capacity.
Finally, each IC features internal
over-temperature protection to prevent silicon damage during elevated
thermal operation.
Recovery from all fault conditions is
under full control of the analog feedLinear Technology Magazine • September 2008
back loops, which guarantees charging
remains suspended until the internal
feedback loops respond coherently and
report the need to supply current to
the load to maintain proper voltage or
current regulation.
Conclusion
The LTC4009 family integrates a full
set of charger building blocks in a small
PCB footprint. The result is a high
power battery charger IC with high
precision and a full set of monitoring
and fault handling features.
The LTC4009 provides adjustable
output voltage control with a simple,
external, user-programmed resistive
voltage divider. As such, it is suitable as
a general purpose charger that works
with multiple battery chemistries and
supercaps. It offers direct control over
the entire charge process, facilitating
implementation of a wide range of
charge termination algorithms with
an external microprocessor.
The LTC4009-1 and LTC4009-2
feature pin-programmable output
voltage for common lithium-ion or
lithium-polymer battery pack conigurations with one to four series cells.
For these chemistries, the number of
precision external application components is reduced without sacriicing
accuracy. Both 4.1V/cell (LTC4009-1)
and 4.2V/cell (LTC4009-2) options are
available, allowing the user to balance
capacity and safety per the demands
of the application. L
33
L DESIGN IDEAS
Complete Power Solution for Digital
Cameras and Other Complex
Compact Portable Applications
by Brian Shaffer
Introduction
Digital cameras, portable GPS systems,
MP3 players and other feature-rich
mobile devices have complicated
power requirements. In these complex
devices, the low of power must be
carefully managed between a number of specialized sources and loads,
including charging/discharging the
battery, current-limited USB power
and a set of multivoltage power supply rails, including negative rails for
CCDs or LCDs. The supply rails must
be sequenced and tracked and faults
must be handled cleanly and communicated to a microcontroller.
When these requirements are
added together, the task of squeezing
an eficient and robust power system
into a handheld device can seem near
impossible. Linear Technology solves
this problem with a family of devices
called PMICs (Power Management Integrated Circuits) that greatly simplify
the design of complex rechargeable
battery power systems.
Some Linear Technology PMICs
use a switching PowerPath controller
topology with the unique Bat-Track
feature, which allows charge currents
above the USB limit (see Figure 1) for
faster battery charging. The power
solution for digital cameras presented
700
CHARGE CURRENT (mA)
600
500
400
VBUS = 5V
RPROG = 1k
RCLPROG = 3k
300
200
100
5x USB SETTING,
BATTERY CHARGER SET FOR 1A
0
3.0
3.3
3.6
3.9
2.7
BATTERY VOLTAGE (V)
4.2
Figure 1. Battery charge current
vs battery voltage
34
The LTC3586 implements
Linear Technology’s unique
Bat-Track™ technology,
which can use more power
from a USB source than
traditional linear chargers,
resulting in faster charging.
here takes advantage of this and other
powerful PMIC features.
Complete Digital Camera
Power System
Figure 2 shows a complete digital
camera power solution using the
LTC3586 PMIC as the power trafic
control center. Its 4mm × 6mm QFN
package includes a USB PowerPath
manager, a battery charger, plus a
boost DC/DC converter, a buck-boost
and two buck converters. The LT3587
in a 3mm × 3mm package is used to
drive a CCD and an LED backlight
for an LCD screen with a high voltage
monolithic inverter and dual boost
converter.
Switching PowerPath
Controller Maximizes
Available Power
The LTC3586 implements Linear
Technology’s unique Bat-Track™
technology, which maximizes the use
of available power from a USB source
for either providing current to the
load or charging the battery at rates
greater than achievable from linear
chargers.
The switching PowerPath controller maintains accurate control of the
average input current for USB applications. The average level of input
current is controlled by the state
of two digital inputs and can be set
to 100mA, 500mA, 1A or suspend
(500µA). The switching PowerPath
controller is highly eficient, which
results in battery charge currents of
well over 600mA from a 500mA USB
source (Figure 1).
The battery charging eficiency is
between 85% and 90% for the entire
battery voltage range. In contrast,
the eficiency of a traditional linear
charger falls as low as 57%, generating
the losses as heat. See Figure 3 for a
graph of the battery charger eficiency
as a function of battery voltage.
Instant-On Operation
The LTC3586 also features instant-on
operation, which allows the camera to
function immediately when external
power is applied even if the battery
voltage is below the system cutoff
voltage. This is achieved by generating
a separate voltage rail, VOUT, which
is decoupled from the battery voltage when the battery is below 3.3V.
When external power is applied, the
PowerPath controller prioritizes load
current over battery charge current
and regulates VOUT to 3.6V, enabling
the system to operate immediately
upon the application of external power.
The instant-on feature is important in
camera applications because important moments do not wait for batteries
to charge.
Fault Handling
The FAULT signals on both of these
devices are designed to work together
for seamless fault handling. By making
the fault signals both an input and
an output, the two chips can communicate fault events to each other.
If either of the devices has a fault then
all the outputs turn off, protecting
the system and battery from damage.
The enable lines and the fault signal
Linear Technology Magazine • September 2008
DESIGN IDEAS L
USB/WALL
4.5V TO 5.5V
L1
3.3µH
C1
22µF
SW
VBUS
VOUT
VOUT
100k
GATE
NTC
T
2k
+
CLPROG
0.1µF
Li-Ion
RED
GND
2.94k
510Ω
C2
22µF
MP1
BAT
PROG
CHRG
3.3V
1A
VOUT3
15k
3.3V, 20mA
324k
22µF
2.2µF
105k
L2
2.2µH
FAULT
2
33pF
330pF
FB3
SWCD3
10k
SYSTEM RAIL/
I/O
121k
VC3
LDO3V3
1µF
PUSHBUTTON
MICROCONTROLLER
10pF
SWAB3
ILIM
VIN3
L3
4.7µH
LTC3586
1.8V
400mA
SW2
MODE
1.02M
I/O/MEMORY
10pF
FB2
4
10µF 1µF
806k
EN
MICROPROCESSOR
VIN2
L4
4.7µH
1.6V
400mA
SW1
CORE
806k
10pF
FB1
806k
10µF 1µF
C1, C2: TDK C2012X5R0J226M
L1: COILCRAFT LPS4018-332LM
L2, L5: TOKO 1098AS-2R2M
L3, L4: TOKO 1098AS-4R7M
MP1: SILICONIX Si2333
VIN1
VIN4
L5
2.2µH
SW4
10µF
5V
800mA
VOUT4
88.7k
AUDIO/
MOTOR DRIVE
10pF
FB4
16.9k
10µH
22µF
15µH
1µF
2.2µF
8.06k
1µF
SW3
CAP3
VIN
SW1
CAP1
IFB3
VOUT1
GND
FB1
10µF
15V
50mA
1M
2.7pF
LT3587
BACKLIGHT LEDs
FLT
LED DRIVER
20mA, UP TO 6 LEDS
VOUT3
VFB3
EN/SS1
EN/SS3
CCD POSITIVE
CCD
1M
100nF
FB2
SW2
100nF
6.8pF
2.2µF
15µH
= IR05H40CSPTR
15µH
22µF
–8V
100mA
CCD NEGATIVE
Figure 2. Complete power solution for portable cameras
Linear Technology Magazine • September 2008
35
L DESIGN IDEAS
100
EFFICIENCY (%)
90
VVIN
2.5V TO 5V
10µH
RCLPROG = 3.01K
RPROG = 1K
IVOUT = 0mA
IR05H40CSPTR
1x CHARGING EFFICIENCY
VIN
80
1µF
SW3
CAP3
5x CHARGING EFFICIENCY
LT3587
VOUT3
IFB3
70
60
2.7
3
3.5
3.9
3.3
BATTERY VOLTAGE (V)
4.2
Figure 3. Battery charging efficiency vs
battery voltage with no external load
(PBAT/PBUS)
should be pulled-up to the same voltage. In Figure 2 the LDO3V3 regulator
is used as the pull-up voltage for the
FAULT signal and the power supply
for the low power microcontroller
used to process pushbutton events
and sequence the power supplies.
The FAULT pin also acts as an input
and hence, must be high before any
outputs are enabled.
DAC
LTC2630
VDAC-OUT
EN/SS3
8.06k
RIFB3
MN1
Si1304BDL
PWM 2.5V
FREQ 0V
Figure 4. Six white LED driver with PWM and analog dimming
the current regulation loop increases
voltage in an attempt to regulate the
current.
The integrated LED driver in the
LT3587 is capable of accepting a direct
PWM dimming signal into its enable
input (EN/SS3) and/or accommodates
analog dimming via an external DAC.
See Figure 4 for a partial application
circuit showing the LED driver with
direct PWM and analog dimming.
LEDs can change color when the
current through them changes, but
PWM dimming maintains color consistency over the dimming range, as the
ON part of the PWM cycle is always the
same current. In PWM dimming, the
brightness of the LEDs is a function of
average current, adjusted by changing
the duty cycle of the PWM signal. In
analog dimming, the constant current
through the LEDs is adjusted, which
causes variations in color.
The LT3587 accepts PWM signals
with frequencies over 60Hz to assure
licker-free operation. High PWM frequencies are achievable because of
the internal disconnect FET between
CAP3 and VOUT3. This FET ensures
that CAP3 maintains its steady-state
value while the PWM signal is low,
resulting in minimal startup delays.
For a 100Hz PWM dimming signal and
allowing for 10% deviation from linearity at the lowest duty cycle, the LT3587
allows for a dimming ratio of 30:1. If
the maximum amount of adjustment
range is desired, an external DAC,
such as the LTC2630, can be used to
feed an adjustment voltage onto the
IFB3 resistor, creating an LED current
range of 20,000:1.
Conclusion
protects against insuficient turn-on
voltage for the top MOSFET.
eficiency vs load for all three modes
of operation with an input voltage of
12V.
3.3V/15A Regulator with
DCR Sensing
1.5V/15A Regulator
Synchronized at 350kHz
Figure 2 shows a 400kHz, 3.3V output
regulator using DCR current sensing.
The DC resistance of the inductor is
used as the current sense element,
eliminating the need for a discrete
sense resistor and thus maximizing
eficiency. Figure 3 shows a plot of the
Figure 4 illustrates a 1.5V output
regulator that is synchronized to
an external clock. The loop ilter
components connected to the FREQ/
PLLFLTR pin are optimized to achieve
a jitter free oscillator frequency and
reduced lock time.
Compact LED Driver
The LT3587 LED driver is designed
to drive up to six LEDs with average
LED currents between 20mA and 1µA.
When the LT3587’s VOUT3 is used as
a current regulated LED driver, the
VFB3 pin can be used as an overvoltage
protection function. By connecting a
resistor between VOUT and VFB3 the
device limits the maximum allowable
output voltage on VOUT3. This feature
is extremely important in LED applications because without it the client
device may be damaged if one of the
LEDs were to open; in such a case,
the output would continue to rise as
LTC85, continued from page 7
36
Conclusion
Two highly integrated devices, the
LTC3586 and LT3587 can be combined
to create a complete USB compatible
power solution for portable cameras
and other feature-rich portable devices. The solution is robust, high
performance and compact, with eficient battery charging, instant-on
capability and LED protection. L
The LTC3851 combines high performance, ease of use and a comprehensive
feature set in a 3mm × 3mm 16-pin
package. DCR current sensing and
Burst Mode® operation keep eficiency
high. With a broad 4V to 38V input
range, strong MOSFET drivers, low
minimum on-time and tracking, the
LTC3851 is ideal for automotive electronics, server farms, datacom and
telecom power supply systems and
industrial equipment. L
Linear Technology Magazine • September 2008
NEW DEVICE CAMEOS L
New Device Cameos
I2C Buffer Level Shifts 2.3V–
5.5V Busses to Low Voltage
Busses as Low as 1V
The LTC4308 is a low voltage, level
shifting hot swappable 2-wire bus
buffer device with output rise time
acceleration and stuck bus recovery.
The LTC4308’s negative offset from
output to input allows communication
between output bus devices with high
VOL and devices on the low voltage
input side, where bus supplies can
be as low as 0.9V. The transparent
level shifting provides reliable communications between new low voltage
supply devices, such as EEPROMs and
microcontrollers, with legacy devices
operating at supply voltages ranging from 2.3V to 5.5V. Bus buffering
provides capacitive isolation between
the upstream and downstream busses, allowing a single large bus to
be broken up into two manageable
smaller busses.
The LTC4308 also provides rise
time acceleration on the output side
and stuck bus recovery. The strong
rise time accelerator pull up currents
reduce long rise times associated
with high bus capacitances, allowing
weaker pull-up resistors to be used,
thereby reducing DC power consumption. Stuck bus recovery automatically
disconnects the input and output
busses when the connected busses
remain low for greater than 30ms.
The LTC4308 attempts to free the
stuck bus by generating up to 16 clock
pulses and a Stop Bit on the output
bus. If the stuck bus recovers to a
logic high the busses are automatically
reconnected.
The LTC4308 is also ideal for output
side data and clock hot-swapping.
During insertion the LTC4308 precharges the SDAOUT and SCLOUT
pins to 1V, minimizing the voltage
differential between its pins and the
live bus. Once inserted, the LTC4308
waits for a Stop Bit or Bus Idle to
occur on both sides, to ensure data
transfers are complete and coherent,
before a connection is made. High
Linear Technology Magazine • September 2008
6kV HBM ESD performance provides
added protection from stresses during
assembly, handling, and insertion. The
combination of low voltage level shifting and Hot Swap features allows low
voltage supply I/O cards to interface
with legacy backplanes.
Available in a small 8-pin DFN
(3mm × 3mm) and MS8 package, the
LTC4308 is the ideal solution for low
voltage level shifting in 2-wire bus
systems.
Low Voltage Hot Swap
Controller with Adjustable
Current Limit
Linear Technology Corporation introduces the LTC4218 Hot Swap
Controller for protecting boards with
load supply voltages ranging from 2.9V
to 26.5V. When a board is plugged into
a backplane, large inrush currents
can create a glitch on the load supply
causing other boards on the bus to
malfunction. The LTC4218 enables
safe board insertion and removal from
a live backplane, using an external Nchannel pass MOSFET in the power
path to limit the inrush current during
power up. An adjustable current limit
allows users to vary the current limit
threshold under various loading conditions, such as disk drive spin-up to
normal operation. The wide operating
voltage and adjustable current limit
differentiate the LTC4218 from other
low voltage Hot Swap controllers on
the market today.
The LTC4218 features are tailored
for use in RAID, server, telecom (i.e.,
ATCA, AMC, µTCA) and industrial
applications. The load current is
monitored using the voltage sensed
across a current sense resister
and adjusting the MOSFET’s gateto-source voltage accordingly. A
separate ISET pin allows users to
adjust the 5% accurate (15mV) current limit threshold during startup
and normal operation as needed.
Meanwhile, current foldback and
power-good circuitry ensure that the
switch is protected from excessive
load current and indicate whether a
power-good condition is maintained.
The LTC4218 also features current
monitor and fault outputs, 2% accurate overvoltage and undervoltage
protection, and an adjustable current limit timer. A dedicated 12V
version (LTC4218-12) is also available, which contains preset 12V
speciic thresholds.
The LTC4218 is available in a
16-lead SSOP, while the LTC421812 is available in a 16-lead 5mm ×
3mm DFN, both of which are RoHS
compliant.
The small size, high integration,
low quiescent curent draw, and low
external component requirements of
the LTC3670 make it ideal for driving
the myriad low voltage rails in Li-ion
powered handheld devices.
Micropower 50mA Linear
Regulator Withstands 80V
Input and Offers PowerGood Status Signaling with
Programmable Delay
Linear Technology Corporation announces the LT3011, a high voltage
micropower, low dropout regulator
that delivers up to 50mA of continuous output current with a low dropout
voltage of only 300mV at full load. The
LT3011 features an input voltage range
of 3V to 80V, delivering output voltages as low as 1.24V and up to 60V.
The device’s power-good lag indicates
output regulation. However, a single
capacitor may be used to program the
delay between this regulated output
level and the lag indication. The
80V input voltage capability makes
it ideal for automotive applications,
48V telecom backup supplies and
industrial control applications. Low
quiescent current of 46µA (operating)
and 1µA (in shutdown) make it an
excellent choice for battery-powered
“keep alive” systems that require optimum run time.
Output noise is minimized at only
100µVRMS over a 10Hz to 100kHz bandwidth, making the LT3011 ideal for
37
L NEW DEVICE CAMEOS
noise-sensitive applications. For high
voltage applications that require large
input-to-output voltage differentials,
the LT3011 provides a very compact
solution. Its thermally enhanced
MSOP and DFN packages offer thermal
resistance equivalent to much larger
conventional packages.
The LT3011 is able to operate with
very small, low cost, ceramic output
capacitors and is stable with a 1µF
output capacitor—far better than the
10µF to 100µF required by most other
linear regulators. These tiny external
capacitors can be used without the
necessary addition of series resistance (ESR) as is common with many
other regulators. Internal protection
circuitry includes reverse-battery
protection, current limiting, thermal
limiting, and no reverse current low
from output to input.
Ideal Diode Controller with
Integrated 5A MOSFET
Replaces Lossy Schottky
Diodes
The LTC4358 is a high voltage ideal
diode controller with an internal 5A
MOSFET. The controller and 20mΩ
internal N-channel MOSFET perform
the function of a low forward voltage diode, making it a simple, low
loss replacement to Schottky diodes
in high current applications. This
provides a lower loss path compared
to the Schottky diode that in high
current applications provides higher
efficiency and preserves precious
board area by eliminating the need
for heat sinking.
The LTC4358 regulates the forward voltage drop across the internal
MOSFET to ensure smooth switchover
from one path to another without
oscillation. A fast pull-down circuit
minimizes reverse current transients
in the event a power supply fails or is
shorted. The LTC4358 can be viewed as
a 3-terminal diode for general purpose
applications such as reverse battery
protection in automotive applications,
or ORing power supplies together in
applications that demand high system
reliability.
The LTC4358 single ideal diode
controller is useful in applications
38
where multiple, redundant power
supplies are paralleled to provide load
sharing. In N+1 redundant systems,
the LTC4358 provides a convenient
method to OR together an additional
supply to safeguard the system in the
event one of the N supplies fail. This
ORing technique provides necessary
isolation for live insertion and removal
of converters onto the power bus and to
provide isolation from the bus during
a hard short. If the power source fails
or is shorted, the LTC4358 ensures a
fast 500ns turn-off to minimize reverse
current transients.
The LTC4358 joins a growing family
of ideal diode-OR controllers, including the LTC4355 positive voltage ideal
diode-OR, LTC4354 negative voltage
ideal diode-OR, and the LTC4357
and LT4352 single ideal diode controllers.
The LTC4358 is offered in 4mm ×
3mm 14-pin DFN and 16-lead TSSOP
packages.
70µA IQ Triple Power Supply
in 3mm × 2mm DFN
The LTC3670 is a triple power supply
in a single IC, integrating a 400mA
synchronous buck regulator with two
150mA low dropout linear regulators
(LDOs) in a 0.75mm proile, 3mm ×
2mm DFN. The input supply range of
2.5V to 5.5V is especially well-suited
for single-cell lithium-ion and lithiumion/polymer applications, and for
powering low voltage ASICs and SoCs
from 3V, 3.3V or 5V rails. To extend
battery life, total quiescent current
with all three regulators running is
only 70µA.
Regulated output voltages are programmed via external resistors, and
can be set as low as 0.8V. Each output
has its own enable pin for maximum
lexibility. An onboard supply monitor
indicates when all enabled outputs are
in regulation.
The 400mA buck regulator features constant-frequency 2.25MHz
operation, allowing the use of small
surface mount inductors and capacitors. Burst Mode operation maintains
high eficiency in light-load and noload conditions. Internal control-loop
compensation simpliies application
design. The 150mA LDOs are stable
with as little as 1µF of external output
capacitance, minimizing application
size, and feature short-circuit protection.
Dual 8A or Single 16A Step
Down DC/DC µModule
Regulator in a 15mm × 15mm
Surface Mount Package
The LTM4616 is a complete dual DC/
DC µModule™ power supply in a tiny
surface mount package. The LTM4616
can regulate two outputs ranging from
0.6V to 5V at 8A each, or it can regulate
one output at 16A by sharing current
from the two outputs in a multiphase
coniguration. The LTM4616 is just as
versatile on the input. It can operate
from two different input supply rails
ranging from 2.375V to 5.5V (6V max)
or from one input supply by tying the
input pins together.
All of the support components
needed for a dual point-of-load regulator—inductors, capacitors, DC/DC
controller, compensation circuitry
and power switches—are encapsulated and protected in the 15mm ×
15mm × 2.8mm plastic surface mount
LGA (land grid array) package. The
package’s low proile allows smooth
airlow for cooling in densely populated
circuit boards. It is a perfect solution
for powering both core and I/O supplies for FPGAs and ASICs.
The LTM4616 is guaranteed to better than ±1.75% total DC output error
over the full operating temperature
range, including the line and load
regulation. As a current mode device with high switching frequency,
the LTM4616 has a fast transient
response to line and load changes
while operating with excellent stability with a variety of output capacitors,
including schemes that use all ceramic
capacitors.
Eficiency is as high as 94%. Frequency synchronization, multiphase
operation, spread spectrum phase
modulation, output voltage tracking
and margining are just a few of the
other features of this versatile part.
Safety features include overvoltage
and overcurrent protection as well as
thermal shutdown. L
Linear Technology Magazine • September 2008
DESIGN TOOLS L
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Linear Technology Magazine • September 2008
39
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San Diego, CA 92122
Tel: (858) 638-7131
Fax: (858) 638-7231
CENTRAL
Chicago
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Schaumburg, IL 60173
Tel: (847) 925-0860
Fax: (847) 925-0878
Cleveland
7550 Lucerne Dr., Ste. 106
Middleburg Heights, OH 44130
Tel: (440) 239-0817
Fax: (440) 239-1466
Columbus
Tel: (614) 488-4466
Detroit
39111 West Six Mile Road
Livonia, MI 48152
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Philadelphia
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Tel: (215) 638-9667
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SOUTHEAST
Atlanta
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Austin, TX 78759
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Dallas
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Dallas, TX 75248
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Fax: (972) 380-5138
Fort Lauderdale
Tel: (954) 473-1212
Houston
1080 W. Sam Houston Pkwy.,
Ste. 225
Houston, TX 77043
Tel: (713) 463-5001
Fax: (713) 463-5009
Huntsville
Tel: (256) 881-9850
Orlando
Tel: (407) 688-7616
Raleigh
15100 Weston Pkwy., Ste. 202
Cary, NC 27513
Tel: (919) 677-0066
Fax: (919) 678-0041
Tampa
Tel: (813) 634-9434
EUROPE
AUSTRALIA / NEW ZEALAND
Linear Technology Corporation
133 Alexander Street
Crows Nest NSW 2065
Australia
Tel: +61 (0)2 9432 7803
Fax: +61 (0)2 9439 2738
JAPAN
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8F Shuwa Kioicho Park Bldg.
3-6 Kioicho Chiyoda-ku
Tokyo, 102-0094, Japan
Tel: +81 (3) 5226-7291
Fax: +81 (3) 5226-0268
CHINA
Linear Technology Corp. Ltd.
Units 1503-04, Metroplaza
Tower 2
223 Hing Fong Road
Kwai Fong, N.T., Hong Kong
Tel: +852 2428-0303
Fax: +852 2348-0885
Linear Technology KK
6F Kearny Place Honmachi Bldg.
1-6-13 Awaza, Nishi-ku
Osaka-shi, 550-0011, Japan
Tel: +81 (6) 6533-5880
Fax: +81 (6) 6543-2588
Linear Technology Corp. Ltd.
Room 2701, City Gateway
No. 398 Cao Xi North Road
Shanghai, 200030, PRC
Tel: +86 (21) 6375-9478
Fax: +86 (21) 5465-5918
Linear Technology Corp. Ltd.
Room 816, 8/F
China Electronics Plaza B
No. 3 Dan Ling Rd., Hai Dian
District
Beijing, 100080, PRC
Tel: +86 (10) 6801-1080
Fax: +86 (10) 6805-4030
Linear Technology Corp. Ltd.
Room 2604, 26/F
Excellence Times Square Building
4068 YiTian Road, Futian District
Shenzhen, 518048, PRC
Tel: +86 755-8236-6088
Fax: +86 755-8236-6008
Linear Technology KK
7F, Sakuradori Ohtsu KT Bldg.
3-20-22 Marunouchi, Naka-ku
Nagoya-shi, 460-0002, Japan
Tel: +81 (52) 955-0056
Fax: +81 (52) 955-0058
KOREA
Linear Technology Korea Co.,
Ltd.
Yundang Building, #1002
Samsung-Dong 144-23
Kangnam-Ku, Seoul 135-090
Korea
Tel: +82 (2) 792-1617
Fax: +82 (2) 792-1619
SINGAPORE
Linear Technology Pte. Ltd.
507 Yishun Industrial Park A
Singapore 768734
Tel: +65 6753-2692
Fax: +65 6752-0108
TAIWAN
Linear Technology Corporation
8F-1, 77, Nanking E. Rd., Sec. 3
Taipei, Taiwan
Tel: +886 (2) 2505-2622
Fax: +886 (2) 2516-0702
FINLAND
Linear Technology AB
Teknobulevardi 3-5
P.O. Box 35
FIN-01531 Vantaa
Finland
Tel: +358 (0)9 2517 8200
Fax: +358 (0)9 2517 8201
FRANCE
Linear Technology S.A.R.L.
Parc Tertiaire Silic
2 Rue de la Couture, BP10217
94518 Rungis Cedex
France
Tel: +33 (1) 56 70 19 90
Fax: +33 (1) 56 70 19 94
GERMANY
Linear Technology GmbH
Osterfeldstrasse 84, Haus C
D-85737 Ismaning
Germany
Tel: +49 (89) 962455-0
Fax: +49 (89) 963147
Linear Technology GmbH
Haselburger Damm 4
D-59387 Ascheberg
Germany
Tel: +49 (2593) 9516-0
Fax: +49 (2593) 951679
Linear Technology GmbH
Jesinger Strasse 65
D-73230 Kirchheim/Teck
Germany
Tel: +49 (0)7021 80770
Fax: +49 (0)7021 807720
ITALY
Linear Technology Italy Srl
Orione 3, C.D. Colleoni
Via Colleoni, 17
I-20041 Agrate Brianza (MI)
Italy
Tel: +39 039 596 5080
Fax: +39 039 596 5090
SWEDEN
Linear Technology AB
Electrum 204
Isafjordsgatan 22
SE-164 40 Kista
Sweden
Tel: +46 (8) 623 16 00
Fax: +46 (8) 623 16 50
UNITED KINGDOM
Linear Technology (UK) Ltd.
3 The Listons, Liston Road
Marlow, Buckinghamshire
SL7 1FD
United Kingdom
Tel: +44 (1628) 477066
Fax: +44 (1628) 478153
CANADA
Calgary, AB
Tel: (403) 455-3577
Montreal, QC
Tel: (450) 689-2660
Ottawa, ON
Tel: (613) 421-3090
Toronto, ON
Tel: (440) 239-0817
Vancouver, BC
Tel: (604) 729-1204
Linear Technology Corporation
1630 McCarthy Blvd.
Milpitas, CA 95035-7417
1-800-4-LINEAR • 408-432-1900 • 408-434-0507 (fax)
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© 2008 Linear Technology Corporation/Printed in U.S.A./43K
Linear Technology Magazine • September 2008