V21N1 - APRIL

April 2011
I N
T H I S
I S S U E
dual input/output 3A
monolithic buck with
3V–36V input range 10
Volume 21 Number 1
Harvest Energy from a
Single Photovoltaic Cell
Nathan Bourgoine
how to drive low power,
1Msps, 16-bit, differential
input ADC from singleended signals 30
maximize output power
from current-limited USB
and PCMCIA sources 38
dual output step-down
controller converts 60V
directly to 3.3V 40
POWER
SUPPLY
CONNECTOR
DROPS
WIRING DROPS
CONNECTOR
DROPS
WIRING DROPS
To simplify the distribution of wireless communications for
instrumentation, monitoring and control applications, power supply
designers strive for device grid-independence. Batteries, the
immediately obvious solution, offer the illusion of grid independence,
but require replacement or recharging, which means eventual
connection to the grid and expensive human intervention and
maintenance. Enter energy harvesting, where energy is collected
from the instrument’s immediate environment, offering perpetual
operation with no connection to the grid and minimal or no
maintenance requirements.
CONNECTOR
DROPS
LOAD
CONNECTOR
DROPS
Figure 1. The simplest model for load regulation over
resistive interconnections.
Caption
A variety of ambient energy sources can be harvested to produce electrical power, including mechanical vibration, temperature differential and
incident light. Linear Technology produces power management solutions
that solve the problems specific to harvesting ambient low energy sources,
including the LTC®3588 for vibration sources, the LTC3108/LTC3109 for
thermal, and now the LTC3105 for photovoltaic energy harvesting applications. Photovoltaic energy harvesting is widely applicable, given that light
is almost universally available, photovoltaic (PV) cells are relatively low
cost and they produce relatively high power compared to other ambient
energy harvesting solutions. Because of its relatively high energy output,
photovoltaic energy harvesting can be used to power wireless sensor
nodes, as well as higher power battery charging applications to extend
battery life, in some cases eliminating tethered charging altogether.
While high voltage stacks of series-connected photovoltaic cells are prolific,
single PV-cell solutions are rare, due to the difficulty of generating useful
power rails from the low voltage produced by a single PV cell under load.
Few boost converters can produce outputs from a low voltage, relatively
(continued on page 2)
w w w. li n ea r.com
…continued from the cover
In this issue...
COVER STORY
Harvest Energy from a Single Photovoltaic Cell
1
DESIGN FEATURES
Protect Mobile Devices from Hot Plug
Transients (to 85V) and from Users Who
Use the Wrong Power Adapter
Kevin Wong
(LTC3105, continued from page 1)
7
Monolithic, Dual 3A Input/Output Buck with
3V–36V Operating Range Simplifies and
Shrinks DC/DC Converters in Automotive,
Industrial and Distributed Power Applications
Jonathan Paolucci
10
3A Output, 96% Efficient Buck-Boost
DC/DC Converter Sets the Standard for
Power Density and Noise Performance
Richard Cook
19
22
Low IQ, Triple Output Boost/Buck/Buck
Synchronous Controller Keeps Electronics
Running Through Battery Transients in Automotive
Start-Stop and Always-On Systems
Joe Panganiban and Jason Leonard
26
DESIGN IDEAS
How to Drive Low Power, 1Msps, ±2.5V
Differential-Input, 16-Bit ADC with a
Variety of Single-Ended Signals
Guy Hoover
30
Easy, Isolated Low Power Telecom Supply:
No Opto-Isolator Required
Mayur Kenia
32
4mm × 5mm, Dual Input/Output,
Synchronous Monolithic Buck Regulator
Converts 12V to 1.2V at 4MHz
Phil Juang
Figure 1. Simple photovoltaic cell model
Two common parameters that characterize a PV cell are the open circuit voltage and the short-circuit current. Typical curves for PV cell current and voltage are shown in Figure 2. Note that the short-circuit current
is the output of the model’s current generator while the open circuit voltage is the forward voltage of the model’s diode. As light levels increase,
the current from the generator increases and the IV curves move up.
To extract maximum power from the PV cell, the input resistance of the power
converter must be matched to the output resistance of the cell, resulting in operation at the maximum power point. Figure 3 shows the power curve for a typical
single photovoltaic cell. To ensure maximum power extraction, the output voltage
of the PV cell should be operated at the peak of the power curve. The LTC3105
adjusts the output current delivered to the load in order to maintain the PV cell
Figure 2. Typical photovoltaic cell IV curve
Figure 3. Typical photovoltaic cell power curve
34
250
36
Jason Leonard and Joe Panganiban
40
product briefs
43
back page circuits
44
90
80
BRIGHTER
200
OUTPUT CURRENT (mA)
38
Low IQ, Dual Output Step-Down Controller
Converts 60V Directly to 3.3V
150
100
60
50
40
30
20
50
0
BRIGHTER
70
DIMMER
0
0.1
0.4
0.2
0.3
CELL VOLTAGE (V)
2 × 1 INCH POLYCRYSTALLINE CELL
2 | April 2011 : LT Journal of Analog Innovation
–
(continued on page 4)
Buck-Boost Converter with Accurate Input
Current Limit Maximizes Power Utilization
from USB and PCMCIA Sources
Michael Munroe
VCELL
Photovoltaic sources can be electrically modeled by a current source connected in parallel with a diode as shown in Figure 1. More complex models
show secondary effects, but for our purposes this model is sufficient.
Isolated Flyback Converters Eliminate Opto-Coupler
Yat Tam
+
UNDERSTANDING PHOTOVOLTAIC CELL SOURCES
Intermediate Bus Buck Regulator Maintains 5V Gate
Drive During Automobile Cold Crank Conditions
Theo Phillips and Tick Houk
high impedance single PV cell. The LTC3105,
however, is designed specifically to meet these
challenges. Its ultralow 250mV start-up voltage and programmable maximum power
point control allow it to generate the typical voltage rails (1.8V–5V) required for most
applications from challenging PV sources.
POWER (mW)
Nathan Bourgoine
The LTC3105 enables autonomous remote sensor nodes,
data collection systems and other applications that
require grid independence and minimal maintenance.
DIMMER
10
0.5
0.6
0
0
0.1
0.4
0.2
0.3
CELL VOLTAGE (V)
2 × 1 INCH POLYCRYSTALLINE CELL
0.5
0.6
Linear in the news
Linear in the News
JAPAN PRESS CONFERENCE
EDN Innovation Award Finalists. EDN magazine
EDN Hot 100 Products. EDN magazine
On December 17, Linear Technology held
a major press conference in Japan to give
27 assembled members of the press an
overview of Linear’s strategy. At the meeting, Executive Chairman Bob Swanson
presented an overview of the company’s
repositioning to focus primarily on three
major market segments—industrial
(including medical, security, factory automation, instrumentation and industrial
control), communications infrastructure
(including cellular base stations to support
worldwide growth in wireless networks,
as well as networking), and automotive electronics (including battery stack
monitors for hybrid/electric vehicles and
LED lighting systems). CEO Lothar Maier
further elaborated on Linear’s strategy
and product focus in Japan and globally.
has announced finalists for their annual
Innovation Awards, to be presented
on May 2. Linear finalists include:
highlighted the Hot 100 Products
of 2010, including:
•Innovator of the Year: Robert Dobkin
and Tom Hack for the LT®4180
Virtual Remote Sense™ Controller.
•LT4180 Virtual Remote Sense Controller
The press conference was covered by
several major Japanese publications,
including Dempa Shimbun, EDN Japan,
EE Times Japan, Electronic Journal and
Nikkei Electronics, among others.
LINEAR RECEIVES
PRODUCT AWARDS
Linear continues to receive numerous awards for its innovative, high
performance products. The awards
are determined by independent
groups of technical editors, representing various print and online publications around the globe. Following
are a few recent award highlights:
•Power ICs: LT4180 Virtual
Remote Sense Controller
•Analog ICs: TimerBlox® IC family
You can read more about the finalists at innovation.edn.com/finalists.
Energy Harvesting Product Awards. Two
UK publications presented Linear
Technology with awards for energy
harvesting products. Electronics Weekly
featured the LTC3109 auto-polarity
energy harvesting power supply as winner of the Renewable Energy Design
Award in their coverage of the Elektra
2010 Awards. In addition, Electronic
Product Design featured the LTC3109
as winner of the Alternative Energy
Award in their 2010 e-Legacy Awards.
EN-Genius Network Awards. Online publica-
tion, En-Genius Network presented
2010 Product of the Year Awards to:
•LT6656 low power voltage reference
•LTC3108 energy harvesting
boost regulator
CONFERENCES & EVENTS
EDN Designing with LEDs Conference & Workshop,
San Jose Marriott Hotel, San Jose, California,
May 4. Linear will showcase LED driver
solutions at the booth and Design
Manager Bryan Legates will conduct a workshop on LED drivers. Info
at ubmelectronicsvirtualconferences.
businesscatalyst.com/LED/index.html
Energy Efficiency & Technology Conference,
Marriott Santa Clara, Santa Clara, California, May
6. Linear will highlight its energy harvest-
ing solutions at the booth, and provide a
speaker at the conference. Info: eetweb.
com/sponsor/energy-efficiency-technology/
Electronic Sourcing Live Exhibition, Regency Park
Hotel, Newbury, UK, May 12. Linear will have a
booth promoting µModule power solutions and Linear Express® product fulfillment. Info: www.es-live.co.uk/index.html
•Best RF Detector: LTC5583
Dual RMS RF Detector
Sensors Expo, Donald E. Stephens Convention
•Best Advance in Remote Load
Power Sensing: LT4180 Virtual
Remote Sense Controller
showcase energy harvesting solutions at
its booth, and designer Jim Noon will give
a presentation on energy harvesting. Info:
www.sensorsmag.com/sensors-expo n
•Best Ultralow Input Voltage
DC/DC Power Module: LTM®4611
1.5V Input 15A µModule® Controller
Center, Rosemont, Illinois, June 6-8. Linear will
April 2011 : LT Journal of Analog Innovation | 3
While high voltage stacks of series-connected photovoltaic
cells are prolific, single PV-cell solutions are rare, due
to the difficulty of generating useful power rails from the
low voltage produced by a single PV cell under load.
90
(LTC3105, continued from page 2)
LTC3105
voltage at the voltage set by the maximum power point control pin. Therefore,
a single programming resistor establishes
the maximum power point and ensures
maximum power extraction from the
PV cell and peak output charging current.
10µA
VIN
RMPPC
+
–
–g
m
+
60
50
40
MPPC VOLTAGE
30
20
DIMMER
10
0
0
0.1
0.4
0.2
0.3
CELL VOLTAGE (V)
0.5
0.6
2 × 1 INCH POLYCRYSTALLINE CELL
Figure 4. Maximum power point control mechanism
Figure 5. Err on the side of a lower voltage when
choosing a maximum power point voltage to avoid
the steep drop-off
inches is sufficient to run many remote
sensors and to trickle charge a battery.
presents design challenges. Even a large
high efficiency crystalline cell with an
area of four square inches generates
only 860µW in typical office lighting.
In contrast, devices operating from indoor
lighting have far less energy available
to them. Common indoor lighting is
roughly 0.25% as strong as full sunlight
(the huge difference in intensity between
indoor lighting and sunlight is hard to
perceive due to the human eye’s ability
to adjust to a wide range of illumination levels). The dramatically lower light
levels available to indoor applications
Figure 6. Li-ion charging circuit
L1
10µH
225mV TO 500mV
L1: COILCRAFT MSS5131-103MX
VIN
+
SW
VOUT
4.1V
VOUT
10µF
LTC3105
1020k
FB
OFF ON
40.2k
LDO
SHDN
AUX
1µF
Li-ION
PGOOD
MPPC
4 | April 2011 : LT Journal of Analog Innovation
POWER (mW)
MPPC
The amount of power that can be generated using a photovoltaic cell depends on
a number of factors. The output power of
the cell is proportional to the brightness of
the light landing on the cell, the total area
of the cell, and the efficiency of the cell.
Most PV cells are rated for use under full
direct sunlight (1000W/m2), but such ideal
conditions are unlikely to occur in most
applications. For devices operating from
sunlight, the peak power available from
the cell can easily change by a factor of ten
from day to day due to weather, season,
haze, dust, and incident angle of the sunlight. Typical output power for a crystalline cell in full sunlight is about 40mW per
square inch depending on cell characteristics. A PV cell with an area of a few square
–
IPEAK
VCC
BRIGHTER
70
BOOST CONVERTER
HOW MUCH POWER IS AVAILABLE?
PHOTOVOLTAIC
CELL
80
332k
2.2V
FBLDO
GND
4.7µF
10µF
CHOOSING THE MAXIMUM
POWER POINT CONTROL VOLTAGE
Figure 4 shows a model of the maximum power point control mechanism
used by the LTC3105. Figure 3 shows
the power curve for a PV cell. Note that
PV cell power declines sharply from its
peak as the cell voltage rises away from
peak power. It is thus generally more
desirable to err on the side of a lowerthan-ideal control voltage, rather than a
higher voltage, because the power curve
rolls off more sharply on the high side.
When selecting the MPPC tracking voltage, various operating conditions must be
considered. Typically, the maximum power
point does not move substantially with
changes in illumination. As a result, it is
design features
The LTC3105’s integrated maximum power point control and low voltage start-up
functionality enable direct operation from a single PV cell and ensure optimal energy
extraction. The LTC3105 can be used to directly power circuitry or for charging
energy storage devices to allow operation through dark or low light periods.
11
SUNNY DAY
OUTPUT CURRENT (mA)
9
BUILDING
SHADOW
7
5
3
1
–1
7:00
RAINY OVERCAST DAY
10:00
13:00
16:00
TIME OF DAY
19:00
2 × 1 INCH POLYCRYSTALLINE CELL
Figure 7. Charging profiles for two square inch
photovoltaic cell
possible to choose a single tracking voltage
that provides operation near the maximum
power point for a wide range of illumination levels. Even though the operating
point will not be precisely at the maximum power point at extreme levels of illumination, the reduction in output power
from the ideal is usually only 5%–10%.
For the power curve shown in Figure 5, an
MPPC voltage of 0.4V yields performance
near the maximum power point at either
illumination extreme. The voltage difference from the maximum power point
is approximately 20mV in both cases,
resulting in a power loss of less than 3%.
As a rule of thumb, the maximum power
point control voltage should be around
75%–80% of the open circuit voltage for
the PV cell. Tracking the cell to this voltage results in a cell output current that is
75%–80% of the short-circuit current.
LI-ION BATTERY CHARGING
IN OUTDOOR LIGHTING
CHOOSING THE RIGHT
ENERGY STORAGE DEVICE
One of the challenges faced by applications using a photovoltaic source is the
lack of input power during darkness and
low light conditions. For most applications this necessitates use of energy
storage elements such as a supercapacitor or rechargeable battery that is large
enough to provide power throughout
the longest expected dark period.
There are many alternatives for storing
harvested energy, including a wide variety
of rechargeable battery technologies and
high energy density capacitors. No one
technology is perfect for all applications.
When selecting the storage element for
your application, consider a number of
factors, including the self-discharge rate,
maximum charge and discharge current,
voltage sensitivity, and cycle lifetime.
Figure 7 shows the measured charging
current profile using a 2” × 1” polycrystalline PV cell to charge a Li-ion battery using the LTC3105 circuit shown in
Figure 6. The upper curve of Figure 7
shows the charging current on a typical
clear day with full sun. The lower curve
shows the charging current observed
over the course of a heavily overcast day.
Even under these low light conditions a
charging current of 250µ A or more was
maintained throughout the day totaling
6mAh of charge delivered to the battery.
The self-discharge rate is particularly
important in photovoltaic applications.
Given the limited amount of charging
current available in most photovoltaic
power applications, a high self-discharge
rate may consume a large portion of the
available energy from the PV source. Some
energy storage elements, such as large
supercapacitors, may have self-discharge
current in excess of 100µ A, which could
dramatically reduce the net charge
accumulated over a daily charge cycle.
Figure 8. A Li-ion trickle charger operates from a single photovoltaic cell
L1
10µH
225mV TO 500mV
PHOTOVOLTAIC
CELL
L1: COILCRAFT MSS5131-103MX
VIN
+
SW
10µF
–
VOUT
VOUT
LTC3105
1020k
ADJ
FB
PGOOD
MPPC
OFF ON
40.2k
LDO
SHDN
AUX
1µF
2.2V
FBLDO
GND
NTC
332k
10µF
BAT
Li-ION
LBSEL
4.7µF
VCC
LTC4071
GND
April 2011 : LT Journal of Analog Innovation | 5
The LTC3105 is a complete single-chip
solution for energy harvesting from low
cost, single photovoltaic cells.
Another key consideration is the rate at
which the energy storage device can be
charged. For example, a lithium coin cell
with a maximum charging current of
300µ A requires a large resistor between
it and the output of the LTC3105 in order
to prevent overcurrent conditions. This
can put a limit on the amount of energy
harvested, decreasing the amount of
energy available to the application.
In many cases the charge rate is proportional to another important factor, cycle
lifetime. The cycle life of a storage element determines how long it can operate in the field without maintenance.
Generally, faster charging and discharging
reduces the operational life of the element. Supercapacitors offer very good
cycle life, while batteries charged with
relatively high currents (charge > 1C)
have degraded lifetimes. In addition to
the charge and discharge rate, the depth
–
VIN
SW
VOUT
CIN
10µF
–
FB
OFF ON
MPPC
PGOOD
SHDN
LDO
CAUX
1µF
AUX
RMPPC
40.2k
of each charge/discharge cycle can affect
the lifetime of batteries, with deeper
cycles leading to shorter life times.
With several battery types, notably lithium
and thin film, the maximum and minimum
voltage must be carefully controlled. The
VOUT
3.3V
COUT
100µF
R2
1.02M
XMTR
I/O
OFF ON
EN
PGOOD
SHDN
LDO
AUX
CAUX
1µF
2.2V
FBLDO
GND
* COILCRAFT MSS5131-103MX
6 | April 2011 : LT Journal of Analog Innovation
+
NiMH
×2
1.8V
FBLDO
GND
COUT
10µF
VOUT
3.2V
+
R4
1.27M
CLDO
4.7µF
Figure 9. Single-cell photovoltaic NiMH trickle charger
R1
2.32M
FB
R2
470k
R3
1M
RMPPC
40.2k
VOUT
LTC3105
R1
1.02M
LTC3105
SW
CIN
10µF
MPPC
2N7000
VIN
+
Figure 10. Single-cell-powered remote wireless sensor
L1*
10µH
+
L1, 10µH
µC
RPG
499k
VDD
CLDO
4.7µF
A/D
GPIO
GND
SENSOR
maximum charge voltage is well controlled in LTC3105 applications since the
converter terminates charging when the
output comes into regulation. To prevent
over-discharge, the LTC3105 can be used
in conjunction with the LTC4071 shunt
battery charger as shown in Figure 8.
CONCLUSION
The LTC3105 is a complete single chip solution for energy harvesting from low cost,
single photovoltaic cells. Its integrated
maximum power point control and low
voltage start-up functionality enable direct
operation from a single PV cell and ensure
optimal energy extraction. The LTC3105
can be used to directly power circuitry
or for charging energy storage devices to
allow operation through dark or low light
periods. The LTC3105 makes it possible
to produce autonomous remote sensor
nodes, data collection systems and other
applications that require grid independence and minimal maintenance. n
design features
Protect Mobile Devices from Hot Plug Transients (to 85V)
and from Users Who Use the Wrong Power Adapter
Kevin Wong
Battery powered mobile gadgets like smart phones,
tablets and digital cameras have become integral parts
of the modern lifestyle. More and more functionality
is squeezed into increasingly small form factors in the
endless quest for more mobility. The proliferation of mobile
devices has spawned a corresponding number of power
adapters to charge batteries and power the devices: from
AC wall outlets, car battery adapters, USB ports and
even mobile solar panels. Although many adapters use
similar plugs, their electrical specifications can be very
different. Product designers are thus forced to employ
protection circuitry to protect the low voltage rated
electronics from transient and steady state overvoltages.
Failures or faults in the power adapters
can cause an overvoltage event. So can
hot-plugging an adapter into the power
input of the mobile device (see Linear
Technology Application Note 88). With the
prevalence of universal connectors, a user
can also unknowingly plug in the wrong
adapter, damaging the device with a high
or even negative power supply voltage.
The LTC4360, LTC4361 and LTC4362 can
protect against the above mentioned fault
situations with minimal components. See
Table 1 for a comparison of these devices.
Table 1. Comparison of overvoltage
protection parts
VIN
5V
The LTC4360 and LTC4361 protect low
voltage electronics from overvoltage
conditions by controlling a low cost
external N-channel MOSFET configured as
a pass transistor. The LTC4362 achieves
an even smaller PCB footprint by incorporating an internal 28V, 71mΩ RDS(ON)
MOSFET and a 31mΩ sense resistor.
The LTC4360 and LTC4361 can withstand
up to 85V at IN, SENSE and GATEP. For all
three parts, there is no requirement for
a high voltage bypass capacitor at IN,
eliminating a potential point of failure.
The low voltage capacitor required at
M2
Si3590DV
RSENSE
0.05Ω
VOUT
5V
0.5A
M1
COUT
10µF
GATE
SENSE
OUT
LTC4361
VIO
5V
IN
R1
1k
GATEP
ON
PWRGD
GND
Figure 1. A 5V system protected from ±24V power
supplies and overcurrent
OUT is also the bypass capacitor to the
downstream circuits. It helps to slow
down the dV/dt at OUT during a fast
overvoltage, allowing time for the protection part to shut off the MOSFET before
VOUT overshoots to a dangerous voltage.
These features make the parts versatile
building blocks for some very robust yet
simple overvoltage protection circuits.
OPERATION
When power is first applied or the part
is activated by pulling ON low, a 130ms
delay cycle starts. Any undervoltage or
overvoltage event at IN (VIN < 2.1V or VIN >
PART
FEATURES
PACKAGE
LTC4360
85V Rated Input, 5.8V Overvoltage Threshold
SC70
LTC4361
85V Rated Input, 5.8V Overvoltage Threshold,
50mV Electronic Circuit Breaker Threshold
SOT23, DFN
(2mm x 2mm)
LTC4362
28V Rated Input, Internal 40mΩ N-Channel MOSFET and
31mΩ R SENSE , 5.8V Overvoltage Threshold, 1.5A Overcurrent
Threshold
DFN
(2mm x 3mm)
April 2011 : LT Journal of Analog Innovation | 7
With the prevalence of universal connectors for mobile
devices, users can easily plug in the wrong adapter,
damaging the device with a high or even negative power
supply voltage. The LTC4360, LTC4361 and LTC4362,
along with a few components, protect valuable downstream
devices against this and other fault situations.
WALL ADAPTER
AC/DC
RIN
LIN
IN
RSENSE
M1
Si1470DH
OUT
MOBILE
DEVICE
ICABLE
+
COUT
GATE
CABLE
SENSE OUT
LOAD
LTC4361
IN
GND
Figure 2. Typical circuit of voltage adapter charging a mobile device
5.7V) restarts the delay cycle. This allows
the MOSFET to isolate the output from
any input transients that occur at startup. When the delay cycle completes, the
MOSFET is turned on by a controlled 3V/ms
gate ramp. The voltage ramp of the output
capacitor follows the slope of the gate
ramp and sets the supply inrush current at:
IINRUSH = COUT • 3 [mA/µF]
As GATE ramps higher, it trips an internal
gate high threshold (7.2V for VIN = 5V)
to start a 65ms delay cycle. After the
delay, PWRGD asserts low to signal that
the MOSFET has fully enhanced. An
internal circuit clamps GATE at 6V above
OUT to protect the MOSFET gate.
When VIN rises above the 2% accurate
overvoltage threshold of 5.8V, a 30m A fast
pull-down on the GATE pin is activated
within 1µs and the PWRGD pull-down
is released. After an overvoltage condition, the MOSFET is held off until VIN once
again remains below 5.7V for 130ms.
In addition to overvoltage protection, the
LTC4361 and LTC4362 have overcurrent
protection to protect the pass MOSFET from
8 | April 2011 : LT Journal of Analog Innovation
excessive current. The LTC4361 implements a 10% accurate 50mV electronic
circuit breaker threshold with a 10µs
glitch filter. A 50mΩ RSENSE connected
between IN and SENSE implements a
1A overcurrent threshold as shown
in Figure 1. The LTC4362 implements
internal current sensing and has a 20%
accurate 1.5A overcurrent threshold with
a 10µs glitch filter. As in an overvoltage, an overcurrent activates a 30m A fast
pull-down on GATE and releases the
PWRGD pull-down. After an overcurrent
fault, the LTC4361-1 and LTC4362-1 latch
off while the LTC4361-2 and LTC4362-2
automatically restart after a 130ms delay.
An optional P-channel MOSFET driven
by the GATEP pin as shown in Figure 1
provides low loss negative input voltage protection down to the BVDSS of the
MOSFET. An internal IN to GATEP Zener
protects the MOSFET gate by clamping
its VGS to 5.8V when VIN goes high.
Another feature is the CMOS-compatible,
active low enable input ON. With
VIN
10V/DIV
VIN
10V/DIV
VOUT
1V/DIV
ICABLE
20A/DIV
ICABLE
20A/DIV
5µs/DIV
RIN = 150mΩ,
LIN = 0.7µH, RSENSE = 25mΩ
LOAD = 10Ω, COUT = 10µF
WITH LTC4361 PROTECTION CIRCUIT
5µs/DIV
RIN = 150mΩ,
LIN = 0.7µH
LOAD = 10Ω, COUT = 10µF
WITHOUT LTC4361 PROTECTION CIRCUIT
Figure 3. Hot-plug waveform with and without the protection of the LTC4361
design features
The LTC4360 and LTC4361 overvoltage protection controllers use small footprint
and low cost external N-channel MOSFETs while the LTC4362 incorporates the
MOSFET into a 2mm × 3mm DFN package. Although these overvoltage protection
circuits occupy very little PCB space, they are rich in features like an 85V rating at
the input side and fast response in the event of overvoltage or overcurrent.
RIN
20V
WALL
ADAPTER
+
–
LIN
ICABLE
IN
D1
B160
5V
USB
+
–
M1
Si1470DH
IN
R1
100k
GATE
OUT
VIN
20V/DIV
OUT
VOUT
5V/DIV
COUT
VGATE
10V/DIV
LOAD
LTC4360
ICABLE
10A/DIV
GND
1µs/DIV
Figure 4. Overvoltage protection waveforms when 20V adapter is plugged into 5V system
ON actively pulled to ground or left
open to pull low with its internal
5µ A pull-down, the device operates
normally. If ON is driven high while
the MOSFET is turned on, GATE is pulled
low with a weak pull-down current
(40µ A) to turn off the MOSFET gradually, minimizing input voltage transients.
The part then goes into a low current
sleep mode and draws only 1.5µ A at IN.
INPUT TRANSIENTS
Figure 2 shows the circuit of a LTC4361
protecting the power input of a mobile
device. LIN and RIN model the accumulated parasitic inductance and resistance
in the wall adapter, adapter cable and
the connector. A 20V wall adapter’s
output is hot-plugged into the device to
simulate an accidental plug-in with the
wrong adapter. To do a before and after
comparison, the LTC4361, RSENSE and
MOSFET are removed and the same hotplug is repeated with IN shorted to OUT.
Figure 3 compares the two hot-plug
waveforms. Due to the low capacitance
at the IN pin, there is little overshoot
and inrush current for the case with the
LTC4361 circuit. A higher voltage rated
MOSFET can be used to protect the system
against even higher transient or DC voltages up to the BVDSS of the MOSFET. For
example, a MOSFET with a 60V BVDSS used
with the LTC4361 is able to withstand
transient and DC voltages up to 60V at IN.
The circuit in Figure 4 illustrates a worst
case overvoltage situation that can occur
at a mobile device power input. In a
device with dual power inputs, a 20V wall
adapter is mistakenly hot-plugged into
the 5V adapter input with the 5V USB input
already live. The LTC4360 detects the
overvoltage at IN quickly and cuts off the
MOSFET. But the large current built up in
LIN causes an inductive spike at IN. The
body diode of the avalanche breakdown
rated MOSFET breaks down to discharge
this energy into COUT, clamping IN at about
40V, well below the 85V that IN can withstand. If the avalanche capability of the
MOSFET is exceeded or the voltage rise at
VOUT due to the discharge of the energy in
LIN into COUT is not acceptable, an additional external clamp like the SMAJ24A can
be placed between IN and GND.
RIN = 150mΩ, LIN = 2µH
LOAD = 10Ω, COUT = 10µF (16V, SIZE 1210)
CONCLUSION
The LTC4360 and LTC4361 overvoltage
protection controllers use small footprint
and low cost external N-channel MOSFETs
while the LTC4362 incorporates this
MOSFET into a 2mm × 3mm DFN package.
Although these overvoltage protection
circuits occupy very little PCB space, they
are rich in features like an 85V rating at
the input side and fast response in the
event of overvoltage or overcurrent. In
addition, there is a PWRGD status flag for
the downstream circuits and a low power
mode enabled by a CMOS compatible
input to save battery power when necessary. The LTC4360, LTC4361 and LTC4362
form a simple yet effective and rugged
barrier between the sensitive electronics inside a mobile device and real life
accidents like faulty, substandard power
adapters or a user’s absent-mindedness
in plugging in the wrong adapter. n
April 2011 : LT Journal of Analog Innovation | 9
Monolithic, Dual 3A Input/Output Buck with 3V–36V
Operating Range Simplifies and Shrinks DC/DC Converters
in Automotive, Industrial and Distributed Power Applications
Jonathan Paolucci
Automotive, industrial, and distributed power supplies often require buck converters
to step down their poorly regulated outputs to produce the plurality of rails used by
low voltage mixed signal systems. These supplies subject the step-down converters
to a vast assortment of supply voltage transients, underscoring the need for rugged
and efficient buck converters that provide tightly regulated outputs from a wide range
of input voltages. The LT3692A, a monolithic dual 3A step-down converter, satisfies
power demands imposed by these systems. Its wide 3V–36V input operating range
and overvoltage transient protection up to 60V, allows it to easily reign in unruly
automotive or industrial sources. Flexible configuration options allow the designer to
power the LT3692A from one or two separate input supplies while producing two
independent outputs, or to parallel the outputs to create one high current supply.
A TRUE DUAL SWITCHER
input voltages, output voltages, current
limits, power good outputs, soft-start,
undervoltage lockouts and even different synchronized switching frequencies.
Independent programmable undervoltage
lockout permits a customizable operating range within 3V to 36V while withstanding up to 60V input transients.
The LT3692A simultaneously offers high
performance, high power, uncompromising features and high voltage operation
in a dual monolithic switching converter.
The two buck channels of the LT3692A
shown in Figure 1 are completely independent. The channels can have different
Figure 1. Compact, dual-output converter produces 5V/2A and 3.3V/2A outputs from a 6V–36V input.
VIN
6V TO 35V
4.7µF
×2
VIN1
VIN2
SHDN1
BST1
VOUT1
3.3V 2A
600kHz
0.22µF
D1
D2
100µF
100pF
L1
5.6µH
24.9k
100k
8.06k
PG1
SHDN2
BST2
SW1
SW2
IND1
IND2
LT3692A
VOUT2
VOUT1
FB1
FB2
CMPI1
CMPI2
CMPO1
CMPO2
SS1
ILIM2
0.1µF
ILIM1
VC1
DIV
47pF
36.5k
D1, D4: CMDSH-4E
D2, D3: B340
15.8k
L1: IHLP2525EZER5R6M01
L2: IHLP2525EZER6R8M01
10 | April 2011 : LT Journal of Analog Innovation
D3
0.22µF
100pF
42.2k
VOUT2
5V 2A
600kHz
D4
8.06k
47µF
100k
PG2
SS2
ILIM2
VC2
RT/SYNC
330pF
L2
6.8µH
CLKOUT
GND
TJ
10nF
CLOCKOUT
600kHz
330pF
0.1µF
33pF
49.9k
39.2k
49.9k
The LT3692A tolerates low line conditions
as well, thanks to an enhanced dropout
scheme, which maintains greater than
95% maximum duty cycles regardless of
switching frequency. Two independent
programmable output current limits
minimize component size and provide
overload protection, while independent
soft-start eliminates input current surges
during start-up. Channel-independent
internal thermal shutdown circuitry lends
additional overload protection by allowing
one switcher to continue operating despite
a brief overload on the other channel.
Programmable power good pins, combined with a die junction temperature
output pin, greatly simplify power
sequencing and the task of monitoring the
LT3692A supply. Adjustable or synchronized fixed-frequency operation spans
250kHz to 2.25MHz and a synchronized
clock output allow multiple regulators to
be synchronized to the LT3692A. A unique
clock divide feature optimizes solution
design features
The two buck channels of the LT3692A are completely
independent. Each can have its own input voltage,
output voltage, current limit, power good output,
soft-start ramp, undervoltage lockout and even
its own synchronized switching frequency.
VINX
R1
(R3)
SHDN1
2.8V
R2
(R4)
1.3V
LT3692A
–
+
+
–
–
+
38V
THERMAL
SHUTDOWN
CHANNEL
DISABLE
Figure 2. Block diagram shows undervoltage and
overvoltage lockout functionality of the LT3692A.
size, efficiency and system cost by permitting channel 1 to operate at a synchronized frequency 1-, 2-, 4- or 8-times
slower than the master clock frequency.
The combination of a wide feature set and
independent channel operation simplifies complex power supply designs.
Referring to Figure 2, the LT3692A enters
shutdown if SHDN1 is below 1.3V or
VIN1 falls below 2.8V, protecting battery-powered systems from excessive
discharge. All internal regulators are
controlled by channel 1, effectively
shutting down the entire IC if channel 1
enters shutdown. With sufficient VIN voltage, Channel 1 is allowed to operate if
SHDN1 exceeds 1.3V. The single voltage
divider composed of the R1/R2 or R3/R4
combination controls the UVLO levels.
A shutdown UV/OVLO or overtemperature
condition causes an internal power-on
reset latch to be enabled, discharging the
soft-start and VC pin capacitors. This latch
remains set until the shutdown condition
terminates, whereupon the LT3692A initiates a full start-up sequence. The soft-start
voltage waveforms in Figure 4 show how
the calculated UV/OVLO limits in Figure 3
protect the LT3692A during undervoltage
and overvoltage power supply transients.
The circuit in Figure 3 shows how
the LT3692A can be configured for
Figure 3. Dual converter with default and programmable UV/OVLO
VIN
6V TO 36V
4.7µF
×2
UVLO = 2.8V
(MINIMUM INPUT VOLTAGE)
UNDERVOLTAGE AND
OVERVOLTAGE LOCKOUT
A switching regulator appears as negative
impedance to the source, potentially causing a latched fault if the source voltage
drops and the regulator draws increasingly
more current. Programmable undervoltage lockout (UVLO) offers an easy way
to avoid this problem by preventing the
buck converter from drawing current if
the input voltage is too low to support
full load operation. Overvoltage lockout (OVLO), on the other hand, prevents
the converter from operating above its
desired range. A default undervoltage
and overvoltage lockout is internally
set to 2.8V and 36V, respectively, but
can be programmed to any value.
programmable UV/OVLO on one channel
while utilizing the default UV/OVLO protection on the other channel.
VOUT1
5V 2A
1MHz
D1
0.22µF
D2
100pF
22µF
100k
L1
4.7µH
42.2k
OVLO = 38V
8.06k
SHDN1
BST1
SHDN2
BST2
SW1
SW2
IND1
IND2
D3
24.9k
FB2
CMPI1
CMPO1
33pF
9.31k
D1, D4: CMDSH-4E
D2, D3: B340
SS2
VC2
RT/SYNC
DIV
0.22µF
D4
100pF
47µF
VOUT2
3.3V 2A
1MHz
8.06k
ILIM2
ILIM1
VC1
220pF
OVLO = 28V
R6
7.15k
CMPO2
SS1
100k
L2
3.3µH
LT3692A
VOUT2
VOUT1
FB1
R5
274k
R4
13.3k
UVLO = 8V
CMPI2
PG
0.1µF
R3
68.1k
VIN2
VIN1
CLKOUT
GND
TJ
33pF
10nF
28.0k
L1: IHLP2525CZER4R7M01
L2: IHLP2525CZER3R3M01
120k
0.1µF
CLOCKOUT
1MHz
220pF
10.2k
Q1
Q1: 2N3904
UNDERVOLTAGE AND OVERVOLTAGE CALCULATIONS
VIN1: (DEFAULT UV AND OV)
UVLO = 2.8V (MINIMUM INPUT VOLTAGE)
OVLO = 38V: R1 = 0Ω, R2 = OPEN (FIG 2)
VIN2:
UVLO = 8V
SHUTDOWN THRESHOLD = 1.3V
R4/(R3 + R4) = 1.3V/8V
R4 = 13.3k, R3 = 68.1k
VIN2:
OVLO = 28V
CMPI THRESHOLD = 0.72V
R6/(R5 + R6) = 0.72V/28V
R6 = 7.15k, R5 = 274k
April 2011 : LT Journal of Analog Innovation | 11
INDEPENDENT START-UP
RATIOMETRIC START-UP
ABSOLUTE START-UP
VOUT1
0.5V/DIV
VOUT1
0.5V/DIV
VOUT2
0.5V/DIV
VOUT2
0.5V/DIV
5ms/DIV
LT3692A
LT3692A
R1
FB1
CMPI1
SS1
+
–
0.1µF
SS2
0.72V
+
–
12µA
PG1
0.72V
SS1
FB2
CMPI2
FB1
CMPI1
2.5V
CMPO1
+
–
0.1µF
PG2
12µA
SS2
0.72V
+
–
The LT3692A provides access to the positive inputs of the power good (PG) comparators via the CMPI pins. Each negative
comparator input is fixed at 0.72V to
allow tying of the input to the feedback
pin (806mV reference) for a standard
90% power good signal. Other inputs
(divided down) could come from the
internal junction temperature pin (TJ) for
overtemperature indication or the input
voltage to indicate input power good. The
comparator output could be tied to one
of the soft-start pins to disable a channel, the DIV pin to change the frequency,
the ILIM pin to reduce the current, or any
external device to communicate information. These comparators are versatile and
allow for custom, compact solutions.
Start-up sequencing and control is vitally
important in modern electronics. Complex
output tracking and sequencing between
channels can be implemented using
the LT3692A’s SS and PG pins. Figure 5
0.72V
+
–
0.1µF
PG1
VOUT2
R4
R5
CMPO2
FB2
CMPI2
2.5V
PG2
12µA
SS2
R8
0.72V
+
–
CMPO2
PG2
R7
adds a user-programmable frequency
foldback function during start-up.
The SS pins also double as independent
channel shutdown pins. Pulling either
channel’s soft-start pin below 115mV disables switching for that channel.
ELIMINATE THE CLOCK
Programming the LT3692A switching frequency could not be easier. The
RT/SYNC pin accurately sources 12µ A, so
only a single resistor (RSET) is required to
set the pin voltage and thus the switching frequency as given by the following:
R6
R5
shows various output start-up waveforms and their associated schematics.
PROGRAMMING THE
SWITCHING FREQUENCY
R3
R2
CMPO1
R6
0.22µF
PROGRAMMABLE POWER GOOD
AND START-UP SEQUENCING
12µA
PG1
R4
2.5V
FB1
CMPI1
2.5V
VOUT2
R5
R1
SS1
FB2
CMPI2
VOUT1
R3
R2
CMPO1
R6
CMPO2
LT3692A
VOUT1
R1
R2
R4
12µA
10ms/DIV
R3
VOUT2
2.5V
PG2
10ms/DIV
VOUT1
2.5V
VOUT2
0.5V/DIV
PG2
PG2
0.72V
PG1
PG1
PG1
12µA
VOUT1
0.5V/DIV
More rails mean more converters. If any
of those converters are operating at different frequencies, then the interference
beat frequencies produce radiated and
conducted EMI in addition to the switching fundamental and harmonic frequencies. For example, a converter switching
at 1.015MHz and a converter switching at
1.005MHz combine for a beat frequency
of 10kHz, right in the audio band.
RSET(kΩ) = 1.86E–6 • fSW2 + 0.0281 • fSW – 1.76
with the switching frequency (fSW)
in kHz for frequencies between
150k Hz and 2.25MHz.
To avoid start-up problems, the
LT3692A limits the minimum switching frequency to a typical value of
110k Hz. This feature, coupled with
adding a small capacitor in parallel with
the frequency-programming resistor,
SS1
500mV/DIV
SS2
500mV/DIV
VIN
5V/DIV
100ms/DIV
Figure 4. Soft-start voltage during UVLO/OVLO
12 | April 2011 : LT Journal of Analog Innovation
design features
OUTPUT SEQUENCING
CONTROLLED POWER UP AND DOWN
VOUT1
0.5V/DIV
VOUT1
0.5V/DIV
PG1/PG2
VOUT2
0.5V/DIV
VOUT2
0.5V/DIV
PG1
SS1/2
PG2
10ms/DIV
LT3692A
10ms/DIV
LT3692A
VOUT1
R1
FB1
CMPI1
2.5V
12µA
0.72V
SS1
+
–
0.1µF
R1
R2
12µA
R5
+
–
0.72V
SS1
+
–
FB2
CMPI2
0.72V
SS2
12µA
PG1
VOUT2
2.5V
FB1
CMPI1
2.5V
CMPO1
R4
+
–
PG1
VOUT2
R4
FB2
CMPI2
2.5V
PG2
R3
R2
CMPO1
R6
R5
CMPO2
Figure 5. Soft-start pin configurations
VOUT1
12µA
SS2
0.72V
+
–
CMPO2
R6
R5
PG2
0.22µF
Beat frequencies can easily interfere with
any signal path with similar frequencies. Traditionally, the solution involves
synchronizing the converters by means of
an external oscillator. The LT3692A outputs a 0-to-2.5V square wave on the
CLOCKOUT pin, which matches its free running internal oscillator or the signal on the
RT/SYNC pin. Since the LT3692A can be used
as an oscillator source, this eliminates the
need for an external oscillator, reducing
cost and solution footprint. The circuit in
Figure 7 shows how the CLOCKOUT signal
2500
CLKOUT FREQUENCY (kHz)
2250
2000
1750
1500
1250
1000
750
500
250
0
0
10
20 30 40 50 60
RT/SYNC RESISTANCE (kΩ)
70
80
can synchronize two LT3692A converters
operating at 1MHz. A single high current 3.3V/10A output rail is created by
connecting the VOUT, FB, SS and VC pins
between the two LT3692As. Additionally,
the finite synchronization signal-toswitch delay allows the four channels to
be synchronized with a 90° phase shift
between each channel (shown in Figure 8),
reducing the output voltage ripple and
bulk input and output capacitances.
LT3692A SYNCHRONIZATION
The LT3692A RT/SYNC input offers a unique
synchronization feature—the duty cycle of
the input synchronization signal controls
the switching phase difference between the
two channels. Channel 1’s rising switch
edge synchronizes to the rising edge of the
signal; channel 2’s rising switch edge synchronizes to the falling edge of the signal.
By varying the synchronization duty cycle,
the LT3692A dual switches can be operated
anti-phase and in some cases non-overlapping, effectively reducing the input current
ripple and required input capacitance.
For example, the input ripple voltage shown in Figure 9 has a peak of
472mV for a typical anti-phase dual
14.4V-to-8.5V and 14.4V-to-3.3V regulator. Figure 10 shows that the input ripple
voltage is decreased to 160mV by driving the LT3692A with a 71% duty cycle
synchronization signal to generate a
256° phase shift between the channels.
DROPOUT ENHANCEMENT
Switching regulator dropout performance
is vitally important in systems where the
input voltage may drop close to, and
sometimes below, the regulated output
voltage. During a low input voltage
condition, the converter should supply
an output voltage as close to the regulation voltage as possible in order to keep
the output running. Ideally, in such cases,
the switching regulator would run at
100% duty cycle, simply passing the input
to the output, but this is not possible
because of the minimum switch off-time,
which limits the switching duty cycle.
Figure 6. Switching frequency vs RT/SYNC resistance
April 2011 : LT Journal of Analog Innovation | 13
The LT3692A tolerates low line conditions as
well, thanks to an enhanced dropout scheme,
which maintains greater than 95% maximum duty
cycles regardless of switching frequency.
VIN
5V TO 28V
37.4k
2.2µF
x4
0.22µF
100k
D1
22µF
x4
D2
L1
2.2µH
150pF
24.9k
SW1
SW2
IND1
IND2
U1
LT3692A
VOUT2
VOUT1
FB1
PG
D1, D4, D5, D8: CMDSH-4E
D2, D3, D6, D7: B340
L1, L2, L3, L4: IHLP2525CZER2R2M01
CLKOUT
RT/SYNC
CMPI1
DIV
28.0k
GND
VIN2
SHDN1
BST1
SHDN2
BST2
FB2
CMPI2
CMPO2
CMPO1
8.06k
VIN1
VIN2
SHDN1
BST1
13k
VOUT1
3.3V
10A
VIN1
L2
2.2µH
D3
0.22µF
CLOCKOUT 1MHz
D4 D5
0.22µF
D6
L1
2.2µH
SW1
SW2
IND1
IND2
U2
LT1692A
VOUT2
VOUT1
VOUT1
FB1
FB2
RT/SYNC
SS1
SS2
ILIM1
SS1
SS2
ILIM1
ILIM2
VC1
VC2
ILIM2
VC1
VC2
0.1µF
1500pF
TJ
10nF
SHDN2
BST2
24.9k
33pF
x4
3.92k
DIV
L2
2.2µH
0.22µF
D7
D8
VOUT1
100k
FB1
120°C
TEMP
FLAG
CMPO2
CMPO1
CLKOUT
CMPI2
TJ
GND
23.7k
CMPI1
10nF
35.7k
Figure 7. 3.3V, 10A 4-phase converter with UVLO, power good, 120°C junction temperature flag, and minimal input current ripple
Because the minimum switch off-time
is a fixed value, the maximum switching duty cycle can be increased simply
by decreasing the switching frequency,
but a lower switching frequency necessitates larger filter components to
achieve low output voltage ripple. The
SW1
10V/DIV
SW2
10V/DIV
LT3692A circumvents dropout limitations by keeping the monolithic high side
switch on for multiple switch cycles, only
terminating the extended switch cycle
when the boost capacitor needs to be
recharged. This unique dropout switching technique allows the LT3692A to
SW1
5V/DIV
SW1
5V/DIV
SW2
5V/DIV
SW2
5V/DIV
SW3
10V/DIV
INPUT RIPPLE
(472mVP–P)
200mV/DIV
SW4
10V/DIV
RT/SYNC
2V/DIV
200ns/DIV
Figure 8. 4-phase converter switch waveforms
14 | April 2011 : LT Journal of Analog Innovation
achieve up to a 95% maximum duty cycle,
independent of switching frequency. The
graph in Figure 11 compares the dropout
performance of a LT3692A to a similar
buck converter at 200kHz and 2MHz.
Both converters show similar dropout
performance at 200kHz; however, at
INPUT RIPPLE
(160mVP–P)
200mV/DIV
RT/SYNC
2V/DIV
400ns/DIV
Figure 9. Dual 14.4V/8.5V, 14.4V/3.3V with standard
180° phase shift between channels
400ns/DIV
Figure 10. Dual 14.4V/8.5V, 14.4V/3.3V with 256°
phase shift between channels shows significant
reduction in input voltage ripple. Phase shift
is programmed by the duty cycle of the input
synchronization signal.
design features
Separate input supply pins (VIN1/VIN2) allow the LT3692A’s
two channels to be operated in cascade, with the output
of one buck powering the input of the other. A cascade
configuration allows high input/output ratios at high
frequencies while simultaneously creating two rails.
2MHz, the LT3692A regulates the output
to 5V at a much lower input voltage.
NEVER SKIP A PULSE
High frequency switching permits smaller
components, but it also means shorter
pulse widths. Buck converters have
inherent minimum on-times that prohibit
high step-down ratios at high frequency.
When the input voltage rises too high,
the converter skips a pulse. Though using
the built-in pulse skipping inherent in
many buck converters sounds appealing, the output voltage ripple suffers
significantly, as shown in Figure 12.
Pulse skipping can be avoided by reducing the switching frequency, but in a dual
converter, one channel may benefit from
switching at a higher frequency than the
other channel. For instance, consider a
dual buck converter with an input voltage range of 7V to 36V and output voltages of 5V and 1.8V. At the high end of
the input voltage range, the switching
frequency required to avoid pulse skipping on the 5V channel is almost three
times greater than that required by the
1.8V channel. By running a dual converter
at the lower frequency—chosen to avoid
pulse skipping on the 1.8V channel—the
5V channel requires inductor and capacitor values that are three times larger than
it would if run at the higher frequency.
The LT3692A avoids this predicament by
adding a DIV pin that divides the clock by
1, 2, 4, or 8, allowing channel 1 to run at
a lower synchronized frequency. Figure 13
shows an application that runs at 250kHz
and 1MHz for the low voltage and higher
voltage channels, respectively. Figure 14
shows the switching waveforms. If channel
1 (VOUT = 1.8V) runs at 1MHz , the maximum input voltage for constant output
voltage ripple is only 15V, but at 250kHz
the maximum voltage for constant output
ripple exceeds the LT3692A overvoltage
limit of 38V. Table 1 shows the maximum
input voltage for constant output voltage
ripple for various switching frequencies.
INDEPENDENT SUPPLY INPUTS
Separate input supply pins (VIN1 /VIN2)
allow the LT3692A’s two channels to
be operated in cascade, with the output of one buck powering the input
of the other. A cascade configuration
allows high input/output ratios at
high frequencies while simultaneously
creating two rails. For instance, the
converter in Figure 15 is designed for
3.3V/2.5A at 550kHz and 1.2V/1A at 2.2MHz
across the full input voltage range.
The benefits of cascading both converters on the same chip are numerous:
•The switching frequency is already
synchronized with anti-phase
switching to reduce ripple
•Custom start-up options
are readily available
•Pulse-skipping mode is easily avoided
6
5
VOUT
fSW = 1MHz
(PULSE
SKIPPING)
20mV/DIV
VOUT (V)
4
3
2
1
0
VOUT
fSW = 250kHz
(FULL FREQ)
20mV/DIV
LT3692A, 200kHz
LT3692A, 2MHz
BUCK, 200kHz
BUCK, 2MHz
3
4
5
6
VIN (V)
7
8
9
Figure 11. The LT3692A dropout enhancement
feature improves dropout performance over
a standard buck regulator at high switching
frequencies.
FREQUENCY
(kHz)
RT/SYNC
(kΩ)
V IN(MAX)
(V)
250
5.90
38
500
13.0
30
1000
28.0
15
1500
44.2
10
2250
69.8
6
2µs/DIV
Figure 12. Many regulators will enter pulse-skipping
mode when they can’t support the large step-down
ratio that occurs when the input voltage rises too
high. The pulse-skipping solution is automatic and
easy, but it significantly increases output noise.
Table 1. Maximum input voltage for constant
output voltage ripple (VOUT = 1.8V)
April 2011 : LT Journal of Analog Innovation | 15
VIN
6V TO 36V
4.7µF
×2
D1
D2
100pF
10k
PG1
IND2
LT3692A
VOUT2
VOUT1
L2
3.3µH
FB1
42.2k
FB2
CMPI1
CMPI2
CMPO1
CMPO2
SS1
DIV
33pF
49.9k
10k
D1, D4: CMDSH-4E
D2, D3: B340
28k
VOUT2
5V 2A
47µF 1MHz
100pF
100k
8.06k
PG2
ILIM2
VC2
RT/SYNC
680pF
0.47µF
SS2
ILIM1
VC1
0.1µF
D3
SW2
IND1
L1
6.8µH
100µF
×2 8.06k
100k
D4
SHDN2
BST2
SW1
0.47µF
VOUT1
1.8V 2A
250kHz
VIN2
VIN1
SHDN1
BST1
CLOCKOUT
1MHz
CLKOUT
TJ
GND
SW1
10V/DIV
ILIM1
330pF
10nF
100k
0.1µF
33pF
SW2
10V/DIV
36.5k
L1: IHLP2525CZER6R8M11
L2: IHLP2525CZER3R3M01
1µs/DIV
Figure 13. The LT3692 can avoid pulse skipping by decreasing the operating frequency of its low voltage
channel, while leaving the higher voltage channel at a higher frequency. By running the higher voltage channel
at a higher switching frequency, one can still use a small inductor and output capacitor for that channel. Here,
channel 2 (5V) runs four times faster than channel 1 (1.8V) by setting the DIV pin to 1.2V.
Figure 14. A 5V and 1.8V dual multi-frequency
converter avoids pulse-skipping mode for each
channel throughout the input range while minimizing
component sizes on each channel.
•The overall solution takes much
less space than multi-IC solutions.
in order to ensure safe functionality. By
sizing the external components for fault
conditions, rather than typical operating
conditions, the overall solution tends to
be oversized and unnecessarily expensive.
such as the inductors and diodes, must be
sized to withstand steady-state overload
conditions as well. If the maximum load
drawn from a buck output is 1A, but the
buck converter’s internal current limit is
set to 4A, then all external components
must be rated for the maximum 4A load
ONE SIZE DOESN’T FIT ALL
Even if a switching regulator, such as the
LT3692A, can safely withstand overload
conditions, all the external components,
Figure 17. Current limit programming with ILIM voltage
Figure 15. A 3.3V and 1.2V dual 2-stage converter
VIN
4.5V TO 35V
D1
4.7µF
24.9k
47µF
100k
SHDN2
BST2
8.06k
PG
IND1
IND2
LT3692A
VOUT2
VOUT1
FB1
CMPI1
CMPI2
CMPO1
CMPO2
0.1µF
100k
13k
ILIM2
VC2
RT/SYNC
33pF
68.1
100k
4.02k
FB1
CLKOUT
GND
TJ
CLOCKOUT
2.2MHz
10nF
D1, D4: CMDSH-4E
D2: B340
D3: PD3S220L-7
16 | April 2011 : LT Journal of Analog Innovation
VOUT2
200mV/DIV
0.1µF
100pF
VOUT2
1.2V 1A
47µF 2.2MHz
8.06k
22pF
0.1µF
220pF
40.2k
IL
1A/DIV
ILIM2
500mV/DIV
10ms/DIV
SS2
ILIM1
VC1
DIV
L2
0.5µH
FB2
SS1
680pF
D3
SW2
SW1
L1
4.7µH
D4
VIN2
VIN1
SHDN1
BST1
D2
0.22µF
VOUT1
3.3V 2.5A
550kHz
1µF
33.2k
L1: IHLP2525CZER4R7M11
L2: XPL2010 0.5µH
design features
A cascaded topology allows for increased operating frequencies
while avoiding undesirable pulse skipping that can occur at high
step-down ratios. Ripple is also reduced. This allows the design
to use significantly smaller inductors and capacitors.
LT3692A in the QFN package
Before
Limiting the output current of
each channel allows for the use
of smaller power components.
After
(layout of the circuit in Figure 15)
LT3692A in
the TSSOP
package
With the LT3692A’s built-in frequency divider, each channel can easily be set to
operate at an independent frequency. In this design, the low voltage channel
operates at high frequency without worrying about pulse skipping. This allows
the use of small-package capacitors and inductors.
Figure 16. Comparison of two designs for a dual-output 3.3V/2.5A and 1.2V/1A converter using the LT3692A. The smaller version of the circuit saves board space by
taking advantage of features that the larger circuit does not, including current-limiting the outputs, cascading the channels and running the two channels at different
switching frequencies.
The LT3692A remedies this problem by
providing an independent current limit
pin (ILIM). If full output current capability is not needed on one or both channels,
the user-selectable current limit allows
the use of smaller, cheaper components.
Each channel’s current limit can be set
from 2A to 4.8A by the ILIM pin voltage. An accurate 12µ A internal current source allows the current limit to
be programmed with a single external
resistor or voltage on the ILIM pin. The
ILIM pin may be grounded as well, limiting the maximum output current to 2A.
This feature allows the user to implement current foldback during start-up
simply by placing a small value capacitor in parallel with the current-limitprogramming resistor. The 12µ A internal
current source charges the optional
ILIM cap from zero volts to its final
steady-state value, allowing the current
limit to gracefully ramp from 2A to 4.8A.
Board space is significantly reduced
by using the ILIM feature, as shown in
Figure 16. By employing the ILIM pin
function, as well as operating the channels in cascade with independent switching frequencies, the power components
from the circuit in Figure 15 reduce
board space 3-fold, underscoring
the usefulness of the ILIM pin.
OVERLOAD CONDITIONS
If the load exceeds the maximum output
current, the output voltage drops below
the normal regulation point. The drop in
output voltage activates the VC pin clamp
and discharges the SS capacitor, lowering the SS voltage. The LT3692A regulates
the feedback voltage to the lowest voltage present at either the SS pin or the
internal 806mV reference. As a result,
the output is regulated to the highest voltage that the maximum output
current can support. Once the overload condition is removed, the output
soft-starts from the temporary voltage
level to the normal regulation point.
Figure 17 shows the output voltage and
inductor current for the 1.2V channel in
Figure 15 when loaded by a 0.2Ω load.
As the ILIM pin voltage is varied from
0V to 1.5V, the output voltage is regulated
between 0.32V and 0.96V, limiting the current between the range of 1.6A and 4.8A.
WATTS FROM HERE
AND WATTS FROM THERE
Ever wanted to draw power from a rail,
but needed just a few more watts? A
last-minute increase in power requirements leaves you stuck in a bind? Now
you can draw power from two different
sources with programmable limits for
each source. The independent VIN and
ILIM pins allow the two independent input
supplies in Figure 18 to be programmed
to different current limits. With the SS,
VC , and VOUT pins tied together, the two
inputs serve a single output rail. The
April 2011 : LT Journal of Analog Innovation | 17
4.7µF
VIN1
47.5k
13k
VIN2
SHDN1
BST2
BST1
D1 0.22µF
VOUT1
3.3V
4A
47µF
D2
L1
2.2µH
8.06k
24.9k
SW1
SW2
IND1
IND2
LT3692A
VOUT2
VOUT1
FB1
CMPI1
CMPI2
CMPO1
CMPO2
SS2
ILIM1
VC1
DIV
33pF
64.9k
21.0k
ILIM2
VC2
RT/SYNC
680pF
61.9k
18
L2
2.2µH
0.1µF
D4
D3
16
VOUT1
14
FB2
SS1
0.1µF
1µF
D5
SHDN1
SHDN2
VIN2
5V
2A MAX
CLKOUT
GND
TJ
FB1
SS1
D1, D4, D5: CMDSH-4E
VC1
D2, D3: B340
CLOCKOUT L1, L2: IHLP2525CZER2R2M01
2MHz
The LT3692A TJ pin outputs a voltage proportional to the internal junction temperature. At a junction temperature of 25°C, the
TJ pin outputs 250mV and has a slope of
10mV/°C. Without the aid of external circuitry, the TJ pin output is valid from 20°C
to 150°C with a maximum load of 100µ A.
To extend the operating temperature
range of the TJ output below 20°C, connect
a resistor from the TJ pin to a negative
supply as shown in Figure 20. The negative rail voltage and TJ pin resistor may be
calculated using the following equations:
VNEG
R1 ≤
PIN(CH1) + PIN(CH2)
10
PIN(CH1)
8
6
4
PIN(CH2)
2
0
10nF
0
0.5
1
1.5 2 2.5
ILOAD (A)
3
3.5
4
Figure 19. Power draw from two sources for single
output
Figure 18. Dual input single 3.3V output converter
ALWAYS KNOW YOUR
JUNCTION TEMPERATURE
12
36.5k
61.9k
power drawn from each rail is shown
in Figure 19. This solution provides
flexibility in rail voltages and utilization of available power, making it easy
to solve power-sharing problems.
INPUT POWER (W)
VIN1
12V
where TEMPMIN is the minimum temperature where a valid TJ pin output is required. VNEG = regulated
negative voltage supply.
For example,
TEMPMIN = −40°C
VNEG ≤ −0.8 V
VNEG = −1
R1 ≤
VNEG
= 30.2k
33µA
The simple charge pump circuit in
Figure 21 uses the CLOCKOUT pin output
to generate a negative voltage, eliminating the need for an external regulated
supply. Surface mount capacitors and
dual-package Schottky diodes minimize the board area needed to implement the negative voltage supply.
CONCLUSION
The LT3692A squeezes two complete
regulators, including dual monolithic
3.8A switches, into a 38-lead exposed
pad TSSOP or a 5mm × 5mm 32-lead
exposed pad QFN package. The two
channels operate independently, making it possible to produce two high
performance buck converters with one
part, thus minimizing circuit size and
simplifying complex designs. Separate
soft-start, current limit, power good, and
UV/OVLO features enable the designer to
address unique power sharing, solution
area, and start-up sequencing requirements. With a wide operating range and a
rich feature set, the LT3692A easily tackles
a wide variety of automotive, industrial
and distributed supply challenges. n
Notes
*Many thanks to Scott McClusky for his assistance in
producing this article
(2 • TEMPMIN(°C))
≤
100
VNEG
33µA
LT3692A
TJ
TJ
R1
LT3692A
VNEG
+
GND
30k
330pF
D4
CLKOUT
GND
D3
0.1µF
D3, D4: ZETEX BAT54S
Figure 20. Circuit to extend TJ operating region
18 | April 2011 : LT Journal of Analog Innovation
Figure 21. Negative rail generated from CLOCKOUT
design features
3A Output, 96% Efficient Buck-Boost DC/DC Converter
Sets the Standard for Power Density and Noise Performance
Richard Cook
High power density has become a primary requirement
for DC/DC converters, as they must keep up with ever
increasing functional density of electronics. Likewise, power
dissipation is a major concern for today’s functionally rich,
tightly packed devices pushing the need for highly efficient
solutions to minimize temperature rise. For applications
where the input voltage source can be above or below
the regulated output voltage, finding an efficient compact
solution can be a challenge, especially at elevated power
levels. Conventional design approaches, such as using
a dual inductor SEPIC converter, produce relatively
low efficiencies and relatively large solution sizes.
The LTC3113 single inductor buck-boost
converter offers a compact, highly efficient alternative. Internal low resistance
switches allow the converter to support
an impressive 3A of load current in a tiny
4mm × 5mm package. The LTC3113 offers
an extended input and output operating voltage range from 1.8V to 5.5V, with
peak efficiencies reaching 96%. The
internal PWM controller is designed for
low noise performance and offers a seamless transition between buck and boost
modes. The combination of these features
allows the LTC3113 to easily meet challenging high density power requirements.
Figure 1 shows an 11mm × 14mm ×
2.5mm LTC3113-based solution that can
supply up to 12W of output power from a
Li-ion battery. This translates to a power
density of 31mW/mm3 (511W/in3). A complete SEPIC design would require twice as
much PCB area, resulting in half the power
density and significantly lower efficiency,
which complicates thermal design.
15mm
Figure 1. A typical application occupies 154mm2
The LTC3113 offers a number of options
to optimize performance for specific
applications, including the ability to
adjust the operating frequency from
300kHz to 2MHz, internal soft-start, user
selectable Burst Mode® operation for
improved efficiency at light load currents,
and a host of fault protection features
including short-circuit protection and
thermal shutdown. The LTC3113 is available in both a 4mm × 5mm DFN and
a 20-pin thermally enhanced TSSOP.
Figure 2. Pulsed load or portable RF power amplifier power supply and typical output response
2.2µH
VIN
3.3V
±10%
SW1
4.7µF
47µF
OFF ON
PWM BURST
SW2
VIN
VOUT
845k
LTC3113
RUN
FB
BURST
VC
RT
SGND
90.9k
10k
200µF
4.7µF
VOUT
3.8V
0A TO 3A
VOUT
200mV/DIV
33pF
68k
220pF
PGND
10pF
158k
ILOAD
2A/DIV
100µs/DIV
April 2011 : LT Journal of Analog Innovation | 19
The LTC3113 single inductor buck-boost converter offers a unique combination of features
to meet challenging high density power requirements. Internal low resistance switches
allow the converter to support an impressive 3A of load current in a tiny 4mm × 5mm
package, with peak efficiencies reaching 96%. The internal PWM controller is designed for
low noise performance and offers a seamless transition between buck and boost modes.
GSM APPLICATION
NOISE PERFORMANCE
30
Many GSM applications require expensive
supercapacitors on the DC/DC output supply rail to support the temporary heavy
loads placed on the output by the power
amplifier during transmission bursts.
In many cases, the high output current
capability of the LTC3113 is sufficient to
support the transmit current without the
need for supercapacitors. Figure 2 shows
such a circuit and associated typical load
transient for an RF power amplifier using
a standard, inexpensive 100µF ceramic
capacitor on the 3.8V output.
Many applications, including RF transmission, are sensitive to noise generated by
switching converters. The LTC3113 uses a
low noise switching architecture to reduce
unwanted subharmonic frequencies, which
occur below the operating frequency and
can interfere with other sensitive circuitry.
These subharmonics usually occur when
VIN and VOUT are approximately equal.
Buck-boost converters operating in this
region typically produce pulse width
and frequency jitter—a result of all four
switches changing state during a single
switching cycle. The LTC3113 minimizes
the magnitude of the jitter or subharmonic frequencies to satisfy the requirements of noise-sensitive RF applications.
20
The oscilloscope photo shows the response
of the 3.8V output when a 3A load pulse
lasting 580µs is applied. For this extreme
case the output voltage undershoots
only 150mV (4.5%) and quickly recovers. The output voltage overshoot when
the load is removed shows a similar
response. For this external load pulse,
the transient response has been optimized by tailoring the compensation to
minimize the effects of the load step.
Figure 3 shows worst-case spectral
comparisons of the LTC3113 and a competitive buck-boost converter without
the low noise architecture of the LTC3113.
The worst-case condition was achieved
by placing a fixed 1A load on the output
and slowly increasing or decreasing the
input voltage until the greatest harmonic content in the converter spectrum
LTC3113 fSW = 2MHz
MAGNITUDE (dB)
10
SUBHARMONICS
0
COMPETITOR
fSW = 2.4MHz
–10
–20
–30
–40
–50
–60
–70
0
0.5
2
1
1.5
FREQUENCY (MHz)
2.5
3
Figure 3. Spectral comparison of the LTC3113 and
typical competitor’s part
was observed. The LTC3113 exhibits an
expected single large magnitude tone
at its switching frequency of 2MHz. In
contrast, the competing buck-boost
exhibits several high magnitude subharmonic and harmonic tones across
the entire frequency range, indicative of
significant pulse width jitter and potential
noise interference issues. Note also that
the overall noise floor of the LTC3113 is
10d B to 20d B lower than the competition across the entire frequency range.
Figure 4. Li-ion to 3.3V supply and efficiency
100
2.2µH
SW1
47µF
OFF ON
PWM BURST
SW2
VIN
VOUT
825k
LTC3113
RUN
FB
BURST
VC
RT
SGND
47pF
49.9k
680pF
182k
80
70
PGND
90.9k
12pF
20 | April 2011 : LT Journal of Analog Innovation
6.49k
VOUT
3.3V
100µF 3A
EFFICIENCY (%)
90
VIN
2.5V TO 4.2V
Li-Ion
60
0.001
VIN = 3V
VIN = 3.7V
VIN = 4.2V
VIN = 3V BURST
VIN = 3.7V BURST
VIN = 4.2V BURST
0.01
0.1
1
LOAD CURRENT (A)
10
design features
The LTC3113 monolithic buck-boost converter breaks
new ground in power density, low noise operation and
high efficiency across a wide range of load currents.
SINGLE LI-ION TO
3.3V, 10W CONVERTER
BACKUP POWER SYSTEMS
Figure 5 shows a supercapacitor-powered
backup power supply system, where the
LTC3113 is used to provide a regulated
3.3V output at a constant 1.5A load. In this
application, two series 30F supercapacitors
have been charged to 4.5V during normal
operation to provide the needed backup
energy when the primary power is lost.
Besides generating bias voltages for
RF power amplifiers, creating a 3.3V rail
from an input source such as a Li-ion
battery is another common application
for a buck-boost converter. The LTC3113
can provide up to 10W (3.3V/3A) of output
power over the Li-ion battery’s usable
range. Figure 4 shows a typical application schematic used to generate 3.3V. Also
shown are the associated efficiency curves
for different battery voltages over a range
of load currents for this application. The
efficiency peaks at 92% and efficiencies
greater than 80% are achieved from loads
ranging from 60m A to 3A. Burst Mode
operation employs a variable frequencyswitching algorithm to maintain highly
efficient conversion at lighter loads.
The scope photo shows that the LTC3113
can regulate the output for 22.5s when
powered only by the two series 30F capacitors. Over this time, the capacitors
discharge from an initial 4.5V to just
below 1.8V—output regulation over this
input range is only possible because of
the LTC3113’s low input voltage capability. In this example, the amount
of energy supplied by the input is:
Setting the BURST pin to a voltage greater
than 1.2V allows the LTC3113 to enter
Burst Mode operation at light loads to
maximize efficiency. For noise sensitive
applications the converter can be forced
into fixed frequency operation by keeping
the voltage on the BURST pin below 0.3V.
EIN =
1
2
2

• C • ( VINITIAL ) − ( VFINAL ) 


2
EIN =
1
• 15F • ( 4.5V 2 − 1.8 V 2 ) = 127.6J
2
The output is regulated to 3.3V with a
constant load of 1.5A for 22.5s, which
yields output energy of:
EOUT = IOUT • VOUT • t
= 1.5A • 3.3V • 22.5s
= 111.4J
This shows that about 87% of the available input energy is converted to output
power. The solution size for this application is about 11mm × 14mm, excluding the area of the supercapacitors.
CONCLUSION
The LTC3113 monolithic buck-boost
converter breaks new ground in power
density, low noise operation and high
efficiency across a wide range of load
currents. The LTC3113 is an ideal solution for battery-powered products,
backup power supply systems and RF or
other noise-sensitive applications. n
Figure 5. Supercapacitor-powered supply and typical output response with 1.5A load
2.2µH
VIN
1.8V TO 4.5V
SW1
30F
30F
VIN
0.1µF
VIN
2V/DIV
SW2
VOUT
825k
LTC3113
OFF ON
PWM BURST
RUN
FB
BURST
VC
RT
SGND
6.49k
47pF
49.9k
680pF
PGND
90.9k
12pF
182k
100µF
VOUT
3.3V
VOUT
2V/DIV
RUN
2V/DIV
5s/DIV
April 2011 : LT Journal of Analog Innovation | 21
Intermediate Bus Buck Regulator Maintains 5V Gate Drive
During Automobile Cold Crank Conditions
Theo Phillips and Tick Houk
DC/DC converters in today’s automobiles often take their inputs from a loosely regulated
5V intermediate bus, instead of directly from the battery voltage (which can vary from 4V
during cold crank to above 24V from double-battery jump-starts and other transients).
Incorporating an intermediate voltage bus has a number of advantages, one of which
is the expanded range of DC/DC converter options available to power downstream
electronics. Power supply designers can choose from a wide variety of low dropout (LDO)
and switching post-regulators that have 6V absolute maximum input voltage ratings.
Because typical post-regulator outputs are substantially lower than 5V—from 3.3V down
to 1V—they can continue to operate even as their inputs drop below the nominal 5V.
With this in mind, the ideal intermediate step-down regulator would continue to provide
power even under cold crank conditions, where the battery voltage can drop below 5V.
A synchronous buck regulator often
makes the best intermediate bus converter for these applications because
of its high efficiency over a wide input
range when compared to linear regulators. In this buck topology, 40V MOSFETs
are necessary to tolerate double battery and high voltage transients, so the
regulator should provide the required
minimum 4.5V gate drive for the power
MOSFETs during cold engine cranking.
Buck regulator controllers have traditionally provided gate drive power through
either an external 5V supply or through
an onboard LDO. Both of these supply
options can only step down an input voltage, so the gate drive potential drops with
the input voltage, limiting the operating
range of the regulator. The ideal controller would require no auxiliary supply
and would provide the required 5V gate
drive voltage even when the input supply
Figure 1. The two core circuit blocks of the LTC3852: a step-down DC/DC controller
block and a charge pump doubler block, which allows the controller to continue running
even when inputs drop below 5V.
voltage drops below the minimum specified VGS rating of the power MOSFETs.
BEST OF BOTH WORLDS
The LTC3852 is a synchronous stepdown DC/DC controller with a low voltage charge pump designed to provide
5V drive to external MOSFETs even when
the input drops below 5V. Figure 1
shows the block diagram of the IC. The
LTC3852 (CHARGE PUMP)
SOFT-START
AND
SWITCH CONTROL
SHDN
ON/OFF
1.2MHz
OSCILLATOR
VPUMP
–
+
VIN1
2.7V to 5.5V
CPUMP
CHARGE
PUMP
C+
VIN1
CFLY
C–
CIN
GND1
22 | April 2011 : LT Journal of Analog Innovation
design features
The ideal intermediate step-down regulator would continue
to provide power even under cold crank conditions,
where the battery voltage can drop below 5V. The
LTC3852 does just that—even when its input drops below
5V, its integrated low voltage charge pump produces
the necessary 5V drive for the external MOSFETs.
LTC3852 contains two core circuit blocks,
a step-down DC/DC controller and a
charge pump doubler. The LTC3852 can
be configured to operate from voltages
as low as 2.7V, as shown in Figure 2,
or as high as 38V, as shown in Figure 3.
Figure 4 shows a DC/DC converter that
operates over a wide input voltage
range and provides 5V gate drive to the
MOSFETs even when VIN falls below 5V.
The charge pump doubler inside the
LTC3852 provides a regulated 5V output
at VPUMP. As the schematic of Figure 4
shows, VPUMP is typically connected to
VIN2, the main supply for the DC/DC buck
MODE/PLLIN
LTC3852 (BUCK REGULATOR)
100k
VIN2
0.8V
MODE/SYNC
DETECT
FREQ/
PLLFLTR
5V REG
+
–
VIN
38V MAX
PLL-SYNC
BOOST
OSC
S
R
Q
–
ICMP
IREV
+
–
TG
PULSE SKIP
ON
5k
+
BURSTEN
CB
M1
SW
SWITCH
LOGIC
AND
ANTISHOOT
THROUGH
SENSE+
DB
L1
VOUT
SENSE–
RUN
+
INTVCC
OV
BG
COUT
M2
CINTVCC
SLOPE COMPENSATION
GND2
PGOOD
1
100k
INTVCC
UVLO
ITHB
+
0.72V
UV
VFB
–
ITH
R1
FAULT
LOGIC
RC
+
VIN2
SLEEP
R2
OV
CC1
RUN
0.4V
+
–
SS
+
–
RUN
–
+
2µA
EA
– + +
–
0.8V
REF
0.64V
1.25V
0.88V
1µA
TRACK/SS
CSS
April 2011 : LT Journal of Analog Innovation | 23
If VIN falls below 5V, the switching regulator enters
dropout operation. VOUT also falls, but remains
high enough for the post-regulator LDOs to
continue regulating their output voltages.
100k
PGO0D
VIN1
TG
MODE/PLLIN
0.1µF
FREQ/PLLFLTR SW
TRACK/SS
0.1µF
BOOST
VIN2
INTVCC
1nF
ITH LTC3852
12.1k
CHARGE PUMP
2.2µF
L1: VISHAY IHLP4040DZ-01
M1, M2: VISHAY SILICONIX SiR438DP
DB: CENTRAL SEMI CMDSH-3
INTVCC
82.5k
100k
0.1µF
0.1µF
M2
SENSE
SHDN
SENSE–
VFB
C+
C–
20k
0.1µF
M1 L1
0.68µH
FREQ/PLLFLTR TG
SW
TRACK/SS BOOST
LTC3852
INTVCC
BG
ITH
L1: VISHAY IHLP5050EZ-01
M1,M2: RENESAS HAT2170H
DB: CENTRAL SEMI CMDSH05-4
SHDN
GND1
converter, and INTVCC , the gate drive supply to the external MOSFETs M1 and M2.
The input to the charge pump (VIN1) draws
its supply current from one of two sources.
At start-up, Q1, D1 and R1 form a simple
linear regulator, supplying current to the
charge pump from the input voltage. Once
VOUT is up and regulating, diode D2 turns
off Q1 and supplies voltage to the charge
pump from the output of the converter.
This bootstrap configuration increases the
24 | April 2011 : LT Journal of Analog Innovation
SENSE–
VFB
30
POWER LOSS
VIN = 3.3V
10
1
22µF
3.01k
DB
4.7µF
VIN
4.5V TO 38V
VOUT
3.3V
15A
330µF
×2
0.1µF
20
0.1
LOAD CURRENT (A)
M2
100
10
95
EFFICIENCY
90
85
1
80
POWER LOSS
75
70
0.1
65
60
SENSE+
MODE/PLLIN
1
40
10
GND2
330pF
50
40.2k
VIN2
RUN
60
0
GND1 VPUMP
PGO0D
70
20
+
RUN
2200pF
15k
×2
POWER LOSS (W)
Figure 3. A similar high
efficiency step-down
converter, configured
without the charge pump,
that operates from a 4.5V
to 38V input range
4.7µF
EFFICIENCY
80
+ 470µF
2.1k
10
90
GND2
100pF
OFF ON
BG
100
VOUT
1.2V
20A
0.1µF
DB
VIN
2.7V TO 5.5V
22µF
POWER LOSS (W)
95.3k
M1 L1
0.36µH
EFFICIENCY (%)
INTVCC
EFFICIENCY (%)
Figure 2. A high efficiency
step-down converter that
provides 5V gate drive over
a 2.7V to 5.5V input range
0.047µF
30.1k
55
154k
50
10m
VIN = 12V
0.1
1
10
LOAD CURRENT (A)
10m
100
48.7k
power supply’s efficiency, since the current required to drive the power MOSFETs
comes from the DC/DC converter itself.
If VIN falls below 5V, the switching regulator in Figure 4 enters dropout operation, keeping M1 on most of the time.
VOUT also falls, but remains high enough
for the post-regulator LDOs to continue regulating their output voltages.
Meanwhile, the charge pump maintains
its 5V output, providing solid gate drive
to the MOSFETs, as shown in Figure 5.
Under normal operating conditions the
converter has a 12V input and the LTC3852
behaves just like a conventional synchronous buck controller. Figure 6 shows the
efficiency vs load current for the converter
in Figure 4. The peak efficiency is 96% at
a load current of 6A and efficiency remains
high over a wide range of load currents.
design features
The LTC3852 can be configured to operate from
voltages as low as 2.7V, with no external gate
drive supply required, or as high as 38V.
MODE
100k
VIN
4V TO 36V
+
FORCED CONTINOUS MODE
Burst Mode OPERATION
PULSE-SKIPPING MODE
CIN1
56µF
50V
INTVCC
1
2
3
4
R1
1k
JP2
2.2µF
10V
0.1µF
Figure 4. No access to a 5V supply
is needed for this automotive
intermediate bus converter that
produces 5V gate drive for input
voltages below 5V
82.5k
MODE/ C+
PLLIN
C–
VIN1
TG
COUT1
47µF
10V
VPUMP
150pF
INTVCC
3V
RUN
0V
0.1µF
CIN1: SUNCON 50HVP56M
CIN5: TDK C3225X7R1H335
COUT1: TDK C3225X5ROJ476
COUT2: SANYO 6TPE220MI
DB: CENTRAL SEMI CMDSH05-4
L1: COILTRONICS HC1-3R6-R
M1: RENESAS RJK0451DPB
M2: RENESAS RJK0453DPB
M2
100Ω
1000pF
PGOOD
SENSE–
100Ω
VFB
SHDN
GND1
GND2
8.06k
42.2k
CONCLUSION
VIN
2V/DIV
VOUT
2V/DIV
100
7V
5V
3.16V
3.04V
0V
5V
INTVCC
2V/DIV
90
EFFICIENCY (%)
The LTC3852 is a synchronous stepdown DC/DC controller with a charge
pump doubler that provides 5V gate
drive, even when VIN drops below 5V.
The application presented here powers
an intermediate 5V bus from an automotive 12V battery input. Strong drive to
the MOSFETs is maintained even during
cold crank events, and high efficiency is
maintained over all operation conditions.
The LTC3852 is offered in a 3mm × 5mm
thermally enhanced QFN package. n
220µF
6.3V
GND
4.7µF
10V
SENSE+
VOUT
5V/10A
(VIN > 5V)
+ COUT2
0.1µF
BG
TRACK/SS
INTVCC
DB
VIN2
LTC3852
100k
0.003Ω
1%
SW
ITH
2200pF
PGOOD
M1 L1
3.6µH
BOOST
3.6k
D2
BAT85
CIN5
3.3µF ×4
50V
4.7µF
10V
FREQ/
PLLFLTR
OFF ON
Q1
2N3904
D1
4.7V
80
70
60
0V
VIN = 7V (NOM)
VOUT = 5V (NOM)
10ms/DIV
Figure 5. Line transient response for the
intermediate bus converter in Figure 4, illustrating
5V gate drive during a cold crank event
50
VIN = 12V
VOUT = 5V
0.1
1
LOAD CURRENT (A)
10
Figure 6. Efficiency vs load current for the
intermediate bus converter in Figure 4
April 2011 : LT Journal of Analog Innovation | 25
Low IQ, Triple Output Boost/Buck/Buck Synchronous
Controller Keeps Electronics Running Through Battery
Transients in Automotive Start-Stop and Always-On Systems
Joe Panganiban and Jason Leonard
Several automotive manufacturers use the concept of a “start-stop” system to improve
fuel economy and reduce emissions for vehicles that spend a significant amount of time
at traffic lights and in heavy, stop-and-go traffic. This system automatically turns off the
internal combustion engine whenever the car is at a complete stop and then restarts it
immediately when the driver wants to go. This reduces the amount of time the engine
spends idling, thus saving fuel. Start-stop systems have been installed in hybrid-electric
vehicles for years, but are now becoming more common in traditional vehicles (with
both manual and automatic transmissions) that lack a hybrid-electric powertrain.
Typically, a central control unit coordinates the start-stop system to ensure that
driver comfort and safety are not compromised. For example, the system is not
activated if the air conditioner has not
brought the cabin to the desired temperature or if the driver moves the steering
wheel. However, there are many systems,
such as navigation, telematics and infotainment systems (CD and DVD players,
audio systems, etc.) that remain active
when the engine is off. These systems often
operate from 5V–10V supplies generated
by step-down (buck) converters from
the nominally 12V car battery. When the
engine starts, the battery voltage can
dip to well below 5V, potentially causing these systems to glitch or reset.
synchronous buck controllers in a single
package. To achieve the wide input voltage range required in the automotive
applications described above, the part
can be configured with the vehicle battery
feeding the input to the boost converter
and the boost converter’s output feeding
the inputs to the buck converters. This
allows the two buck outputs to maintain
regulation whether the battery is above or
below the buck outputs. The outputs can
stay regulated through the entire input
range presented by the vehicle battery,
handling transients as low as 2.5V during
engine restart or cold crank and transients as high as 38V during load dump.
In a vehicle with a start-stop system,
the engine by definition restarts frequently. While it may not present a
safety risk if your DVD or CD player
restarts every time you stop at a traffic
light, it certainly is annoying, especially
for a parent relying on the DVD player
to babysit the kids in the back seat.
In this configuration, the LTC3859 can
be thought of as a dual output buckboost controller, in that it produces two
regulated outputs that can be above or
below the input voltage. When the input
is low, the boost converter operates and
steps up the voltage to an intermediate rail that provides enough headroom
for the buck converters to operate.
When the input voltage is high enough,
the boost converter stops switching
Fortunately, Linear Technology has
the solution. The LTC3859 combines a
synchronous boost controller with two
26 | April 2011 : LT Journal of Analog Innovation
THINK OF IT AS A
DUAL BUCK-BOOST
and simply turns on the top switch to
pass the input voltage through to the
intermediate rail to feed the bucks.
BOOST CONTROLLER
The LTC3859’s boost controller is based on
Linear Technology’s new LTC3788/LTC3787/
LTC3786 family of high voltage, constant
frequency, current-mode synchronous
boost controllers that drive all N-channel
MOSFET power stages. It can boost to
output voltages as high as 60V from a
4.5V to 38V (40V abs max) input voltage. If the LTC3859 is biased from VOUT or
another supply, the boost converter can
operate from an input voltage as low as
2.5V after start-up. Synchronous rectification eliminates both the high power loss
in the catch diode and the need for a
heat sink at high output currents. Strong
internal gate drivers reduce switching losses at high output voltages.
The control architecture senses current
at the input supply using a sense resistor
in series with the inductor (or by using
inductor DCR sensing). The inductor
current is constantly monitored and no
blanking is required, enabling it to achieve
very low bottom MOSFET duty cycles with
a very small 110ns minimum on-time.
design features
VOUT1
RB1
357k
RA1
68.1k
VFB1
LTC3859
SENSE1–
C1
1nF
CITH1A
100pF
SENSE1+
RITH1
15k
CITH1
1500pF
CSS1
0.1µF
ITH1
PGOOD1
FREQ
PLLIN/MODE
SW1
RUN1
RA2
68.1k
RB2
649k
10pF
RUN3
VOUT1
5V
5A
COUT1
220µF
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
2.2nF
CINT1
1µF
RITH2
15k
ITH2
CITH2A
68pF
CINT2
4.7µF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
6.5µH
RSENSE2
8mΩ
SW2
VOUT3
RB3
499k
RA3
68.1k
COUT2
68µF
MBOT2
BG2
VOUT2
8.5V
3A
VFB3
CITH3
0.01µF
RITH3
3.6k
SENSE2+
C2
1nF
ITH3
CITH3A
820pF
SENSE2–
CSS3
0.1µF
VOUT3
10V*
D3
SS3
VOUT2
EXTVCC
MTOP3
TG3
SW3
BOOST3
CB3
0.1µF
BG3
MTOP1, MTOP2: BSZ097NO4LS
MBOT1, MBOT2: BSZ097NO4LS
MTOP3: BSC027NO4LS
MBOT3: BSCO1BN04LS
L1: WÜRTH 744314490
L2: WÜRTH 744314650
RSENSE1
6mΩ
MBOT1
BG1
RUN2
VOUT2
L1
4.9µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
MTOP1
TG1
TRACK/SS1
Figure 1. Typical automotive
application using the LTC3859
100k
L3: WÜRTH 744325120
COUT1: SANYO 6TPB220ML
COUT2: SANYO 10TPC68M
CIN, COUT3: SANYO 50CE220LX
D1, D2: CMDH-4E
D3: BAS140W
In a boost converter, the duty cycle gets
smaller as the input voltage approaches
the programmed output voltage and
equals 0% when VIN = VOUT. Traditional
non-synchronous boost controllers
that sense the bottom FET current do
not smoothly handle the transition as
VIN approaches the programmed VOUT,
often having excessive, unpredictable, low
frequency ripple that begins when the minimum on-time is reached. Most of those
controllers have relatively long minimum
on-times (often greater than 200ns), which
means that high ripple can occur over a
relatively wide band of input voltages.
MBOT3
SENSE3–
C3
1nF
SENSE3+
In contrast, the LTC3859 boost controller gracefully handles the transition as
VIN moves up or down through the programmed output voltage without creating
excessive ripple. Because of the small minimum on-time, constant frequency operation is maintained until VIN is just below
VOUT, at which point the part skips bottom
FET on cycles as needed until it is off continuously (0% duty cycle) and the synchronous top FET is on continuously (100%
duty cycle). Unlike most boost converters,
the LTC3859’s ripple during this transition
region in substantially smaller than it is
at lower VIN during “normal” boosting.
L3
1.2µH
RSENSE2
2mΩ
COUT3
220µF
VIN
2.5V TO 38V
(START-UP ABOVE 5V)
CIN
220µF
* VOUT3 IS 10V WHEN VIN < 10V,
FOLLOWS VIN WHEN VIN > 10V
The LTC3859 is able to keep the synchronous MOSFET on continuously by integrating a small charge pump inside its driver.
This charge pump maintains the voltage
on the bootstrap capacitor that serves as
the floating supply (BOOST3-SW3 voltage) for the top driver. Otherwise, the
voltage on this capacitor might decay
due to board or diode leakage current.
DUAL BUCK CONTROLLERS
Along with the single boost controller, the LTC3859 also integrates a pair of
synchronous buck (step-down) controllers based on the LTC3857/58 family of
low quiescent buck controllers. They
April 2011 : LT Journal of Analog Innovation | 27
drive all N-channel MOSFETs and feature
a precision 0.8V reference. They accept
inputs up to 38V (40V abs max) and the
outputs can be programmed between
0.8V to 24V (28V abs max). The 95ns
minimum on-time allows high frequency operation at low duty cycles.
OTHER FEATURES
The LTC3859 shares many of the same
popular features of the LTC3788 and
LTC3857 families on which it was based.
The MOSFET drivers and control circuits
are powered by INTVCC , which by default
is generated from an internal low dropout
(LDO) regulator from the main bias supply
pin (VBIAS). To reduce power dissipation
due to MOSFET gate charge losses and
improve efficiency, a supply between
5V and 14V (abs max) may be connected to
the EXTVCC pin. When a supply is detected
on EXTVCC , the VBIAS LDO is disabled
and another LDO between EXTVCC and
INTVCC is enabled. EXTVCC is commonly
connected to one of the output voltages generated by the buck controllers.
The switching frequency can be programmed between 50kHz and 900kHz
using the FREQ pin, or synchronized
via the PLLIN/MODE pin to an external clock between 75kHz and 850kHz
using an integrated phase-locked loop.
The buck controllers (channels 1 and
2) operate 180° out-of-phase to minimize the capacitance required on their
All outputs have independent enable
(RUN1,2,3) and soft-start (TRACK/SS1,2 and
SS3 pins). The TRACK/SS pins on the buck
controllers can also be used to track other
supplies during start-up. The PGOOD1 and
OV3 are open-drain pins that respectively
indicate whether buck channel 1 is in regulation and whether the boost channel is
in overvoltage (VIN > programmed VOUT +
10%). Protection features include shortcircuit and overvoltage protection for the
bucks and overtemperature protection. At
light loads, the user can select from three
modes of operation—Burst Mode operation, pulse-skipping mode, or forced continuous mode—using the PLLIN/MODE pin.
5V AND 8.5V OUTPUTS
FROM AUTOMOTIVE BATTERY,
EVEN DURING COLD CRANK
LOW I Q FOR ALWAYS-ON SYSTEMS
The LTC3859 VBIAS pin is powered from
the output of the boost converter. The
EXTVCC pin is connected to the 8.5V (or
alternatively the 5V supply) to improve efficiency, particularly at high battery voltage.
When Burst Mode operation is selected,
the LTC3859 features an ultralow operating quiescent current (55µ A with one
buck on, 65µ A with one buck and the
boost on, or 80µ A with all three channels on). This makes the LTC3859 ideal
for always-on systems, where one or
more outputs are always enabled and
low quiescent current is required to
extend run-times and preserve battery
life. Automobiles have an increasing
number of these systems (regardless of
whether they also have start-stop systems)
that remain on even when the vehicle is
parked for days or weeks. Examples of
VOUT2 = 8.5V
95
80
EFFICIENCY (%)
80
75
70
65
60
ILOAD = 2A
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
35
40
Figure 2. Efficiency vs input voltage for Figure 1.
28 | April 2011 : LT Journal of Analog Innovation
1
70
60
50
0.1
POWER LOSS (W)
85
55
10
90
VOUT1 = 5V
90
EFFICIENCY (%)
these include telematics systems, antitheft systems, and keyless-entry systems.
100
100
50
input. The boost controller (channel 3)
operates in phase with channel 1.
FCM EFFICIENCY
10m
PULSE-SKIPPING
EFFICIENCY
BURST LOSS
20
BURST EFFICIENCY 1m
FCM LOSS
10
PULSE-SKIPPING
LOSS
0.1m
0
10m
1
10
0.1m
1m
0.1
OUTPUT CURRENT (A)
VIN = 10V, VOUT = 5V
40
30
Figure 3. Efficiency and power loss vs load current
of 5V output for Figure 1.
Figure 1 shows a highly integrated solution that utilizes the unique features of
the LTC3859 to efficiently solve the design
challenges associated with automotive
start-stop and always-on systems. In this
circuit, the boost controller input is connected directly to the car battery, and the
boost output, which is programmed to
10V, serves as the input to the two buck
controllers, which generate the 5V and
8.5V outputs. The 5V supply might typically
be used to power an always-on system
and the 8.5V supply for a DVD player.
Normally, the battery sits around 12V–14V,
so the input to the boost converter is
higher than its programmed 10V output.
Under these conditions, the control loop
forces the top MOSFET on continuously.
The internal charge pump maintains the
supply voltage (BOOST3-SW3) for the top
MOSFET driver (TG3 to ensure 100% duty
cycle operation). With the top MOSFET on
continuously, the boost converter simply
passes the battery voltage directly through
to the buck inputs, minimizing power loss.
During engine start-up, when the battery voltage can dip to 5V or lower, the
boost converter starts switching when
the battery voltage drops below 10V and
keeps the buck inputs pinned at 10V. This
prevents the buck converters from ever
going into dropout, allowing the bucks
to maintain output regulation at 5V and
8.5V although the car battery can fall
below these voltages. The LTC3859 boost
controller’s very low 2.5V input common mode range allows for a regulated
design features
VOUT1
RB1
28.7k
56pF
RA1
115k
CITH1A
200pF
LTC3859
SENSE1–
C1
1nF
SENSE1+
RITH1
3.93k
CITH1
1000pF
VFB1
ITH1
CSS1
0.1µF
PGOOD1
FREQ
PLLIN/MODE
SW1
RUN1
RB2
57.6k
56pF
RA2
115k
RUN3
VOUT1
1V
8A
COUT1
220µF
×2
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
1000pF
CINT1
1µF
RITH2
3.93k
ITH2
CITH2A
200pF
CINT2
4.7µF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
0.47µH
RSENSE2
3.5mΩ
VOUT2
1.2V
8A
SW2
VOUT3
RB3
232k
RA3
12.1k
COUT2
220µF
×2
MBOT2
BG2
VFB3
CITH3
15nF
RITH3
8.66k
SENSE2+
SENSE2–
CSS3
0.1µF
D3
SS3
TG3
EXTVCC
SW3
MTOP1, MTOP2: RENESAS RJK0305
MBOT1, MBOT2: RENESAS RJK0328
MTOP3, MBOT3: RENESAS HAT2169H
L1, L2: SUMIDA CDEP105-0R4
L3: PULSE PA1494.362NL
COUT1, COUT2: SANYO 2R5TPE220M
CIN, COUT3: SANYO 50CE220AX
D1, D2: CMDH-4E
D3: BAS140W
boost output voltage, and thus stable
buck output voltages, even through some
of the harshest cold crank transients.
C2
1nF
ITH3
CITH3A
220pF
Figure 2 shows the total efficiency at
2A load with VIN spanning from 2.5V to
38V. The low quiescent current allows
the 5V supply to remain on at all times
without significantly deteriorating the
vehicle battery life. Figure 3 shows
the efficiency and power loss of the
5V output over a broad load range.
RSENSE1
3.5mΩ
MBOT1
BG1
RUN2
VOUT2
L1
0.47µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
MTOP1
TG1
TRACK/SS1
Figure 4. General purpose triple
output application using the
LTC3859
100k
BOOST3
MTOP3
CB3
0.1µF
BG3
MBOT3
L3
3.3µH
RSENSE2
4mΩ
COUT3
220µF
VOUT3
24V
5A
VIN
12V
CIN
220µF
SENSE3–
C3
1nF
SENSE3+
GENERAL PURPOSE
TRIPLE OUTPUT CONTROLLER
As impressive as the LTC3859 is in
these dual buck-boost applications, it
of course can also be configured as a
simple triple output converter. Figure 4
shows a circuit generating 24V, 1V and
1.2V outputs from a 12V input.
CONCLUSION
The LTC3859 is a low IQ triple output
controller that offers a compelling, compact solution to the demanding design
challenges in modern automotive electronics. Configuring the synchronous boost
controller in front of the two synchronous
buck controllers provides dual supply
voltages that maintain regulation over
the entire voltage range of the car battery.
This makes the LTC3859 ideal for high
efficiency power conversion in alwayson and start-stop systems. The LTC3859
packs all of this and more into small,
thermally enhanced 38-pin 5mm × 7mm
QFN or 38-lead TSSOP packages. n
April 2011 : LT Journal of Analog Innovation | 29
How to Drive Low Power, 1Msps, ±2.5V Differential-Input,
16-Bit ADC with a Variety of Single-Ended Signals
Guy Hoover
It can be difficult to match a sensor’s output to an ADC
input range, especially when faced with the variety of
output voltage swings produced by today’s sensors. This
article presents easy, but high performance, ADC input
driver solutions for differential, single-ended, unipolar and
bipolar signals covering a variety of spans. All circuits in
this article achieve 92dB SNR, with the LTC2383-16 ADC
acting alone or in conjunction with the LT6350 ADC driver.
The LTC2383-16 is a low noise, low power,
1Msps , 16-bit ADC with a ±2.5V fully
differential input range. The LT6350 is a
rail-to-rail input and output low noise,
low power single-ended to differential
converter/ADC driver featuring fast settling
time. Using the LT6350, single-ended input
ranges of 0V–2.5V, 0V–5V and ±10V can be
easily converted to the ±2.5V fully differential input range of the LTC2383-16.
reduce the effect of the ADC input
sampling spike that can disturb the
sensor or ADC driver outputs.
Figure 1 shows the building block that
is used for all of the circuits described
here. It serves a DC-coupled fully differential signal to the LTC2383-16 analog
inputs. Resistors R1, R2 and capacitor C1
limit the input bandwidth to approximately 500kHz. Resistors R3 and R4
R3
100Ω
C1
3300pF
NPO
AIN–
0V TO 2.5V
R2
49.9Ω
R4
100Ω
R5
1k
C2
10µF
AIN+
0V TO 2.5V
R1
49.9Ω
R3
100Ω
C1
3300pF
NPO
AIN–
0V TO 2.5V
This circuit is useful for sensors with
low impedance differential outputs.
The common mode voltage driving
AIN+ and AIN– needs to be VREF /2 to
satisfy the common mode input range
requirements of the LTC2383-16.
R2
R6 49.9Ω
1k
C3
10µF
R4
100Ω
component choice is essential for maintaining performance. All of the resistors
used in these circuits are relatively low
values. This keeps the noise and settling
time low. Metal film resistors are recommended to reduce distortion caused by
self-heating. An NPO capacitor is used
for C1 because of its low voltage coefficient, which minimizes distortion.
Figure 4. FFT of circuit of Figure 3
0V to 2.5V
C2
15pF
NPO
+
–
–IN1
RINT
+IN2
VCM = VREF/2
OUT1
R1
49.9Ω
RINT
–
+
+
–
30 | April 2011 : LT Journal of Analog Innovation
–40
R3
100Ω
C1
3300pF
NPO
OUT2
V–
–5V
R2
49.9Ω
2.5V to
0V
R4
100Ω
IN+
LTC2383-16
IN–
AMPLITUDE (dB)
VIN
V
fS = 1MHz
fIN = 20kHz
SNR = 92dB
THD = –107dB
–20
5V
SHDN
LTC2383-16
IN–
Figure 2. AC-coupled fully differential drive circuit
0
LT6350
IN+
VCM
Figure 3. Single-ended to differential converter
0V to
2.5V
LTC2383-16
IN–
VCM
When driving a low noise, low distortion ADC such as the LTC2383-16, proper
+
IN+
Figure 1. Fully differential drive circuit
The circuit in Figure 1 can be AC-coupled
to match the common mode voltage of the
ADC input to the sensor if necessary. Simply
bias AIN+ and AIN– to VCM (VCM = VREF /2)
through a 1k resistor and couple the sensor output to AIN+ and AIN– through a
10µ F capacitor. This is shown in Figure 2.
FULLY DIFFERENTIAL DRIVE
R1
49.9Ω
AIN+
0V TO 2.5V
–60
–80
–100
–120
–140
–160
0
100
200
300
fIN (kHz)
400
500
design ideas
5V
VREF
R9
2.49k
C4
150pF
NPO
LT6350
VIN
C3
10µF
R10
4.99k
V+
+
–
R8
1.65k
–IN1
0V to
2.5V
SHDN
OUT1
RINT
RINT
+IN2
–
+
R5
2.49k
R7
4.99k
+
–
R3
100Ω
C1
3300pF
NPO
OUT2
V–
R9
499Ω
0V to 5V
R1
49.9Ω
–5V
R2
49.9Ω
R4
100Ω
IN+
LTC2383-16
IN–
2.5V to
0V
Of course, not all sensor outputs are differential. Here are some ways to drive the
LTC2383-16 from single-ended signals.
0V–2.5V Single-Ended Input
The circuit of Figure 3 converts a singleended 0V-to-2.5V signal to a fully differential ±2.5V signal. This circuit also has a
high impedance input so that most sensor
outputs should be able to drive it directly.
The common mode voltage at VIN can be
matched to the ADC by AC-coupling VIN as
shown in Figure 2. The common mode
voltage of the second amplifier is set at
the +IN2 pin of the LT6350. The 32k-point
FFT in Figure 4 shows the performance
of the LTC2383-16 combined with the
LT6350 using the circuit shown in Figure 3.
The measured SNR of 92dB and THD of
–107dB match closely with the typical
data sheet specs for the LTC2383-16. This
indicates that little, if any, degradation
of the ADC’s specifications result from
inserting the single-ended to differential converter into the signal path.
0V–5V Single-Ended Input
If a wider input range is required, the
minus input of the LT6350 can be driven,
allowing the input voltage to be attenuated
by the first stage of the LT6350. The circuit
of Figure 5 converts a single-ended 0V to
5V signal to a differential ±2.5V signal that
drives the inputs of the LTC2383-16. The
input impedance of this circuit is equal to
R7. Increasing R7 results in higher input
impedance, making it easier to drive. This
is done at the expense of slightly increased
Figure 6. A ±10V single-ended driver
VIN
C3
10µF
R10
4.99k
R8
549Ω
V+
LT6350
+
–
–IN1
R7
4.99k
R5
619Ω
SNR
(dB)
THD
(dB)
2
1
0.665
92
–100
4.99
2.49
1.65
92
–101
10
4.99
3.32
91
–100
49.9
24.9
16.5
91
–97
100
49.9
33.2
91
–94
noise and distortion if R7 is increased
above 4.99k, as shown in Table 1.
±10V Single-Ended Input
Some sensors provide an output voltage that goes above and below ground.
The circuit of Figure 6 converts a
±10V ground-referred single-ended signal
to a differential ±2.5V signal that drives
the inputs of the LTC2383-16. Again, the
input impedance is set by R7. Table 2
shows noise and distortion vs input
impedance for the circuit of Figure 6.
CONCLUSION
The LTC2383-16 is a low power, low
noise, 16-bit ADC that can be easily interfaced with a wide variety of
sensor outputs, including unipolar,
bipolar, differential and single-ended
signals over a wide range of spans. n
Table 2. Noise and distortion vs input resistance for
±10V driver
5V
R9
619Ω
R8
(kΩ)
Table 1. Noise and distortion vs input resistance for
0V-to-5V driver
Figure 5. A 0V-to-5V single-ended driver
SINGLE-ENDED TO
DIFFERENTIAL CONVERSION
VREF/2
R5,R9
(kΩ)
VCM = VREF/2
C2
75pF, NPO
C4
200pF
NPO
R7,R10
(kΩ)
0V to
2.5V
SHDN
OUT1
RINT
R1
49.9Ω
RINT
–
+
+IN2
+
–
VCM = VREF/2
R3
100Ω
C1
3300pF
NPO
OUT2
V–
–5V
R2
49.9Ω
2.5V to
0V
R4
100Ω
IN+
LTC2383-16
IN–
R7,R10
(kΩ)
R5,R9
(kΩ)
R8
(kΩ)
SNR
(dB)
THD
(dB)
3.24
0.402
0.357
92
–97
4.99
0.619
0.549
92
–97
10
1.24
1.10
92
–96
49.9
6.19
5.49
91
–96
100
12.4
11.0
91
–97
–10V to 10V
C2
200pF, NPO
April 2011 : LT Journal of Analog Innovation | 31
Easy, Isolated Low Power Telecom Supply:
No Opto-Isolator Required
Mayur Kenia
Isolated power delivery in telecom systems is traditionally a challenging design problem. High
input voltages and isolation requirements lead to complex schemes involving complicated
magnetics and opto-couplers in the feedback loop. The LT3511 and LT3512 monolithic
switching regulators bring simplicity to isolated power supplies with a non-traditional
approach. Figure 1 shows the significant difference in complexity between traditional
designs versus a new simplified approach. The LT3511 and LT3512 are specifically
designed for the isolated flyback topology with no third winding or opto-isolator required for
regulation—the isolated output voltage is sensed directly from the primary-side waveform.
INTEGRATION
4.5V to 100V. In addition, internal isolated
sensing circuitry allows programming
of the output voltage with two external
resistors and the transformer turns ratio.
An accurate internal threshold offers
a programmable undervoltage lockout
threshold using the EN/UV pin. Finally,
a resistor from the TC pin to ground
A high level of integration simplifies the
overall application solution. The LT3511
features an integrated 240m A, 150V power
switch while the LT3512 features a 420m A,
150V power switch. A pre-regulator is
integrated in both parts, allowing a
wide input operating voltage range from
VIN+
18V TO 72V
VOUT+
•
VIN–
provides adjustable temperature compensation to compensate for the temperature coefficient of the output diode.
48V-TO-5V ISOLATED POWER WITH
EXCELLENT LOAD REGULATION
Figure 2 shows an isolated 5V application,
from a 36V-to-72V input, using the LT3511.
A Zener in series with a diode placed
VIN
36V TO 72V
•
EN/UVLO
GND
VOUT–
LT3511
VCC
Before
After
•
ITH/RUN
GND
VFB
VOUT+
VIN
RFB
RREF
SW
TC
VC
GND
BIAS
GATE
VCC
VOUT+
SENSE
VCC
VIN
OPTO
GND COMP
OC
32 | April 2011 : LT Journal of Analog Innovation
FB
VOUT+
Figure 1. The traditionally complex approach
to isolated power supply design gives way to
a new, simpler way using the LT3511/LT3512.
design ideas
The LT3511/LT3512 infers the isolated output voltage by
examining the primary-side flyback pulse waveform. In this
manner, neither an opto-isolator nor an extra transformer
winding is required to maintain regulation, and the output
voltage is easily programmed with two resistors.
4:1:1
C1
1µF
R1
1M
Z1
VIN
R2
43.2k
EN/UVLO
LT3511
TC
RFB
RREF
D3
R3
169k
T1
300µH
R5
69.8k
GND
BIAS
R6
16.9k
C2
3.3nF
D2
5.15
5.10
VIN = 48V
5.05
VIN = 36V
5.00
VIN = 72V
4.95
4.90
4.85
4.80
4.75
L1C
19µH
C3
4.7µF
5.20
C4
22µF
19µH
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM32ER71C226KE18B
D1, D2: DIODES INC. SBR140S3
D3: DIODES INC. BAV19W
T1: WÜRTH 750311558
Z1: ON SEMI MMSZ5266BT1G
R4
10k
5.25
VOUT+
5V
0.3A
VOUT–
SW
VC
D1
VOUT (V)
VIN
36V TO 72V
0
50
150
200
100
LOAD CURRENT (mA)
250
300
Figure 3. Load regulation for application in Figure 2
OPTIONAL THIRD
WINDING IMPROVES
HV EFFICIENCY
HIGH VOLTAGE PIN SPACING
Figure 2. A 48V-to-5V isolated flyback converter using the LT3511 requires remarkably few components
24V-TO-5V ISOLATED POWER
from the SW pin to the VIN pin ensures that
the switch node stays below 150V across
all operating conditions and transients.
The application produces excellent load
regulation across its input voltage range.
Figure 3 shows load regulation at 36V,
48V and 72V at the input. The LT3512 is
pin-compatible with the LT3511 and delivers 500m A at 5V with a similar scheme.
Figure 4 shows the LT3512 generating an
isolated 5V from 24V at VIN . The lower
input voltage allows the use of a larger
turns ratio. The application can deliver
450m A of output current, while the LT3511
can be used in a similar configuration
to deliver 250m A of output current.
Figure 4. A 24V to 5V isolated flyback converter using the LT3512
VIN
20V TO 30V
C1
4.7µF
6:1
R1
1M
Z1
VIN
R2
80.6k
EN/UVLO
LT3512
SW
TC
VC
R5
69.8k
RFB
RREF
GND
R6
6.49k
C2
4.7nF
BIAS
C3
4.7µF
R3
249k
R4
10k
D2
T1
200µH
D1
5.5µH
The LT3511 and LT3512 are available in an MS16 package with pins
removed to provide adequate spacing for high voltage operation.
Figure 5 shows the pins removed.
CONCLUSION
The LT3511 and LT3512 provide a simple, elegant solution to isolated power
delivery. A high level of feature integration minimizes the number of external
components and lowers the overall
solution cost. Most importantly, the
LT3511 and LT3512 ease the design of
isolated high voltage power supplies. n
VOUT+
5V
0.45A
C4
47µF
Figure 5. LT3511/LT3512 MS16 high voltage pin spacing
VOUT–
C1: TAIYO YUDEN UMK316AB7475KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: TAIYO YUDEN LMK32587476MM-TR
D1: DIODES INC. SBR2A30P1
D2: DIODES INC. DFLS1100
T1: SUMIDA 10396-T027
Z1: ON SEMI MMSZ5270BT1G
TOP VIEW
EN/UVLO 1
16 SW
VIN 3
14 RFB
GND
BIAS
NC
GND
5
6
7
8
12
11
10
9
RREF
TC
VC
GND
MS PACKAGE
16(12)-LEAD PLASTIC MSOP
April 2011 : LT Journal of Analog Innovation | 33
4mm × 5mm, Dual Input/Output, Synchronous Monolithic
Buck Regulator Converts 12V to 1.2V at 4MHz
Phil Juang
Systems that are powered from a battery stack or a single
12V supply typically incorporate a number of relatively low
power point-of-load power supplies. These supplies are
typically low enough power that using DC/DC controllers with
external power MOSFETs would make them unnecessarily
large and complex. In such applications, a monolithic DC/DC
converter can save significant space and design time.
The LTC3633 is a dual output 3A/channel synchronous monolithic step-down
converter capable of operating from
input supply voltages anywhere between
3.6V and 15V. It uses a patented controlled on-time architecture, which
allows very large step-down ratios at
high switching frequencies and provides
extremely fast transient response.
shift between channels. Synchronous
switching eliminates external Schottky
diodes and maintains efficiencies above
90% over a wide range of input and
load conditions. An internal compensation option further simplifies designs. A
10m A linear regulator provides a fixed
2.5V output that can be used as a bias
voltage or a low power supply rail.
The LTC3633 also includes a number of
important features, including high efficiency Burst Mode operation, resistorprogrammable switching frequency,
external clock synchronization, output
tracking capability, programmable input
undervoltage lockout, short-circuit protection and selectable 0°/180° degrees phase
The LTC3633 is offered in a tiny, thermally enhanced 4mm × 5mm 28-lead
QFN package as well as a thermally
enhanced 28-lead TSSOP package.
SHORT ON-TIMES ALLOW FAST
SWITCHING FREQUENCIES AND
SMALLER COMPONENT SIZE
One limitation of many current mode
control schemes for step-down converters is the minimum on-time of the main
power switch, typically in the range of
60ns–100ns. The minimum on-time is
the amount of time needed to accurately
sense the peak inductor current through
the main power MOSFET, commonly used
in current-mode control architectures.
Unfortunately, the minimum on-time
constrains the duty cycle and maximum
switching frequency. This is typically not a
problem when the duty cycle is more than
25%, which is usually the case if the input
voltage is 5V or less, but when using a
12V input supply or a Li-ion battery stack,
output voltages of 1.8V and 2.5V often run
into the minimum on-time constraint at
high switching frequencies. In this case,
the only remedy is to operate at a lower
Figure 2. The LTC3633 configured as a 6A single-output buck converter
VIN
3.6V TO 15V
47µF
×2
RUN1
RUN2
Figure 1. Switch waveform with 30ns on-time
ITH2
VIN2
VIN1
V2P5
PHMODE
MODE/SYNC
10pF
(OPT)
IL
1A/DIV
ITH2
680pF
VIN = 12V
VOUT = 1.2V
100ns/DIV
fSW = 4MHz
ILOAD = 3A
34 | April 2011 : LT Journal of Analog Innovation
L2
0.47µH
0.1µF
L1
0.47µH
SW1
SW1
VON1
VON2
RT
R5
162k
0.1µF
BOOST1
10pF
(OPT)
SW
5V/DIV
2.2µF
BOOST2
LTC3633
3.01k
INTVCC
SGND PGND
VFB1
VFB2
10k
20k
VOUT
1.8V AT 6A
COUT
47µF
×2
design ideas
The LTC3633 control loop solves this
problem by sensing the inductor current during the off-time, which allows
the minimum on-time to be extremely
low (20ns typical) and frees the switching converter of the minimum duty cycle
constraint. Although this control scheme
imposes a minimum off-time constraint
and limits the maximum duty cycle of
the converter, the off-time constraint is
often less critical when operating from
high input supply voltages. The result
is a step-down converter that can easily
handle extremely low duty cycles while
operating well above 1MHz. Figure 1
shows a scope capture of the LTC3633
configured to step down from 12V to
1.2V with a switching frequency of 4MHz.
The ability to operate at such high
frequencies is advantageous for spaceconstrained designs which require smaller
bypass capacitors as well as applications designed to operate the switching
converter above the AM band to prevent
electromagnetic interference (EMI).
UNIQUE CONTROL ARCHITECTURE
PROVIDES SHORT ON-TIMES
AND CONTROLLED SWITCHING
FREQUENCY
One commonly used method of sensing
inductor current during the off-time is
“constant on-time” current-mode control,
in which the top switch is forced on for a
fixed period of time. One drawback of this
method is that the switching frequency
can change with input/output voltage,
temperature and load current conditions.
This is a problem in systems that require
a well controlled switching frequency
to avoid EMI or satisfy other noise concerns. It also precludes synchronizing the
switch edges to a known clock signal.
Linear Technology’s patented “controlled
on-time” architecture solves this problem
by incorporating a phase-locked loop that
servos the on-time to match the switching frequency to a known clock signal.
Thus, the LTC3633 steady-state switching
frequency is constant over all temperature
and load current conditions. A program
resistor can be used to set the frequency of
an on-chip oscillator or the LTC3633 can be
synchronized to an external clock signal.
DUAL OUTPUTS AND INPUTS
The LTC3633’s dual inputs and outputs
allow it to meet the specifications of
a wide variety of power supplies. For
instance, the two outputs can be paralleled to form a single output capable
of sourcing 6A, as shown in Figure 2.
Connecting the ITH pins together forces
the LTC3633 to share the load current
equally between channels. For lower input
and output voltage ripple, the PHASE pin
can set the converters to switch 180°
out-of-phase. Although only one compensation network is needed to stabilize the
converter, optional 10pF capacitors are
recommended to bypass each ITH pin and
prevent parasitic board capacitances from
injecting noise into the ITH signal path.
The LTC3633’s two separate inputs allow
each step-down channel to draw power
from different supplies. Multiple inputs
can also be used when the outputs are
tied together, as in Figure 2. This is useful
when a single supply rail is not sufficient
to satisfy the output power requirements.
When operating from different supplies,
it is important to note that VIN1 must
always be powered, as it supplies the gate
drive for the internal power MOSFETs.
THERMALLY ENHANCED PACKAGES
REDUCE HEAT DISSIPATION
When running at the maximum allowable output loads (3A per channel), heat
dissipation must be managed properly. The
LTC3633 is offered in thermally enhanced
packages—at extremely high power levels,
the PCB should be designed to sink as
much heat as possible by tying the exposed
3.5
CHANNEL 1 LOAD CURRENT (A)
switching frequency, which typically
requires larger input and output capacitors to meet voltage ripple requirements.
3.0
2.5
2.0
1.5
1.0
CH2 LOAD = 0A
CH2 LOAD = 1A
CH2 LOAD = 2A
CH2 LOAD = 3A
0.5
0
0
25
75
100
50
MAXIMUM ALLOWABLE AMBIENT
TEMPERATURE (°C)
125
Figure 3. Thermal derating curve for LTC3633 demo
circuit (DC1347)
pad of the package to a ground plane and
flooding any unused areas of the PCB with
copper tied to ground. A top layer ground
plane is preferred, but if it is not available, the PCB should use as many vias as
possible to connect to the ground plane.
At high ambient temperatures, full
output loads can cause excessive heating
of the die. Although the LTC3633 uses a
thermal shutdown circuit to prevent the
die temperature from exceeding 160°C,
prolonged temperatures above 125°C may
impact long-term reliability of the IC.
Figure 4 shows the thermal derating curve
for the tiny 4mm × 5mm QFN package,
as measured on the LTC3633 demo circuit
(DC1347). Although larger in size, the
TSSOP package’s thermal performance is
nearly 40% better than the QFN, allowing
the designer to trade off improved thermal performance for more board space.
CONCLUSION
With its unique control loop that allows
extremely low duty cycles, the LTC3633 is
well suited to operate from a wide range
of input supply voltages and provides two
efficient low voltage power supplies at
the point of load. It offers a multitude of
features, including the ability to parallel
the outputs and handle inputs from different supply voltages. It is easy to use, with
the basic application requiring only seven
external components for each channel. n
April 2011 : LT Journal of Analog Innovation | 35
Isolated Flyback Converters Eliminate Opto-Coupler
Yat Tam
The flyback converter is widely used in isolated DC/DC applications because it solves
the problem of isolation, not because it is the favorite topology of switch-mode
power supply designers. Traditionally, flyback converters demand careful attention to
a number of parameters, including complex trade-offs involving transformer design,
control loop analysis and power device selection. Fortunately, this is not the case
with Linear’s family of breakthrough monolithic ICs—including the LT3573, LT3574
and LT3575—which make it easy to build flyback converters (see Table 1).
MSOP package for LT3573 and LT3574,
or 16-lead TSSOP package for LT3575.
SENSING OUTPUT VOLTAGE
WITHOUT OPTO-COUPLERS
Traditional flyback converters normally
require opto-couplers or an additional
transformer winding to feed back output voltage information used to maintain load regulation across the isolation
barrier. Opto-couplers consume output
power and obviously add to the cost and
physical size of the design. In addition,
opto-couplers introduce nonlinearities,
VIN
12V TO 24V
10µF
357k
0.22µF
VIN
T1
3:1
2k
50µH
SHDN/UVLO
PMEG6010
51.1k
LT3574
RFB
B340A
•
5.6µH
•
80.6k
RREF
TC
6.04k
RILIM
SS
SW
VC
28.7k
GND
TEST BIAS
10nF
4.7µF
1nF
T1: PULSE PA3019NL OR WÜRTH ELEKTRONIK 750370041
Figure 1. A 12V–24V input, 5V/350mA output flyback converter
36 | April 2011 : LT Journal of Analog Innovation
Figure 1 shows how a typical LT3574
application performs without an optocoupler or extra transformer winding.
Figure 2 shows the efficiency of the
application in Figure 1. Figure 3 shows
an alternate configuration that employs
90
VOUT–
VIN = 24V
80
VIN = 12V
70
60
50
40
30
20
59k
10k
VOUT+
5V
0.35A
22µF
and have unit-to-unit variations and
aging over life that complicates loop
compensation. Opto-couplers also
require a secondary side voltage reference and error amplifier, adding two or
three packages to the layout. Circuits
that employ extra transformer windings
can increase transformer size and cost,
sacrifice the transient response performance and degrade load regulation.
EFFICIENCY (%)
The LT3574 operates over input ranges of
3V to 40V at output power levels of up to
3W, and can be used in a wide variety of
industrial, medical, datacom and automotive applications requiring isolated power.
The LT3573 and LT3575 are higher power
versions, extending the output power
delivery up to 7W and 14W, respectively.
This family of ICs packs many popular
features—such as programmable soft
start, undervoltage lockout, adjustable
current limit and output voltage temperature compensation—in a 16-lead
10
0
0
100
200 300 400 500
LOAD CURRENT (mA)
600
Figure 2. Efficiency of the circuit in Figure 1
700
design ideas
an optional third winding on the BIAS pin
for higher input voltage operation. Use
the third winding to boost efficiency
by 3%–4% at high input voltages.
The LT3574 derives its information about
the output voltage by examining the
primary-side flyback pulse waveform. The
output voltage, programmed by resistors R3 and R4, is accurately measured
from the switch node waveform during the off-time of the power switch.
BOUNDARY MODE SAVES SPACE
AND IMPROVES LOAD REGULATION
The LT3573, LT3574 and LT3575 utilize
boundary mode operation, which is a
variable frequency current mode switching scheme. When the internal switch
turns on, the primary current increases
VIN
12V TO 24V
(30V*)
V IN RANGE
POWER SWITCH
POWER LEVEL
PACKAGE
LT3573
3V to 40V
1.25A, 60V
Up to 7W
16-Lead MSOP
(thermally enhanced)
LT3574
3V to 40V
0.65A, 60V
Up to 3W
16-Lead MSOP
LT3575
3V to 40V
2.5A, 60V
Up to 14W
16-Lead TSSOP
(thermally enhanced)
until its controlled current limit is reached.
The switch node voltage rises above the
input by an amount equal to the output
voltage divided by the secondary-toprimary turns ratio. When the secondary
current through the diode falls to zero,
the switch node voltage falls back to the
input voltage. An internal discontinuous conduction mode (DCM) comparator
detects this latter event and turns the
switch back on and repeats the cycle.
Boundary mode returns the secondary
current to zero every cycle, so parasitic
resistive voltage drops do not cause
load regulation errors. This results in a
tight regulation band over a wide input
voltage range and output load current
range, as shown in Figure 4. Here, total
regulation is better than ±1%, while
3:1:1
C1
10µF
R1
499k
R2
71.5k
R8
1k
C6
0.22µF
VIN
SHDN/UVLO
T1
24µH
D1
2.6µH
VOUT +
5V, 1.4A
R3
80.6k
RFB
RREF
1.0
R4
6.04k
SW
SS
VC
R6
28.7k
R5
10k
C2
10nF
GND TEST BIAS
R7
11.5k
C3
4700pF
D1: PDS835L
D2: CMDSH-3
D3: PMEG6010
The LT3573, LT3574 and LT3575 significantly simplify the design of an isolated
flyback DC/DC converter by eliminating
the need for an opto-coupler, external
power switch, secondary-side reference
voltage and error amplifier, and extra
third winding off the power transformer.
They provide space saving, cost efficient
solutions without sacrificing overall
performance, while maintaining primary
to secondary isolation with only one
component crossing the isolation barrier.
This family of ICs operates from 3V to
40V input and delivers up to 3W, 7W and
14W of continuous output power. n
VOUT –
TC
RILIM
CONCLUSION
C5
47µF
D3
LT3575
±5% load regulation is easily achievable
for most applications. Boundary mode
permits the use of a smaller transformer
compared to continuous conduction
mode (CCM) and eliminates concerns
related to subharmonic oscillation.
D2
C4
4.7µF
L1C
2.6µH
* OPTIONAL THIRD
WINDING FOR
30V OPERATION
LOAD REGULATION (%)
Table 1. This family of flyback converters eliminates
the need for an opto-coupler.
PRODUCT
VIN = 12V
0
–0.5
–1.0
C5: MURATA, GRM32ER71A476K
T1: PULSE PA2454NL OR
WÜRTH ELEKTRONIK 750310471/750311675
Figure 3. A 12V–24V input, 5V/1.4A output flyback converter
VIN = 24V
0.5
0
100
200 300 400 500
LOAD CURRENT (mA)
600
700
Figure 4. Tight load regulation is achieved from the
circuit in Figure 1.
April 2011 : LT Journal of Analog Innovation | 37
Buck-Boost Converter with Accurate Input Current Limit
Maximizes Power Utilization from USB and PCMCIA Sources
Michael Munroe
Input power source current limitations, such as those
imposed by USB or PCMCIA, pose a problem for
applications demanding high peak power—such as charging
capacitors to support the pulsed load currents required by
GSM modems. The LTC3127 buck-boost DC/DC converter
simplifies powering such applications with an accurate
programmable input current limit. High accuracy (±4% at
1A) allows the designer to push the current draw to just
below the capability of the input source. This capability,
along with the LTC3127’s high efficiency over a wide input
voltage range, maximizes the available output current. The
LTC3127 also eliminates inrush current during start-up,
maintaining control of the current seen by the input supply
when low ESR output reservoir capacitors are charged.
The LTC3127 is offered in either a
3mm × 3mm × 0.75mm DFN or 12-lead
MSOP package (see Figure 1). The 1.35MHz
switching frequency and integrated low
resistance, low gate charge switches
produce an efficient, compact and low
profile solution for pulsed load applications. Additional features include
Figure 2. USB (5.0V/500mA max) to a 3.8V
output, Class 10 GPRS pulsed load
anti-ringing control for EMI suppression, selectable Burst Mode operation,
output disconnect and thermal overload
protection. Together with an external
bulk or reservoir capacitor, the LTC3127
can satisfy pulsed loads with peak currents that far exceed the capabilities of
the source, without overloading it.
SW2
VOUT
C1, C2, C3: VISHAY TANTAMOUNT
TANTALUM, LOW ESR CAPACITORS
L1: COILCRAFT XPL4020-472ML
VOUT
3.8V
LTC3127
OFF ON
SHDN
10µF
PROG
SGND
32.4k
The LTC3127 solves the problem of using
current-limited power sources to drive
high current pulsed load applications or
to charge high density ultra-capacitors.
For instance, a USB or PCMCIA powered
GSM/GPRS modem typically requires a
3.8V bias to produce a transmission
modulated to 2A pulses, which occupy one
IIN
500mA/DIV
2.15M
MODE
PWM BURST
POWERING GSM/GPRS MODEMS
FROM USB OR PCMCIA
Figure 3. Waveforms of input current and VOUT for a
pulsed load current of circuit in Figure 2
L1
4.7µH
SW1
VIN
VIN
3.3V OR
5V
9mm
Figure 1. A complete, low profile power converter
solution for pulsed load applications
FB
VC
PGND
100pF
C3
2.2mF
1M
499k
VOUT
200mV/DIV
C2
2.2mF
C1
2.2mF
ILOAD
1A/DIV
1ms/DIV
38 | April 2011 : LT Journal of Analog Innovation
design ideas
or more of the eight available 577µs time
slots. During the remaining time slots,
the required load is reduced to less than
100m A, typically. Producing 2A directly
from the DC/DC converter would overload the input source, since the pulsed
load would draw well in excess of the
500m A guaranteed by USB and PCMCIA.
The solution is to use the DC/DC to charge
output capacitors to store energy that
can be released as needed to satisfy the
modem’s pulsed-load current draw.
capacitors supply energy to the load and
maintain the output voltage within specified limits during the 2A pulses. Using the
LTC3127 configured with proportional gain
provides loop stability with any output
capacitor value greater than 1000µF.
Figure 2 shows the LTC3127 powered
from a standard 5.0V/500m A USB port.
The 500m A input current limit is set by
the 32.4k resistor tied to the PROG pin.
Given the magnitude and duration of
the pulsed load current, the capacitors
must be chosen to meet the output voltage droop specification of the transmitter, typically 300mV. The total output
voltage droop can be calculated by:
The advantage of using the LTC3127
with its high accuracy input current
limit, is that the current draw from
the limited source can be maximized.
More available input power results in
faster charging, which allows the output
capacitors to be minimally sized, reducing overall solution cost and size.
VDROOP =


 VIN • IIN(MAX ) • η
  D • T
+ RESR 
− ISTANDBY  • 
IPULSE − 
VOUT



  COUT
For a GPRS Class 10 standard, a 2A load
pulse is on for two 577µs cycles with the
transmitter idle for the remaining six
cycles. In Figure 2, the supply must be able
to handle a pulse of 2A for 1.15ms out of
every 4.6ms, for an average load current
of 575m A. With properly sized output
capacitors, operating from a 5V input, the
LTC3127 can supply this load current at
3.8V without exceeding an input current
of 500m A for USB2.0/PCMCIA applications or 900m A for USB3.0 applications.
where VDROOP is the change in output
voltage, IPULSE is the pulsed load current, VIN is the input voltage, IIN(MAX)
is the programmed input current limit,
η is the fractional converter efficiency
(η = 1 is 100% efficiency), VOUT is the
programmed output voltage, ISTANDBY is
the idle output current of the converter
between pulses, D is the duty cycle of
the pulsed load, T is the period of the
pulsed load, COUT and RESR are the total
output capacitance and capacitor’s
equivalent series resistance, respectively.
In Figure 2, three 2200µF, low profile,
Vishay TANTAMOUNT solid tantalum
For a given pulsed load application,
regardless of how much capacitance is
Figure 4. High density capacitor charger
SW1
VIN
PWM BURST
MODE
SW2
VOUT
SHDN
10µF
SGND
FB
VC
PROG
32.4k
VOUT
5V
374k
LTC3127
OFF ON
ILOAD(MAX ) =
VIN • IIN(MAX ) • η
D • VOUT
The minimum capacitance required
for a desired VOUT droop during
the load pulse (assuming that
RESR = 0) can be calculated by:
COUT(MIN) =

 D • T
 VIN • IIN(MAX ) • η
− ISTANDBY  •
IPULSE − 
VOUT

 VDROOP

The typical pulsed load response for the
circuit in Figure 2 is shown in Figure 3.
CHARGING HIGH DENSITY
CAPACITORS
When charging high density ultra-capacitors for backup purposes, the converter
operates at full current for minutes, even
hours, depending on the size of the ultracapacitor. The LTC3127 accurately limits
the input current to the programmed value
through the entire charging cycle. Once
charged, the ultra-capacitors can provide power to other circuitry in the event
of a power source failure or removal.
Figures 4 and 5 show the schematic
and the response of the LTC3127 charging stacked 400F, 2.5V supercapacitors
to a VOUT of 5.0V with a USB input.
(continued on page 43)
Figure 5. Waveforms of input current and VOUT—charging a
high density capacitor from 0V using circuit of Figure 4.
L1
4.7µH
VIN
4V to 5.5V
on the output of the converter, there
is a maximum average load that can
be supported for a given input current. The maximum pulsed load that
the converter can support with a programmed input current limit is given by:
100pF
C1
400F
IIN
500mA/DIV
121k
C2
400F
PGND
499k
VOUT
2V/DIV
SHDN
5V/DIV
L1: COILCRAFT XPL4020-472ML
500s/DIV
April 2011 : LT Journal of Analog Innovation | 39
Low IQ, Dual Output Step-Down Controller
Converts 60V Directly to 3.3V
Jason Leonard and Joe Panganiban
The LTC3890 is a versatile low quiescent current, 2-phase dual output synchronous
step-down DC/DC controller ideal for high input voltage applications. It operates from a
4V-to-60V (65V abs max) input supply and regulates two outputs ranging from 0.8V to 24V.
Its 50µA no-load quiescent current extends operating life in battery-powered systems.
Many high step-down-ratio DC/DC converter designs use a transformer-based
topology, external high side drivers,
and/or external bias supplies to operate at high input voltages. Others
require 2-stage conversion because of
duty cycle or on-time limitations. The
LTC3890, however, trumps them all with
an easy to use, high performance solution that fits in a small footprint.
The LTC3890 uses a proven synchronous
buck DC/DC converter topology, but with a
very low 95ns minimum on-time, enabling
it to directly convert high input voltages
to low output voltages. It has integrated
high and low side N-channel MOSFET drivers and an integrated LDO, which operates
directly from VIN, to power the drivers and
the IC. To improve efficiency and minimize
power dissipation in the IC, the EXTVCC pin
can be used to bypass the LDO at VIN, thus
allowing the IC to derive power from one
of the outputs after it has started up.
FEATURES
The LTC3890 uses a constant frequency
current mode control architecture for fast
transient response and easy loop compensation. The two channels run 180°
out-of-phase. The switching frequency can
be programmed from 50kHz to 900kHz
or synchronized to an external clock
from 75kHz to 850kHz with the internally compensated phase-locked loop.
40 | April 2011 : LT Journal of Analog Innovation
RB1
100k
RA1
31.6k
C1
1nF
SENSE1+
INTVCC
SENSE1–
PGOOD1
VFB1
PGOOD2
CITH1A 100pF
CITH1 1000pF
100k
BG1
RITH1
34.8k
MBOT1
SW1
LTC3890
TRACK/SS1
ILIM
PHASMD
CLKOUT
PLLIN/MODE
VOUT2
RFREQ
41.2k
SGND
EXTVCC
RUN1
RUN2
FREQ
CSS2 0.01µF
TRACK/SS2
RITH2
34.8k
ITH2
RA2
10.5k
VFB2
RB2
100k
L1
4.7µH
BOOST1
ITH1
CSS1 0.01µF
CITH2 470pF
100k
TG1
CB1
0.1µF
RSENSE1
8mΩ
COUT1
470µF
MTOP1
D1
VIN
INTVCC
CIN
220µF
CINT
4.7µF
PGND
VOUT1
3.3V
5A
VIN
4.2V TO 60V
D2
TG2
BOOST2
CB2 0.1µF
MTOP2
L2
8µH
SW2
BG2
MBOT2
RSENSE2
10mΩ
VOUT2
8.5V
3A
COUT2
330µF
SENSE2–
C2
1nF
SENSE2+
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-472KL
L2: COILCRAFT SER1360-802KL
COUT1: SANYO 6TPE470M
COUT2: SANYO 10TPE330M
D1, D2: DFLS1100
Figure 1. High efficiency dual output 3.3V/8.5V step-down converter operates from 4.2V to 60V inputs and can
operate down to 4V after start-up. The 99% maximum duty cycle allows the 8.5V output to follow VIN when VIN
is less than 8.5V.
design ideas
The LTC3890 uses a venerable synchronous buck DC/DC
converter topology, but with a with a very low 95ns minimum
on-time, allowing it to directly convert high input voltages
to low output voltages. It has integrated high and low side
N-channel MOSFET drivers and an integrated LDO, which
operates directly from VIN to power the drivers and the IC.
VOUT
100mV/DIV
(AC-COUPLED)
VOUT
100mV/DIV
(AC-COUPLED)
VOUT
100mV/DIV
(AC-COUPLED)
IL
2A/DIV
IL
2A/DIV
IL
2A/DIV
50µs/DIV
VIN = 12V
VOUT = 3.3V
Burst Mode OPERATION
50µs/DIV
VIN = 12V
VOUT = 3.3V
PULSE-SKIPPING MODE
50µs/DIV
VIN = 12V
VOUT = 3.3V
FORCED CONTINUOUS MODE
Figure 2. Transient response of Figure 1 for each of the three different modes of operation. At light load, the inductor current is allowed to reverse in forced continuous
operation. Burst Mode operation has slightly higher ripple to achieve its much higher efficiency.
The low 95ns minimum on-time allows
the LTC3890 to easily handle high stepdown ratios (high VIN to low VOUT), even
at high frequencies. High frequency
capability enables the use of small inductors and capacitors, reducing cost and
footprint over slower controllers. The
LTC3890 can also run up to 99% duty
cycle, providing low dropout when the
input falls close to the output in high
VOUT applications. Adjustable soft-start/
tracking input and enable pins are provided for each channel. At light loads,
one of three modes—forced continuous,
pulse-skipping, or Burst Mode operation—
can be selected with the PLLIN/MODE pin
to trade off constant frequency operation,
output ripple, and quiescent current.
The LTC3890 is the latest in Linear
Technology’s growing family of low
IQ DC/DC controllers. Its closest relative
is the pin-compatible LTC3857, which
shares the same core features. The main
differences are that the LTC3890 can
handle a higher input voltage (65V versus
40V maximum) and the LTC3890’s Burst
Mode operation significantly improves
mid-range efficiency while maintaining
very low no-load quiescent current.
LOW I Q, HIGH VOLTAGE DUAL
OUTPUT CONVERTER
VOUT2
2V/DIV
VOUT2
2V/DIV
VOUT1
2V/DIV
2ms/DIV
VOUT1
2V/DIV
2ms/DIV
Figure 3. A soft-start capacitor on the TRACK/SS pin can program the ramp time (left), or a resistor divider
from one output to the TRACK/SS pin of the other channel can enable the output voltages to track one
another during start-up (right).
Figure 1 shows the LTC3890 in an application that converts a 4V to 60V input into
3.3V/5A and 8.5V/3A outputs. This circuit
needs about 4.2V to get started but can
operate down to below 4V. The transient responses for the three modes of
operation are shown in Figure 2. Figure 3
shows start-up waveforms using standard soft-start and voltage rail tracking.
April 2011 : LT Journal of Analog Innovation | 41
100
90
VIN = 12V
90 VOUT = 3.3V
VOUT = 3.3V
80
EFFICIENCY (%)
60
50
40
30
10
70
FCM LOSS
0.1
60
50
BURST LOSS
PULSE-SKIPPING
LOSS
40
10m
30
FCM EFFICIENCY
20
Burst Mode OPERATION
VIN = 12V
0
0.0001 0.001
0.01
0.1
1
OUTPUT CURRENT (A)
1
80
70
20
10
BURST EFFICIENCY
PULSE-SKIPPING
EFFICIENCY
10
0
0.1m
10
1m
10m
0.1
1
OUTPUT CURRENT (A)
POWER LOSS (W)
EFFICIENCY (%)
100
VOUT = 8.5V
1m
0.1m
10
Figure 4. Efficiency in Burst Mode operation for
the two outputs. Due to its low IQ in Burst Mode
operation, the LTC3890 maintains high efficiency and
low power loss operation down to very light loads.
Figure 5. Efficiency and power loss for the three
modes of operation. At heavy loads, there is no
difference.
LOW I Q AND HIGH EFFICIENCY
BURST MODE EXTENDS BATTERY
RUN-TIME
informative. Note the significantly lower
(by orders of magnitude) power loss in
Burst Mode operation at very light load.
In many applications, one or more supplies remain active at all times, often in a
standby mode where little or no load current is drawn. In these always-on systems,
the quiescent current of the power supply
circuit represents the vast majority of
the current drawn from the input supply
(battery). Having a low IQ power supply
is crucial to extending battery run-times.
In Burst Mode operation, the LTC3890
draws only 50µ A when one output is
active with no load, and only 60µ A when
both outputs are enabled. It consumes
only 14µ A when both outputs are shut
down. Figures 4 and 5 show the efficiency
curves for the circuit in Figure 1. At light
loads, the power loss curves are most
MULTIPHASE SINGLE OUTPUT
APPLICATIONS
The LTC3890 is normally configured for
two independent outputs that run 180°
out-of-phase. Operating the channels
out-of-phase minimizes the required
input capacitance. The LTC3890 can
be configured with both power stages
driving a single output. In this configuration, both channels’ compensation (ITH), feedback (VFB), enable (RUN)
and soft-start (TRACK/SS or SS) pins are
tied together. Since the channels operate out-of-phase, the effective switching
frequency is doubled, reducing not only
the input but also the output capacitance,
while improving transient response. The
LTC3890 provides inherently fast, accurate
cycle-by-cycle current sharing due to its
peak current mode control architecture.
Multiple LTC3890s can be used in designs
with three or more phases. The CLKOUT pin
can drive the PLLIN/MODE pin of other
controllers, while the PHASMD pin adjusts
the relative phases of each controller. This
allows 3-, 4-, 6-, and 12-phase operation.
VERSIONS
The LTC3890 is available in two versions.
The standard LTC3890 is the full-featured
version available in a small 32-pin
5mm × 5mm QFN package. The LTC3890-1
has slightly fewer features and is offered
in a 28-lead narrow SSOP package. These
differences are summarized in Table 1.
CONCLUSION
The LTC3890 brings a new level of performance to high voltage step-down converters. This low quiescent current, 2-phase,
dual output DC/DC controller enables a
highly efficient, easy to design, compact
solution for power supplies with a wide
range of input and output voltages. n
Table 1. Key differences between the LTC3890 and LTC3890-1
LTC3890
LTC3890-1
CURRENT SENSE VOLTAGE
(SENSE RESISTOR OR INDUCTOR DCR SENSING)
Adjustable 30mV/50mV/75mV
(Set via ILIM pin)
Fixed 75mV
POWER GOOD OUTPUT VOLTAGE MONITOR
Independent Monitors for Each Channel
(PGOOD1 and PGOOD2 pins)
Monitor for Channel 1 Only (PGOOD1 Pin)
CLKOUT/PHASMD PINS FOR THREE OR MORE PHASES
Yes
No
PACKAGE
5mm × 5mm QFN
28-Lead Narrow SSOP
42 | April 2011 : LT Journal of Analog Innovation
product briefs
Product Briefs
16-BIT 125Msps ADCs
REDUCE POWER TO 185mW
Three new families of low power
16-bit, 25Msps to 125Msps ADCs dissipate
approximately half the power of competing 16-bit solutions. The LTC2165 and
LTC2185 families are single- and twochannel simultaneous-sampling parallel
ADCs, respectively, offering a choice of
full-rate CMOS, or double data rate (DDR)
CMOS/LVDS digital outputs with programmable digital output timing, programmable LVDS output current and optional
LVDS output termination. The LTC2195
family includes 2-channel, simultaneous
sampling ADCs with serial LVDS outputs.
Each ADC family offers a choice of pincompatible converters, sampling from
25Msps up to 125Msps and optimized
for the lowest power dissipation at the
rated speed. They include such popular
features as Linear Technology’s digital
output randomizer and alternate bit
polarity (ABP) mode that minimize digital feedback. These low power 16-bit
ADCs enable designers to upgrade performance while maintaining portability in
such applications as handheld test and
instrumentation, radar/LIDAR, portable
medical imaging, PET/SPECT scanners,
smart antenna systems and a variety of
low power communication systems.
The dual LTC2185/LTC2195 and single
LTC2165 consume 185mW/channel at
125Msps and offer signal to noise ratio
(SNR) performance of 76.8dB and SFDR of
90dB at baseband. Pin-compatible speed
grade options include 25Msps, 40Msps,
65Msps, 80Msps and 105Msps with approximate power dissipation of just 1.5mW/Msps
per channel. Further power savings can
be achieved by placing the devices in
standby (20mW) or shutdown (1mW).
Analog full power bandwidth of 550MHz
and ultralow jitter of 0.07psRMS allows
undersampling of IF frequencies with
excellent noise performance. n
Table 1. Complete family of 16-bit parallel and serial interface ADCs
SINGLE
CHANNEL
DUAL
CHANNEL
POWER
25Msps
40Msps
65Msps
80Msps
105Msps
125Msps
7 × 7 QFN
1.8V Single ADCs, Parallel Outputs
2160
2161
2162
2163
2164
2165
9 × 9 QFN
1.8V Dual ADCs Parallel Outputs
2180
2181
2182
2183
2184
2185
7 × 8 QFN
1.8V Dual ADCs, Serial LVDS Outputs
2190
2191
2192
2193
2194
2195
(mW/Ch)
40
60
80
100
155
185
(LTC3127, continued from page 39)
CONCLUSION
The compact LTC3127 1.0A buck-boost
DC/DC converter with ±4% accurate,
programmable average input current
limit is an optimal power supply solution for charging capacitors that supply
energy in pulsed load or emergency load
applications. The accuracy of the LTC3127
allows the designer to set the input current
limit close to the source’s maximum, thus
minimizing capacitor charge time, and
by extension, minimizing capacitor size
and cost. The LTC3127’s high efficiency
through all operating modes makes it
an ideal fit for a wide variety of power
sources including PCMCIA and USB. These
features, when combined with low profile
supercapacitors, elegantly solve the pulsed
load problem in a compact footprint. n
April 2011 : LT Journal of Analog Innovation | 43
highlights from circuits.linear.com
CCOMP*
2-TERMINAL 3A CURRENT SOURCE
The LT3083 is a 3A low dropout linear regulator that can be paralleled to increase output current or spread heat on surface-mounted
boards. Architected as a precision current source and voltage
follower, this new regulator finds use in many applications requiring
high current, adjustability to zero with no heat sink. The device
brings out the collector of the pass transistor to allow low dropout
operation down to 310mV when used with multiple supplies.
www.linear.com/3083
IN
*CCOMP
R1 ≤ 10Ω 10µF
R1 ≥ 10Ω 2.2µF
LT3083
VCONTROL
IOUT =
+
–
1V
R1
R1
SET
20k
VIN
4.5V TO 36V
3.3V STEP-DOWN CONVERTER
The LT3690 is an adjustable frequency monolithic buck switching regulator
that accepts input voltages up to 36V. A high efficiency 90mΩ switch is
included on the device along with the boost diode and the necessary oscillator, control, and logic circuitry. The internal synchronous power switch
of 30mΩ increases efficiency and eliminates the need for an external
Schottky catch diode. The low ripple Burst Mode operation maintains high
efficiency at low output currents, reducing quiescent current to less than
75uA while keeping output ripple below 15mV in typical applications.
www.linear.com/3690
BIAS
VIN
10µF
UVLO
ON OFF
PG
EN
0.68µF
LT3690
SS
BST
VC
1nF
0.47µF
316k
FB
VCCINT
22k
L
3.3µH
SW
RT
SYNC
GND
680pF
VOUT
3.3V
4A
100µF
40.2k
102k
ƒ = 500kHz
(FIXED FREQUENCY AT VIN < 31V)
BACKPLANE RESIDENT DIODE-OR
APPLICATION WITH INRUSH CURRENT
LIMITING AT 12V SUPPLY INPUTS
The LTC4227 offers ideal diode and
Hot Swap™ functions for two power
rails by controlling external N-channel
MOSFETs. MOSFETs acting as ideal
diodes replace two high power
Schottky diodes and the associated
heat sinks, saving power and board
area. A Hot Swap control MOSFET
allows a board to be safely inserted
and removed from a live backplane
by limiting inrush current. The supply
output is also protected against shortcircuit faults with a fast acting current
limit and internal timed circuit breaker.
www.linear.com/4227
MD1
Si7336ADP
VIN1
12V
BULK
SUPPLY
BYPASS
CAPACITOR
VIN2
12V
CCP1
0.1µF
MD2
Si7336ADP
RS
0.008Ω
MH
Si7336ADP
+
BULK
SUPPLY
BYPASS
CAPACITOR
RH
10Ω
CCP2
0.1µF
12V
5A
CL
1000µF
RHG
47Ω
CHG
15nF
R2
137k
R1
20k
CPO1
CF
10nF
IN1 DGATE1 CPO2
IN2 DGATE2 SENSE+
SENSE– HGATE
OUT
FAULT
PWRGD
ON
LTC4227
INTVCC
D2ON
C1
0.1µF
GND
EN
TMR
CT
0.1µF
BACKPLANE
PLUG-IN
CARD
L, LT, LTC, LTM, Linear Technology, the Linear logo, Linear Express, Burst Mode, TimerBlox and µModule are registered trademarks, and Hot Swap, and Virtual Remote Sense are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2011 Linear Technology Corporation/Printed in U.S.A./50.1K
Linear Technology Corporation
1630 McCarthy Boulevard, Milpitas, CA 95035
(408) 432-1900
www.linear.com
Cert no. SW-COC-001530