Application Note AN-1144 IRS20957S Functional Description By Jun Honda, Xiao-chang Cheng Table of Contents Floating PWM Input .................................................................................................................................................. 2 Over-Current Protection (OCP) ................................................................................................................................ 3 Protection Control ..................................................................................................................................................... 5 Self Reset Protection................................................................................................................................................ 5 Designing Ct.............................................................................................................................................................. 5 Shutdown Input......................................................................................................................................................... 6 Latched Protection.................................................................................................................................................... 6 Interfacing with System Controller............................................................................................................................ 6 Programming OCP Trip Level .................................................................................................................................. 7 Low-side Over-Current Sensing ............................................................................................................................... 7 Low-Side Over-Current Setting ................................................................................................................................ 8 High-Side Over-Current Sensing.............................................................................................................................. 9 High-Side Over-Current Setting.............................................................................................................................. 10 Choosing the Right Reverse Blocking Diode ......................................................................................................... 10 Deadtime Generator ............................................................................................................................................... 11 How to Determine Optimal Deadtime..................................................................................................................... 11 Programming Deadtime.......................................................................................................................................... 11 Supplying VDD ......................................................................................................................................................... 12 Charging VBS Prior to Start ..................................................................................................................................... 13 Start-up Sequence (UVLO) .................................................................................................................................... 14 Power-down Sequence .......................................................................................................................................... 14 Power Supply Decoupling ...................................................................................................................................... 14 VSS Negative Bias Clamping................................................................................................................................... 14 Junction Temperature Estimation........................................................................................................................... 15 www.irf.com AN-1144 1 IRS20957S General Description Note: The IRS20957S is an improved version of the IRS20955S. The IRS20955S is no longer recommended for new designs. For details, refer to application note AN-1141, IRS20955S and IRS20957S Comparison. The IRS20957 is a high voltage, high speed MOSFET driver with a floating PWM input designed for Class D audio amplifier applications. Bi-directional current sensing detects over current conditions during positive and negative load currents without any external shunt resistors. A built-in protection control block provides a secure protection sequence against over-current conditions and a programmable reset timer. The internal dead-time generation block enables accurate gate switching and optimum dead-time setting for better audio performance, such as lower THD and lower audio noise floor. For the convenience of half bridge configuration, the PWM input and protection logic are constructed on a floating well. Typical Implementation The following explanations are based on a typical application circuit with self-oscillating PWM topology shown in Figure 1. For further information, refer to the IRAUDAMP4 reference design. Figure 1. IRS20957 Typical Application Circuit Floating PWM Input The IRS20957 accepts floating inputs, enabling easy half-bridge implementation. VDD, CSD and IN refer to VSS. As a result, the PWM input signal can directly feed into IN while referencing VSS, which is typically the midpoint between the positive and negative DC bus voltages in a half-bridge configuration. The IRS20957 also accepts a non-floating input when VSS is tied to COM. www.irf.com AN-1144 2 VDD HV LEVEL SHIFT CSD 10.2V IN PROTECTION VSS Floating Input Isolation Floating Bias 0V – 200V COM Figure 2. Floating PWM Input Structure Over-Current Protection (OCP) The IRS20957 features over-current protection to protect the power MOSFETs during abnormal load conditions. The IRS20957 starts a sequence of events when it detects an over-current condition during either high-side or low-side turn on of a pulse. As soon as either the high-side or low-side current sensing block detects over-current: 1. The OC Latch (OCL) flips logic states and shutdowns the outputs LO and HO. 2. The CSD pin starts discharging the external capacitor Ct. 3. When VCSD, the voltage across Ct, falls below the lower threshold Vth2, an output signal from COMP2 resets OCL. 4. The CSD pin starts charging the external capacitor Ct. 5. When VCSD goes above the upper threshold Vth1, the logic on COMP1 flips and the IC resumes operation. As long as the over-current condition exists, the IC will repeat the over-current protection sequence at a repetition rate dependent upon capacitance in CSD pin. www.irf.com AN-1144 3 Figure 3. Over-Current Protection Timing Chart VDD Vth1 ` COMP1 CSD OC S Q UVLO(VB) ` COMP2 Ct R OC DET (H) Vth2 VSS HV LEVEL SHIFT FLOATING INPUT HV LEVEL SHIFT HV LEVEL SHIFT FLOATING HIGH SIDE LOW SIDE OC DET (L) UVLO(VCC) SD PWM HO DEAD TIME ` LO Figure 4. Shutdown Functional Block Diagram www.irf.com AN-1144 4 Protection Control The internal protection control block dictates the operational mode, normal, or shutdown, using the input of the CSD pin. In shutdown mode, the IC forces LO and HO to output 0V with respect to COM and VS respectively to turn off the power MOSFETs. The CSD pin provides five functions. 1. Power up delay timer 2. Self-reset timer 3. Shutdown input 4. Latched protection configuration 5. Shutdown status output (host I/F) Self Reset Protection By putting a capacitor between CSD and VSS, the IRS20957 resets itself after entering the shutdown mode. Figure 5. Self Reset Protection Configuration Designing Ct The timing capacitor, Ct, is used to program tRESET and tSU. • tRESET, is the amount of time that elapses from when the IC enters the shutdown mode to the time when the IC resumes operation. tRESET should be long enough to avoid over heating the MOSFET from the repetitive sequence of shutting down and resuming operation during over-current conditions. In most applications, the minimum recommended time for tRESET is 0.1 second. • tSU is the amount of time between powering up the IC in the shutdown mode to the moment the IC releases shutdown to begin normal operation. The values chosen for tRESET and tSU will determine the capacitance of Ct using the given equations: The Ct determines tRESET and tSU as following equations: t RESET = t SU = Ct ⋅ VDD 1.1 ⋅ I CSD Ct ⋅ VDD 0.7 ⋅ I CSD [s] [s] where ICSD = the charge/discharge current at the CSD pin www.irf.com AN-1144 5 VDD = the floating input supply voltage with respect to VSS. Shutdown Input The IRS20957 can be shut down by an external shutdown signal SD. Figure 6 shows how to add an external discharging path to shutdown the PWM. Figure 6. Shutdown Input Latched Protection Connecting CSD to VDD through a 10k Ω or less resistor configures the over-current protection latch. The latch locks the IC in shutdown mode after over-current is detected. An external reset switch can be used to bring CSD below the lower threshold Vth2 for a minimum of 200 ns to properly reset the latch. After the power up sequence, a reset signal to the CSD pin is required to release the IC from the latched shutdown mode. <10k SD 1 VDD CSH 16 2 CSD VB 15 3 IN HO 14 4 VSS VS 13 5 NC NC 12 VCC 11 LO 10 COM 9 6 VREF 7 OCSET 8 DT Figure 7. Latched Protection Configuration Interfacing with System Controller The IRS20957 can communicate with an external system controller through a simple interfacing circuit shown in Figure 8. A generic PNP transistor U1 detects the sink current at the CSD pin during an OCP event and outputs a shutdown signal to an external system controller. Another generic NPN transistor U2 can then reset the internal protection logic by pulling the CSD voltage below the lower threshold Vth2 for a minimum of 200 ns. Note that the CSD pin is configured to operate in latched OCP. After the power up sequence, a reset signal to the CSD pin is required to release the IC from the shutdown mode. www.irf.com AN-1144 6 U1 SD <10k RESET 1 VDD CSH 16 2 CSD VB 15 3 IN HO 14 4 VSS VS 13 5 NC NC 12 6 VREF VCC 11 7 OCSET LO 10 8 DT COM 9 U2 Figure 8. Interfacing with Host Controller Programming OCP Trip Level In a Class D audio amplifier, the direction of the load current alternates with the audio input signal. An overcurrent condition can therefore occur during either a positive current cycle or a negative current cycle. The IRS20957 uses the RDS(ON) of the output MOSFETs as current sensing resistors. Due to the structural constraints of high voltage ICs, current sensing is implemented differently for high side and low side. If the measured current exceeds a predetermined threshold, the OCP block outputs a signal to the protection block, focing HO and LO low and protecting the MOSFETs. D1 R2 +B CSH UV DETECT R1 VB UV HIGH SIDE CS Q Dbs R3 HO Q1 Cbs OUT HV LEVEL SHIFT FLOATING HIGH SIDE 5V REG VS HV LEVEL SHIFT Vcc VCC UV DETECT DEAD TIME Q2 LO SD -B COM R5 LOW SIDE CS OCSET R4 VREF Figure 9. Bi-Directional Over-Current Protection Low-side Over-Current Sensing www.irf.com AN-1144 7 For negative load currents, low-side over-current sensing monitors the load condition and shuts down switching operation if the load current exceeds the preset trip level. Low-side current sensing is based on the measurement of VDS across the low side MOSFET during low-side turn on. In order to avoid triggering OCP from overshoot, a blanking interval inserted after LO turn on disables overcurrent detection for 450 ns. The OCSET pin is used to program the threshold for low-side over-current sensing. When the VDS measured across the low-side MOSFET exceeds the voltage at the OCSET pin with respect to COM, the IRS20957 begins the OCP sequence described earlier. Note that programmable OCSET range is 0.5V to 5.0V. To disable low side OCP, connect OCSET to VCC directly. To program the trip level for over current, the voltage at OCSET can be calculated using the equation below. VOCSET = VDS(LOW SIDE) = ITRIP+ x RDS(ON) In order to minimize the effect of the input bias current at the OCSET pin, select resistor values for R4 and R5 such that the current through the voltage divider is 0.5 mA or more. * Note: Using VREF to generate an input to OCSET through a resistive divider provides improved immunity from fluctuations in VCC. +B Q1 OC REF OCREF 5.1V R4 R5 OUT VS 0.5mA - OCSET OC + OC Comparator COM LO LO Q2 IRS20957 -B Figure 10. Low-Side Over-Current Sensing Low-Side Over-Current Setting Assume that the low side MOSFET has RDS(on) of 100mΩ. VOCSET to set the current trip level at 30A is given by: VOCSET = ITRIP+ x RDS(ON) = 30 A x 100 mΩ = 3.0 V Choose R4+R5=10 kΩ to properly load the VREF pin. R5 = = VOCSET ⋅ 10 kΩ VREF 3.0 V ⋅ 10 kΩ 5.1 V = 5.8 kΩ where VREF = 5.1 V Based on the E-12 series of resistor values, choose R5 to be 5.6 kΩ and R4 to be 3.9 kΩ to complete the design. www.irf.com AN-1144 8 In general, RDS(ON) has a positive temperature coefficient that needs to be considered when setting the threshold level. Also, variations in RDS(ON) will affect the selection of external or internal component values. High-Side Over-Current Sensing For positive load currents, high-side over-current sensing also monitors the load condition and shuts down the switching operation if the load current exceeds the preset trip level. High-side current sensing is based on the measurement of VDS across the high-side MOSFET during high-side turn on through pins CSH and VS. In order to avoid triggering OCP from overshoot, a blanking interval inserted after HO turn on disables over-current detection for 450 ns. In contrast to low-side current sensing, the threshold at which the CSH pin engages OC protection is internally fixed at 1.2V. An external resistive divider R2 and R3 can be used to program a higher threshold. An external reverse blocking diode, D1, is required to block high voltages from feeding into the CSH pin while the high-side is off. Due to a forward voltage drop of 0.6V across D1, the minimum threshold required for high-side over-current protection is 0.6V. VCSH = R3 ⋅ (V DS ( HIGHSIDE ) + V F ( D1) ) R 2 + R3 where VDS(HIGH SIDE) = the drain to source voltage of the high-side MOSFET during high-side turn on VF(D1) = the forward drop voltage of D1 Since VDS(HIGH SIDE) is determined by the product of drain current ID and RDS(ON) of the high-side MOSFET. VCSH can be rewritten as: VCSH = R3 ⋅ (RDS ( ON ) ⋅ I D + VF ( D1) ) R 2 + R3 The reverse blocking diode D1 is forward biased by a 10 kΩ resistor R1. www.irf.com AN-1144 9 Figure 1. Programming High-Side Over-Current Threshold High-Side Over-Current Setting Figure 11 demonstrates the typical circuitry used for high-side current sensing. In the following example, the over-current protection level is set to trip at 30A using a MOSFET with an RDS(ON) of 100 mΩ. The component values of R2 and R3 can be calculated using the following formula: Let R2 + R3=10 kΩ. VthOCH R3 = 10 kΩ ⋅ VDS + VF where Vth,OCL = 1.2V VF = the forward voltage of reverse blocking diode D1 = 0.6V. VDS@ID=30A = the voltage drop across the high-side MOSFET when the MOSFET current is 30 A. Therefore, VDS@ID=30A = ID x RDS(ON) = 30A x 100 mΩ = 3V Based on the formulas above, R2 = 6.8 kΩ and R3 = 3.3 kΩ. Choosing the Right Reverse Blocking Diode The selection of the appropriate reverse blocking diode D1 depends on its voltage rating and speed. To effectively block bus voltages, the reverse voltage must be higher than the voltage difference between +B and -B and the reverse recovery time must be as fast as the bootstrap charging diode. A diode such as the NXP BAV21 W, a 200V, 50 ns high-speed switching diode, is more than sufficient. www.irf.com AN-1144 10 Deadtime Generator Deadtime is the blanking period inserted between either high-side Turn-OFF and low-side TurnON, or low-side Turn-OFF and high-side Turn-ON. Its purpose is to prevent shoot through, or a rush of current through both MOSFETs. In the IRS20924(S), an internal deadtime generation block allows the user to select the optimum deadtime from a range of preset values. Selecting a preset deadtime through the DT pin voltage can easily be done through an external voltage divider. This way of setting deadtime prevents outside noise from modulating the switching timing, which is critical to the audio performance. How to Determine Optimal Deadtime The effective deadtime in an actual application differs from the deadtime specified in this datasheet due to the switching fall time, tf.. The deadtime value in this datasheet is defined as the time period between the beginning of turn-off on one side of the switching stage and the beginning of turn-on on the other side as shown in Figure 12. The fall time of the MOSFET gate voltage must be subtracted from the deadtime value in the datasheet to determine the effective deadtime of a Class D audio amplifier. (Effective deadtime) = (Deadtime in datasheet) – tf. 90% HO (or LO) Effective dead - time 10% tf LO (or HO) Dead-time in datasheet 10% Figure 12. Effective Deadtime A longer deadtime period is required for a MOSFET with a larger gate charge value because of the longer tf.. Although a shorter effective deadtime setting is beneficial to achieving better linearity in Class D amplifiers, the likelihood of shoot-through current increases with narrower dead-time settings. Negative values of effective dead-time may cause excessive heat dissipation in the MOSFETs, leading to potentially serious damage. To calculate the optimal deadtime in a given application, the fall time tf for both HO and LO in the actual circuit needs to be taken into account. In addition, variations in temperature and device parameters could also affect the effective deadtime in the actual circuit. Therefore, a minimum effective deadtime of 10 ns is recommended to avoid shoot-through current over the range of operating temperatures and supply voltages. Programming Deadtime www.irf.com AN-1144 11 The IRS20957 selects the deadtime from a range of preset deadtime values based on the voltage applied at the DT pin. An internal comparator translates the DT input to a predetermined deadtime by comparing the input with internal reference voltages. These internal reference voltages are set in the IC through a resistive voltage divider using VCC. The relationship between the operation mode and the voltage at DT pin is illustrated in the Figure13 below. Dead- time 15nS 25nS 35nS 45nS 0.23 xVcc 0.36 xVcc 0.57 xVcc Vcc VDT Figure 2. Deadtime vs. VDT Table 1 suggests pairs of resistor values used in the voltage divider for selecting deadtime. Resistors with up to 5% tolerance are acceptable when using these values. IRS20957 >0.5mA Vcc R1 DT R2 COM Figure 3. External Voltage Divider Table 1 Recommended Resistor Values for Deadtime Selection Deadtime Mode DT1 DT2 DT3 DT4 R1 <10 kΩ 5.6 kΩ 8.2 kΩ Open R2 Open 4.7 kΩ 3.3 kΩ <10 kΩ DT Voltage VCC 0.46(VCC) 0.29(VCC) COM Supplying VDD VDD is designed to be supplied with an internal Zener diode clamp. IDD, the supply current for VDD, can be estimated by: IDD ≈ 1.5 mA x 300 x 10-9 x switching frequency + 0.5 mA + 0.5 mA (Dynamic power consumption) (Static) (Zener bias) The value of RDD used to supply IDD should meet the following requirement: www.irf.com AN-1144 12 RDD ≤ V+ B − 10.2 V I DD [Ω] Example: In the case where the average PWM switching frequency is 400kHz, the required IDD is 1.18 mA. Based on this calculation, a 50V power supply voltage would require RDD to be 33 kΩ or less. Furthermore, make sure IDD is below the maximum Zener diode bias current, IDDZ, during static state conditions. I DDZ ≥ V+ B − 10.2 V − 0.5 mA Rdd Figure 4. Supplying VDD Charging VBS Prior to Start The high-side bootstrap capacitor can be charged through a resistor from the positive supply bus to the VB pin by utilizing an internal 15.3V Zener diode between VB and VS. This scheme provides proper PWM start-up with self-oscillating topologies. The value of this charging resistor is subject to several constraints: - The minimum value of RCHARGE is limited by the leakage current of the bootstrap voltage supply through RCHARGE, which would limit the maximum PWM modulation index of the system. - The maximum value of RCHARGE is limited by the current charge capability of the resistor during startup: I CHARGE > I QBS where ICHARGE = the current through RCHARGE IQBS = the high side quiescent current. www.irf.com AN-1144 13 Figure 16. Boot Strap Supply Pre-charging Start-up Sequence (UVLO) The protection control block in the IRS20957 monitors the status of VDD and VCC to ensure that both voltage supplies are above the UVLO (under- voltage lockout) threshold before beginning normal operation. If either VDD or VCC is below the under voltage threshold, LO and HO are disabled in shutdown mode until both VDD and VCC rise above the voltage threshold. Power-down Sequence As soon as VDD or VCC falls below the UVLO threshold, protection logic in the IRS20957 turns off LO and HO, shutting off the power MOSFETs. Figure 5. IRS20957 UVLO Timing Chart Power Supply Decoupling Ceramic capacitors of 0.1 µF or more should be placed close to the power supply pins of the IC on the board. Please refer to the application note AN-978 for general design considerations of a high voltage gate driver IC. VSS Negative Bias Clamping VSS can go below COM when a negative supply is missing in a dual supply configuration. In this case, excessive negative VSS voltage with respect to COM could damage the IRS20957. Having www.irf.com AN-1144 14 a diode to clamp potential negative biases to VSS is recommended to protect the IC. A standard recovery diode with a current rating of 1A such as the 1N4002 is sufficient for this purpose. 1 VDD CSH 16 2 CSD VB 15 3 IN HO 14 4 VSS VS 13 5 NC NC 12 6 VREF VCC 11 7 OCSET LO 10 8 DT COM 9 -B Figure 6. Negative VSS Clamping Junction Temperature Estimation The power dissipation in the IRS20957 is dominated by the following items: - PMID: Power dissipation of the floating input logic and protection circuitry - PLSM: Power dissipation of the input level shifter - PLOW: Power dissipation in low-side - PLSH: Power dissipation of the high-side level shifter - PHIGH: Power dissipation in high-side 1. PMID: Power Dissipation of the Floating Input Logic and Protection Circuitry The power dissipation of the floating input section is given by: PMID = PZDD + PLDD ≈ V+ BUS − VDD ⋅VDD RDD where PZDD = the power dissipation from the internal Zener diode clamping VDD PLDD = the power dissipation from the internal logic circuitry V+BUS = the positive bus voltage feeding VDD RDD = the resistor feeding VDD from V+BUS *For obtaining the value of RDD, refer to the section “Supplying VDD.” 2. PLSM: Power Dissipation of the Input Level Shifter PLSM = 2 nC x fsw x VSS,BIAS where www.irf.com AN-1144 15 fSW = the PWM switching frequency VSS,BIAS = the bias voltage of VSS with respect to COM 3. PLOW: Power Dissipation in Low-Side The power dissipation in low-side comes from the losses of the logic circuitry and the losses of driving LO. PLOW = PLDD + PLO ⎛ ⎞ RO ⎟ = (I QCC ⋅ VCC ) + ⎜Vcc ⋅ Qg ⋅ f SW ⋅ ⎜ ⎟ + + R R R O g g (int) ⎝ ⎠ where PLDD = the power dissipation from the internal logic circuitry PLO = the power dissipation from the gate drive stage to LO RO = the output impedance of LO, typically 10 Ω for the IRS20957 Rg(int) = the internal gate resistance of the low side MOSFET driver, typically 10 Ω for the IRS20957 Rg = the external gate resistance of the low side MOSFET Qg = total gate charge of the low side MOSFET 4. PLSH: Power Dissipation of the High-Side Level Shifter PLSH = 0.4 nC x fsw x VBUS where fSW = the PWM switching frequency VBUS = the difference between the positive bus voltage and negative bus voltage 5. PHIGH: Power Dissipation in High-side The power dissipation in high-side comes from the losses of the logic circuitry and the losses of driving LO. PHIGH = PLDD + PHO ⎛ ⎞ RO ⎟ = (I QBS ⋅ VBS ) + ⎜VBS ⋅ Qg ⋅ f SW ⋅ ⎜ RO + Rg + Rg (int) ⎟⎠ ⎝ where PLDD = the power dissipation from the internal logic circuitry www.irf.com AN-1144 16 PLO = the power dissipation from the gate drive stage to HO RO = equivalent output impedance of HO, typically 10 Ω for the IRS20957 Rg(int) = the internal gate resistance of the high-side MOSFET driver, typically 10 Ω for the IRS20957 Rg = external gate resistance of the high-side MOSFET Q g = total gate charge of the high- side MOSFET Total power dissipation, Pd, is given by Pd = PMID + PLSM + PLOW + PHSM + PHIGH . Tj: Junction Temperature Given Rth,JA, the thermal resistance between the ambient and junction temperature, TJ, the junction temperature, can be calculated from the formula provided below. TJ = Rth , JA ⋅ Pd + TA < 150 °C www.irf.com AN-1144 17 Revision History Date Xx/xx/2007 September 16th, 2008 www.irf.com Change Initial online release Updated for IRS20957S. IRS20955S is not recommended for new design. Charging VBS Prior to Start: Vbs Zener diode clamping voltage from 20.4V to 15.3V. Other minor language corrections. AN-1144 18