A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Features and Benefits
Description
• AEC-Q100 Grade 0 qualified
• Internal buck pre-regulator followed by LDO outputs
• 5.5 to 36 V VIN operating range (50 V maximum);
for start/stop, cold crank, and load dump requirements
•2.2 MHz constant on-time buck regulator
•50 V absolute maximum input voltage for surges
•Valley current sensing achieves shortest buck on-times
•5.5 to 46 V input voltage range
•−40ºC to 150ºC junction temperature range
•Power-on reset (NPOR pin) with adjustable rising delay
•5 V (V5P pin) internal low dropout tracking
linear regulator with both overcurrent foldback and
short-to-battery protection
•3.3 V (V33 pin) external linear regulator DMOS driver
with a programmable current limit (up to 500 mA) and
The A4406 is an automotive power management IC that uses a
high frequency constant on-time 5.45 V pre-regulator to supply
an internal 5 V linear regulator and a 3.3 V linear DMOS driver.
Designed to supply CAN and microprocessor power supplies
in high temperature environments, the A4406 is ideal for under
hood applications. Efficient operation is achieved by using a
buck pre-regulator to efficiently drop the input voltage before
supplying the linear regulators; this reduces power dissipation
and increases overall regulator efficiency.
The switching regulator is capable of operating at a nominal
switching frequency of 2.2 MHz. The high switching frequency
enables the customer to select low value inductors and ceramic
capacitors while avoiding EMI in the AM frequency band.
Protection features include undervoltage lockout and thermal
shutdown. The V5P output is protected from short-to-battery
events. In case of a shorted load, all regulators feature
Continued on the next page…
Package: 20-pin TSSOP with exposed
thermal pad (suffix LP)
Continued on the next page…
Applications:
Automotive Control Modules, such as:
• Electronic power steering (EPS)
• Transmission control (TCU)
• Antilock braking (ABS)
• Emissions control
Not to scale
Typical Application Circuit
0.22 µF
RTON
412 kΩ
VIN
VCP
CP2
CP1
3.3 V
CIN2
4.7 µF
CIN1
4.7 µF
VBAT
DIN
B240A
TON
0.22 µF
4.7 kΩ
LX
NPOR
3.3 V
4.7 kΩ
CPOR
CCPOR
0.22 µF
DBUCK
B240A
ISEN–
A4406
RSENSE
300 mΩ
1/ W
4
ISEN+
L1
10 µH, 1.3 A
65 mΩ MAX
ENBATS
1 kΩ
VREG
100 Ω
ENBAT
100 nF
Enable
VIN(Pin1)
D2
A
5V
B240A
Protected
CV5P
1 to 2.2 µF
CVREG
10 µF
PAD
CLV33
0.47 µF
ENB
V5P
D1
B240A
QV33
G33
V5P
B
GND
VIGN
V33
RCL
390 mΩ
1/4 W
5.0 V / 400 mA operation, see page 15).
40 °C/W
175°C MAX
CV33
3.3 V, 400 mA
(500 mA MAX)
1 to 4.7 µF
Option with external LDO set to 3.3 V / 400 mA,
(Add R1 and R2 and remove RDROP to set external LDO to 5.0 V / 400 mA)
A4406-DS, Rev. 2
when the V5P pin is driving a wiring harness
(or excessively long PCB trace) where
parasitic inductance will cause the voltage at
the V5P to momentarily transition above VIN
or below ground during a fault condition.
B R1 and R2 should be ≤0.5% (used only for
RDROP
1.2 Ω
1/2 W
R1
R2
A Protection diodes D1 and D2 are required
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Features and Benefits (continued)
Description (continued)
overcurrent foldback protection
•Logic enable input (ENB pin)
•Ignition enable input (ENBAT pin)
•Ignition status indicator (ENBATS pin)
•Buck pulse-by-pulse overcurrent protection
•Buck LX short circuit protection (latched)
•Missing asynchronous diode protection (latched)
•Switcher (VREG pin), 3.3 V (V33 pin), and charge pump
(VCP pin) undervoltage lockout protection (UVLO)
•Thermal shutdown protection (TSD)
overcurrent protection. The A4406 also features power-on-reset
with adjustable delay for the microprocessor output.
The A4406 is supplied in a low profile (1.2 mm max) 20-pin TSSOP
package with exposed thermal pad (suffix LP). The package is lead
(Pb) free with 100% matte-tin leadframe plating
Selection Guide
Part Number
Packing
A4406KLPTR-T
4000 pieces per 13-in. reel
Absolute Maximum Ratings1
Characteristic
VIN Pin
Symbol
Notes
VIN
LX Pin
VLX
VCP, CP1, and CP2 Pins
Rating
Unit
–0.3 to 50
V
–0.3 to 50
V
t < 250 ns
VVCP , VCPx
–1.5
V
–0.3 to 60
V
V
ISEN– Pin
VISEN–
–0.5 to 1
ISEN+ Pin
VISEN+
–0.5 to 0.5
V
ENBAT Pin2
VENBAT
–0.3
V
ENBAT Pin Current
IENBAT
–50 to 50
mA
VREG Pin
VVREG
–0.3 to 8
V
V33 Pin
VV33
–0.3 to 7
V
G33 Pin3
VG33
–0.3
V
CLV33 Pin
VCLV33
–0.3 to 10
V
VV5P
–0.3 to VIN+0.5
V
TON Pin
VTON
–0.3 to 50
V
NPOR and CPOR Pins
VxPOR
–0.3 to 7
V
ENB and ENBATS Pins
VEN , VENBATS
V5P Pin
Operating Ambient Temperature
Junction Temperature
Storage Temperature Range
–0.3 to 7
V
–40 to 135
°C
TJ(max)
–40 to 150
°C
Tstg
–40 to 150
°C
TA
Range K
1Absolute
maximum ratings are limiting values that should not be exceeded under worst case operating conditions or damage may occur.
2The ENBAT pin is internally clamped to approximately 8.5 V due to an ESD protection device.
3The G33 pin is internally clamped by an ESD protection device. Clamp voltages range from 10 V (min) to 15 V (max).
Thermal Characteristics may require derating at maximum conditions, see application information
Characteristic
Package Thermal Resistance
Symbol
RθJA
Test Conditions*
Estimated on 4-layer PCB based on JEDEC standard
Value
Unit
32
ºC/W
*Additional thermal information available on the Allegro website.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Table of Contents
Specifications
2
4
5
6
Characteristic Performance
9
Functional Block Diagram
Pin-out Diagram and Terminal List
Electrical Characteristics
Functional Description
11
11
11
11
11
11
11
12
12
13
13
13
14
14
15
15
15
Timing Diagrams
16
Application Information
18
18
18
18
19
19
19
19
20
21
23
26
Basic Operation
Overcurrent Protection
Dropout Mode
Soft Start
Buck Pulse Width ( tON )
ISEN+ and ISEN–
Switcher Overcurrent Protection
LX Short Circuit Protection
Missing Asynchronous Diode Protection
Thermal Shutdown
Power-On Reset (NPOR)
V5P Tracking Regulator
3.3 V Linear Regulator
Charge Pump
ENBATS
ENB
Switcher On-Time and Switching Frequency
Low Voltage Operation
Inductor Selection
Output Capacitor
Input Capacitor
Rectification Diode
External MOSFET Selection
3.3 V Dropping Resistor (RDROP)
PCB Layout
Application Circuit Performance
Package Outline Drawing
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Functional Block Diagram
CIN1
4.7 µF
CCPOR
V33 UVLO
Fault
VIN
PWM Control
TSD
ENBAT
+
ISEN+
Microcontroller
Enable
ENB
4.0 VH
2.2 VL
ENB
VREF V33 VREG
V5 Control and
V33 to V5P
Tracking Control
VIN(Pin1)
A
Regulator VREF
VREF
CVREG
10 µF
0.47 µF
V33 FET
Driver
VREG
L1
10 µH, 1.3 A
65 mΩ MAX
RCL
390 mΩ
1/ W
4
RDROP
1.2 Ω
1/ W
2
QV33
G33
VREF
3.3 V, 400 mA
(500 mA MAX)
V33
CV33
V5P
1 to 4.7 µF
V5P
DV5P
B240A
RSENSE
300 mΩ
1/ W
4
CLV33
–
–
ENBATS
VREG
Soft Start
Ramp
Generator
+
1 to 2.2 µF
ISEN–
–
8.5 V
100 nF
CV5P
Switch
Disable
+
100 Ω
4.7 kΩ
D2
DBUCK
B240A
VREG UVLO
3.3 V
(From V33)
5V
B240A
Protected
VBAT
LX
CPOR
0.22 µF
1 kΩ
DIN
B240A
NPOR
Charge Pump
VIGN
CIN2
4.7 µF
TON
RTON
412 kΩ
VCP
CP1
4.7 kΩ
Microcontroller
Reset
0.22 µF
CP2
0.22 µF
3.3 V
(From V33)
GND
Short to Supply
Protection
PAD
A
Protection diodes D1 and D2 are required when the V5P pin is driving a wiring harness (or excessively long
PCB trace) where parasitic inductance will cause the voltage at the V5P to momentarily transition above
VIN or below ground during a fault condition.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Pin-out Diagram
VIN 1
20 VCP
GND 2
19 CP2
TON 3
18 CP1
ENBAT 4
ENB 5
ENBATS 6
17 LX
PAD
16 ISEN+
15 ISEN–
NPOR 7
14 VREG
CPOR 8
13 CLV33
V5P 9
12 G33
V5P 10
11 V33
Terminal List Table
Name
Number
Function
CLV33
13
3.3 V current sense and limit input
CP1
18
Charge pump capacitor terminal 1
Charge pump capacitor terminal 2
CP2
19
CPOR
8
NPOR delay programming pin
ENB
5
Logic enable input from the microcontroller
ENBAT
4
Ignition enable input from the key or switch via a 1 kΩ resistor
ENBATS
6
Open drain ignition status output
G33
12
Gate driver to the external MOSFET for 3.3 V regulation
GND
2
Ground terminal
ISEN–
15
Buck negative current sense pin, sense resistor and diode node
ISEN+
16
Buck positive current sense pin, sense resistor and ground node
LX
17
Buck regulator switching node
NPOR
7
Active low, open-drain fault indication output
PAD
–
Exposed pad for enhanced thermal dissipation
TON
3
Buck regulator on-time programming pin
3.3 V regulator output
V33
11
V5P
9,10
VCP
20
Charge pump reservoir capacitor terminal
VIN
1
Input voltage
VREG
14
Buck regulator DC output, and input to the 3.3 V external regulator
5 V tracking, protected regulator output
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
ELECTRICAL CHARACTERISTICS Valid at –40°C ≤ TJ ≤ 150°C, 5.5 V ≤ VIN ≤ 36 V; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
General
Function Input Voltage
VIN(f)
Operating Input Voltage
VIN(op)
Supply Quiescent
Current1
Device functional, parameters not guaranteed
5.5
−
46
V
5.5
13.5
36
V
IQ
VIN = 13.5 V, VIGN > VIGN(H) or VENB > VENB(H),
no load on VREG
–
10
–
mA
IQ(SLEEP)
VIN = 13.5 V, VIGN < VIGN(L), VENB < VENB(L), no
load on VREG
–
–
10
µA
ENB = high , VINSW(L) < VIN < 27 V,
25 mA < IVREG < 600 mA
5.25
5.45
5.60
V
ENB = high , VINSW(NOM) < VIN < 27 V,
25 mA < IVREG < 750 mA
5.30
5.45
5.60
V
ENB = high , 5.5 V < VIN < 6.5 V,
LX 100% on, 100 mA < IVREG < 600 mA
5.15
–
6.46
V
–
1.6
–
µs
Buck Regulator
Switcher Output
Switcher
Period2
VVREG
TSW(L)
VINSW(L) < VIN < VINSW(NOM), RTON = 412 kΩ
TSW(NOM)
VINSW(NOM) < VIN < VINSW(H), RTON = 412 kΩ
–
450
–
ns
VINSW(H) < VIN < 36 V, RTON = 412 kΩ
–
1.6
–
µs
VIN = 7 V, RTON = 412 kΩ
1070
1335
1600
ns
VIN = 13.5 V, RTON = 412 kΩ
160
200
240
ns
TSW(H)
Switcher On-Time
tON
VINSW(L)
Switcher Period VIN Threshold
80
118
135
ns
VIN = 35 V, RTON = 412 kΩ
220
275
330
ns
VIN falling, TSW changes from TSW(L) to
100% duty cycle
5.9
6.2
6.5
V
7.7
8.3
8.9
V
VIN rising, TSW changes from TSW(NOM) to TSW(H)
28
31
34
V
VINSW(L) and VINSW(NOM) comparators, relative to
the VIN voltage that initially caused the switcher
period to change
–
250
–
mV
VINSW(H) comparator, relative to the VIN voltage
that initially caused the switcher period to
change
–
700
–
mV
TJ = 25°C, IDS = 0.1 A
–
275
300
mΩ
TJ = 150°C, IDS = 0.1 A
–
400
470
mΩ
VINSW(NOM) VIN falling, TSW changes from TSW(NOM) to TSW(L)
VINSW(H)
Switcher Period VIN Hysteresis
VIN = 27 V, RTON = 412 kΩ
VINSW(HYS)
Buck Switch On-Resistance
RDS(on)
Minimum On-time
ton(min)
VIN = 13.5 V, RTON = 49.9 kΩ
–
65
90
ns
Minimum Off-time
toff(min)
VIN = 13.5 V
85
110
140
ns
VISEN
VISEN+ – VISEN–
175
220
265
mV
–
733
–
mA
2.5
5.0
–
A
ISEN Voltage Threshold
VREG Valley Current Limit
VREG Peak Current Limit
ILIM(VALLEY) RSENSE = 300 mΩ, VIN > VINSW(L)
ILIM(PEAK)
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
ELECTRICAL CHARACTERISTICS (continued) Valid at –40°C ≤ TJ ≤ 150°C, 5.5 V ≤ VIN ≤ 36 V; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
5 V Linear Regulator
V5P Accuracy and Load Regulation
V5P/V33 Tracking Ratio
V5P/V33 Tracking Accuracy
VV5P
V5Ptrack
ErrV5Ptrack
10 mA < IV5P < 270 mA, VVREG ≥ 5.25 V
4.9
5.0
5.1
V
VV5P / VV33
1.507
1.515
1.523
–
2.69 V < VV33 < 3.37 V, IV5P = 75 mA,
5.5 V < VIN < 27 V
−0.5
–
+0.5
%
Linear Regulator and FET Driver
V33 Accuracy
ErrV33
10 mA < IV33 < 500 mA
3.23
3.30
3.37
V
IG33(SRC)
VV33 = 3.0 V, VG33 = VG33(MAX) – 1 V
−175
−250
−400
µA
G33 Sink Current1
IG33(SINK)
VV33 = 3.6 V, VG33 = 6 V
0.5
3
–
mA
G33 Maximum Voltage
VG33(MAX)
VV33 = 3.0 V
9
–
15
V
G33 Minimum Voltage
VG33(MIN)
VV33 = 3.6 V
G33 Source Current1
–
0.7
1.0
V
G33 Output Impedance2
ROUT
–
175
–
Ω
External FET Gate Capacitance2
CISS
250
–
5200
pF
Charge Pump (VCP Pin)
Output Voltage
ΔVVCP
Switching Frequency
fSW(CP)
VVCP − VIN
4.1
6.6
–
V
–
100
–
kHz
Logic Enable Input (ENB Pin)
ENB Logic Input Threshold
ENB Logic Input
Current1
ENB Pull-Down Resistance
VENB(H)
VENB rising
–
–
2.0
V
VENB(L)
VENB falling
0.8
–
–
V
IENB(IN)
VENB = 3.3 V
RENB
–
–
100
µA
–
60
–
kΩ
Ignition Enable Input (ENBAT and ENBATS Pins)
ENBAT and ENBATS Thresholds
ENBAT Input Current1
ENBAT Input Resistance
VIGN(H)
VIGN rising via a 1 kΩ series resistance,
measure VIGN when IQ occurs
–
–
4.0
V
VIGN(L)
VIGN falling via a 1 kΩ series resistance,
measure VIGN when IQ(SLEEP) occurs
2.2
–
–
V
VIGN = 5.5 V via a 1 kΩ series resistance
–
50
100
µA
VIGN = 0.8 V via a 1 kΩ series resistance
0.5
–
5
µA
–
650
–
kΩ
IENBAT(IN)
RENBAT
Ignition Status Output (ENBATS Pin)
ENBATS Output Voltage
VENBATS(L) IENBATS = 4 mA
–
–
400
mV
ENBATS Leakage Current1
IENBATS
VENBATS = 3.3 V
–
–
1
µA
ENBATS Turn-On Delay
tENBATS
Sleep mode to VENBATS = 3.3 V
–
11
–
ms
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
ELECTRICAL CHARACTERISTICS (continued) Valid at –40°C ≤ TJ ≤ 150°C, 5.5 V ≤ VIN ≤ 36 V; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
CPOR = 0.22 µF
–
20
–
ms
ENB = high or ENBAT = high,
VVREG < VREGNPOR(L) or VV33 < V33NPOR(L) ,
INPOR ≤ 4 mA
–
–
400
mV
ENBAT = low, ENB transitioning to low,
VVREG = 5.45 V, INPOR ≤ 0.3 mA,
0.8 V < VV33 < ErrV33 , 0°C ≤ TJ ≤ 150°C
–
350
800
mV
ENBAT = low, ENB transitioning to low,
VVREG = 5.45 V, INPOR ≤ 0.3 mA,
1.0 V < VV33 < ErrV33 , −40°C ≤ TJ ≤ 150°C
–
–
800
mV
VNPOR = 3.3 V
–
–
1
µA
NPOR Pin Output and Timing
NPOR Power-Up Delay
NPOR Output Voltage
NPOR Leakage Current1
tNPOR
VNPOR(L)
INPOR(LEAK)
CPOR Pin Characteristics
CPOR Charge Current1
ICPOR(SRC)
CPOR Threshold
VCPOR(H)
–
−13
–
µA
1.0
1.2
1.4
V
–
10
–
ms
VREGNPOR(H) VVREG rising, NPOR transitioning to high
4.80
5.00
5.20
V
VREGNPOR(L) VVREG falling, NPOR transitioning to low
4.75
4.94
5.14
V
–
60
–
mV
V33NPOR(H) VV33 rising, NPOR transitioning to high
2.80
2.95
3.10
V
V33NPOR(L) VV33 falling, NPOR transitioning to low
2.69
2.83
2.97
V
–
125
–
mV
VCPOR rising
VREG Pin Soft Start Timing
Soft Start
tSS
Protection Circuitry
VREG Pin NPOR Thresholds
VREG Pin NPOR Hysteresis
V33 Regulator NPOR Thresholds
VREG(HYS)
V33 Regulator NPOR Hysteresis
V33(HYS)
V33 Regulator Overcurrent Threshold
V33OCP
VVREG – VCLV33
175
200
245
mV
V33 Regulator Current Limit
IV33ILIM
RCL = 620 mΩ
–
323
–
mA
V33 Regulator Foldback Threshold
V33IFB
VV33 = 0 V, VVREG – VCLV33
35
55
75
mV
V33 Regulator Foldback
Current Limit
IV33IFB
RCL = 620 mΩ
–
89
–
mA
V5P Pin Current Limit1
IV5PILIM
VV5P = 5 V
−300
−405
–
mA
V5P Pin Foldback
Current1
IV5PIFB
VV5P = 0 V
−70
−110
−150
mA
Thermal Shutdown Threshold
TJTSD
TJ rising
155
170
–
ºC
Thermal Shutdown Hysteresis
TJTSD(HYS)
–
20
–
ºC
1For
input and output current specifications, negative current is defined as coming out of (sourcing) the specified pin.
by design and systems characterization. Not production tested.
2Determined
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
8
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Characteristic Performance
tON versus Temperature
VREG Output versus Temperature
1,400
1,300
1,200
1,100
1,000
900
800
700
600
500
400
300
200
100
0
5.50
5.49
5.47
VIN = 7.5 V
tON Pulse Width (ns)
VREG Output Voltage (V)
5.48
5.46
5.45
5.44
5.43
5.42
5.41
5.40
-40
-20
0
20
40
60
80
100
120
VIN = 35 V
VIN = 13.5 V
VIN = 27 V
-40
140
-20
0
20
40
60
80
100
120
140
120
140
Temperature (°C)
Temperature (°C)
V5P Output versus Temperature
V33 Output versus Temperature
5.05
3.33
5.04
5.03
V5P Output Voltage (V)
V33 Output Voltage (V)
3.32
3.31
3.30
3.29
5.02
5.01
5.00
4.99
4.98
4.97
3.28
4.96
4.95
3.27
-40
-20
0
20
40
60
Temperature (°C)
80
100
120
140
-40
-20
0
20
40
60
80
100
Temperature (°C)
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
9
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
ENBAT Start / Stop Thresholds versus Temperature
ENB Start / Stop Thresholds versus Temperature
4.0
2.0
3.8
1.8
3.6
1.6
Start
1.4
3.2
ENB Threshold (V)
ENBAT Threshold (V)
3.4
3.0
Stop
2.8
2.6
1.2
0.6
0.4
2.2
0.2
-40
-20
0
20
40
60
80
100
120
Stop
0.8
2.4
2.0
Start
1.0
0.0
140
-40
-20
0
Temperature (°C)
60
80
100
120
140
ENBATS (Low) Voltage versus Temperature
16
400
350
ENBATS Voltage (mV)
15
CPOR Charging Current (uA)
40
Temperature (°C)
CPOR Charging Current versus Temperature
14
13
12
11
300
IENBATS = 4 mA
250
200
150
100
50
10
0
-40
-20
0
20
40
60
80
100
120
-40
140
-20
0
Temperature (°C)
20
40
60
80
100
120
140
Temperature (°C)
VREG Valley Current Limit versus Temperature
V33 Overcurrent Threshold versus Temperature
300
210
V33 Over current Threshold (mV)
250
VREG Valley Limit (mV)
20
200
150
100
50
205
200
195
190
185
0
-40
-20
0
20
40
60
Temperature (°C)
80
100
120
140
-40
-20
0
20
40
60
80
100
120
140
Temperature (°C)
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
10
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Functional Description
Basic Operation
The A4406 contains a fixed on-time, buck switching pre-regulator with valley sensing current mode control, an integrated
5 V linear regulator, and an N-channel FET driver for a 3.3 V
linear regulator. The constant on-time (COT) converter maintains
a constant output frequency because the on-time is inversely
proportional to the supply voltage. As the input voltage decreases
the on-time is increased, which maintains a relatively constant
period. Valley mode current control allows the converter to
achieve very short on-times because current is measured during
the off-time.
With very low input voltages the buck switch maintains a 100%
duty cycle. This turns the buck switch on 100% of the time (no
switching) and allows the regulator to operate in dropout mode.
The device is enabled via logic level ENB or high voltage ignition ENBAT input. When the device is enabled the converter
starts up under the control of an internal soft start routine. The
two enable inputs are logically ORed together internally so either
of the inputs can be used to enable the device.
Under light load conditions the switch enters pulse-skipping
mode to ensure regulation is maintained. In order to maintain a
wide input voltage range the switcher period is extended when
the minimum on- or off-time is reached, or when the input supply
is at either end of its range.
Overcurrent Protection
The A4406 features overcurrent protection on all regulators
including the VREG pre-regulator. The buck switch current limit
is determined by the selection of the sense resistor at the ISENx
pins. Output current is also monitored on the 5VP and V33 linear
regulators, and if shorted the outputs fold back. The external FET
driver has a current limit tap that can be used with a sense resistor
to trigger a current limit based on an external resistor and trip
voltage.
Dropout Mode
The topology of a COT timer is ideal for systems that have high
input voltages. Because current is measured during the off-time,
very short on-times can be achieved. With low input voltages
the switcher must maintain very short off-times. To prevent
the switcher from reaching its minimum off-time, the switcher
is designed to enter a 100% duty cycle mode. This causes the
switcher to stop acting as a buck switch. The voltage at VREG
then becomes the simply the supply voltage minus the drop
across the buck switch and inductor. In this mode the maximum
available current may be lower, depending on ambient temperature and supply voltage, while in dropout mode.
Soft Start
An internal ramp generator and counter allow the output voltage
to ramp-up. This limits the maximum demand on the external
power supply by controlling the inrush current required to charge
the external capacitor and any DC load at startup. Internally, the
ramp is set to 10 ms nominal.
The following conditions are required to trigger a soft start:
• ENBAT or ENB transition to high, and
• There is no thermal shutdown, and
• V33 voltage is below its UVLO threshold, and
• VREG voltage is below its UVLO threshold.
Buck Pulse Width ( tON )
A resistor from the TON input to VIN sets the on-time of the
converter for a given input voltage. When the supply voltage is
between 8.3 and 31 V, the switcher period remains constant based
on the selected value of RTON . At voltages lower than 6.5 V the
switch is in dropout mode. For reasonable input voltage ranges
the period of the converter is held constant resulting in a constant
operating frequency over the input supply range. More information on how to choose RTON can be found in the Application
Information section.
The formula to calculate the value for the on-time resistor is:
ton = ( RTON / VIN ) × 6.36 × 10 –12 + 5 × 10 –9 (ns).
(1)
ISEN+ and ISEN–
The sense inputs are used to sense the current in the buck, freewheeling diode during the off-time cycle. The value for RSENSE is
obtained by the formula:
RSENSE = 220 (mV) / IVALLEY ,
(2)
where IVALLEY is the lowest current measured through the inductor during the off-time cycle. It is recommended that the current
sense resistor be sized so that, at peak output current, the voltage
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
11
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
at the ISEN– pin does not exceed –0.75 V during PWM operation (that is, a transient condition). Because the diode current is
measured when the inductor current is at the valley, the average
output current is greater than the IVALLEY value. The value for
IVALLEY should be:
IVALLEY = IOUT(AVG) – 0.5 × IRIPPLE + K ,
(3)
where:
IOUT(AVG) is the average of the output currents of all
the regulators,
IRIPPLE is the inductor ripple current, and
K is a design margin allowing for component tolerances.
The peak current in the switch is simply:
IPEAK = IVALLEY + IRIPPLE .
(4)
Information on how to calculate the ripple current is included in
the Application Information section.
Switcher Overcurrent Protection
The converter utilizes pulse-by-pulse valley current limiting,
which is activated when the current through the sense resistor
(that is, the buck output current) is high enough to create 220 mV
between the ISEN pins. During an overload condition, the switch
is turned on for a period determined by the constant on-time
circuitry. The switch off-time is extended until the current decays
to the current limit value set by the selection of the sense resistor,
at which point the switch is allowed to turn on again. Because no
slope compensation is required in this control scheme, the current
limit is maintained at a reasonably constant level across the input
voltage range.
Figure 1 illustrates how the current is limited during an overload
condition. The current decay (period with switch off) is proportional to the output voltage. As the overload is increased, the output voltage tends to decrease and the switching period increases.
LX Short Circuit Protection
If the LX node is shorted to ground there will be a relatively high
peak current in the buck MOSFET within a very short time. The
A4406 protects itself by detecting the unusually high current,
turning off the buck MOSFET, and latching itself off. To avoid
false tripping, the current required to activate the peak current
protection (ILIM(PEAK), nominally 5 A) is set well above the
normal range of currents. When the peak current limit is activated the A4406 is latched off until either VIN is cycled below
its UVLO threshold or the A4406 is disabled (both ENBAT and
ENB must be brought low) and re-enabled. NPOR is not directly
activated (pulled low) by the peak current protection circuitry.
However, NPOR will certainly be in the correct state depending
on VREG and V33.
Inductor current, operating at maximum load
Current Limit level
Current
Maximum load
Constant On-Time
Constant Period
Time
Inductor current, operating a soft overload
Current
Overload
Current Limit level
Constant On-Time
Extended Period
Time
Figure 1. Current limiting during overload conditions
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115 Northeast Cutoff
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12
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Missing Asynchronous Diode Protection
In most high voltage asynchronous buck regulators, if the asynchronous diode is missing or damaged the LX pin will transition
to a very high negative voltage when the upper MOSFET turns
off, resulting in damage to the regulator. The A4406 includes protection circuitry to detect when the asynchronous diode is missing
or damaged. If the LX pin becomes more negative than 1.2 V at
25°C for more than 157 ns, the A4406 will latch itself in the off
state to prevent damage. After a missing diode fault occurs, the
latch must be reset by either cycling VIN or ENBAT or ENB. See
figure 2 for the missing diode voltage threshold and time filtering
versus temperature.
after both VREG and V33 transition above their upper UVLO
thresholds. The rising edge delay allows time for the regulators to
be within specification when the DSP or microcontroller begins
processing. The amount of the rising edge delay is determined by
the value of the external capacitor from the CPOR pin to ground.
The rising delay can be calculated from the following formula:
tNPOR = 92.3 × 103 × CCPOR (seconds).
(5)
Any of the following conditions will force NPOR to transition to
low immediately (within a few microseconds):
• V33 voltage falls below its UVLO threshold, or
• VREG voltage falls below its UVLO threshold, or
If thermal shutdown occurs, PWM switching will terminate,
VVREG and/or VV33 will decay below the UVLO threshold, and
NPOR will transition to low. Thus, a thermal shutdown event
indirectly causes NPOR to transition to low.
Power-On Reset (NPOR)
The NPOR output is an open drain pin that can be used to signal
a reset event to a DSP or microcontroller. The NPOR function
actively monitors ENBAT, ENB, V33, and VREG. During powerup, NPOR is held low for a programmable amount of time, tNPOR,
When the A4406 is disabled (either both ENB and ENBAT are
low, or VIN is removed) the NPOR output is held low until the
voltage from the 3.3 V regulator (VV33) falls below 1.0 V. This
assumes maximum initial current (4 mA) in the NPOR open drain
DMOS. The NPOR voltages would be somewhat lower for lower
values of INPOR . See figure 3.
1.60
210
1.50
200
1.40
190
Time Filtering
1.30
180
1.20
170
1.10
160
1.00
150
Voltage Threshold
0.90
140
0.80
130
0.70
120
-50
-25
0
25
50
75
100
125
150
• ENBAT and ENB are both low, or
• Charge pump voltage is too low. or
• Internal IC regulation (VRAIL) is too low.
ENB, ENBAT
3 .3 V
Time Filtering (ns)
Negative Voltage Threshold (V)
Thermal Shutdown
If the A4406 junction temperature becomes too high, a thermal
shutdown circuit disables the VREG output, thus protecting the
A4406 from damage. When a thermal shutdown occurs, the buck
regulator stops switching and the VREG voltage will decay.
When VVREG crosses its UVLO threshold, the NPOR signal is
pulled low. Thermal shutdown is not a latched condition so, when
the junction temperature cools to an acceptable level, the A4406
will automatically restart.
VV33
≤ 1.0 V
≤ 4 mA
I NPOR
VNPOR
≤ 0 .3 mA
350 mV(typ)
800 mV
400 mV
Junction Temperature (°C)
Figure 2. Missing diode protection versus device junction temperature
Figure 3. NPOR and V33 characteristics when the A4406 is disabled
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
13
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
V5P Tracking Regulator
The 5VP linear tracking regulator is provided to supply remote
circuitry such as off-board sensors. The V5P pin is monitored and
if a short to ground or a short to battery occurs the V5P output
is disabled and/or disconnected and the other outputs (VREG
and V33) remain active until the short is removed. The regulator
can deliver 375 mA (typ), 270 mA (min). When a direct short is
applied to this regulator the output the current folds back to 0 V
at approximately 112 mA (typ) (figure 4).
The V5P regulator is designed to track the V33 output during
power-up and when the device is completing a soft start ramp.
The V5P regulator tracks the 3.3 V output to within ±0.5% under
normal steady state operating conditions. If the V33 regulator is
decreasing, the V5P regulator accurately tracks the V33 output
down to the point at which a V33 undervoltage fault (2.825 V
nominally: 2.95 V – 125 mV) results in the NPOR output
going active.
The figures 5 and 6 show A4406 operation when the V5P pin
is shorted to ground and VIN (battery). In both cases, the V5P
output is disabled and/or disconnected while the other outputs
(VREG and V33) remain active.
3.3 V Linear Regulator
An additional 3.3V linear regulator can be implemented using
an external MOSFET. In the event the 3.3V regulator output is
shorted to ground, the A4406 protects the external MOSFET
by folding back when the programmed current limit, ICL , is
exceeded. The current limit is determined by the voltage devel-
6
4
Mi
nim
um
Output Voltage (V)
5
3
2
l
ca
pi
Ty
um
im
ax
M
1
0
50
100
150
200
250
300
350
Output Current (mA)
400
450
500
Figure 4. Foldback current limit of the 5VP regulator
30 V
VVREG
VIN pin
VV33
25 V
C1
VV5
VREG
C2
Ringing due to parasitics from a long wire
V5P is clamped to a safe level above VIN
by D2 (see application schematic)
VV33
C3
VV5P
C4
t
Figure 5. V5P is shorted to ground in 5 µs (DV5P is populated); shows
VVREG (ch1, 2 V/div.), VV33 (ch2, 1 V/div.), VV5 (ch3, 2 V/div.), VV5P (ch4,
2 V /div.), t = 10 µs/div.
All
VV5P
5V
t
Figure 6. V5P is shorted to a 25 V battery; shows VVREG (ch1, 2 V/div.),
VV33 (ch2, 2 V/div.), VIN pin (ch3, 5 V/div.), VV5P (ch4, 5 V /div.),
t = 10 µs/div.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
14
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
oped across the external sense resistor, RCL , shown in the Typical
Application Circuit schematic. The 3.3V current limit can be
calculated using the following formula:
ICL = VCLV33 / RCL ,
(6)
where VCLV33 is as documented in the Electrical Characteristics table, nominally 200 mV. Typically RCL will be a fairly low
value so it will not dissipate significant power, 1/4 W should be
adequate, but the tolerance should be 1% or less.
When ICL is exceeded, the maximum load current through the
external MOSFET is typically folded back to 48% of ICL as
shown in figure 7.
Some applications require 5.0 V instead of 3.3 V. The external
LDO controller will produce 5.0 V if a resistor divider is inserted
between the controller output (that is, the source of the external
MOSFET) and the V33 pin, as shown on page 1 of this datasheet.
In this case, the 1.2 Ω/0.5 W power dropping resistor in series
with the drain of the external MOSFET must be removed from
the design.
Allegro recommends using resistors with ≤0.5% tolerance for two
reasons: (1) the 5.0 V output will have the best accuracy, and (2)
to maintain a low tracking error between the V5P and the 5.0 V
output. A comparison of two sets of resistor values at 0.1% and
0.5% tolerance are shown in the following table.
Min.
Typ.
Max.
V5P/V5
Tracking
Accuracy (%)
100 Ω and 196 Ω
±0.1%
4.887
4.999
5.112
±0.6
100 Ω and 196 Ω
±0.5%
4.874
4.999
5.125
±0.9
1.02 kΩ and 2.18 kΩ
±0.1%
4.867
5.002
5.141
–1.1, +1.2
1.02 kΩ and 2.18 kΩ
±0.5%
4.855
5.002
5.154
–1.4, +1.5
5V Output Range (V)
If an external resistor and capacitor are used to form a low-pass
filter to the ENBAT pin, then a 100 Ω resistor must be used to
prevent the external capacitor from discharging into and damaging the ENBAT pin. See the Typical Application Circuit schematic for connection of these 3 components.
ENBATS
When a logic high is sensed on the ENBAT input, the ENBATS
output will go high, signaling to the user that the ignition input
is high. When a logic low is sensed on the ENBAT input, then
ENBATS will also transition to low. The ENBATS input logic
levels are identical to the ENBAT input logic levels.
ENB
This pin can be used as an enable input from either a DSP or from
a microcontroller. This input has an internal pull-down resistor so
it may be left unconnected if not used.
3.5
3.0
im
1.5
um
2.0
ax
Charge Pump
The charge pump is used to generate a supply above VIN .
A 0.22 µF ceramic monolithic capacitor should be connected
between VCP and VIN to act as a reservoir to run the DMOS
switch. The VCP voltage is internally monitored to ensure that
the charge pump is disabled in the case of a fault condition.
A 0.22 µF ceramic monolithic capacitor should be connected
between CP1 and CP2.
2.5
M
R1 and R2 Values
and Tolerances
This is a level-triggered enable input, use for enabling the device
based on a high voltage ignition or battery switch (via a 1 kΩ
resistor). The ENBAT comparator thresholds are VIGN(L) =
2.2 V(min)and VIGN(H) = 4.0 V (max). ENBAT is used only as a
momentary switch to enable or wake up the A4406. After ENBAT
is removed, ENB must be high to keep the A4406 enabled. The
ENB and ENBAT signals are logically ORed together internally
so either one may wake up the A4406 and both must be low to
disable the A4406. Only one of the two inputs must be pulled
high in order to enable the part. If there is no requirement for an
ignition switch, then ENBAT can be pulled low, which makes
ENB a single reset control.
M
in
Ty imu
m
pi
ca
l
ENBAT
Output Voltage (V)
A4406
1.0
05
0
50
10
20
30
40
50
60
70
80
90
100 110 120
Percentage of Normal Current Setting (%)
Figure 7. Foldback current limit of the V33 regulator
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115 Northeast Cutoff
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15
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Timing Diagrams
13.5 V
VIN
ENBAT
VH=4.0 V
Clamped at ≈8.5 V via 1 kΩ
ENBATS is open-drain, pulled up to V33
ENBATS
VH =2 V
ENB
Internal
VRAIL
or VCP
VREG
VL= 2.2 V
Internal
UVLO
Internal
UVLO
VH=5.00V
Decay rates of VREG, V5P,
and V33 depend on output
capacitances and loading
V5P and V33 ramp at
approximately the same
rate as VREG
VH =2.95 V
VL= 2.83 V
V33
NPOR
VL=4.94V
10 ms
V5P
CPOR
VL= 0.8 V
1.0V
1.2 V
VREG > 5.00 V and
V33 > 2.95 V
20 ms
NPOR is open-drain, pulled up to V33
0.8 V MAX
ENB < 0.8V or
VREG < 4.94V or
V33 < 2.83V or
VCP low or
Internal VRAIL low
Typical power-up and power-down by ENBAT and ENB with VIN = 13.5 V; ENBATS is assumed to be connected to V33 via a pull-up resistor
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
16
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
13.5V
VIN
6.5 V
5.5 V
5.2 V
VENBAT = 0V
ENBAT
ENBATS
ENB
Internal
VRAIL
or VCP
ENBATS is not connected
VENB ≥ 2V prior to
VIN ramping up
Internal regulators
collapse
Internal
UVLO
100 %Duty
Cycle
VH = 5.00V
VREG
Internal
UVLO
VL= 4.94V
10 ms
V5P and V33 ramp
at approximately the
same rate as VREG
V5P
V 33
4.9 V
V5P tracks V33 until
VV33NPOR(L) or VIN < 5.5 V
VH=2.95 V
Decay rates of VREG,
V5P, and V33 depend on
output capacitances and
loading
VL=2.83 V
1.0V
1.2V
CPOR
VREG > 5.00 V and
V33 > 2.95 V
NPOR is open-drain, pulled up to V33
0.8V MAX
NPOR
20 ms
ENB < 0.8V or
VREG < 4.94V or
V33 < 2.83V
VCP low or
Internal VRAIL low
Typical power-up and power-down via VIN with ENB always logic high; ENBAT and ENBATS are not used
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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17
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Application Information
Switcher On-Time and Switching Frequency
In order for the switcher to maintain regulation, the energy that is
transferred to the inductor during the on-time must be transferred
to the capacitor during the off-time. Because of this relationship, the load current and IR drops, as well as input and output
voltages, affect the on-time of the converter. The formula that
governs switcher on-time is shown below:
tON =
TSW × {VVREG + [(RL + RSENSE) × IPEAK] + Vf }
VIN – RDS(on) × IPEAK + Vf
.
(7)
where Vf is the forward voltage on the diode DBUCK in the Typical Application Circuit schematic.
The effects of the voltage drop on the inductor and trace resistance will affect the switching frequency. However, the frequency
variation due to these factors is small and is covered in the variation of the switcher period, which is ±25% of the target. Removing these current dependant terms simplifies the formula:
tON =
(1/ fSW) × (VVREG + RSENSE × IPEAK ) + Vf
VIN – RDS(on) × IPEAK + Vf
.
(8)
Be sure to use the worst-case sense voltage and forward voltage
of the diode DBUCK , including any effects due to temperature.
For an example: assume a 1 A converter with a supply voltage of
13.5 V. The output voltage is 5.45 V, Vf is 0.45 V, RSENSE × IPEAK
is 0.20 V, RDS(ON) × IPEAK is 0.15 V, and the required frequency
is 2.2 MHz. Substituting into equation 8, we can solve for tON:
tON = 1 / 2.2 (MHz) ×
[(5.45+0.20+0.45) / (13.5 – 0.15 + 0.45)]
= 201 (ns) .
The formulas above describe how tON changes based on input
and load conditions. Because load changes are minimal and the
output voltage is fixed, the only factor that will affect the on-time
is the input voltage. The converter is able to maintain a constant
period over a varying supply voltage because the on-time changes
based on the input voltage. The current into the TON terminal is
derived from a resistor tied to VIN, which sets the on-time proportional to the supply voltage. Selecting the resistor value based
on the tON calculated above is done using the following formula:
RTON = [VIN × ( tON − 5 (ns) )] / 6.36 × 10 –12 .
(9)
After the resistor is selected and a suitable tON is found, it must
be demonstrated that tON does not, under worst-case conditions, exceed the minimum on-time or minimum off-time of
the converter. The minimum on-time occurs at maximum input
voltage and minimum load. The maximum off-time occurs at
minimum supply voltage and maximum load. For supply voltages
below 8.3 V and above 6.5 V, refer to the Low Voltage Operation section.
Low Voltage Operation
The converter can run at very low input voltages; with a 5.25 V
output the minimum input supply can be as low as 5.5 V. When
operating at high frequencies the on-time of the converter must
be very short because the available period is short. At high input
voltages the converter should not violate the minimum on-time,
tON(min), while at low input voltages the converter should not
violate the minimum off-time, tOFF(min). Rather than limit
the supply voltage range, the converter solves this problem by
automatically increasing the period. With the period extended
the converter will not violate the minimum on-time or off-time
specifications. If the input voltage is between 8.3 and 31 V, the
converter maintains a constant period. When calculating worst
case on-times and off-times, make sure to use the highest switching frequency if the supply voltage is in that range.
When operating at voltages below 8.3 V, additional care must
be taken when selecting the inductor and diode. At low voltages
the maximum current may be limited due to the IR drops in the
current path. When selecting external components for low voltage operation, the IR drops must be considered for determining
on‑time, so the complete equation (formula 8) should be used to
make sure the converter does not violate the timing specification.
Inductor Selection
Choosing the right inductor is critical to the correct operation of
the switcher. The converter is capable of running at frequencies
above 2 MHz, this makes it possible to use small inductor values,
which reduces cost and board area.
The inductor value is what determines the ripple current. It is
important to size the inductor so that under worst-case conditions ITRIP equals IAVG , minus half of the ripple current, plus a
reasonable margin. If the ripple current is too large, the converter
will activate the current limit function. Typically peak-to-peak
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115 Northeast Cutoff
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18
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
ripple current should be limited to a range of 20% to 25% of the
maximum average load current.
Worst-case ripple current occurs at maximum supply voltage.
After calculating the duty cycle for this condition, the ripple current can be calculated:
VVREG + ( RSENSE × IPEAK ) + Vf .
(10)
D =
VIN(max)– RDS(on) × IPEAK + Vf
Using the duty cycle, a ripple current can be calculated using the
formula below:
VIN – VVREG
1
,
(11)
L =
×D ×
IRIPPLE
fSW(min)
where IRIPPLE is 25% of the maximum load current, and fSW(min)
is the minimum switching frequency, nominal frequency minus
25%. For the example used above, a 1 A converter with a supply
voltage of 13.5 V was the design objective. The supply voltage
can vary by ±10%. The output voltage is 5.45 V, Vf is 0.5 V,
VSENSE is 0.20 V and the required frequency is 2.2 MHz. The
duty cycle is calculated to be 36.45%. The worst-case frequency
is 1.76 MHz, 2.2 MHz minus 20%. Using these numbers in
formula 11 shows that the minimum inductance for this converter
is 9.6 µH.
Output Capacitor
The converter is designed to operate with a low value ceramic
output capacitor on VREG (CVREG ). When choosing a ceramic
capacitor make sure the rated voltage is at least 3 times the
maximum output voltage of the converter. This is because the
capacitance of a ceramic decreases as it operates closer to the
capacitor rated voltage. It is recommended that the VREG output
be decoupled with a 10 µF X7R ceramic capacitor. Greater
capacitance may be required on the output if load surges dramatically influence the output voltage.
Output ripple is determined by the output capacitance and the
effects of ESR and ESL can be ignored assuming recommended
layout techniques are followed. The output voltage ripple is
approximated by:
VRIPPLE = IRIPPLE / (8 × fSW × CVREG )(12)
Input Capacitor
The value of the input capacitance affects the amount of current ripple on the input. This current ripple is usually the source
of supply-side EMI. The amount of interference will depend on
the impedance from the input capacitor and the bulk capacitance
located on the supply bus. Placing a 0.1 µF ceramic capacitor
very close to the input supply pin will help reduce EMI effects.
The small capacitor will help reduce high frequency transient currents on the supply line. If further filtering is needed it is recommended that two ceramic capacitors be used in parallel to further
reduce emissions.
Rectification Diode
The diode conducts the current during the off cycle. A Schottky
diode is required to minimize the forward drop and switching
losses. In order to size the diode correctly it is necessary to find
the average diode conduction current using the formula below :
Idiode(avg) = I load × (1 – D(min ))(13)
where D(min) is the minimum duty cycle, defined as:
D(min ) = (VVREG + Vf ) / (VIN + Vf ) (14)
where VIN is the maximum input voltage and Vf is the maximum
forward voltage of the diode.
The average power dissipation in the diode is:
PDdiode(avg) = Iavg × D(min ) × Vf (15)
The power dissipation in the sense resistor must also be considered using I2R and the minimum duty cycle.
External MOSFET Selection
To choose an external MOSFET for the 3.3 V linear regulator
consider the maximum of: drain-to-source voltage (VDS), continuous drain current (ID ), threshold voltage (VGSTH), on-resistance
(RDS(on)), and thermal resistance (RθJC ).
The buck switcher pre-regulates the voltage to the external
MOSFET, so even under worst case conditions, the MOSFET
will not have to support more than 7 V from drain to source. Also,
the 3.3 V current limit will usually be set from 200 to 500 mA
using the external current setting resistor, RCL . Numerous
MOSFETs are available with VDS ratings of at least 20 V that
can support much more than 1 A. These two goals should not be
difficult to achieve.
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19
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
The A4406 gate drive circuitry is guaranteed to pull the G33
voltage down to 1 V, maximum. Therefore, Allegro recommends
using a MOSFET with a VGS threshold (VGSTH) higher than 1 V.
Do not use a MOSFET that will conduct significant current when
VGS is at 1 V and the system is at the highest expected ambient
temperature.
One of the more critical specifications is the MOSFET onresistance, RDS(on) . If the on-resistance is too high, then the 3.3 V
regulator will not be able to maintain 3.3 V at the maximum
required load current, ILIM(V33) . Calculate the typical RDS(on) (at
25°C) using the following formula:
RDS(on)25°C < 0.6 × 1.56 (V) / (ILIM(V33) – RDROP)(16)
where ILIM(V33) is the maximum required 3.3 V output current,
and RDROP is the value of the resistor connected from the CLV33
pin to the drain of the MOSFET.
The multiplier of 0.6 in the following formula allows a 66%
increase in RDS(on) when the MOSFET is very hot:
RDS(on)25°C < 0.6 ×
1.56 (V)
– RDROP
ILIM(V33)
.
(17)
where ILIM(V33) is the maximum required 3.3 V output current.
The necessity and value of RDROP is closely related to the thermal
resistance (RθJC) of the MOSFET. For a medium size MOSFET
(such as an SOT-223) including RDROP in the PCB layout is
highly recommended. For a large size MOSFET with a very
low thermal resistance (such as a DPAK) RDROP is probably not
necessary.
MOSFET thermal resistance is a function of die size, package
size, and cost. So, choosing RDROP and RθJC together should
result in optimal performance, minimal component sizes, and
lowest system cost. Determining the value and power dissipated
by the series dropping resistor and MOSFET thermal resistance
are addressed in detail in the 3.3 V Dropping Resistor section.
3.3 V Dropping Resistor (RDROP)
In the Typical Application Circuit schematic, there are two resistors, RCL and RDROP , from the output of the buck regulator to
the drain of the external MOSFET. RCL must always be present because it sets the 3.3 V regulator current limit threshold.
However, RDROP , if used, prevents the external MOSFET from
dissipating too much power during certain conditions. In particular, when the battery voltage is extremely low (VBAT ≤ 6.5 V) and
the buck regulator transitions to dropout mode (100% duty cycle)
then VVREG will be approximately 1 V higher than normal. In this
situation, without RDROP , the MOSFET could dissipate too much
power.
The value of RDROP depends on the maximum PCB temperature,
the maximum current load on the 3.3 V regulator, ILIM(V33) , the
maximum allowable junction temperature of the MOSFET, and
the thermal resistance of the MOSFET. As the thermal resistance
of the MOSFET decreases, the required value of RDROP will also
decrease. If the MOSFET is relatively large and has a very low
thermal resistance, then RDROP is not required (0 Ω).
Figure 8 shows recommended values of RDROP versus MOSFET
thermal resistance at various 3.3 V regulator maximum current
settings ( ILIM(V33) ). This graph assumes a PCB temperature of
135°C, a maximum MOSFET junction temperature of 145°C,
VBAT of 6.5 V, and 3.23 V from the linear regulator. This graph
takes into account the voltage drop across the 3.3 V current limit
resistor, RCL .
6.5
6.0
5.5
230 mA
5.0
4.5
320 mA
4.0
RDROP (Ω)
A4406
410 mA
3.5
500 mA
3.0
2.5
140 mA
2.0
1.5
1.0
0.5
0.0
10
15
20
25
30
35
40
45
50
55
60
MOSFET Thermal Resistance (°C/W)
Figure 8. Value of RDROP versus MOSFET thermal resistance at various
V33 regulator maximum current settings, ILIM(V33)
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20
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
After a value for RDROP is determined, the designer should calculate its maximum power dissipation (I2 × R) and select an appropriate component, allowing adequate design margin. Assuming
the RDROP value was chosen referencing figure 8, then figure 9
can be used to determine the power dissipated by RDROP versus
MOSFET thermal resistance at various 3.3V regulator current
settings. The exact value of RDROP is not critical, so a component
with 1% or 5% tolerance could be used.
PCB Layout
The board layout will have a large impact on the performance of
the device. It is important to isolate high current ground returns
to minimize ground bounce that could produce reference errors in
the device. The method used to isolate power ground from noise
sensitive circuitry is to use a star ground. This approach ensures
that the high current components such as the input capacitor, output capacitor, and diode have very low impedance paths to each
other. Figure 10 illustrates the technique.
The ground connections from each of the components should be
very close to each other and be connected on the same surface as
the components. Internal ground planes should not be used for
the star ground connection, because vias add impedance to the
current path.
In order to further reduce noise effects on the PCB, noise sensitive traces should not be connected to internal ground planes.
The feedback network from the switcher output should have an
independent ground trace that goes directly to the exposed pad
underneath the device. The exposed pad should be connected
to internal ground plans and any exposed copper used for heat
dissipation. If the ground connections from the device are also
connected directly to the exposed pad, the ground reference from
the feedback network will be less susceptible to noise injection or
ground bounce.
1.0
0.9
500 mA
0.8
410 mA
0.7
PRDROP (W)
A4406
0.6
320 mA
0.5
0.4
230 mA
0.3
0.2
140 mA
0.1
0.0
10
15
20
25
30
35
40
45
50
55
60
MOSFET Thermal Resistance (°C/W)
Figure 9. RDROP Dissipation versus MOSFET thermal resistance at various
V33 regulator maximum current settings, IV33ILIM
Current path (on-cycle)
VIN
L1
LX
CIN
Q1
A4406
DBUCK
RSENSE
Current path
(off-cycle)
CVREG
RLOAD
Star Ground
Figure 10. Illustration of star ground connection
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21
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
To reduce radiated emissions from the high frequency switching
nodes, it is important to have an internal ground plane directly
under the LX node. The plane should not be broken directly
under the node because the lowest impedance path back to the
star ground would then be directly under the signal trace. If
another trace does break the return path, the energy will have to
find another path, which is through radiated emissions.
For accurate current sensing, the current sense pins, ISEN+ and
ISEN–, and the internal differential amplifier comprise a differential signal receiver, and a balanced pair of traces should be
routed from the pins of the buck current sense resistor, RSENSE ,
as shown in figure 11 (upper panel). The ISEN+ pin and the sense
resistor ground should not be separated by simply using local via
connections to the ground plane (figure 16 lower panel). Incorrect
routing of the ISEN+ pin would likely add an offset error to the
buck current sense signal.
Differential
Amplifier
–
+
L1
LX
ISEN–
DBUCK
(Asynchronous)
ISEN+
RSENSE
A4406
Correct routing of ISEN+ and ISEN– traces
(direct on same plane)
Differential
Amplifier
–
+
A4406
L1
LX
ISEN–
ISEN+
DBUCK
(Asynchronous)
RSENSE
Ground plane
Incorrect routing of ISEN+ and ISEN– traces
(using vias to a ground plane)
Figure 11. Comparison of routing paths for the traces between the A4406
ISEN+ and ISEN– traces and the sense resistor, RSENSE
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22
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Application Circuit Performance
(Refer to Typical Application Circuit diagram.)
Bill of Materials for Critical Components
This design is capable of full load, 135°C ambient, and 5.5 VBAT indefinitely with an adequate thermal solution
Component
Description
Infineon
IPD90N04S3-04
Murata
GCM32ER71H475KA55L
MOSFET, 40 V, 90 A, 4.3 mΩ, TJ 175°C
DPAK
Resistor, 0.300 Ω, 1/4 W, 1%
1206
RCL
Resistor, 0.390 Ω, 1/4 W, 1%
1206
Resistor, 1.2 Ω total, 1/2 W, 1%
1210
CIN1 , CIN2
Capacitor, Ceramic, 4.7 µF, 50 V, 10%, X7R
1210
CVREG
Capacitor, Ceramic, 10 µF, 16 V, 10%, X7R
1206
Kemet
C1206C106K4RACTU
CV33, CV5P
Capacitor, Ceramic, 2.2 µF, 16 V, 10%, X7R
1206
Murata
GRM31MR71C225KA35L
DBUCK , DIN, DV5P
Diode, Schottky, 2 A, 40 V
L1
SMA
Inductor, 10 µH, 64 mΩ, 2.39 Asat , 165°C
Buck Regulator (VREG) Efficiency
85
70
12
0
-24
60
-36
55
-48
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
120
Phase 215 mA
80
40
Phase Margin 0.8 A (59°)
0
Phase Margin 215 mA (55°)
-40
Gain 0 dB
(215 mA: at 106 kHz
0.8 mA: at 104 kHz)
-80
-120
Gain Margin 12 dB
-60
10–1
1
10
Output Current, I OUT (A)
-160
-200
103
100
Frequency (kHz)
Buck Regulator (VREG) Load Regulation
0.0
-0.05
Linear Regulator Load Regulation
-0.1
VOUT Percentage Drop (%)
-0.10
VOUT Percentage Drop (%)
160
Phase 0.8 A
-12
65
50
Gain 0.8 A
24
Gain (dB)
75
200
Gain 215 mA
36
VIN = 16 V
80
At ILOAD = 215 mA and 0.8 A
48
VIN = 12 V
90
B240A-13-F
Cooper/Bussman DRA73-100-R
Buck Regulator Bode Plots
VIN = 8 V
95
Diodes, Inc.
7.6 x 7.6 mm
60
100
0.00
Part Number
RSENSE RDROP
Efficiency (%)
Manufacturer
Phase (°)
QV33
Package
-0.15
-0.20
-0.25
VIN = 8 V
-0.30
-0.35
VIN = 12 V
-0.40
-0.50
0.10
0.20
0.30
0.40
0.50
Output Current, I OUT (A)
0.60
0.70
V5
-0.3
V5P
-0.4
-0.5
VIN = 16 V
-0.45
V33
-0.2
0.80
-0.6
0.000
0.050
0.100
0.150
0.200
0.250
0.300
Output Current, I OUT (A)
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23
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
VVREG
VVREG
VV33
VV5P
C1
C1
C2
NPOR
C3
NPOR
C2
C3
t
t
Startup at VIN = 13.5 V; shows VVREG (ch1, 2 V/div.), VV33 (ch2, 2 V/div.),
NPOR (ch3, 2 V /div.), t = 5 ms/div.
Startup at VIN = 13.5 V; shows VVREG (ch1, 2 V/div.), VV5P (ch2, 2 V/div.),
NPOR (ch3, 2 V /div.), t = 5 ms/div.
VVREG
VVREG
VV33
VV5P
C1
C1
C2
NPOR
C3
C3
t
Startup at VIN = 6.5 V; shows VVREG (ch1, 2 V/div.), VV33 (ch2, 2 V/div.),
NPOR (ch3, 2 V /div.), t = 5 ms/div.
NPOR
C2
t
Startup at VIN = 6.5 V; shows VVREG (ch1, 2 V/div.), VV5P (ch2, 2 V/div.),
NPOR (ch3, 2 V /div.), t = 5 ms/div.
VVREG
C1
VREG
VLX
C1
C2
VLX
IL
C2
IL
C3
t
PWM at VBAT = 12 V with a VREG 25 mA load; shows VVREG (ch1,
5 V/div.), VLX (ch2, 5 V/div.), IL (ch3, 100 mA/div.), t = 2 µs/div.
C3
t
PWM at VBAT = 12 V with a VREG 0.8 A load; shows VVREG (ch1,
5 V/div.), VLX (ch2, 5 V/div.), IL (ch3, 500 mA/div.), t = 500 ns/div.
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24
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
VV33
C1
VV5P
C1
IV5P
IV33
C2
C2
t
t
VV5P Transient Response: 125 mA to 250 mA; shows VV5P (ch1,
50 mV/div.), IV5P (ch2, 100 mA/div.), t = 50 µs/div.
VV33 Transient Response: 125 mA to 250 mA; shows VV33 (ch1,
50 mV/div.), IV33 (ch2, 100 mA/div.), t = 50 µs/div.
Before Overcurrent
After Overcurrent
IL
IL
VVREG
C1
VVREG
IL
VVREG
C2
t
VREG Short Circuit Operation, VIN = 12 V; shows VVREG (ch1, 2 V/div.),
IL (ch2, 500 mA/div.), t = 5 µs/div.
C1
C1
C2
C2
t
VREG Normal and Overloaded Operation, VIN = 12 V; shows IL (ch1,
250 mA/div.), VVREG (ch2, 2 V/div.)
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25
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
A4406
Package LP, 20-Pin TSSOP
with Exposed Thermal Pad
0.45
6.50±0.10
8º
0º
20
0.65
20
0.20
0.09
1.70
C
3.00
4.40±0.10
6.40±0.20
3.00
6.10
0.60 ±0.15
A
1
1.00 REF
2
4.20
0.25 BSC
20X
SEATING
PLANE
0.10 C
0.30
0.19
C
SEATING PLANE
GAUGE PLANE
1
2
4.20
B
PCB Layout Reference View
1.20 MAX
0.65 BSC
0.15
0.00
For Reference Only; not for tooling use (reference MO-153 ACT)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
A Terminal #1 mark area
B Reference land pattern layout (reference IPC7351
SOP65P640X110-21M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as
necessary to meet application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal vias at the
exposed thermal pad land can improve thermal dissipation (reference
EIA/JEDEC Standard JESD51-5)
C Exposed thermal pad (bottom surface)
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26
A4406
Constant On-Time Buck Regulator
With One External and One Internal Linear Regulator
Revision History
Revision
Revision Date
Rev. 2
February 11, 2013
Description of Revision
Update typical application and asynchronous
diode description
Copyright ©2012-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
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27