A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR FEATURES AND BENEFITS • • • • • • • • • • • • • • • • • • • • • • AEC-Q100 Qualified Operating voltage range: 2.5 to 5.5 V UVLO stop threshold: 2.25 V (max) Dual outputs with up to 2 A output current per regulator Adjustable output voltage as low as 0.8 V Internal 80 mΩ high-side switching MOSFET Internal 55 mΩ low-side switching MOSFET Adjustable switching frequency ( fSW ): 0.35 to 2.2 MHz Synchronizes to external clock: 1.2 × to 1.5 × fOSC 180° phase shift between switching regulators Sleep mode supply current less than 5 μA Soft start time externally set via the SS pin Pre-biased startup capable Externally adjustable compensation Stable with ceramic output capacitors Independent enable inputs and NPOR output pins NPOR delay of 7.5 ms (A8651) or 120 µs (A8651-1) Adjustable current limiting (OCP) for each regulator Hiccup mode short-circuit protection (HIC) Overvoltage and overtemperature protection Open-circuit and adjacent pin short-circuit tolerant Short-to-ground tolerant at every pin Package: 20-pin TSSOP with exposed thermal pad (suffix LP) DESCRIPTION The A8651 is an adjustable frequency, high output current, PWM regulator that integrates a high-side, P-channel MOSFET and a low-side, N-channel MOSFET. The A8651 incorporates current-mode control to provide simple compensation, excellent loop stability, and fast transient response. The A8651 utilizes external compensation to accommodate a wide range of power components to optimize transient response without sacrificing stability. The A8651 regulates input voltages from 2.5 to 5.5 V, down to output voltages as low as 0.8 V and is able to supply up to 2 A of load current per regulator. The A8651 features include an externally adjustable and synchronizable switching frequency, an externally-set soft start time to minimize inrush currents, independent EN inputs, and independent NPOR outputs with either 7.5 ms (A8651) or 120 µs (A8651-1) delay. The sleep mode current of the A8651 control circuitry is less than 5 μA. Protection features include VIN undervoltage lockout (UVLO), cycle-by-cycle overcurrent protection (OCP), hiccup mode short-circuit protection (HIC), overvoltage protection (OVP), and thermal shutdown (TSD). In addition, the A8651 provides open-circuit, adjacent pin short-circuit, and short-to-ground protection at every pin to satisfy the most demanding automotive applications. Continued on the next page… APPLICATIONS: • • • • Not to scale 5 SYNCin CSYNC GPS/Infotainment Automotive audio Home audio Network and telecom FSET/SYNC CIN1 EN1 R Z1 C SS1 C P1 R Z2 C SS2 C P2 C Z2 A8651 RFB2 RPU1 NPOR1 LO2 VOUT2 CO2 FB2 14 RFB3 RFB4 RPU2 6 NPOR2 RSET2 Typical Application Diagram A8651-DS, Rev. 6 NPOR1 15 VIN2 9 PGND2 8 EN2 13 COMP2 12 SS2 7 ISET2 EN2 FB1 RFB1 SW2 11 CIN2 CO1 18 10 RSET1 C Z1 VOUT1 SW1 20 VIN1 16 GND 2 PGND1 3 EN1 17 COMP1 19 SS1 4 ISET1 VIN LO1 1 RFSET NPOR2 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 DESCRIPTION (CONTINUED) The A8651 device is available in a 20-pin TSSOP package with exposed thermal pad for enhanced thermal dissipation (suffix LP). It is lead (Pb) free, with 100% matte tin leadframe plating. Selection Guide Part Number Operating Ambient Temperature Range TA, (°C) Package Packing* Leadframe Plating –40 to 125 20-pin TSSOP with exposed thermal pad 4000 pieces per 13-in. reel 100% matte tin A8651KLPTR-T A8651KLPTR-T-1 *Contact Allegro™ for additional packing options. Specifications Absolute Maximum Ratings Thermal Characteristics Top Level Functional Block Diagram Detailed Functional Block Diagram Pin-out Diagram and Terminal List Top Level Functional Block Diagram Electrical Characteristics Characteristic Performance Timing Diagram Functional Description Overview Reference Voltage Oscillator/Switching Frequency (RFSET, fOSC) Transconductance Error Amplifier Slope Compensation ENX, VINX, and Sleep Mode Synchronization (FSET/SYNC) Power MOSFETs Pulse Width Modulation (PWM) Current Sense Amplifier Table of Contents 2 3 3 4 5 6 6 7 10 12 13 13 13 13 13 13 14 14 14 14 14 Soft Start (Startup) and Inrush Current Control Pre-Biased Startup Active Low Power-On Reset (NPORx) Protection Features Undervoltage Lockout (UVLO) Thermal Shutdown (TSD) Overvoltage Protection (OVP) Pulse-by-Pulse Overcurrent Protection (OCP) Output Short Circuit (Hiccup Mode) Protection Design and Component Selection 14 15 16 16 16 16 16 16 17 20 20 20 21 22 22 23 24 26 Power Dissipation and Thermal Calculations 28 PCB Component Placement and Routing 30 Package Outline Diagram 33 Setting the Output Voltage (VOUT1 , RFBx ) PWM Switching Frequency (RFSET) Output Inductor (LO ) Output Capacitors Input Capacitors Soft Start and Hiccup Mode Timing (CSS1) Compensation Components (RZ, CZ, CP) A Generalized Tuning Procedure Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 2 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR SPECIFICATIONS Absolute Maximum Ratings1 Characteristic VIN1 and VIN2 to GND SW1 and SW2 to GND2 All Other Pins Symbol Notes Rating VIN VSW Unit –0.3 to 6.0 V Continuous –0.3 to VIN + 0.3 V t < 50 ns –1.0 to VIN +2.0 V –0.3 to 6.0 V – Operating Ambient Temperature TA –40 to 125 ºC Maximum Junction Temperature TJ(max) 150 ºC Tstg –55 to 150 ºC Storage Temperature K temperature range 1Operation at levels beyond the ratings listed in this table may cause permanent damage to the device. The Absolute Maximum ratings are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the Electrical Characteristics table is not implied. Exposure to Absolute Maximum-rated conditions for extended periods may affect device reliability. 2SW1 and SW2 have internal clamp diodes to GND and V . Applications that forward bias these diodes should take care not to exceed the A8651 package power IN dissipation limits. Thermal Characteristics may require derating at maximum conditions, see application information Characteristic Package Thermal Resistance Symbol RθJA Test Conditions* On 4-layer PCB based on JEDEC standard Value Unit 32 ºC/W *Additional thermal information available on the Allegro website. Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 3 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 A8651 VIN 1 VIN1 UVLO GND VIN1 START VIN1 STOP EN1 ISENSE1 SWITCHER #1 (SW1) COMP1 ADJUSTABLE VOUT SS1 ISET1 SYNCHRONOUS BUCK VREF & POR & VREF POR 2 80mΩ SW1 55mΩ PGND1 FB1 CLK0° TSD Oscillator 180° shift w/ SYNC FSET / SYNC NPOR1 VIN2 VIN2 UVLO VIN2 START VIN2 STOP EN2 ISENSE2 SWITCHER #2 (SW2) COMP2 ADJUSTABLE VOUT SS2 ISET 2 2 SYNCHRONOUS BUCK 80mΩ SW2 55mΩ PGND2 FB2 CLK180° TSD TSD NPOR2 Note: VIN1 supplies the bandgap(V REF ), oscillator , TSD , and other critical circuits Top Level Functional Block Diagram Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 4 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 A8651 VINx ISETx GAIN CURRENT VIN ADJ Oscillator CLKx NON-OVERLAP Slope Comp CLKin 80m OFF OCP ADJ1 Fsw CLAMP ACTIVE RESET DOM ADJ2 PWM COMP ADJ3 S Q R Q SWx VIN - IF VFB > 400mV, Fsw = CLKx IF VFB < 400mV, Fsw = CLKx/2 IF VFB < 400mV & CLAMP ACTIVE & OCP, Fsw = CLKx/4 55m + OFF MIN_Toff PWMoffset 380mV VREF PGNDx VSSoffs 200mV 150nA ERROR AMP Clamp CLAMP ACTIVE + COMPx + FBx - 1.5K TSD POR OFF EN ENx CURRENT + 10ns 100K ENx OCP - LATCHED 1.65Vtyp 1.25Vtyp FBx UVLO HICCUP HIC RST SSx PROTECTIONS 10uA FAULT FAULT = 1 IF: EN = 0 or UVLO = 1 EN_OCP_COUNT UVLOx 2K OCP EN / CLR 20uA HICCUP HICCUP = 1 IF: FB < 700mV and 7 OCP EVENTS GNDx NPORx OFF OFF + 125ns UV / OV OVP COMP OV UV COMP UV 115% x VREF 111% x VREF FBx + Falling Delay 92% x VREF 88% x VREF Pad 1 of 2 regulators shown Detailed Functional Block Diagram Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 5 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 Pin-out Diagram and Thermal Characteristics SW1 1 20 VIN1 PGND1 2 19 SS1 EN1 3 18 FB1 ISET1 4 FSET/SYNC 5 NPOR2 6 17 COMP1 PAD ISET2 7 EN2 8 PGND2 9 SW2 10 16 GND 15 NPOR1 14 FB2 13 COMP2 12 SS2 11 VIN2 Package LP, 20-Pin TSSOP with Exposed Thermal Pad Pin-out Diagram Terminal List Table Number Name Function 1, 10 SW1, SW2 2, 9 PGND1, PGND2 3, 8 EN1, EN2 4, 7 ISET1, ISET2 Pulse–by-pulse current limit setting pins. 5 FSET/SYNC A resistor, RFSET, from this pin to GND sets the base PWM switching frequency (fOSC). If an external clock is ACcoupled to this pin by a 22 pF capacitor, the switching frequency of the regulator can be increased higher than fOSC. 6, 15 NPOR2, NPOR1 11, 20 VIN2, VIN1 12, 19 SS2, SS1 13, 17 COMP2, COMP1 14, 18 FB2, FB1 16 GND Ground. – PAD Exposed pad of the package providing enhanced thermal dissipation. This pad must be connected to the ground plane(s) of the PCB with at least 6 vias, directly in the pad. The drains of the internal high-side P-channel MOSFETs. The output inductors should be connected to these pins. The output inductors should be placed as close as possible to these pins and be connected with relatively wide traces. Power ground pins for switcher 1 and switcher 2. Inputs to enable switcher 1 and/or enable switcher 2. Active low, open-drain fault indication outputs, with fixed delay. Power inputs for the control circuits and the sources of the internal high-side P-channel MOSFETs. VIN1 is the primary supply and must be present for the A8651 to operate. At least one high quality ceramic capacitor must be placed very close to these pins. Soft-start pins. Connect a capacitor, from these pins to GND to set the soft-start time. These capacitors also determine the hiccup period during an overcurrent condition. Outputs of the error amplifiers and compensation nodes for the current mode control loops. Connect a series RC network from these pins to GND for loop compensation. See the Design and Component Selection section of this datasheet for further details. Feedback (negative) inputs to the error amplifiers. Connect a resistor divider from the converter output nodes to these pins to program the output voltages. Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 6 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR ELECTRICAL CHARACTERISTICS: valid at VIN1 = VIN2 = 5 V, –40°C ≤ TA = TJ ≤ 125°C; unless otherwise specified Characteristic Symbol Test Conditions Min. Typ. Max. Unit 2.5 − 5.5 V Input Voltage Specifications Operating Input Voltage Range VIN Undervoltage Lockout (UVLO) Start Threshold VINSTART VIN1 = VIN2, rising 2.00 2.22 2.45 V Undervoltage Lockout (UVLO) Stop Threshold VINSTOP VIN1 = VIN2, falling 1.80 2.02 2.25 V Undervoltage Lockout (UVLO) Hysteresis VUVLO(HYS) − 200 − mV VEN1 = VEN2 = 5 V, VFB1 = VFB2 = 1.0 V, no PWM switching − 3 6 mA VINx = VSWx = 5 V, VEN1 = VEN2 ≤ 0.4 V − 0.2 5 μA 792 800 808 mV – –150 –300 nA Input Currents Input Quiescent Current Input Sleep Supply Current IQ IQSLEEP Reference Voltage Reference (Feedback) Voltage VREF 2.5 V < VIN1 = VIN2 < 5.5 V, VFBx = VCOMPx Error Amplifier Feedback Input Bias Current(1) Open Loop Voltage Gain(2) Transconductance Source Current Sink Current Maximum Output Voltage COMP Pull Down Resistance IFB VCOMPx = 1.5 V, VFBx regulated so that ICOMPx = 0 A AVOL gm ICOMPx = 0 μA, VSSx > 500 mV − 65 − dB 550 750 950 μA/V 0 V < VSSx < 500 mV – 250 – μA/V IEA(SRC) VFBx < 0.8 V, VCOMPx = 1.5 V − –50 − μA IEA(SINK) VFBx > 0.8 V, VCOMPx = 1.5 V VEAVO(max) RCOMP FAULT = 1, HICCUP = 1 or VEN1 = VEN2 ≤ 0.4 V − +50 − μA 1.00 1.25 1.50 V − 1.5 − kΩ − 380 − mV − 65 105 ns − 50 100 ns Pulse Width Modulation (PWM) PWM Ramp Offset VPWMOFFSET VCOMPx for 0% duty cycle High-Side MOSFET Minimum Controllable On-Time tON(MIN) Low-Side MOSFET Minimum On‑Time tOFF(MIN) Gate Driver Non-Overlap Time(2) COMP to SW Current Gain Slope Compensation(2) Does not include total gate driver non-overlap time, 2 x tOFF tOFF − 15 − ns gmPOWER − 4.5 − A/V SE RSETx = 41.2 kΩ, fSW = 2.0 MHz 2.1 2.5 2.9 A/μs RSETx = 41.2 kΩ, fSW = 0.35 MHz 0.36 0.44 0.51 A/μs RSETx = 30.9 kΩ, fSW = 2.0 MHz 1.0 1.4 1.9 A/μs RSETx = 30.9 kΩ, fSW = 0.35 MHz 0.17 0.25 0.35 A/μs Note 1: For input and output current specifications, negative current is defined as coming out of the node or pin (sourcing), positive current is defined as going into the node or pin (sinking). Note 2: Ensured by design and characterization, not production tested. Continued on the next page… Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 7 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 ELECTRICAL CHARACTERISTICS (continued): valid at VIN1 = VIN2 = 5 V, –40°C ≤ TA = TJ ≤ 125°C; unless otherwise specified Characteristic Symbol Test Conditions Min. Typ. Max. Unit − 80 − mΩ − 12 − ns VENx ≤ 0.4 V, VSWx = 0 V, VINx = 5 V, –40˚C < TA = TJ < 85˚C(3) − − 4 μA VENx ≤ 0.4 V, VSWx = 0 V, VINx = 5 V, TA = TJ = 125˚C − − 25 μA IDSx = 100 mA − 55 − mΩ VENx ≤ 0.4 V, VSWx = 5 V, –40˚C < TA = TJ < 85˚C (3) − − 1 μA VENx ≤ 0.4 V, VSWx = 5 V, TA = TJ = 125˚C − − 10 μA RFSET = 10.2 kΩ 1.98 2.20 2.45 MHz RFSET = 24.9 kΩ 0.90 1.00 1.10 MHz RFSET = 82.5 kΩ 325 375 425 kHz − 180 − deg. MOSFET Parameters High-Side MOSFET On-Resistance SW Node Rise Time(2) High-Side MOSFET Leakage Current Low-Side MOSFET ON Resistance Low-Side MOSFET Leakage Current RDS(on)HS IDSx = 100 mA tr(SW) IDSS(HS) RDS(on)LS IDSS(LS) Oscillator Frequency Oscillator Frequency SW1 to SW2 Phase Delay(2) fOSC Φ1,2 FSET/SYNC Input FSET/SYNC High Threshold VFSETSYNC(H) − − 1.8 V FSET/SYNC Low Threshold VFSETSYNC(L) 0.4 − − V FSET/SYNC Pin Voltage VFSETSYNC Without external SYNCin signal − 0.8 − V FSET/SYNC Pin Current IFSETSYNC 9 − 90 µA fSYNCM − − 2.5 MHz fSYNC 1.2 × fOSC − 1.5 × fOSC – Maximum SYNC Frequency SYNC Frequency Range(2) SYNC Duty Cycle (2) Without external SYNCin signal DSYNC 50 60 70 % Synchronization Minimum On-Time tONSYNC 150 − − ns Synchronization Minimum Off-Time tOFFSYNC 150 − − ns Enable Inputs EN High Threshold VENIH VENx rising − − 1.8 V EN Low Threshold VENIL VENx falling 0.8 − − V EN Hysteresis EN Input Resistance EN Shutdown Delay(2) VENHYS VENIH – VENIL REN tdEN(SD) From ENX transitioning low to SWx switching stops − 200 − mV 50 100 − kΩ 0 5 10 µs Note 2: Ensured by design and characterization, not production tested. Note 3: Specifications at 25°C or 85°C are ensured by design and characterization, not production tested at these temperatures. Continued on the next page… Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 8 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR ELECTRICAL CHARACTERISTICS (continued): valid at VIN1 = VIN2 = 5 V, –40°C ≤ TA = TJ ≤ 125°C; unless otherwise specified Characteristic Symbol Test Conditions Min. Typ. Max. Unit Overcurrent Protection (OCP) and Hiccup Mode Pulse-by-Pulse Current Limit ILIM RSET = 41.2 kΩ, duty cycle = 5% 3.5 4.1 4.7 A RSET = 41.2 kΩ, duty cycle = 90%(2) 2.2 3.0 3.8 A RSET = 30.9 kΩ, duty cycle = 5% 1.9 2.4 2.9 A RSET = 30.9 kΩ, duty cycle = 90%(2) 1.1 1.8 2.3 A Hiccup Disable Threshold VHICDIS VFBx rising − 740 − mV Hiccup Enable Threshold VHICEN VFBx falling − 700 − mV OCPLIMIT HICCUP enabled (see Functional Block diagram), OCP pulses − 7 − counts Soft Start Offset Voltage VSSOFFS VSSx rising due to ISSSU 100 200 270 mV Soft Start Fault/Hiccup Reset Voltage VSSRESET VSSx falling due to ISSHIC − 120 185 mV −10 –20 −30 μA OCP / HICCUP Count Limit2 Soft Start (SS pin) Soft Start Startup (Source) Current ISSSU VSSx = 1 V, HICCUP = FAULT = 0 (see Functional Block diagram) Soft Start Hiccup (Sink) Current ISSHIC VSSx = 0.5V, HICCUP = 1 (see Functional Block diagram) 5 10 20 μA FAULT (see Functional Block diagram) = 1 or ENx = 0 − 2 − kΩ tSS(DELAY) CSSx = 10 nF − 85 − μs tSS CSSx = 10 nF − 400 − μs 0 V < VFBx < 400 mV, VCOMPx = VEAVO(max) , IDSx > ILIMx (2) − fOSC / 4 − − 0 V < VFBx < 400 mV − fOSC / 2 − − VFBx > 400 mV − fOSC − − Percentage of VREF , VFBx rising 89 92 95 % Soft Start Input Resistance Soft Start to VOUT Delay Time VOUT Soft Start Ramp Time Soft Start Switching Frequency RSS fSW(SS) NPOR Outputs NPOR Undervoltage Threshold NPOR Undervoltage Hysteresis NPOR Overvoltage Threshold NPOR Overvoltage Hysteresis NPOR Rising Delay (A8651) NPOR Rising Delay (A8651-1) VNPORUV VNPORUVhys Percentage of VREF , VFBx falling VNPOROV Percentage of VREF , VFBx rising VNPOROVhys Percentage of VREF , VFBx falling tNPOR NPOR Low Output Voltage VNPOR(L) NPOR Leakage Current(1) INPOR(LEAK) 2.5 V < VIN1 = VIN2 < 5 V, INPOR = 4 mA 2 4 6 % 112 115 118 % 2 4 6 % 4.0 7.5 11 ms 65 120 175 µs — — 400 mV VIN1 = VIN2 =1.2 V, INPOR = 2 mA — — 800 mV VNPORx = 3.3 V — — 1 µA Thermal Protection (TSD) Thermal Shutdown Threshold(2) TSD(th) Temperature rising 155 170 185 ºC Thermal Shutdown Hysteresis(2) TSD(HYS) Temperature falling − 20 − ºC Note 1: For input and output current specifications, negative current is defined as coming out of the node or pin (sourcing), positive current is defined as going into the node or pin (sinking). Note 2: Ensured by design and characterization, not production tested. Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 9 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 Characteristic Performance Oscillator Frequency versus Temperature 806 804 802 800 798 796 794 792 -50 -25 0 25 50 75 100 125 150 Oscillator Frequency, fOSC (MHz) Reference Voltage, VREF (mV) Reference Voltage versus Temperature 808 2.75 2.50 2.25 f OSC = 2.2 MHz 2.00 1.75 1.50 1.25 f OSC = 1 MHz 1.00 0.75 -50 -25 0 Temperature (°C) 25 50 75 100 125 150 Temperature (°C) NPOR Overvoltage and Undervoltage Thresholds versus Temperature VIN UVLO Start and Stop Thresholds versus Temperature 2.5 120 2.4 115 2.3 Threshold (% of VFB) Input Voltage, VIN (V) VNPOROV, VF B rising UVLO Start Threshold, VINSTART 2.2 2.1 UVLO Stop Threshold, VINSTOP 2.0 1.9 1.8 1.7 110 VNPOROV, VF B f alling 105 100 VNPORUV, VF B rising 95 90 VNPORUV, VF B f alling 85 -50 -25 0 25 50 75 100 125 80 150 -50 -25 0 Temperature (°C) 75 100 125 150 125 150 950 Transconductance, gm (µA/V) 4.5 4.4 Current Limit , ILIM (A) 50 Error Amplifier Transconductance versus Temperature Pulse-by-Pulse Current Limit at tON(MIN) versus Temperature 4.3 4.2 4.1 4.0 3.9 3.8 3.7 3.6 3.5 25 Temperature (°C) -50 -25 0 25 50 75 Temperature (°C) 100 125 150 900 850 800 750 700 650 600 550 -50 -25 0 25 50 75 100 Temperature (°C) Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 10 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 Enable High and Low Thresholds versus Temperature Soft Start Startup and Hiccup Currents versus Temperature 25 1.8 23 1.7 SS Pin Current (µA) EN Pin Threshold (V) EN High Threshold, VENIH 1.6 1.5 1.4 1.3 EN Low Threshold, VENIL 1.2 1.1 1.0 SS Startup Current, ISSSU 21 19 17 15 13 11 9 SS Hiccup Current, ISSHIC 7 -50 -25 0 25 50 75 Temperature (°C) 100 125 5 150 -50 75 100 125 150 825 Hiccup Threshold (mV) NPOR Low Output Voltage, VNPOR(L) (mV) 50 850 0.35 0.30 0.25 INPOR = 4 mA 0.20 0.15 INPOR = 2 mA 0.10 0.05 800 775 Hiccup Disable Threshold, VHICDIS 750 725 700 675 Hiccup Enable Threshold,VHICEN 650 625 600 575 -25 0 25 50 75 Temperature (°C) 100 125 150 550 Input Sleep Supply Current versus Temperature -50 -25 0 25 50 75 Temperature (°C) 100 125 150 High- and Low-Side MOSFETs Leakage Current versus Temperature 4.0 40 3.5 35 Leakage Current (µA) Input Sleep Supply Current, IQSLEEP (µA) 25 Hiccup Enable and Disable Thresholds versus Temperature 0.40 3.0 2.5 2.0 1.5 1.0 30 25 20 15 High-Side MOSFET Leakage Curent, IDSS(HS) 10 Low-Side MOSFET Leakage Curent, IDSS(LS) 5 0.5 0 -50 0 Temperature (°C) NPOR Low Output Voltage versus Temperature 0 -50 -25 0 -25 0 25 50 75 Temperature (°C) 100 125 150 -50 -25 0 25 50 75 100 125 150 Temperature (°C) Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 11 NPORx COMPx SSx VOUTx SWx VINx OCx HICCUPx HIC _ENx TSD ENx MODE F / 2 SS NPOR set below VINSTOP OFF/UVLO tNPOR HICDIS Fsw VINSTART PWM F / 4 x7 O C F / 4 x7 S O S C Vout shorted to GND PWMOFFSET EAVO( max) VSS,OFFS VSS,RESET HICCUP HICCUP F / 2 SS F / 2 SS EN glitches low for more than tdEN,SD HICEN Fsw ENVIL PWM OFF/DISABLED Fsw VIN dropout VINSTOP PWM OFF/UVLO F / 2 SS Fsw PWM TSD OFF/TSD F / 2 SS Fsw PWM tdEN,SD OFF/DISABLED A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR Timing Diagram (one of two regulators shown) Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 12 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR FUNCTIONAL DESCRIPTION Overview The A8651 is a dual synchronous PWM regulator that incorporates all the control and protection circuitry necessary to satisfy a wide range of applications. The A8651 employs current mode control to provide fast transient response, simple compensation, and excellent stability. The features of the A8651 include, for each of the two regulators: a precision reference, an adjustable switching frequency, a transconductance error amplifier, an enable/synchronization input, integrated high-side and low-side MOSFETs, adjustable Soft Start, pre-bias startup, low current sleep mode, and a Power-On Reset output (NPOR). The protection features of the A8651 include undervoltage lockout (UVLO), pulse-by-pulse overcurrent protection (OCP), hiccup mode short-circuit protection (HIC), overvoltage protection (OVP), and thermal shutdown (TSD). In addition, the A8651 provides opencircuit, adjacent pin short-circuit, and pin-to-ground short circuit protection. Reference Voltage The A8651 incorporates an internal reference that allows output voltages as low as 0.8 V. The accuracy of the internal reference is ±1% across the operating temperature range. The output voltage for each of the regulators is adjusted by connecting a resistor divider (RFB1‑RFB2 and RFB3‑RFB4 in the Typical Application diagram) from VOUTx to the corresponding FBx pin of the A8651. Oscillator/Switching Frequency (RFSET, fOSC) The PWM switching frequency of the A8651 is adjustable from 350 kHz to 2.2 MHz and has an accuracy of about ±10% across the operating temperature range. Connecting a resistor ( RFSET ) from the FSET/SYNC pin to GND, as shown in the Typical Application diagram, sets the base switching frequency, fOSC . An FSET/SYNC resistor with ±1% tolerance is recommended. A graph of switching frequency versus RFSET resistor value is shown in the Design and Component Selection section of this datasheet. inputs are used for soft start and regulation. The error amplifier performs an “analog OR” selection between its two positive inputs. The error amplifier regulates to either the soft start pin voltage minus 200mV or the A8651’s internal reference, whichever is lower. To stabilize the regulator, a series RC compensation network (RZ and CZ ) must be connected from the error amplifier output (COMPx pin) to GND as shown in the Typical Applications diagram. In some applications, an additional, low value capacitor (CP ) may be connected in parallel with the RZ-CZ compensation network to reduce the loop gain at higher frequencies. However if the CP capacitor is too large, the phase margin of the converter may be reduced. If the regulator is disabled or a fault occurs, the corresponding COMPx pin is immediately pulled to GND via approximately 1.5 kΩ and PWM switching is inhibited. During startup (VSSx < 500 mV) the transconductance of the error amplifier is reduced to approximately one-third of the normal operating level to minimize transients when the system is requesting on-times less than or equal to the minimum controllable on-time. Slope Compensation The A8651 incorporates internal slope compensation to allow PWM duty cycles above 50% for a wide range of input/output voltages, switching frequencies, and inductor values. As shown in the Detailed Functional Block diagram, the slope compensation signal is added to the sum of the current sense and PWM Ramp Offset (VPWMOFFSET ). The amount of slope compensation is scaled directly with the switching frequency. 200 m V SSx + + Transconductance Error Amplifier The transconductance error amplifier’s primary function is to regulate the converter’s output voltage. The error amplifier for one of the regulators is shown in Figure 1. It is shown as a threeterminal input device with two positive and one negative input. The negative input is simply connected to the FBx pin and is used to sense the feedback voltage for regulation. The two positive Error Amplifier VREF 800 m V COMPx - FBx Figure 1: The A8651 Error Amplifier (for one regulator) Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 13 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR ENX, VINX, and Sleep Mode Power MOSFETs The A8651 provides two independent inputs, VIN1 and VIN2, to sequence the output voltages after the A8651 is powered up. However, VIN1 is the primary supply input. VIN1 must be greater than VINSTART or the A8651 will not come out of sleep mode. VIN2 can start or stop regulator 2 but cannot wake up the A8651. The A8651 regulators each include an 80 mΩ, high-side P-channel MOSFET capable of delivering up to 4.1 A at a 5% duty cycle. The A8651 regulators also each include a 55 mΩ, low-side N-channel MOSFET to provide synchronous rectification. If the voltage at EN1 or EN2 is driven below VENIL (800 mV) for more than tdEN(SD) (approximately 5 µs) the regulator stops switching. In sleep mode (EN1< VENVIL ) for more than tdEN(SD) the control circuits are de-biased and draw less than 5 μA from VIN. However, the total current drawn by the VIN pin will be the sum of the current drawn by the control circuitry ( IQSLEEP ) plus any leakage due to the high-side MOSFETs ( IDSS(HS) ). Synchronization (FSET/SYNC) By using a 22 pF capacitor (CSYNC ) to AC-couple an external clock to the FSET/SYNC pin, as shown in Figure 2, the switching frequency of the A8651 can by increased from 1.2 × fOSC to 1.5 × fOSC. In these equations, fOSC is the typical frequency determined by the RFSET resistor. The applied SYNC waveform must satisfy the conditions shown in the Electrical Characteristics table. The SYNC waveform must simultaneously satisfy three conditions for all operating frequencies: duty cycle (DSYNC), minimum on-time (tONSYNC), and minimum off-time (tOFFSYNC). At relatively low frequencies (<1 MHz), the duty cycle will be the primary specification to satisfy; however, at higher frequencies (>2 MHz), the minimum on-time and minimum off-time will be the primary specifications to satisfy. SYNCin CSYNC 22 pF FSET/SYNC RFSET Figure 2: FSET/SYNC AC-Coupling The low-side MOSFET continues to conduct when the inductor current crosses zero to maintain constant conduction mode (CCM). This helps minimize EMI/EMC for noise sensitive applications by eliminating the SW high-frequency ringing associated with discontinuous conduction mode (DCM). When the A8651 is disabled, via the ENx input or a fault condition, the A8651 output stage is tri-stated by turning off both the high-side and low-side MOSFETs. Pulse Width Modulation (PWM) A high-speed PWM comparator, capable of pulse widths less than 105 ns, is included in each A8651 regulator. The inverting input of the comparator is connected to the output of the error amplifier. The non-inverting input is connected to the sum of the current sense signal, the slope compensation, and the PWM Ramp Offset (VPWMOFFSET ). At the beginning of each PWM cycle, the CLK signal sets the PWM flip-flop and the upper MOSFET is turned on. When the summation of the DC offset, the current sense signal, and the slope compensation rises above the error amplifier voltage, the comparator resets the PWM flip-flop and the high-side MOSFET is turned off. If the output voltage of the error amplifier drops below the PWM Ramp Offset (VPWMOFFSET ) then a zero percent PWM duty cycle (pulse skipping) operation is achieved. Current Sense Amplifier A high-bandwidth current sense amplifier monitors the current in the high-side MOSFETs. The PWM comparator, the pulseby-pulse current limiter, and the hiccup mode up/down counter require the current signal. Soft Start (Startup) and Inrush Current Control Inrush currents to the converter are controlled by the soft start function of the A8651. When the A8651 is enabled and all faults are cleared, the soft start (SSx) pins source approximately 20 μA (ISSSU ) and the voltage on the soft start capacitors (CSSx ) ramp Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 14 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 upward from 0 V. When the voltage on a soft start pin exceeds the Soft Start Offset Voltage (VSSOFFS , typically 200 mV measured at the SSx pin) the output of the error amplifier is released and shortly thereafter the high-side and low-side MOSFETs begin switching. As shown in figure 3, there is a short delay (tSS(DELAY) ) between when the enable (ENx) pin transitions high and when the soft start voltage reaches 200 mV to initiate PWM switching. When the A8651 begins PWM switching, the error amplifier regulates the voltage at the FBx pin to the soft start (SSx) pin voltage minus the soft start offset voltage (VSSOFFS ). When PWM switching starts, the voltage at the SSx pin rises from 200 mV to 1000 mV, a difference of 800 mV, the voltage at the FBx pin rises from 0 V to 800 mV, and the regulator output voltage rises from 0 V to the target setpoint determined by the feedback resistor divider (RFB1-RFB2 or RFB3-RFB4). When the voltage at the soft start pin reaches approximately 1000 mV the error amplifier begin regulating using the A8651 internal reference, 800 mV. The voltage at the soft start pin continues to rise to approximately VIN . The soft start functionality is shown in Figure 3. If the A8651 is disabled or a fault occurs, the internal fault latch is set and the soft start (SSx) pin is pulled to ground via approximately 2 kΩ. The A8651 clears the internal fault latch when the voltage at the SSx pin decays to approximately 120 mV (VSSRESET). If the A8651 enters hiccup mode, the capacitor (CSSx ) on the soft start pin is discharged by a 10 μA current sink (ISSHIC ). Therefore, the soft start pin capacitor value (CSSx ) controls the time between soft start attempts. Hiccup mode operation is discussed in more detail in the Output Short Circuit (Hiccup Mode) Protection section of this datasheet. When FBx > 400 mV the PWM switching frequency is fSW . If FBx < 300 mV the PWM switching frequency is reduced to fSW / 2 to provide the low duty cycles and accurate, stable control required during initial startup (when VOUT ≈ 0 V). Also, if FBx < 400 mV and COMPx = VEAVO(max) , it can be assumed the regulator output is shorted to ground. In this case the PWM switching frequency is further reduced to only fSW / 4 to allow more off-time between PWM pulses. This is done to prevent stair-casing of the output inductor current, which could result in damage to the inductor or the A8651. This is especially important when the input voltage is relatively high and the output of the regulator is either shorted or soft starting a relatively high output capacitance. Pre-Biased Startup If the output capacitors are pre-biased to some voltage, the A8651 modifies the normal startup routine to prevent discharging the output capacitors. Normally, the COMPx pin becomes active and PWM switching starts when the voltage at the soft start (SSx) pin reaches 200 mV. With pre-bias at the output, the pre-bias voltage is sensed at the FBx pin. The A8651 does not start switching until the voltage at the soft-start pin increases to approximately VFBx + 200 mV. At this soft start pin voltage, the error amplifier output is released, the voltage at the COMPx pin rises, PWM 1.2 V 1.2 V VOUT tSS VOUT 1000 mV VCOMP VSS VEN 200 mV Switching starts when VCOMP > 350 mV tSS(DELAY) Figure 3: Startup to VOUTx = 1.2 V, 2.0 A, with CSS = 22 nF VOUT rises from 0.6 V COMP pin released at VSS = VFB+200 mV 1000 mV VCOMP VSS Switching starts when VCOMP > 350 mV 200 mV VEN Figure 4: Startup to VOUTx = 1.2 V, 2.0 A, with VOUT prebiased to 0.6 V Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 15 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR switching starts, and VOUT ramps upward, starting from the pre-bias level. Figure 4 shows startup when the output voltage is pre-biased to 0.9 V. Active Low Power-On Reset (NPORx) The NPORx pins are open drain outputs, so an external pull-up resistor must be connected to each. An internal comparator monitors the voltage at the FBx pin and controls the open drain device at the NPORx pins. NPORx is pulled high by the external resistor approximately 7.5 ms after VOUTx is within regulation. The NPORx output is pulled low if: • VFBx(RISING) < 92% of the reference voltage, or • VFBx(RISING) > 115% of the reference voltage, or • EN is low, or • VIN UVLO occurs, or • Thermal shutdown (TSD) occurs. If the A8651 is running and VINx transitions low, then NPORx transitions low and remains low only as long as the internal circuitry is able to enhance the open-drain output device. When VIN fully collapses, the NPORx pin returns to the high impedance state. The NPOR comparator incorporates hysteresis to prevent chattering due to voltage ripple at the FBx pin. Protection Features UNDERVOLTAGE LOCKOUT (UVLO) An undervoltage lockout (UVLO) comparator monitors the voltage at the VIN pin and keeps the regulator disabled if the voltage is below the lockout threshold (VINSTART ). The UVLO comparator incorporates enough hysteresis (VUVLO(HYS) ) to prevent on-off cycling of the regulator due to IR drops in the VIN path during heavy loading or during startup. THERMAL SHUTDOWN (TSD) The A8651 protects itself from overheating with an internal thermal monitoring circuit. If the junction temperature exceeds the upper thermal shutdown threshold (TSD(th) , nominally 170°C), the voltages at the soft start (SSx) and COMPx pins is pulled to GND and both the high-side and low-side MOSFETs are turned off. The A8651 stops PWM switching, but it does not enter the shutdown or sleep mode supply current levels. The A8651 automatically restarts when the junction temperature decreases more than the thermal shutdown hysteresis (TSD(HYS) , 20°C (typ) ). OVERVOLTAGE PROTECTION (OVP) The A8651 uses the FBx pins to provide a basic level of overvoltage protection. An overvoltage condition could occur if the load decreases very quickly or the COMPx pin or the regulator output are pulled high by some external voltage. When an overvoltage condition is detected, (1) NPORx is pulled low, and (2) PWM switching stops (the SWx node becomes high impedance). The COMPx and SSx pin voltages are not affected by OVP. If the regulator output decreases back to the normal operating range, NPORx transitions high and PWM switching resumes. PULSE-BY-PULSE OVERCURRENT PROTECTION (OCP) The A8651 monitors the current in the high-side P-channel MOSFET and if the current exceeds the pulse-by-pulse current overcurrent threshold (ILIM), then the high-side MOSFET is turned off. Normal PWM operation resumes on the next clock pulse from the oscillator. The A8651 includes leading edge blanking to prevent false triggering of the pulse-by-pulse current limit when the high-side MOSFET is turned on. Pulse-by-pulse current limiting is always active. A key feature of the A8651 is the ability to adjust the peak switch current limit. This can be useful when the full current capability of the regulator is not required for a given application. A smaller current limit may allow the use of power components with lower current ratings, thus saving space and reducing cost. A single resistor between the ISET pin and ground controls the current limit. Resistor values should be set in the range between 30.9 kΩ (for the lowest current limit setting) and 41.2 kΩ (for the highest current limit setting). The maximum switch current is affected by slope compensation via the duty cycle. The A8651 is conservatively rated to deliver 2.0 ADC for most applications. However, the exact current the Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 16 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 A8651 supports is heavily dependent on duty cycle, ambient temperature, thermal resistance of the PCB, airflow, component selection, and nearby heat sources. The A8651 is designed to deliver more current at lower duty cycles and slightly less current at higher duty cycles. For example, the pulse-by-pulse current limit at 20% duty cycle is typically 3.85 A, but at 80% duty cycle the pulse limit is typically 3.10 A. Use Table 1a and Figure 5a, and Table 1b and Figure 5b to determine the real current limit given the duty cycle required for each application. Take care to do a careful thermal solution or thermal shutdown can occur. Table 1a. Pulse-by-Pulse Current Limit versus Duty Cycle RSET = 41.2 kΩ, fSW = 2 MHz OUTPUT SHORT CIRCUIT (HICCUP MODE) PROTECTION Hiccup mode protects the A8651 when the load is either too high or when the output of the converter is shorted to ground. When the voltage at the FBx pin is below the Hiccup Enable Threshold (VHICEN , 700 mV (typ)), hiccup mode protection is enabled. When the voltage at the FBx pin is above the Hiccup Disable Threshold (VHICDIS, 740 mV (typ)) hiccup mode protection is disabled. Hiccup mode overcurrent protection monitors the number of overcurrent events using an up/down counter. An overcurrent pulse increments the counter by 1 and a PWM cycle without an Table 1b. Pulse-by-Pulse Current Limit versus Duty Cycle RSET = 30.9 kΩ, fSW = 2 MHz Pulse-by-Pulse Current Limit (A) Duty Cycle (%) Min. Typ. Max. 5 3.42 4.04 20 3.17 40 Duty Cycle (%) Min. Typ. Max. 4.65 5 1.85 2.37 2.87 3.86 4.51 20 1.69 2.27 2.77 2.83 3.61 4.31 40 1.49 2.13 2.64 60 – 3.37 4.12 60 1.28 2.00 2.52 80 – 3.12 3.92 80 – 1.86 2.39 90 – 3.00 3.83 90 – 1.80 2.32 4.8 3.0 4.6 2.8 4.4 2.6 4.0 2.4 3.8 2.2 3.6 ILIM (A) ILIM (A) 4.2 3.4 3.2 2.0 1.8 1.6 3.0 2.8 Max., fSW = 350 kHz Max., fSW = 2.00 MHz 1.4 2.4 Typ., fSW = 350 kHz Typ., fSW = 2.00 MHz 1.2 2.2 Min., fSW = 350 kHz Min., fSW = 2.00 MHz 2.6 2.0 Pulse-by-Pulse Current Limit (A) 5 Max., fSW = 350 kHz Max., fSW = 2.00 MHz Typ., fSW = 350 kHz Typ., fSW = 2.00 MHz 1.0 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100 Duty Cycle (%) Figure 5a: Current Limit versus Duty Cycle, with RSET = 41.2 kΩ 0.8 Min., fSW = 350 kHz Min., fSW = 2.00 MHz 5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100 Duty Cycle (%) Figure 5b: Current Limit versus Duty Cycle, with RSET = 30.9 kΩ Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 17 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR overcurrent pulse decrements the counter by 1. If more than 7 consecutive overcurrents are detected then the Hiccup latch is set and PWM switching is stopped. The Hiccup signal causes the COMPx pin to be pulled low with a relatively low resistance (1.5 kΩ). Hiccup mode also enables a current sink connected to the soft start (SSx) pin (ISSHIC,10 µA), so when hiccup initially occurs, the voltage at the soft start pin ramps downward. Hiccup mode operation is shown in Figure 6. When the voltage at the soft start pin decays to a low level (VSSRESET , 120 mV (typ) ), the hiccup latch is cleared and the 10 µA soft start pin current sink is turned off. The soft start pin resumes charging the soft start capacitor with 20 µA, and the voltage at the soft start pin ramps upward. When the voltage at the soft start pin exceeds the soft start offset voltage (VSSOFFS , 200 mV (typ)) the low resistance pull-down at the COMPx pin is turned off. The error amplifier forces the voltage at the COMPx pin to ramp up quickly, and PWM switching begins. If the short circuit at the converter output remains, another hiccup cycle occurs. Hiccup cycles repeat until the short circuit is removed or the converter is disabled. If the short circuit is removed the A8651 soft starts normally and the output voltage ramps to the operating level, as shown in Figure 6. Short removed VOUT VCOMP VSS ≈ 4.1 A 200 mV 120 mV IL Figure 6: Hiccup Mode Operation and Recovery Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 18 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR Table 2: Summary of A8651 Fault Modes and Operation Fault Mode VSS VCOMP High-Side Switch Low-Side Switch NPOR Reset Condition Active during tOFF , off during hiccup Depends on VOUT Automatic, remove the short Output hard short to ground (VOUT and VFB = 0 V) Hiccup after VCOMP ≈ 1.25 V and 7 OC faults Clamped to ≈1.25 V for ILIM , then pulled low during hiccup Controlled by VCOMP, fSW / 2 if 0 < VFB< 400 mV, fSW / 4 if COMP ≈1.25 V and ILIM Output overcurrent and VFB < VHICDIS Hiccup after VCOMP ≈ 1.25 V and 7 OC faults Clamped to ≈1.25 V for ILIM , then pulled low during hiccup Controlled by VCOMP, fSW / 2 if 0 < VFB < 400 mV, fSW / 4 if VCOMP ≈1.25 V and ILIM Active during tOFF, off during hiccup Depends on VOUT Automatic, decrease the load current SW hard short to ground Ramps to VIN , hiccup may occur when the short is removed Clamped to ≈1.25 V, pulled low if hiccup occurs Controlled by VCOMP, turn off if VSW ≈ 0 V and blanking time expires, fSW / 4 Active during tOFF , off if hiccup occurs when the short is removed Depends on VOUT Automatic, remove the short SW soft short to ground Hiccup after VCOMP ≈1.25 V and 7 OC faults Clamped to ≈1.25 V for ILIM, then pulled low during hiccup Controlled by VCOMP, fSW / 2 if 0 < VFB < 400 mV, fSW / 4 if VCOMP ≈1.25 V and ILIM Active during tOFF , off during hiccup Depends on VOUT Automatic, remove the short FB pin open (VFB floats high due to negative bias current) Not affected Transitions low via loop response as VFB floats high Off, fSW / 2 if 0 < VFB < 400 mV, fSW if 300 mV < VFB Off, disabled if VCOMP < 200 mV Pulled low whenVFB > 115% × VREF Automatic, connect the FB pin Output overvoltage (VFB > 115% × VREF) Not affected Transitions low via loop response because VFB > VREF Off, fSW / 2 if 0 < VFB < 400 mV, fSW if 300 mV < VFB Off, disabled if VCOMP< 200 mV Pulled low when VFB > 115% × VREF Automatic, VFB returns to the normal range Active during tOFF Pulled low when VFB < 92% × VREF Automatic, VFB returns to the normal range Off Pulled low Automatic, after the junction cools down Output undervoltage Not affected Transitions high via loop response Controlled by VCOMP , fSW / 2 if 0 < VFB < 400 mV, fSW if 300 mV < VFB Thermal shutdown (TSD) Pulled low and latched until VSS < VSSRESET Pulled low and latched until VSS > VSS(RELEASE) Off Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 19 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR DESIGN AND COMPONENT SELECTION This section shows how to design and select external component values. For simplicity, the naming convention used here refers only to regulator 1, but the same design methods can be used for regulator 2. RFB1 FB1 PIN RFB2 Setting the Output Voltage (VOUT1 , RFBx ) The output voltage of the A8651 is determined by connecting a resistor divider from the output node (VOUT1) to the FB1 pin as shown in Figure 7. There are trade-offs when choosing the value of the feedback resistors. If the series combination (RFB1+RFB2 ) is relatively low then the light load efficiency of the regulator is reduced. So, to maximize the efficiency it is best to choose high values of resistors. On the other hand, if the parallel combination (RFB1 // RFB2 ) is too high, then the regulator may be susceptible to noise coupling into the FB1 pin. Figure 7: Connection for the Feedback Divider Table 3. Recommended Feedback Resistors In general, the feedback resistors must satisfy the ratio shown in equation 1 to produce an output voltage, VOUT1: (1) Table 3 shows the most common output voltages and recommended feedback resistors, assuming less than 0.2% efficiency loss at a light load of 100 mA and a parallel combination of 4 kΩ presented to the FB1 pin. For optimal system accuracy, it is recommended that the feedback resistors have ≤1% tolerances. PWM Switching Frequency (RFSET) The PWM switching frequency is set by connecting a resistor from the FSET/SYNC pin to ground. Figure 8 is a graph showing the relationship between the typical switching frequency (y-axis) and the FSET resistor (x-axis). VOUT1 (V) RFB1 (VOUT1 to FB1 pin) (kΩ) RFB2 (FB1 pin to GND) (kΩ) 1.2 6.04 12.1 1.5 7.50 8.45 1.8 9.09 7.15 2.5 12.4 5.76 3.3 16.5 5.23 2250 2000 Oscillator Frequency (kHz) RFB1 VOUT1 –1 = RFB2 0.8 (V) V OUT1 1750 1500 1250 1000 750 500 250 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 RFSET (kΩ) Figure 8. PWM Switching Frequency versus RFSET Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 20 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 To set a specific oscillator frequency (fosc ), the RFSET resistor can be calculated as follows: RFSET = ( fOSC 15456 ) – 1.186 (2) where fosc is in kHz and RFSET is in kΩ. When the PWM switching frequency is chosen, the designer should be aware of the minimum controllable on-time (tON(MIN) ) of the A8651. If the application system required on-time is less than the A8651 minimum controllable on-time, then switch node jitter occurs and the output voltage has increased ripple or oscillations. The PWM switching frequency should be calculated as follows: fSWMAX = VOUT1 tON(MIN) ×VIN1(MAX) SE = (0.054 × RSET – 0.96) × fSW (4b) where RSET is in kΩ, fSW is in MHz, and the calculated SE is in A/µs. Another limitation is shown in equation 5. This is based on a formula to calculate the amount of slope compensation required to critically damp the double poles at half the PWM switching frequency (this approach includes the duty cycle (D), which should be calculated at the minimum input voltage to insure optimal stability): LO1 ≥ VOUT1 VIN1(MIN) 1 – 0.18 × SE VOUT1 (5) (3) where VOUT1 is the output voltage, tON(MIN) is the minimum controllable on-time of the A8651 (worst case is 105 ns), and VIN1(MAX) is the maximum required operational input voltage to the A8651 (not the peak surge voltage). If the A8651 synchronization function is employed, then the base switching frequency should be chosen such that jitter does not result at the maximum synchronized switching frequency according to equation 3: 1.5 × fSW < fSWMAX calculated by equation 3. Output Inductor (LO ) To avoid dropout (saturation of the buck regulator), VIN1(MIN) must be approximately 0.75 to 1.0 V above VOUT1 when calculating the inductor value with equation 5. If equations 4a or 5 yield an inductor value that is not a standard value then the next closest available value should be used. The final inductor value should allow for 10% to 20% of initial tolerance and 10% to 20% of inductor saturation. The saturation current of the inductor should be higher than the peak current capability of the A8651. Ideally, for output short circuit conditions, the inductor should not saturate given the highest pulse-by-pulse current limit at minimum duty cycle (ILIM(5%)); 4.7 A. This may be too costly. At the very least, the inductor should not saturate given the peak operating current according to the following equation: For a peak current mode regulator it is common knowledge that, without adequate slope compensation, the system becomes unstable when the duty cycle is near or above 50%. However, the slope compensation in the A8651 is a fixed value (SE ). Therefore, it is important to calculate an inductor value such that the falling slope of the inductor current (SF) works well with the A8651 slope compensation. Equations 4a and 4b can be used to calculate a range of values for the output inductor based on the well known approach of providing slope compensation that matches 50% to 100% of the down slope of the inductor current. VOUT1 VOUT1 ≤ LO1 ≤ 2 × SE SE where LO is in µH and the slope compensation (SE ) is a function of switching frequency, as follows: (4a) IPEAK = 4.1 – SE × VOUT1 1.15 × fSW × VIN1(MAX) (6) where VIN1(MAX) is the maximum continuous input voltage, such as 5.5 V. Starting with equation 6 and subtracting half of the inductor ripple current provides us with an interesting equation to predict the typical DC load capability of the regulator at a given duty cycle (D = VOUT1 / VIN1): SE × D VOUT1 × (1 – D ) – IOUT1(DC) ≤ 4.1 – (7) fSW 2 × fSW × LO1 Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 21 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 After an inductor is chosen it should be tested during output short circuit conditions. The inductor current should be monitored using a current probe. A good design should ensure the inductor or the regulator are not damaged when the output is shorted to ground at maximum input voltage and the highest expected ambient temperature. Output Capacitors The output capacitors filter the output voltage to provide an acceptable level of ripple voltage and they store energy to help maintain voltage regulation during a load transient. The voltage rating of the output capacitors must support the output voltage with sufficient design margin. The output voltage ripple (ΔVOUT1 ) is a function of the output capacitors parameters: COUT1, ESRCOUT1, and ESLCOUT1 : ∆VOUT1 = ∆IL1 × ESRCOUT1 V – VOUT1 + IN1 × ESLCOUT1 LO1 ∆IL1 + 8 fSW C OUT1 (8) ∆IL1 8 fSW C OUT1 (9) To reduce the voltage ripple of a design using ceramic output capacitors simply: increase the total capacitance, reduce the inductor current ripple (that is, increase the inductor value), or increase the switching frequency. For electrolytic output capacitors the value of capacitance will be relatively high, so the third term in equation 8 will be very small and the output voltage ripple will be determined primarily by the first two terms of equation 8: ∆VOUT1 = ∆IL1 × ESRCOUT1 + VIN1 LO1 × ESLCOUT1 The ESR of some electrolytic capacitors can be quite high so Allegro recommends choosing a quality capacitor for which the ESR or the total impedance is clearly documented in the capacitor datasheet. Also, the ESR of electrolytic capacitors usually increases significantly at cold ambients, as much as 10 X, which increases the output voltage ripple and, in most cases, reduces the stability of the system. The transient response of the regulator depends on the quantity and type of output capacitors. In general, minimizing the ESR of the output capacitance will result in a better transient response. The ESR can be minimized by simply adding more capacitors in parallel or by using higher quality capacitors. At the instant of a fast load transient (di/dt), the output voltage changes by the amount: ∆VOUT1 = ∆IL1 × ESRCOUT1 + The type of output capacitors determines which terms of equation 8 are dominant. For ceramic output capacitors the ESRCOUT1 and ESLCOUT1 are virtually zero, so the output voltage ripple will be dominated by the third term of equation 8: ∆VOUT1 ≤ by using a high(er) quality capacitor, or add more capacitors in parallel, or reduce the inductor current ripple (that is, increase the inductor value). COUT1 (10) To reduce the voltage ripple of a design using electrolytic output capacitors simply: decrease the equivalent ESRCOUT1 and ESL- di dt × ESLCOUT1 (11) After the load transient occurs, the output voltage will deviate from its nominal value for a short time. The length of this time depends on the system bandwidth, the output inductor value, and output capacitance. Eventually, the error amplifier brings the output voltage back to its nominal value. The speed at which the error amplifier brings the output voltage back to the setpoint depends mainly on the closed-loop bandwidth of the system. A higher bandwidth usually results in a shorter time to return to the nominal voltage. However, a higher bandwidth system may be more difficult to obtain acceptable gain and phase margins. Selection of the compensation components (RZ , CZ , and CP ) are discussed in more detail in the Compensation Components section of this datasheet. Input Capacitors Three factors should be considered when choosing the input capacitors. First, the capacitors must be chosen to support the maximum expected input surge voltage with adequate design margin. Second, the capacitor rms current rating must be higher than the expected rms input current to the regulator. Third, the capacitors must have enough capacitance and a low enough ESR to limit the input voltage dV/dt to something much less than the hysteresis of the UVLO circuitry (nominally 200 mV for the A8651) at maximum loading and minimum input voltage. Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 22 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR The input capacitor(s) must limit the voltage deviations at the VIN1 pin to something significantly less than the device UVLO hysteresis during maximum load and minimum input voltage. The following equation allows us to calculate the minimum input capacitance: CIN1 ≥ IOUT1 × D × (1– D) 0.85 × fSW × ∆VIN1(MIN) (12) where ΔVIN1(MIN) is chosen to be much less than the hysteresis of the VIN1 UVLO comparator (ΔVIN1(MIN) ≤ 100 mV is recommended), and fSW is the nominal PWM frequency. The D × (1–D) term in equation 12 has an absolute maximum value of 0.25 at 50% duty cycle. So, for example, a very conservative design based on IOUT1 = 2.0 A, fSW = 85% of 2 MHz, D × (1–D) = 0.25, and ΔVIN1 = 100 mV yields: 2.0 (A) × 0.25 = 2.9 µF 1.7 (MHz) × 100 (mV) The input capacitors must deliver an rms current (IRMS) according to the following formula: CIN ≥ IRMS = IOUT1 D × ( 1 – D) (13) where the duty cycle (D) is defined as: D = VOUT1 / VIN1(14) Figure 9 shows the normalized input capacitor rms current versus duty cycle. To use this graph, simply find the operational duty cycle (D) on the x-axis and determine the input/output current multiplier on the y-axis. For example, at a 20% duty cycle, the input/output current multiplier is 0.40. Therefore, if the regulator is delivering 2.0 A of steady-state load current, the input capacitor(s) must support 0.40 × 2.0 A or 0.8 Arms. A good design should consider the DC-bias effect on a ceramic capacitor: as the applied voltage approaches the rated value, the capacitance value decreases. This effect is very pronounced with the Y5V and Z5U temperature characteristic devices (as much as 90% reduction), so these types should be avoided. The X5R and X7R type capacitors should be the primary choices due to their stability versus both DC bias and temperature. For all ceramic capacitors, the DC-bias effect is even more pronounced on smaller sizes of device case, so a good design uses the largest affordable case size (such as 1206 or 1210). Also, it is advisable to select input capacitors with plenty of design margin in the voltage rating to accommodate the worst case transient input voltage. Soft Start and Hiccup Mode Timing (CSS1) The soft start time of the A8651 is determined by the value of the capacitance at the soft start pin, CSS1. When the A8651 is enabled the voltage at the soft start pin (SS1) starts from 0 V and is charged by the soft start current, ISSSU1 . However, PWM switching does not begin instantly because the voltage at the soft start pin must rise above 200 mV. The soft start delay (tSS(DELAY)) can be calculated using the following equation: tSS(DELAY) = CSS1× 200 (mV) ISSSU (15) If the A8651 is starting into a very heavy load a very fast soft start time may cause the regulator to exceed the cycle-by-cycle overcurrent threshold. This occurs because the total of the full load current, the inductor ripple current, and the additional current required to charge the output capacitors: IRMS / IOUT1 A8651 0.55 0.50 0.45 0.40 0.35 0.30 0.25 0.20 0.15 0.10 0.05 0.00 0 10 20 30 40 50 60 Duty Cycle, D (%) 70 80 90 100 Figure 9: Normalized Input Capacitor Ripple versus Duty Cycle Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 23 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 ICOUT1 = COUT1 × VOUT1 / tSS(16) is higher than the cycle-by-cycle current threshold, as shown in figure 10. This phenomena is more pronounced when using high value electrolytic type output capacitors. To avoid prematurely triggering hiccup mode the soft start capacitor, CSS1 , should be calculated according to: I V C CSS1 ≥ SSSU × OUT1 × OUT1 0.8 (V) × ICOUT1 (17) where VOUT1 is the output voltage, COUT1 is the output capacitance, ICOUT1 is the amount of current allowed to charge the output capacitance during soft start (recommend 0.1 A < ICOUT1 < 0.3 A). Higher values of ICOUT1 result in faster soft start times. However, lower values of ICOUT1 ensure that hiccup mode is not inappropriately triggered. Allegro recommends starting the design with an ICOUT1 of 0.1 A and increasing it only if the soft start time is too slow. If a non-standard capacitor value for CSS1 is calculated, the next larger value should be used. The output voltage ramp time, tSS , can be calculated by using either of the following methods: tSS = VOUT1 × or COUT1 (18) ICOUT1 tSS = 0.8 (V) × CSS1 ISSSU (19) When the A8651 is in hiccup mode, the soft start capacitor is used as a timing capacitor and sets the hiccup period. The soft start pin charges the soft start capacitor with ISSSU during a startup attempt, and discharges the same capacitor with ISSHIC } I LIM I LOAD1 Output capacitor current, I COUT1 t SS Figure 10. Output Current (ICOUT1) During Startup between startup attempts. Because the ratio ISSSU / ISSHIC is approximately 2:1, the time between hiccups is about two times as long as the startup time. Therefore, the effective duty-cycle of the A8651 is very low and the junction temperature is kept low. Compensation Components (RZ, CZ, CP) To compensate the system, it is important to understand where the buck power stage, load resistance, and output capacitance form their poles and zeros in frequency. Also, it is important to understand that the (Type II) compensated error amplifier introduces a zero and two more poles, and where these should be placed to maximize system stability, provide a high bandwidth, and optimize the transient response. First, consider the power stage of the A8651, the output capacitors, and the load resistance. This circuitry is commonly referred as the control-to-output transfer function. The low frequency gain of this circuitry depends on the COMP1 to SW1 current gain ( gmPOWER ), and the value of the load resistor (RL1 ). The DC gain (GCO(0HZ) ) of the control-to-output is: GCO(0Hz) = gmPOWER × RL1(20) The control-to-output transfer function has a pole (fP1), formed by the output capacitance (COUT1) and load resistance (RL1 ), located at: fP1 = 1 2� × RL1× COUT1 (21) The control-to-output transfer function also has a zero (fZ1) formed by the output capacitance (COUT1) and its associated ESR: fZ1 = 1 2� × ESRCOUT1× COUT1 (22) For a design with very low-ESR type output capacitors (such as ceramic capacitors), the ESR zero, fZ1, is usually at a very high frequency so it can be ignored. On the other hand, if the ESR zero falls below or near the 0 dB crossover frequency of the system (as happens with electrolytic output capacitors), then it should be cancelled by the pole formed by the CP capacitor and the RZ resistor (discussed and identified later as fP3 ). A Bode plot of the control-to-output transfer function for the schematic shown in figure 15, with VOUT1 = 1.2 V, IOUT1 = 1.5 A, Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 24 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 and RL1 = 0.8 Ω, is shown in Figure 11. The pole at fP1 can easily be seen at 8.8 kHz while the ESR zero, fZ1 , occurs at a very high frequency, 4 MHz (this is typical for a design using ceramic output capacitors). Note, there is more than 90° of total phase shift because of the double-pole at half the switching frequency. Next, consider the feedback resistor divider, (RFB1 and RFB2), the error amplifier (gm), and its compensation network RZ-CZ-CP . It greatly simplifies the transfer function derivation if RO >> RZ, and CZ >> CP . In most cases, RO > 2 MΩ, 1 kΩ < RZ <100 kΩ, 220 pF < CZ < 47 nF, and CP < 50 pF, so the following equations are very accurate. The low frequency gain of the control section (GC(0Hz) ) is formed by the feedback resistor divider and the error amplifier. It can be calculated as: GC(0Hz) = = = RFB2 RFB1 +RFB2 VFB1 VOUT1 VFB1 × gm × RO fP2 = fZ2 = (23) VFB is the reference voltage (0.8 V), RO is the error amplifier output impedance (AVOL/gm ). 80 f P1 = 8.8 kHz fz1 = 4 MHz Gain (dB) Gain (dB) 40 (26) 40 GCO(0Hz) = 58 dB fz2 = 16 kHz f P2 = 45 Hz fP2 ≈ 1.1 MHz -40 180 90 Phase (°) Phase (°) 1 2� × RZ × CP 0 -80 180 Double Pole at 1 MHz 0 -90 -180 10 (25) Finally, consider the combined Bode plot of both the control-tooutput and the compensated error amplifier (Figure 13). Careful examination of this plot shows that the magnitude and phase of the entire system (red curve) are simply the sum of the error gm is the error amplifier transconductance (750 µA/V ), and -40 1 2� × RZ × CZ A Bode plot of the error amplifier and its compensation network is shown in Figure 12, where fP2, fP3, and fZ2 are indicated on the Gain plot. Notice that the zero (fZ2 at 16 kHz) has been placed so that it is just above the pole at fP1 previously shown at 8.8 kHz in the control-to-output Bode plot (Figure 11). Placing fZ2 just above fP1 results in excellent phase margin, but relatively slow transient recovery time, as we will see later. VOUT is the output voltage, GCO(0Hz) = 12 dB (24) Lastly, the transfer function of the Type-II compensated error amplifier has a (very) high frequency pole (fP3) dominated by the RZ resistor and the CP capacitor: fP3 = where 0 1 2� × RO × CZ The transfer function of the Type-II compensated error amplifier also has frequency zero (fZ2) dominated by the RZ resistor and the CZ capacitor: × gm × RO × AVOL VOUT1 The transfer function of the Type-II compensated error amplifier has a (very) low frequency pole (fP2 ) dominated by the output error amplifier output impedance (RO) and the CZ compensation capacitor: 135 90 -45 100 103 10×103 Frequency (Hz) 100×103 1.06 Figure 11: Control-to-Output Bode Plot 106 0 10 100 103 10×103 Frequency (Hz) 100×103 1.06 106 Figure 12: Type-II Compensated Error Amplifier Bode Plot Allegro MicroSystems, LLC 25 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 amplifier response (blue curve) and the control to output response (green curve). As shown in Figure 13, the bandwidth of this system (fc ) is 72 kHz, the phase margin is 73 degrees, and the gain margin is 27 dB. A Generalized Tuning Procedure This section presents a methodology to systematically apply design considerations provided above. 1. Choose the system bandwidth (fC ). This is the frequency at which the magnitude of the gain crosses 0 dB. Recommended values for fC, based on the PWM switching frequency, are in the range fSW / 20 < fC < fSW / 7.5. A higher value of fC generally provides a better transient response, while a lower value of fC generally makes it easier to obtain higher gain and phase margins. 2. Calculate the RZ resistor value. This sets the system bandwidth (fC): 2� ×COUT1 RZ = fC × VOUT1 × (27) VFB1 gmPOWER × gm 3. Determine the frequency of the pole (fP1). This pole is formed by COUT and RL. Use equation 21 (repeated here): fP1 = 4. Calculate a range of values for the CZ capacitor. Use the following: 4 1 < CZ < (28) 2� × RZ × fC 2� × RZ × 1.5 × fP1 To maximize system stability (that is, to have the greatest gain margin), use a higher value of CZ . To optimize transient recovery time, although at the expense of some phase margin, use a lower value of CZ . 5. Calculate the frequency of the ESR zero (fZ1 ) formed by the output capacitor(s) by using equation 22 (repeated here): fZ1 = 1 2� × ESRCOUT1× COUT1 If fZ1 is at least 1 decade higher than the target crossover frequency (fC ) then fZ1 can be ignored. This is usually the case for a design using ceramic output capacitors. Use equation 26 to calculate the value of CP by setting fP3 to either 5 × fC or fSW / 2, whichever is higher. Alternatively, if fZ1 is near or below the target crossover frequency (fC ), then use equation 26 to calculate the value of CP by setting fP3 equal to fZ1. This is usually the case for a design using high ESR electrolytic output capacitors. 1 2� × RL1× COUT1 (V) 1.840 100 Gain (dB) 1.820 fC = 72 kHz 0 GM = 27 dB 1.800 -100 1.780 Phase (°) 180 PM = 73 deg 90 1.760 0 -90 -180 1.740 10 100 103 10×103 Frequency (Hz) 100×103 1.06 Figure 13: Bode Plot of the Complete System (red curve) 106 1.725 240 250 260 270 280 290 (µs) 300 310 320 330 Figure 14: Transient Recovery Comparison for fZ2 at 16 kHz/69° and 50 kHz/51° Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 26 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 Csync22pF SYNCin 5 Vin 20 RFSET 11.3K CIN1 3.3uF 1206 CIN2 0.1uF 0805 CIN3 10nF 0603 EN1 CP1 68pF VIN1 CSS1 22nF GND PGND1 SW1 FB1 11 CIN5 0.1uF 0805 CZ2 1.8nF CO2 0.1uF 0805 CO3 10nF 0603 RPU1 10K NPOR1 15 LO2 1.5uH 10 1 RFB3 6.04K VIN2 1.2V / 1.5A 2 14 CO5 10uF 1206 CO6 10uF 1206 CO7 10uF 1206 CO8 0.1uF 0805 CO9 10nF 0603 3.3V 8 13 12 7 RZ2 5.62K CO1 10uF 1206 3.3V FB2 9 CP2 68pF 18 RFB2 5.23K NPOR1 CIN6 10nF 0603 EN2 3.3V / 1.5A 2 RFB1 16.5K SW2 CIN4 3.3uF 1206 1 EN1 COMP1 SS1 ISET1 RSET1 34.8K CZ1 1.8nF U1 1 A8651 16 2 3 17 19 4 RZ1 4.99K LO1 3.3uH FSET/SYNC CSS2 22nF RFB4 12.1K PGND2 EN2 COMP2 SS2 ISET2 RPU2 10K NPOR2 6 NPOR2 RSET2 34.8K Figure 15: Typical Application Circuit for VIN = 5 V at TA = 125°C: VOUT 3.3 V/1.5 A and 1.2 V /1.5 A at 2 MHz CSYNC are only required if synchronizing to an external clock Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 27 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 POWER DISSIPATION AND THERMAL CALCULATIONS The power dissipated in the A8651 is the sum of the power dissipated from the VIN supply current (PIN ) and the power dissipated by the two regulators. The regulator power dissipation is composed of: the power dissipated due to the switching of the high-side power MOSFET (PSW(HS) ), the power dissipated due to the rms current being conducted by the high-side and low-side MOSFETs (PCOND(HS) and PCOND(LS) ), and the power dissipated by the low-side body diode (PNO ) during the non-overlap time. The power dissipated from the VIN supply current can be calculated using the following equation: PIN = VIN1 × IQ+ (VIN1 + VIN2 ) × (QG(HS) + QG(LS) ) × fSW (29) where VINx are the input voltages, IQ is the input quiescent current drawn by the device (nominally 2 mA), tion: 2 PCOND(HS) = Irms(FET) × RDS(on)HS = Note: The calculation after this point refers only to regulator 1. The power dissipated by the internal high-side MOSFET during PWM switching can be calculated using the following equation: PSW1 = VIN × IOUT × (tr + tf ) × fSW 2 (30) where VIN is the input voltage, IOUT is the output current, fSW is the PWM switching frequency, and tr and tf are the rise and fall times measured at the SW node. The exact rise and fall times at the SW node depend on the external components and PCB layout so each design should be measured at full load. Approximate values for both tr and tf range from 10 to 15 ns. The fall time is usually about 50% faster than the rise time. The conduction losses dissipated by the high-side MOSFET while it is conducting can be calculated using the following equa- (31) where IOUT is the regulator output current, ΔIL is the peak-to-peak inductor ripple current, and RDS(on)1 is the on-resistance of the high-side MOSFET. The conduction losses dissipated by the low-side MOSFET can be calculated as: 2 PCOND2 = Irms(FET) × RDS(on)2 QG(HS) and QG(LS) are the internal high- and low-side MOSFET gate charges (approximately 3.3 nC and 1.4 nC, respectively), and fSW is the PWM switching frequency. VOUT ∆IL2 2 × IOUT + 12 × RDS(on)HS VIN = 1– VOUT ∆IL2 2 × IOUT + 12 × RDS(on)2 VIN (32) where IOUT is the regulator output current, ΔIL is the peak-to-peak inductor ripple current, and RDS(on)1 is the on-resistance of the high-side MOSFET. The RDS(on)of the MOSFETs has some initial tolerance plus an increase from self-heating and elevated ambient temperatures. A conservative design should accommodate an RDS(on) with at least a 15% initial tolerance plus 0.39%/°C increase due to temperature. The power dissipated by the low-side MOSFETs body diode during the non-overlap time can be calculated as: PNO = VSD × IOUT × 2 × tNO × fSW (33) where The deadtime is the same for the rising and falling edges of SW, VSD is the source-to-drain voltage of the low-side MOSFET (typically 0.60 V), and tNO is the non-overlap time (typically 15 ns), Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 28 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR Finally, the total power dissipated by the device (PTOTAL) is the sum of the previous equations: PTOTAL = PIN + PREGULATOR1 + PREGULATOR2(35) where PREGULATOR1 = PSW + PCOND(HS) + PCOND(LS) + PNO(36) The average junction temperature can be calculated with the following equation: TJ = PTOTAL × RθJA + TA (37) where PTOTAL is the total power dissipated as described in equation 35, TA is the ambient temperature. The maximum junction temperature is dependent on how efficiently heat can be transferred from the PCB to ambient air. It is critical that the thermal pad on the bottom of the IC should be connected to a at least one ground plane using multiple vias. As with any regulator, there are limits to the amount of heat that can be dissipated before risking thermal shutdown. There are trade-offs among: ambient operating temperature, input voltage, output voltage, output current, switching frequency, PCB thermal resistance, airflow, and other nearby heat sources. Even a small amount of airflow will reduce the junction temperature considerably. RθJA is the junction-to-ambient thermal resistance (48°C/W on a 4-layer PCB), and Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 29 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR PCB COMPONENT PLACEMENT AND ROUTING A good PCB layout is critical if the A8651 is to provide clean, stable output voltages. Follow these guidelines to insure a good PCB layout. Figure 16 shows a typical buck converter schematic with the critical power paths/loops. Figure 17 shows an example PCB component placement and routing with the same critical power paths/loops from the schematic. 1. Place the ceramic input capacitors as close as possible to the VINx pins and ground the capacitors at the PGNDx pins. The ceramic input capacitors and the A8651 must be on the same layer. Connect the input capacitors, the VINx pins, and the PGNDx pins with a wide trace. This critical loop is shown as a red trace in Figures 16 and 17. 2. Place the output inductor (LOx ) as close as possible to the SWx pins. The output inductor and the A8651 must be on the same layer. Connect the SWx pins to the output inductor with a relatively wide trace or polygon. For EMI/EMC reasons, its best to minimize the area of this trace/polygon. This critical trace is shown as a green trace in Figure 16. Also, keep low level analog signals (like FB and COMP) away from the SW metal. 3. Place the output capacitors relatively close to the output inductor and the A8651. Ideally, the output capacitors, output inductor and the A8651 should be on the same layer. Connect the output inductor and the output capacitors with a fairly wide trace. The output capacitors must use a ground plane to make a very low inductance connection back to the PGND pin. These critical connections are shown in blue in Figures 16 and 17. 4. Place the feedback resistor dividers (RFB1-RFB2 ) very close to the FB pin. Orient RFB2 such that its ground is as close as possible to the A8651. 5. Place the compensation components (RZ, CZ, and CP) as close as possible to the COMP pin. Orient CZ and CP such that their ground connections are as close as possible to the A8651. 6. Place and ground the FSET resistor as close as possible to the FSET pin. 7. The output voltage sense trace (from VOUT to RFB1) should be connected as close as possible to the load to obtain the best load regulation. 8. The thermal pad under the IC should be connected a ground plane (preferably on the bottom layer) with as many vias as possible. Allegro recommends vias with approximately a 10-15 mil hole and a 5-7 mil ring. 9. Place the soft start capacitor (CSS) as close as possible to the SS pin. Place a via to the GND plane as close as possible to this component. 10. When connecting the input and output ceramic capacitors to a power or ground plane, use multiple vias and place the vias as close as possible to the component’s pads. Do not use thermal reliefs (spokes) around the pads for the input and output ceramic capacitors. 11. EMI/EMC issues are always a concern. Allegro recommends having locations for an RC snubber from SW to ground. The snubber components can be placed on the back of the PCB and populated only if necessary. The resister should be 0805 or 1206 size. 12. Allegro strongly recommends the use of current steering (a cut in the ground plane) to prevent current from SW1 from disturbing SW2 and vice versa. Notice the horizontal cut in the ground plane as shown in Figure 17. Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 30 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 VIN RFB1 A8651 SW1 FB CIN RFB2 SS FSET COMP RFSET CP Lo RZ Cout LOAD Css SW CZ GND PGND SINGLE POINT GROUND Could be the therm al/ground pad under the IC Figure 16: Typical Synchronous Buck Converter with Critical Paths/Loops Shown Notice the cut in the GND plane Figure 17: Example PCB Component Placement and Routing Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 31 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 VIN = 5V, fSW = 2 MHz 2.5 2 2 1.5 3.3Vo 1 1.5Vo Current Rating (A) Current Rating (A) VIN = 5V, fSW = 400 kHz 2.5 1.5 3.3Vo 1 1.5Vo 0.5 0.5 0 0 75 80 85 90 95 100 105 110 115 120 75 125 80 85 90 VIN = 5V, fSW = 400 kHz 2.5 2.5 2 2 1.5 3.3Vo 1 1.5Vo Current Rating (A) Current Rating (A) 95 100 105 110 115 120 125 Ambient Temperature (°C) Ambient Temperature (°C) VIN = 5V, fSW = 2 MHz 1.5 3.3Vo 1 1.5Vo 0.5 0.5 0 0 75 80 85 90 95 100 105 Ambient Temperature (°C) 110 115 120 125 75 80 85 90 95 100 105 110 115 120 125 Ambient Temperature (°C) Figure 16: Current Derating Curve versus Output Voltages, Switching Frequency, and Ambient Temperature Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 32 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR A8651 PACKAGE OUTLINE DIAGRAM For Reference Only – Not for Tooling Use (Reference MO-153 ACT) Dimensions in millimeters – NOT TO SCALE Dimensions exclusive of mold flash, gate burrs, and dambar protrusions Exact case and lead configuration at supplier discretion within limits shown 6.50 ±0.10 0.45 8º 0º 4.20 0.65 20 20 0.20 0.09 1.70 C 3.00 4.40 ±0.10 6.40 ±0.20 3.00 6.10 A 0.60 ±0.15 1.00 REF 1 2 1 20X C 1.20 MAX 0.10 C 0.30 0.19 SEATING PLANE 2 4.20 0.25 BSC SEATING PLANE GAUGE PLANE B PCB Layout Reference View 0.65 BSC 0.15 0.00 A Terminal #1 mark area B Reference land pattern layout (reference IPC7351 SOP65P640X110-21M); all pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary to meet application process requirements and PCB layout tolerances; when mounting on a multilayer PCB, thermal vias at the exposed thermal pad land can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5) C NNNNNNN YYWW LLLLLLL 1 Exposed thermal pad (bottom surface) B Standard Branding Reference View N = Device part number = Supplier emblem Y = Last two digits of year of manufacture W = Week of manufacture L = Lot number Figure 17: Package LP, 20-Pin TSSOP with Exposed Thermal Pad Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 33 A8651 Low Input Voltage, Adjustable Frequency Dual Synchronous 2 A / 2 A Buck Regulator with Synchronization, 2x EN, and 2x NPOR Revision History Revision Revision Date 1 July 2, 2014 Description of Revision 2 September 25, 2014 Revised equation 2 3 September 29, 2014 Revised Switching Frequency values 4 February 20, 2015 Removed references to DSYNC Revised equation 4b 5 April 17, 2015 Added -1 option 6 June 10, 2015 Revised soft start fault/hiccup reset voltage values Copyright ©2013-2015, Allegro MicroSystems, LLC Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that the information being relied upon is current. Allegro’s products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of Allegro’s product can reasonably be expected to cause bodily harm. The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its use; nor for any infringement of patents or other rights of third parties which may result from its use. For the latest version of this document, visit our website: www.allegromicro.com Allegro MicroSystems, LLC 115 Northeast Cutoff Worcester, Massachusetts 01615-0036 U.S.A. 1.508.853.5000; www.allegromicro.com 34