Offline LED Driver Intended for ENERGY STAR® Residential LED Luminaire Applications

TND371/D
Offline LED Driver Intended
for ENERGY STAR[
Residential LED Luminaire
Applications
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TECHNICAL NOTE
Overview
The bulk capacitor fills in the missing power providing a
more constant input to the switching regulator maintaining
power flow to the load.
This configuration comes at the expense of poor
utilization or power factor of the input line waveform. Line
current is drawn in high amplitude narrow pulses near the
peaks of the voltage waveform introducing disruptive high
frequency harmonics. Passive solutions are well
documented but typically introduce many additional
components. One approach is the valley-fill type rectifier
where a collection of electrolytic capacitors and diodes
increases the line frequency conduction angle resulting in
improved power factor. In effect, this process charges the
series-connected capacitors from the high line voltage at low
current and discharges them to the switching regulator at a
lower voltage with higher current. A typical application uses
two capacitors and three diodes or, for enhanced power
factor performance, three capacitors and six diodes.
This reference document describes a built-and-tested,
GreenPoint solution for an isolated 8 W constant current
LED driver which is intended to support the residential
power factor requirements of the DoE ENERGY STAR
Standard for Solid State Lighting Luminaires (Version 1.1
− 12/19/08). Some of the typical products in this category
include portable desk lamps, under-cabinet lights, and
outdoor porch lights.
One of the most common power supply topologies for low
power offline LED drivers is an isolated flyback topology.
Unfortunately standard design techniques used for these
supplies typically result in a power factor in the range of
0.5−0.6. This design note describes why the power factor is
low and discusses techniques to improve the power factor.
Finally it illustrates how an existing design was modified to
substantially improve the power factor and easily comply
with the residential power factor requirements.
Background
Two Capacitor
Valley−Fill
The NCP1014LEDGTGEVB evaluation board has been
optimized to drive 1−8 high power high brightness LEDs
such as the Cree XLAMP XR−E/XP−E, Luxeont Rebel,
Seoul Semiconductor Z−POWER, or OSRAM Golden
Dragont. The design is built around the NCP1014, a
compact fixed frequency PWM converter which integrates
a high voltage power switch with internal current limiting.
Since the converter is limited to a maximum power of
approximately 8 W with a universal AC input (90 –
265 Vac), the number of LEDs which can be driven is a
function of the drive current. Specifically for this design
note, the load will be one Cree XLAMP MC−E driven at
630 mA where all LEDs are connected in series. The MC−E
is comprised of 4 LEDs mounted in a single package and the
maximum rated current per LED is 700 mA. The evaluation
board can be modified for other LED drive currents by
making slight modifications to the bill of materials.
A typical off-line flyback power converter utilizes a full
wave bridge rectifier and substantial bulk capacitance
preceding the switching regulator. This configuration is
chosen because twice every line cycle the line power reduces
and ultimately reaches zero before rising to the next peak.
 Semiconductor Components Industries, LLC, 2013
April, 2013 − Rev. 1
Three Capacitor
Valley−Fill
Figure 1. Valley Fill Circuits
While the valley-fill rectifier improves the utilization of
the line current, it does not provide a constant input to the
switching regulator. Power delivered to the load will have
significant ripple at twice the line power frequency. Note
that the 4 diodes rectifying the line power are still needed
bringing the total number of diodes for this solution to 7 or
1
Publication Order Number:
TND371/D
TND371/D
The fixed feedback level represents the current in the
power switch corresponding to the point where the proper
average energy is transferred to the LED over a complete
half cycle of line input. Achieving this fixed feedback level
requires nothing more than increasing the feedback
capacitor C6 to the point that any correction made by
optocoupler U2 is averaged below the line frequency
allowing only compensation for LED voltage and RMS line
voltage variations and not ripple present on the output. The
schematic is shown in Figure 2.
The single stage converter is not without caveats. As
mentioned, energy is transferred to the secondary in a
sine-squared shape. The flyback transformer must couple
this energy and therefore be capable of processing peak
power ~1.4x the average delivered power. The core may be
larger than a conventional flyback transformer design
approach. Moreover, the peak ripple must be below the
maximum rating of the LEDs. Increasing the filter capacitor
integrates the pulsating power delivered to the secondary
and provides more constant current level to the LED load.
The capacitance can be tailored to limit ripple current. In this
case, 2000 mF is sufficient to limit ripple to less than 25%
which is 2x what the non-power factor corrected demo board
design (NCP1014LEDR2GEVB) utilizes.
Note that while the power supply was designed to meet
agency requirements, it has not been submitted for
compliance. Standard safety practices should be used when
this circuit is energized and in particular when connecting
test equipment. During evaluation, input power should be
sourced through an isolation transformer.
10. These diodes and multiple electrolytic capacitors add
cost, degrade reliability and consume considerable circuit
board area.
Another solution is an active power factor boost stage
such as the NCP1607B situated before the flyback converter.
This approach provides superior power factor with typical
performance > 0.98, but it comes with increased parts count,
reduced efficiency and increased complexity. This approach
is most suitable at power levels well above the modest power
level of this application.
Approach
High power factor requires generally sinusoidal line
current and minimal phase displacement between the line
current and voltage. The first step is to have minimal
capacitance before the switching stage to allow a more
sinusoidal input current. This allows the rectified voltage to
follow the line voltage resulting in a more desirable
sinusoidal input current flow. The input voltage to the
flyback converter now follows a rectified sine shape at twice
the line frequency. If the input current is kept to the same
shape, the power factor will be high. The energy delivered
to the load will follow the product of voltage and current
which is a sine-squared shape. As a result of this
sine-squared energy transfer, the load will experience ripple
at twice the line frequency similar in nature to the ripple seen
with a valley fill circuit.
As mentioned above, the input current must be kept to a
nearly sinusoidal shape to achieve high power factor. The
key to this is not allowing the control loop to correct for
output ripple by holding the feedback input at a constant
level with respect to the line frequency. One option is to
significantly increase the output capacitance to reduce the
amount of 120 Hz ripple, an approach which some
applications may require. LEDs for general lighting are
more tolerant to ripple provided the frequency is above the
visible optical perception range. The more compact and less
costly way is to filter the feedback signal going back to the
PWM converter establishing a nearly constant level. This
level fixes the maximum current in the power switch. The
current in the power switch is determined by the applied
instantaneous input voltage divided by the transformer
primary inductance times the length of time the power
switch is conducting. Since the NCP1014 operates at a fixed
frequency, the current cannot rise beyond a certain point as
determined by the input voltage and primary inductance
before the end of the switching period or conduction time.
As a result of the conduction time limitation, the input
current will follow the shape of the input voltage providing
improved power factor.
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1
Neutral
J1−2
C1
100 nF
D2
D4
C4
100 nF
MRA4007 MRA4007
D1
D3
D5
MURA160
3.3k
R3
1
Figure 2. Schematic of NCP1014LEDGTGEVB
C5
2.2 mF
NCP1014
VCC
U1
MMBD914LT1
D6
C2
220 nF
C3
1.5 nF
3
2
4
C6
47 mF
T1C
T1A
R2
47K
2
3
4
1
4
U2
3
2
1
C7
2.2 nF
T1B
FL2
Fly Leads
FL1
D7
MURS320T3
R4
200
89
1000 mF
L1
2.7 mH
Q1
BC857
R5
2.2k
+
+
R15
1k
D9
5.1 V
R12
1K
Optional Dimming Components
R14
820
10k
R13
D8
18 V
R11
100
R9 NF
R8 10R
R7 1R8
R6 1R8
R10
10K
Off Board
MRA4007 MRA4007
1
J2−1
LED Anode
J2−2
E2
−TESTPOINT
LED Cathode
1 J2−6
1 J2−5
1
1
+TESTPOINT
E1
1
Line
C9
1000 mF
Q2
BC846
C10
10 nF
1 R1 4R7
1
J1−1
TND371/D
TND371/D
Design Procedure
the reflected primary voltage is 22*(105/20) = 115.5 V.
Adding the peak input to the reflected voltage yields
489.5 V. Allow ~10 V for inductive voltage spike or 500 V
total. This is well below the 700 V maximum rating of the
internal switch. Peak current is limited to the NCP1014
current limit which is 450 mA. The fast recovery MURA160
is rated for 1 A at 600 V and is a good choice for the diode.
Capacitor C3 must absorb the leakage energy with little
increase in voltage. 1.5 nF is adequate for this low power
application. Resistor R3 must dissipate the leakage energy
but not unnecessarily degrade efficiency. This resistor was
empirically selected as 47 kW. Note this resistor and
capacitor must be rated for 115.5 + 10 = 125.5 V.
Higher switching frequency reduces transformer size but
at the same time increases switching losses. These factors
create a minimum loss point depending on exact transformer
design and selected semiconductors. In this case, the
100 kHz version of NCP1014 was chosen as the balance
point. Efficiency for this monolithic converter is expected to
be about 75%; therefore input power of 10.6 W is expected
for the 8 W output. Input operating range is 90 to 265 Vac.
The NCP1014 includes ON Semiconductor’s DSS or
Dynamic Self Supply circuit which simplifies start up by
reducing parts count. Thermal considerations of this
integrated controller determine the maximum output power.
A copper area on the circuit board will dissipate heat and
reduce the temperature. A bias winding on the flyback
transformer disables the DSS when the converter is running
and reduces dissipation in the converter. Lower operating
temperature enables more power to be delivered to the load.
Bias Supply
D6 rectifies the power delivered by the bias winding.
Voltage stress on D6 is set by the peak input voltage times
the transformer turns ratio plus the primary bias voltage. The
peak input was previously determined as 374 V. The turns
ratio is 105:13 therefore the voltage due to primary input is
374*(13/105) = 46.3 V. The maximum output voltage of
22 V is reflected to set the primary bias voltage. 22*(13/20)
= 14.3 V. D6 reverse voltage is the sum of these two or 60.6
V. MMBD914 is rated 100 V at 200 mA and is a good choice
for this rectifier as the operating current of the NCP1014 is
less than 1.2 mA.
Primary bias is filtered by C4, R3, and C5. Since the sine
squared power transfer of this flyback regulator does not
provide constant energy to the primary bias, the DSS circuit
can activate and introduce visible flicker. To avoid this, the
primary bias must be allowed to discharge partially each half
cycle. C5 was chosen as 2.2 mF to allow this voltage
movement. C4 acts as a peak filter with 0.1 mF. R3 limits the
maximum voltage presented to the NCP1014. This
converter has a protection mode which monitors the primary
bias for excessive voltage. Selecting R3 as 1.5 kW avoids
activating this protection feature which is not needed in this
application.
EMI Filter
Switching regulators draw pulsing current from the input
source. Requirements on harmonic content (see references)
restrict the high frequency content of power supply input
current. Typically a filter comprised of capacitors and
inductors attenuates undesirable signals. Capacitors
connected across the input lines conduct a current which is
90 out of phase with the input voltage. This shifted current
degrades power factor by displacing the phase between the
input voltage and current. A balance must be reached
between the need for filtering and maintaining high power
factor.
Given the nature of electromagnetic interference and
complex characteristics of filter components a starting point
of 100 nF for C1 and C2 was chosen. The differential
inductor L1 was chosen to provide an L−C filter frequency
of about one tenth the switching frequency. The following
formula for inductor value was used:
L+
ǒ1ń2p * 0.1 * fSWǓ
C
2
+
ǒ 1ń2p * 0.1 * 100000 Ǔ 2
100 nF
+ 2.5 mH
Output Rectifier
The output rectifier must carry peak currents well in
excess of the average output current of 630 mA. A rectifier
with low forward voltage and fast recovery time will
minimize losses. Maximum reverse voltage will occur at the
peak of maximum input voltage. This voltage is scaled by
the turns ratio of the transformer. The output voltage is
added to this peak switching voltage resulting in peak
reverse voltage stress. The maximum output voltage is 22 V.
Therefore the peak rectifier voltage is 374*(20/105) + 22 V
= 93.2 V. The MURS320 is a 3 A, 200 V, 35 nS rectifier
providing low forward drop and fast switching. As
mentioned, the output capacitance of 2000 mF will limit the
output ripple current to 25% or 144 mA peak-to-peak.
Select 2.7 mH which is a standard value. From this starting
point, the filter was adjusted empirically to meet conducted
emissions limits. C2 was increased to 220 nF providing
margin on emissions limit. R1 limits the inrush current and
provides a fusible element in the event of a fault. A fuse may
be required to meet safety requirements depending on the
application environment. Note the inrush current is low
given the small total primary capacitance.
Primary Clamp
D5, C3, and R2 form a clamp network to control voltage
spiking due to leakage inductance of the flyback
transformer. D5 should be a fast recovery device rated for
peak input voltage plus the output voltage reflected to the
transformer primary. Maximum input voltage is 265 V ac or
374 V peak. The transformer turns ratio is 105:20. Given a
maximum output voltage of 22 V (the open circuit voltage),
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TND371/D
Current Control
Constant average current output is maintained by
monitoring the voltage drop across Rsense, a resistor in series
with the output. Resistor 11 connects the sense resistor to the
base-emitter junction of a general purpose PNP transistor
Q1. Current through R11 biases Q1 on when the voltage
drop on Rsense is ~0.6 V. Q1 establishes a current flow
through the LED of an optical coupler and is limited by R4.
The transistor of the optical coupler U2 provides feedback
to the NCP1014 converter controlling the output current.
Q1 regulates the peaks of the load current. With a ripple
current of 25% peak to peak the peak current is ~12% higher
than the average. As a result, the LED current will follow
this relationship:
I out +
ǒ
Q1 Vbe
R sense
Ǔ
Ip +
Np +
ń1.12
ǒ 0.75 * 126 V Ǔ
( Ac * B )
Nb + Ns *
+ 1858 mH
+
ǒ 0.1 * 1858 mH * 0.339 A Ǔ
ǒ 0.2 cm 2 * 3 kG Ǔ
+ 105 T
ǒ Ǔ
V sec
V pri
+ 105 *
ǒ
Ǔ
33 V
176 V
[ 20 Turns
ǒ Ǔ
V bias
V sec
+ 20 *
ǒ
8.1 V
Ǔ
12.5 V
[ 13 Turns
Transformers intended to meet safety isolation often
include insulated margins in the windings. This amounts to
physical spacers keeping primary and secondary windings
separated. These spacers severely limit the winding volume
in small transformers. An alternate approach to safety
isolation is triple-insulated wire. This type of wire is
approved for direct contact from primary to secondary
circuits. While not as thin as conventional magnet wire,
using this on a winding with a small number of turns often
provides a design which allows larger wire and therefore less
loss for a given size core. The secondary winding in this
transformer is based on triple-insulated wire resulting in a
compact low-loss design.
The bobbin pins on small transformers are often very
close to the core. As such, some safety agency guidelines do
not accept the spacing as adequate for proper isolation.
Flying leads are used in this design to avoid this potential
issue. The triple-insulated wire exits the winding area and
This LED driver is required to run at a minimum input of
90 V ac which is 126 V peak. In this flyback application, the
peak switching current follows the equation below where Po
= 8 W output, h = efficiency = 0.75, and Vin = 126 V:
ǒh * V inǓ
ǒ 0.339 A * 100 Ǔ
The NCP1014 needs a minimum of 8.1 V to keep the DSS
feature from activating while the converter is running,
which helps reduce dissipation as previously mentioned.
Minimum LED voltage is designed for 12.5 V. The primary
bias voltage will then follow the formula below for number
of turns.
Transformer
ǒ4 * 8 W Ǔ
ǒ0.1 * Lb * IpkǓ
Ns + Nb *
Maintaining high power factor in this circuit relies on a
slow feedback response time allowing only a slight change
in feedback level over a given half cycle of input power. For
this current mode control device that means maximum peak
current will be almost constant over a half cycle. This
improves power factor compared to a traditional feedback
system which attempts to minimize output ripple by
increasing switching current as input voltage decreases and
reducing current when input voltage increases. Capacitor C6
provides the slow loop response by working against the
internal 18 kW pullup resistor of the NCP1014 and the
current drawn from the feedback optical coupler transistor.
C6 was empirically determined in the range of 22 mF to
47 mF.
+
ǒ 500 * 126 V Ǔ
There is considerable latitude in selecting the turns ratio
for the secondary winding. The limitations are the maximum
voltage applied to the power switch and limiting the duty
cycle to 50%. The NCP1014 is rated for 700 V and applying
a derating factor of 80% nets a maximum allowable stress of
560 V. Maximum input voltage was previously established
as 374 V which leaves 560 V – 374 V = 186 V. Subtracting
the 10 V for spike leaves 176 V maximum across the
primary winding.
The output voltage is limited to 22 V for protection in the
event of an open load condition. This value is increased by
50% to 33 V providing some margin on output voltage and
lower duty cycle. Minimum number of secondary turns will
then follow this formula:
Power Factor Control
(4 * P o )
ǒIpk * fSWǓ
+
An E16 core with an area Ac = 0.2 cm2 is a good choice
for this power level. Maximum flux density is set at 3 kG to
minimize losses in a high temperature environment. Primary
turns can be calculated using the following formula:
Given Vbe = 0.6 V then Rsense = 0.536/Iout
Setting Iout = 630 mA requires Rsense = 0.85 W. Rsense is
comprised of four paralleled elements, R6−R9 which is a
lower cost solution than a single power resistor and allows
the value to be easily modified to support other LED current
values if needed. Selecting R6 and R7 as 1.8 W, R8 as 10 W
and leaving R9 open yields 0.83 W. Note that the output
current is temperature sensitive as Q1 base-emitter voltage
varies varies −2 mV/C so the output current should be set
based on typical operating conditions.
I pk +
ǒ500 * VinǓ
+ 0.339 A
From this peak current the primary inductance is
calculated using 100 kHz for fSW.
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TND371/D
Q2 and surrounding components form a constant current
source to bias R11 for dimming. Zener D9 establishes a set
voltage at the top of dimming potentiometer R13. This
voltage is divided by the potentiometer and applied to the
base of Q2. The gain of Q2 is high therefore the base current
is negligible and consequently the base voltage will be set by
the potentiometer. A nearly constant base−emitter voltage
on Q2 means the emitter voltage also tracks the
potentiometer voltage and is impressed on R14. This voltage
divided by the resistance of R14 establishes a current
through R14 which is a function of the potentiometer setting.
Given the assumption of high gain, the collector of Q2 will
have nearly the same current as R14. The result is the
collector of Q2 draws a nearly constant current through R11
as set by the potentiometer.
The current through R11 and thus Q2 must be bounded to
provide optimal performance. The LED output current is
reduced by Q2 current therefore setting the maximum Q2
current will establish the minimum output current. The
minimum output current is established when the
potentiometer delivers the full D9 voltage of 5.1 V to the
base of Q2. R14 will be 0.6 V less due to the Q2 emitter base
voltage resulting in 4.5 V impressed across it. Selecting
50 mA as the minimum LED current results in 50 mA *
0.83 W = 42 mV V across Rsense. Subtracting 42 mV from
the Q1 base−emitter voltage of 600 mV leaves 558 mV
across R11 representing a required current of 558 mV /
100 W = 5.58 mA. R14 maximum voltage of 4.5 V /
5.58 mA = 806 W. Select 820 W for R14.
To ensure maximum brightness, no offset current should
flow through R11. In other words, Q2 should be completely
shut off. The lowest setting of the potentiometer should be
below the minimum base-emitter voltage of Q2, but not so
low as to create a significant portion of the adjustment travel
with no visible change in LED brightness. The formula
below defines the minimum control voltage at the base of
Q2.
connects to the PCB in a location far enough away from the
transformer to satisfy safety spacing requirements.
Special bobbin designs are available on the market from
suppliers like Wurth-Midcom which provide the required
safety spacing without the need for flying leads. This type of
bobbin could be a benefit in some applications.
Open Load Protection
A zener diode provides open load protection. Should the
output voltage increase beyond the knee voltage of D8 plus
the forward voltage of the LED in the optical coupler,
current will flow issuing a feedback signal to the NCP1014
protecting against excessive output voltage. Open load
voltage is set by the sum of voltage of D8, drop across R4,
and the LED in the opto-coupler. D8 was selected as 18 V
establishing about 22 V maximum output to maintain output
current control over the forward voltage range of the MC−E
LED array with some margin within the capabilities of the
NCP1014. Higher output voltages are possible with
selection of higher voltage rated capacitors and output
rectifier but will require a reduction in output current to
maintain about an 8 W maximum load.
Bleeder and Filter
R10 and C10 provide a small discharge path and noise
filtering of the output. The LEDs should always be
connected before the circuit is energized as the output
capacitors can store a significant amount of energy and this
would be discharged through the LEDs at the moment of
connection. The output capacitors store a significant amount
of energy which will be discharged through the LEDs at the
moment of connection. The resultant current surge could
damage the LEDs immediately or result in latent damage
which ultimately will shorten the anticipated long life of the
LEDs.
Analog Dimming
This reference design has an optional control section
which implements analog current adjustment for dimming.
Components R12, R14, R15, D9, Q2, and a connection to
potentiometer R13 can be added to the board for this
purpose. When the voltage drop of Rsense plus the drop
across R11 equals the base−emitter threshold, Q1 turns on
and provides the feedback signal to the NCP1014.
Introducing a bias across R11 will reduce the required
voltage and therefore current through Rsense and the LED
load. The LED dimming control scheme is designed to
introduce an offset in R11. Since the common mode voltage
present on R11 changes with temperature and LED
configuration a simple resistor biasing network will perform
poorly as a dimming control. This can be addressed by
passing a controlled current through R11 to develop a
voltage drop independent of the common mode voltage
resulting in more predicable dimming behavior. Varying the
bias current changes the voltage drop across R11. In this
way, varying the controlled current through R11 will have an
inverse effect on the current delivered to the load.
R15 +
ǒ
R13 * V base
VD9 * V base
Ǔ
Setting the voltage to 0.5 V is a good starting point.
R15 +
* 0.5
ǒ10000
Ǔ + 1.09 kW
5.1 * 0.5
Rounding R15 down (R15 = 1 kW) will ensure no loss of
LED current as the transistors heat up.
Capacitor Lifetime
One of the considerations of LED lighting is that the driver
and LEDs should have a comparable operating life. While
power supply reliability is influenced by multiple factors,
the electrolytic capacitors are critical to the overall
reliability of any electronic circuit. Analyzing the capacitors
in the application and selecting the proper electrolytic
capacitors is essential for long operating life. Useful life of
an electrolytic capacitor is strongly affected by ambient
temperature and internal temperature rise due to ripple
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TND371/D
current acting on internal resistance. The manufacturer’s
rated life of an electrolytic capacitor is based on exposure to
maximum rated temperature with maximum rated ripple
current applied. A typical capacitor rated lifetime might be
5000 hours at 105C. Operating stresses lower than the rated
levels will exponentially increase the useful life of the
capacitor. While choosing capacitors with long rated life and
high temperature rating enhances operating life, capacitors
rated for lower temperature and shorter life may still be
suitable depending on the stress and operating temperature
resulting in a lower cost solution. Formulae for useful life
can be found on manufacturer’s websites.
Useful capacitor life follows this equation:
ǒ
L + Lr * 2^
*
ǒ
(T max * T san)
ǒ ǒǒ
K^
10
1*
ǓǓ *
ǒ Ǔ Ǔ
I rpl
I max
^ 2
*
80
EFFICIENCY (%)
75
2
3
4
5
6
7
OUTPUT POWER (W)
8
9
Figure 3. Efficiency Across Output Load
with Vin = 115 Vac
ǓǓ
80
75
EFFICIENCY (%)
70
65
60
55
50
90
The capacitors selected for this application are the
Panasonic ECA−1EM102 which is rated 1000 mF, 25 V,
850 mA, 2000 hr, and 85C. Assign a 50C ambient as Tsan.
Worst case measured ripple for the pair of output capacitors
is 740 mA rms or 370 mA per capacitor. Substituting
parameters in the equation above yields:
ǒ
60
50
Where:
L = calculated useful life
Lr = rated life
Tmax = rated temperature
Tsan = surrounding normalized temperature
Irpl = measured ripple current, Imax = rated ripple current
Tcr = capacitor core temperature rise
K is equal to 2 for applications where ripple current is
below rated current. For an ambient temperature of 55C,
set
Tcr = 30C.
L + 2000 * 2^
65
55
T cr
10
70
115
140
165
190
215
LINE VOLTAGE (Vac)
240
265
Figure 4. Efficiency Across Line Voltage
with Pout = 8.5 W
Output power of the LED driver is characterized from
open circuit condition through short circuit on the output.
The intended operating range is defined by the LED forward
voltage where output current is controlled at a nearly
constant level. Figure 5 below shows this output
characteristic and illustrates the three regions of operation:
constant voltage, constant power and finally constant
current.
Ǔ ^ 2 Ǔ * 30
ǒ(85 10* 50)ǓǓ * ǒ2^ ǒǒ1 * ǒ0.37
ǓǓ
0.85
10
+ 122 069 hours
The capacitor lifetime is in line with the requirements for
LED lighting based on the expected operating temperature.
Test Results
Data was collected on the NCP1014LEDGTGEVB with
a load of 4 LEDs operating at approximately 630 mA unless
otherwise stated. Tests were conducted in prevailing lab
environment after a one hour warm up period. Figures 3
and 4 show efficiency as a function of output load and
applied input voltage respectively.
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TND371/D
400
Constant Voltage
390
20
380
Constant
Power
LED CURRENT (mA)
LED FORWARD VOLTAGE (Vdc)
25
15
10
Constant
Current
5
370
360
350
340
330
320
310
0
0
100
200
300
400
500
600
700
300
800
90
LED CURRENT (mA)
115
140
165
190
215
240
265
LINE VOLTAGE (Vac)
Figure 5. Output Current/Voltage Transfer Function,
115 Vac
Figure 7. Current Regulation Across Line with
Nominal Current set 350 mA, Vf = 12.6 Vdc
LED output current as a function of input voltage is shown
in Figure 6. Dimmer circuit was adjusted for maximum
output. At this point, the controller is operating at peak
power and some reduction in output current occurs as the
efficiency is lower.
Power factor as a function of line voltage is shown in
Figure 8 below. Note the power factor is greater than 0.8 for
the input voltage range of 90 Vac to 135 Vac which well
exceeds the ENERGY STAR requirements for residential
luminaires.
0.95
725
0.90
0.85
POWER FACTOR
LED CURRENT (mA)
700
675
650
625
0.75
0.70
0.65
0.60
600
575
90
0.80
0.55
115
140
165
190
215
240
0.50
265
90
LINE VOLTAGE (Vac)
115
140
165
190
215
240
265
LINE VOLTAGE (Vac)
Figure 6. Current Versus Line Voltage, Vf = 13.1 V at
653 mA Current
Figure 8. Power Factor versus Line Voltage
Lighting products in some regions are required to meet
International Standard IEC 61000−3−2 Class C limits for
harmonic content of input current. In this case, the
applicable limits are for fixtures drawing less than 25 W.
Table 1 shows the results for this LED driver.
Since many high brightness power LEDs are
characterized at 350 mA, the dimming control was adjusted
to provide 350 mA load current at 115 Vac input voltage.
Figure 7 depicts the load current as the line voltage is varied
from 90 Vac to 265 Vac.
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TND371/D
Conducted emissions data was collected using a load of 4
LEDs (1 Cree MC−E). Figure 9 below shows EN55022
Class B limit as the red line and a 6 dB margin as the dashed
line below that. These results show significant margin on
emissions.
Table 1.
Harmonic
115 Vac
60 Hz
230 Vac
50 Hz
Class
C Limit
Third
52.4%
65.0%
86.0%
Fifth
23.0%
47.9%
61.0%
80
70
60
EN 55022; Class B Conducted, Average
50
dBuV
40
Line Average
30
20
10
0
−10
−20
(Start = 0.15, Stop = 30.00) MHz
1
10
Figure 9. Conducted Emissions with Class B Limits, Vin = 115 Vac
The oscilloscope images below were collected using an
isolated differential probe for primary connected
components, a 10X probe for secondary components, and an
isolated current probe for current measurements. Figure 10
below is the NCP1014 drain voltage at 230 Vac input. Peak
voltage is well below the maximum part rating of 700 V.
Figure 10. Drain Switching Voltage Waveform
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TND371/D
approximately 11% at 115 Vac and drops at higher ac line.
This is the main reason that the current slightly increases at
high line as illustrated in Figure 6.
The output current for 115 Vac and 230 Vac input is shown
in Figures 11 and 12. This image shows the DC as well as
AC components. Scale factor is 150 mA per division.
Peak-to-peak ripple current is 144 mA which is
Figure 11. LED Ripple Current with 115 Vac Input
Figure 12. LED Ripple Current with 230 Vac Input
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TND371/D
Figure 13 shows the start up current characteristic after
application of 115 Vac input. Scale factor 150 mA per
division. Output current rise time is ~810 ms.
Figure 13. Startup at 115 Vac
Summary
quantitative requirements which ensure that customers can
be confident in their decisions. These new standards do add
new requirements to LED drivers such as power factor
correction that require novel solutions to meeting the
requirements without adding increased complexity or cost.
Solid state lighting brings great promise in reducing
global energy consumption while at the same time providing
customers with a long lifetime product which can bring new
lighting capabilities to the market. ENERGY STAR
standardization for Solid State Luminaires also establishes
Reference Materials:
[1] ENERGY STAR SSL Luminaire Specification, Version 1.1
http://www.ENERGYSTAR.gov/index.cfm?c=new_specs.ssl_luminaires
[2] Cree XLAMP MC−E Specification
http://www.cree.com/products/xlamp_mce.asp
[3] Fraen Reflector Optics for Cree MC−E
http://www.fraensrl.com/prodinfo.html
[4] ON Semiconductor Design Note DN06051: Improving the Power Factor of Isolated Flyback Converters for
Residential ENERGY STAR LED Luminaire Power Supplies
http://www.onsemi.com/pub_link/Collateral/DN06051−D.PDF
[5] LED Desk Lamp Conversion White Paper TND358:
http://www.onsemi.com/pub_link/Collateral/TND358−D.PDF
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TND371/D
APPENDIX
Table 2. BILL OF MATERIALS
Value
Description
Part
Reference
Manufacturer
Manufacturer Part Number
100 nF
CAP, 100 nF, 275 Vac
C1
Panasonic
ECQ−U2A104ML
220 nF
CAP, 220 nF, 275 Vac
C2
Panasonic
ECQ−U2A224ML
1.5 nF
CAP, 1.5 nF, 1 kV ceramic
C3
Murata
DEBB33A152KA2B
100 nF
CAP 100 nF 25 V 10% 0603 SMD
C4
Panasonic
ECJ−1VF1H104Z
2.2 mF
CAP 2.2 mF 50 V 5 kHr 105C 5x11
Radial
C5
Panasonic
EEU−EB1H2R2S
47 mF
CAP 47 mF 16 V 2 kHr 85C 5x11
C6
Panasonic
ECA−1CM470
2.2 nF
CAP 2.2 nF ”Y1” 250 Vac
C7
TDK Corp
CD12−E2GA222MYNS
C8 C9
Panasonic
ECA−1EM102
1000 mF
10 nF
CAP 1000 mF 25 V 2 kHr 85C
10x20
C10
Panasonic
ECJ−1VB1H103K
MRA4007
CAP 10 nF 50 V 10% 0603 SMD
Rectifier, 1000 V, 1 A, SMA
D1 D2 D3
D4
ON Semiconductor
MRA4007T3
MURA160
Rectifier, 600 V, 1 A, SMA
D5
ON Semiconductor
MURA160T3
MMBD914LT1
Rectifier,100 V, 200 mA, SOT23
D6
ON Semiconductor
MMBD914LT1
MURS320T3
Rectifier, 200 V, 3 A, SMC
D7
ON Semiconductor
MURS320T3
Zener Diode, 18 V, 225 mW, SOT23
D8
ON Semiconductor
BZX84C18LT1G
MMBZ5231
Zener Diode, 5.1 V, 500 mW,
SOT23
D9
ON Semiconductor
MMBZ5231BLT1G
TESTPOINT
Terminal Test point
E1 E2
Kobiconn
151−103−RC
BZX84C18LT1
Conn
Screw terminal
J1
Weidmuller
1716020000
Conn
Header, 1 row, 6 pin
J2
Tyco
535676−5
2.7ĂmH
IND, PWR, 2.7 mH
L1
Coilcraft
RFB0810−272
BC857
TRAN, PNP, 45 V, 100 MA, SOT23
Q1
ON Semiconductor
BC857BLT1G
BC846
TRAN, NPN, 65 V, 100 MA, SOT23
Q2
ON Semiconductor
BC846BLT1G
4R7
Fusible resistor, 4R7, 0.5 W
R1
Vishay
NFR25H0004708JR500
47k
RES, 47k, 1 W
R2
Vishay
PR01000104702JR500
1.5k
RES, 1.5k, 1/10 W, SMD 0603
R3
Panasonic
ERJ−3EKF1501V
200
RES, 200R, 1/10 W, SMD 0603
R4
Panasonic
ERJ−3EKF2000V
2.2k
RES, 2.2k, 1/10 W, SMD 0603
R5
Panasonic
ERJ−3EKF2201V
1R8
RES, 1R8, 1/4 W, SMD 1206
R6 R7
Vishay
CRCW12061R80FKEA
10R
RES, 10R, 1/4 W, SMD 1206
R8
Vishay
CRCW120610R0FKEA
Not Used
R9
10k
RES, 10k, 1/4 W, SMD 1206
R10
Rohm
MCR18EZPJ103
100
RES, 100R, 1/10 W, SMD 0603
R11
Panasonic
ERJ−3EKF1000V
1k
RES, 1k, 1/10 W, SMD 0603
R12 R15
Panasonic
ERJ−3EKF1001V
10k
Potentiometer, 10k
R13
CTS
026TB32R103B1A1
820
RES, 820R, 1/10 W, SMD 0603
R14
Panasonic
ERJ−3EKF8200V
T1
Transformer
T1
ICE Components
Wurth−Midcom
TO09035
750811041
IC, CUR MODE CONT, 100 kHz,
SOT−223
U1
ON Semiconductor
NCP1014ST100T3G
OPTOCOUPLER, TRAN O/P, SMT4
U2
NEC ELECTRONICS
PS2561L−1−A
NCP1014
PS2561
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TND371/D
TRANSFORMER DESIGN SPECIFICATION
Project / Customer: ON Semiconductor − Aspen Greenpoint Desk Lamp (6 Feb 09)
Part Description: 8 watt flyback transformer, 100 kHz, 24 V / 360 mA
Schematic ID: T1
Inductance: 1.8 mH 5%
Bobbin Type: 8 pin horizontal mount for EF16
Core Type: EF 16 (E16/8/5); 3C90 or similar material
Core Gap: Gap for 1.8 mH
Table 3. WINDING DETAIL (in order of assembly)
Operation
Primary Bias
Start
Finish
Pin 3
Pin 2
Insulate
Primary winding
Pin 4
Pin 1
Insulate
24 V Secondary
Fly1
1.5”
Fly2
1.5”
Insulate
Details
Notes
13T #32HN
Spread across bobbin in one layer
1T Mylar tape
Lap ~0.1 inch
105T #32HN
Wind in 3 layers, 35 turns/layer
1T Mylar tape
Lap ~0.1 inch
20T #26 TEX-E
Spread across bobbin in one layer
Mark start lead with tape
3T Mylar tape
Assemble core
Gap for 1.8 mH
Vacuum varish
Mark with part number
Hipot: 3 kV from Primary to Secondary for 1 Minute
1
4
3
Fly 1
Fly 2
tape
4
5
6
3
7
2
8
1
(Bottom View − Face Pins)
Fly 2
Pins 1−4
2
Figure 14. Schematic
Fly 1
Breakout
at top of
coil
Pins 5−8
Figure 15. Lead Breakout / Pinout
SUPPORT INFORMATION
IEC 61000−3−2: International Standard for Electromagnetic Compatibility
DoE ENERGY STAR Standard for Solid State Lighting Luminaires (Version 1.1 − 12/19/08)
GreenPoint is a registered trademark of Semiconductor Components Industries, LLC (SCILLC).
ENERGY STAR and the ENERGY STAR mark are registered U.S. Marks; XLamp is a registered trademark of Cree, Inc.; Z−POWER is a registered trademark of Seoul Semiconductor Co., Ltd.; Golden Dragon is a trademark of Osram; Luxeon is a trademark of Lumileds Lighting.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC owns the rights to a number of patents, trademarks,
copyrights, trade secrets, and other intellectual property. A listing of SCILLC’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent−Marking.pdf. SCILLC
reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any
particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without
limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications
and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC
does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for
surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where
personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and
its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly,
any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture
of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
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TND371/D