MCP3909 / dsPIC33FJ128GP206
3-Phase Energy Meter
Reference Design
© 2009 Microchip Technology Inc.
DS51823A
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Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART,
rfPIC and UNI/O are registered trademarks of Microchip
Technology Incorporated in the U.S.A. and other countries.
FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor,
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Solutions Company are registered trademarks of Microchip
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Analog-for-the-Digital Age, Application Maestro, CodeGuard,
dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN,
ECONOMONITOR, FanSense, HI-TIDE, In-Circuit Serial
Programming, ICSP, Mindi, MiWi, MPASM, MPLAB Certified
logo, MPLIB, MPLINK, mTouch, Octopus, Omniscient Code
Generation, PICC, PICC-18, PICDEM, PICDEM.net, PICkit,
PICtail, PIC32 logo, REAL ICE, rfLAB, Select Mode, Total
Endurance, TSHARC, UniWinDriver, WiperLock and ZENA
are trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
All other trademarks mentioned herein are property of their
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© 2009, Microchip Technology Incorporated, Printed in the
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Printed on recycled paper.
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Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
DS51823A-page 2
© 2009 Microchip Technology Inc.
MCP3909 / dsPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Table of Contents
Preface ........................................................................................................................... 7
Introduction............................................................................................................ 7
Document Layout .................................................................................................. 8
Conventions Used in this Guide ............................................................................ 9
Recommended Reading...................................................................................... 10
The Microchip Web Site ...................................................................................... 10
Customer Support ............................................................................................... 10
Document Revision History ................................................................................. 10
Chapter 1. Meter Overview
1.1 Introduction ................................................................................................... 11
1.2 Meter Design Parameters ............................................................................ 11
1.3 Power Calculations ....................................................................................... 12
1.4 Getting Started ............................................................................................. 13
Chapter 2. Hardware Description
2.1 Overview ...................................................................................................... 17
2.2 Analog Front End Circuitry ........................................................................... 18
2.3 Analog-To-Digital Conversion ...................................................................... 20
2.4 dspic33f Hardware Connection And Peripheral Usage ................................ 22
2.5 Power Supply ............................................................................................... 25
Chapter 3. Firmware
3.1 Overview ...................................................................................................... 27
3.2 Main Loop ..................................................................................................... 27
3.3 Calculation() - Calculating Electrical Parameters ......................................... 29
3.4 ADC Sampling Scheme For Calculations .................................................... 33
3.5 ReadING A/D Data Of The MCP3909 Device .............................................. 35
3.6 Communication Of UART Interface .............................................................. 37
3.7 Resource Configuration ................................................................................ 37
3.8 Description Of Project Files .......................................................................... 38
Chapter 4. Meter Calibration
4.1 Introduction ................................................................................................... 39
4.2 Current/voltage Calibration .......................................................................... 39
4.3 Apparent Power Calibration ......................................................................... 40
4.4 Phase Lag Calibration ................................................................................. 41
© 2009 Microchip Technology Inc.
DS51723A-page 3
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
Chapter 5. PC Software
5.1 Overview And Installation ............................................................................. 43
5.2 Establish Communication ............................................................................. 44
5.3 Basic Parameters Output Screen ................................................................. 45
5.4 Phase A/B/C Harmonic Output Screen ........................................................ 45
5.5 Distortion Rate .............................................................................................. 46
5.6 Harmonic Power ........................................................................................... 46
5.7 Energy Accumulation ................................................................................... 47
5.8 Calibration Step 1 - Reset All Calibration ..................................................... 47
5.9 Linearity Calibration ...................................................................................... 48
5.10 Apparent Power Calibration ....................................................................... 49
5.11 Phase Lag Calibration ................................................................................ 50
Chapter 6. Meter Communications Protocol
6.1 Introduction ................................................................................................... 51
6.2 Test Connection Command .......................................................................... 52
6.3 Total Data Request ...................................................................................... 52
6.4 Status Register ............................................................................................. 54
6.5 Harmonic Content Command ....................................................................... 54
6.6 Total Harmonic Distortion (THD) Command ................................................ 55
6.7 Start Energy Measurement Command ......................................................... 56
6.8 Stop Energy Measurement Command ......................................................... 56
6.9 Harmonic Power Command ......................................................................... 57
6.10 Calibrate Meter Voltage/current Command ................................................ 58
6.11 Calibrate Phase Lag Command ................................................................. 59
6.12 Calibrate Apparent Power Command ......................................................... 59
6.13 Calibrate Energy Pulse Command ............................................................. 60
6.14 Reset All Meter Calibration Values Command ........................................... 60
6.15 Calibrate Meter Constant (Energy Pulse Output Constant) ....................... 61
Appendix A. Schematics and Layouts
A.1 Introduction .................................................................................................. 63
A.2 Schematics And Pcb Layout ........................................................................ 63
Appendix B. Bill Of Materials (BOM)
Appendix C. Power Calculation Theory
C.1 Overview ...................................................................................................... 75
C.2 Synchronous Sampling And Quasi-synchronous Sampling ........................ 75
C.3 The Harmonic Analysis Algorithm Of Quasi-synchronous Sampling ........... 82
C.4 Measuring The Voltage/current Rms Value And Power Using Quasi-synchronous Sampling Algorithm ........................................................................ 84
C.5 Measuring Frequency .................................................................................. 87
C.6 Improving Measurement Precision Of Quasi-synchronous Sampling Algorithm
................................................................................................................. 89
C.7 Measuring Secondary Parameters .............................................................. 91
C.8 Apparent Power Of Each Phase And Total Apparent Power ....................... 91
C.9 Power Factor Of Each Phase And Total Power Factor ............................... 91
DS51723A-page 4
© 2009 Microchip Technology Inc.
C.10 Active Energy And Reactive Energy .......................................................... 92
C.11 Positive/negative Active Energy, Positive/negative Reactive Energy And
Four-quadrant Reactive Energy ............................................................. 92
C.12 Harmonic Components Of Current, Voltage And Total Harmonic Distortion
................................................................................................................. 94
C.13 Compensation For Ratio Error And Phase Lag ......................................... 95
C.14 Relationship Between Error And Current ................................................... 96
C.15 Ratio Error Compensation ......................................................................... 97
C.16 Phase Lag Compensation ......................................................................... 98
Appendix D. 50/60 Hz Meter Operation
D.1 Firmware Versions ..................................................................................... 103
Worldwide Sales and Service .................................................................................. 104
© 2009 Microchip Technology Inc.
DS51723A-page 5
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
NOTES:
DS51723A-page 6
© 2009 Microchip Technology Inc.
MCP3909 / dsPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Preface
NOTICE TO CUSTOMERS
All documentation becomes dated, and this manual is no exception. Microchip tools and
documentation are constantly evolving to meet customer needs, so some actual dialogs
and/or tool descriptions may differ from those in this document. Please refer to our web site
(www.microchip.com) to obtain the latest documentation available.
Documents are identified with a “DS” number. This number is located on the bottom of each
page, in front of the page number. The numbering convention for the DS number is
“DSXXXXXA”, where “XXXXX” is the document number and “A” is the revision level of the
document.
For the most up-to-date information on development tools, see the MPLAB® IDE on-line help.
Select the Help menu, and then Topics to open a list of available on-line help files.
INTRODUCTION
This chapter contains general information that will be useful to know before using the
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design. Items discussed in
this chapter include:
•
•
•
•
•
•
Document Layout
Conventions Used in this Guide
Recommended Reading
The Microchip Web Site
Customer Support
Document Revision History
© 2009 Microchip Technology Inc.
DS51823A-page 7
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
DOCUMENT LAYOUT
This document describes how to use the MCP3909 / dsPIC33F 3-Phase Energy Meter
Reference Design as a development tool to emulate and debug firmware on a target
board. The manual layout is as follows:
This document describes how to use the MCP3909 / dsPIC33F 3-Phase Energy Meter
Reference Design as a development tool. The manual layout is as follows:
• Chapter 1. “Meter Overview” - Summarizes the meter specifications and a quick
getting started section
• Chapter 2. “Hardware Description” - A detailed explanation of the different
circuit blocks, their function, and implementation
• Chapter 3. “Firmware” - All the calculations performed by the dsPIC33F are
described here
• Chapter 4. “Meter Calibration” - Explains how the meter is calibrated to
accuracy
• Chapter 5. “PC Software” - Includes screen shots of the viewer/calibration
software included with the system
• Chapter 6. “Meter Communications Protocol” - The UART commands used to
communicate to the meter
• Appendix A. “Schematics and Layouts” - Both PCB, SCH files are located here
for the 2 board system
• Appendix B. “Bill Of Materials (BOM)” - Part number and ordering information
for all components of the energy meter
• Appendix C. “Power Calculation Theory” - A detailed explanation of the theory
behind the calculations described in Chapter 3. “Firmware”
• Appendix D. “50/60 Hz Meter Operation” - Instructions on converting the meter
for use in a 60 Hz line frequency environment
DS51823A-page 8
© 2009 Microchip Technology Inc.
Preface
CONVENTIONS USED IN THIS GUIDE
This manual uses the following documentation conventions:
DOCUMENTATION CONVENTIONS
Description
Arial font:
Italic characters
Represents
Examples
Referenced books
Emphasized text
A window
A dialog
A menu selection
A field name in a window or
dialog
A menu path
MPLAB® IDE User’s Guide
...is the only compiler...
the Output window
the Settings dialog
select Enable Programmer
“Save project before build”
A dialog button
A tab
A number in verilog format,
where N is the total number of
digits, R is the radix and n is a
digit.
A key on the keyboard
Click OK
Click the Power tab
4‘b0010, 2‘hF1
Italic Courier New
Sample source code
Filenames
File paths
Keywords
Command-line options
Bit values
Constants
A variable argument
Square brackets [ ]
Optional arguments
Curly brackets and pipe
character: { | }
Ellipses...
Choice of mutually exclusive
arguments; an OR selection
Replaces repeated text
#define START
autoexec.bat
c:\mcc18\h
_asm, _endasm, static
-Opa+, -Opa0, 1
0xFF, ‘A’
file.o, where file can be
any valid filename
mcc18 [options] file
[options]
errorlevel {0|1}
Initial caps
Quotes
Underlined, italic text with
right angle bracket
Bold characters
N‘Rnnnn
Text in angle brackets < >
Courier New font:
Plain Courier New
Represents code supplied by
user
© 2009 Microchip Technology Inc.
File>Save
Press <Enter>, <F1>
var_name [,
var_name...]
void main (void)
{ ...
}
DS51823A-page 9
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
RECOMMENDED READING
This user's guide describes how to use MCP3909 / dsPIC33F 3-Phase Energy Meter
Reference Design. Other useful documents are listed below. The following Microchip
documents are available and recommended as supplemental reference resources.
MCP3909 Data Sheet, “Energy Metering IC with SPI Interface and Active Power
Pulse Output“ (DS22025)
This data sheet provides detailed information regarding the MCP3909 device.
AN994 Application Note “IEC61036 Meter Design using the MCP3905/6 Energy
Metering Devices” (DS00994)
This application note documents the design decisions associated with using the
MCP390X devices for energy meter design and IEC compliance.
THE MICROCHIP WEB SITE
Microchip provides online support via our web site at www.microchip.com. This web
site is used as a means to make files and information easily available to customers.
Accessible by using your favorite Internet browser, the web site contains the following
information:
• Product Support – Data sheets and errata, application notes and sample
programs, design resources, user’s guides and hardware support documents,
latest software releases and archived software
• General Technical Support – Frequently Asked Questions (FAQs), technical
support requests, online discussion groups, Microchip consultant program
member listing
• Business of Microchip – Product selector and ordering guides, latest Microchip
press releases, listing of seminars and events, listings of Microchip sales offices,
distributors and factory representatives
CUSTOMER SUPPORT
Users of Microchip products can receive assistance through several channels:
•
•
•
•
Distributor or Representative
Local Sales Office
Field Application Engineer (FAE)
Technical Support
Customers should contact their distributor, representative or field application engineer
(FAE) for support. Local sales offices are also available to help customers. A listing of
sales offices and locations is included in the back of this document.
Technical support is available through the web site at: http://support.microchip.com
DOCUMENT REVISION HISTORY
Revision A (November 2009)
• Initial Release of this Document.
DS51823A-page 10
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 1. Meter Overview
1.1
INTRODUCTION
The MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design is a fully
functional energy meter with many advanced features such as harmonic analysis, per
phase distortion information, voltage sag detection, four quadrant energy measurement, and active and reactive power calculation. It uses Microchip’s powerful 16-bit
dsPIC33F Microcontroller Unit (MCU).
The MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design is unique in the
fact that all calculations take advantage of the dsPIC33F DSP engine, and all output
quantities are calculated in the frequency domain through the use of direct fourier
transforms (DFT). This approach yields a large volume of outputs for a variety of meter
designs, from simple active power only energy meters, to advanced energy meters
requiring harmonic analysis.
Another significant advantage of this design, is that the dsPIC firmware implements a
quasi-synchronous sampling algorithm, eliminating the need for external zero-crossing
detection and PLL (Phase Locked Loop) circuit for the synchronization of ADC samples
to line frequency. The line frequency is measured in software and corrected for measurement errors caused by frequency fluctuations in the power grid. This additional
processing on the dsPIC reduces the overall meter cost by eliminating the requirement
for a PLL circuit.
1.2
METER DESIGN PARAMETERS
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Accuracy Class: 0.2S
Rated Current Ib: 3 X 5(20)A
3-phase 4-wire System
Line Frequency Range: 47-53 Hz or 57-63 Hz
(firmware option, see Appendix D. “50/60 Hz Meter Operation”)
ADC Sampling Rate: 12.8 ksps to 3.2 ksps
Voltage Input:
- 3 x 220/380V
- 3 x 57.7/100V (3-Phase, 4-Wire)
Starting Current: 0.001 IB
Active Power Measurement Range: 0-13200W, Precision Class: 0.2.
Reactive Power Measurement Range: 0-13200VAR, Precision Class: 0.2.
Power Factor (PF) Precision Class: 0.2.
Frequency Measurement: Precision Class: 0.2, Max. Error 0.1 Hz
Harmonic Component Measurement of Voltage Input: 2ND-31ST Harmonic
Harmonic Component Measurement of Current Input: 2ND-31ST Harmonic
Creeping: Anti-creeping Design (<0.0008 IB)
Two Pulse Outputs: Total Phase Active Power, Total Phase Reactive Power
Pulse Constant: 3200 Imp/kWh
© 2009 Microchip Technology Inc.
DS51723A-page 11
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
1.3
POWER CALCULATIONS
A summary of all the calculations performed by this energy meter are summarized
below.
Chapter 3. “Firmware” provides an explanation on the firmware implementation,
Appendix C. “Power Calculation Theory” is included to show the theory behind this
firmware.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
DS51723A-page 12
Power Grid Frequency
RMS Voltage Of Each Phase
RMS Current Of Each Phase
RMS Neutral Current
Active Power Of Each Phase
Reactive Power Of Each Phase
Apparent Power Of Each Phase
Power Factor Of Each Phase
Fundamental Active Power Of Each Phase
Fundamental Reactive Power Of Each Phase
Harmonic Active Power Of Each Phase
Harmonic Reactive Power Of Each Phase
Total Active Power:
- The Algebraic Sum Of Active Power Of Three Phases
Total Reactive Power:
- The Algebraic Sum Of Reactive Power Of Three Phases
Total Apparent Power:
- The Algebraic Sum Of Apparent Power Of Three Phases
Total Power Factor
Phase Missing / Line voltage sag detection and alarm
Total Active Energy:
- The Algebraic Sum Of Positive/negative Active Energy
Positive/negative Active Energy
Positive/negative Reactive Energy
Four-quadrant Reactive Energy
Voltage/current Harmonic Content Of Each Phase
© 2009 Microchip Technology Inc.
Meter Overview
.
Current Transformer
UART Interface
ICD Interface
dsPIC33
FIGURE 1-1:
1.4
MCP3909
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design.
GETTING STARTED
To describe how to use the MCP3909 / dsPIC33F 3-Phase Energy Meter Reference
Design, the following example is given using a 4-wire, 3-phase, 220VAC line voltage
and connection. The rated current of the energy meter is 5(20)A.
The energy meters are not shipped fully calibrated, and a full calibration should be
performed to show the true meter accuracy. See Chapter 4. “Meter Calibration” for
more information.
All connections described in this section are dependent on the choice of current
sensing element and a secondary external transformer may be required in higher
current meter designs.
© 2009 Microchip Technology Inc.
DS51723A-page 13
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
Step 1: Connect the meter to 220V line and load
The diagram below shows where the voltage and current connections should be made.
It is not required to connect all 3 phases for the meter to be operational.
220VAC should be placed between either VA, VB, VC and NIN, NOUT.
The AC load for a given phase should then be connected to the IIN and IOUT of a given
phase.
IAIN VA IAOUT
FIGURE 1-2:
IBIN VB IBOUT
ICIN VC ICOUT
Meter Case Bottom.
Step 2: Turn On Line/Load Power to the Meter
Turn on the power to the energy meter. D1 should be lit showing the meter has power.
At this point, if a load is connected and the meter is measuring power, the power LED,
D1, should be blinking.
DS51723A-page 14
© 2009 Microchip Technology Inc.
Meter Overview
Step 3: Connect the RS-232 Cable
1. Connect the RS-232 cable from the energy meter to a Personal Computer (PC),
using either COM1, COM2, or COM6.
Step 4: Run the PC Calibration Software
After installing and running the PC energy meter software on a PC running a
Windows™ Operating System, and selecting the proper comm port for RS-232
communication, the following screen should show real-time meter results. The
following chapters include more detail on the firmware, calculation, and PC software.
.
FIGURE 1-3:
“PM_Viewer” or Power Meter Viewer PC Software.
© 2009 Microchip Technology Inc.
DS51723A-page 15
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
NOTES:
DS51723A-page 16
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 2. Hardware Description
2.1
OVERVIEW
Figure 2-1 is the basic hardware block diagram of the MCP3909 / dsPIC33F 3-Phase
Energy Meter Reference Design. The hardware includes the dsPIC33F and ICD 2
interface, analog signal conditioning for 3-phase voltage/current inputs and current
using the MCP3909 Energy Meter IC, neutral current measurement using external
op-amp on-board dsPIC33F ADC, UART interface to PC, ICD2 interface for MCU
programming, and power supply circuits. Note there are two PCBs comprising this
energy meter, the power supply PCB, and the MCU/AFE PCB. Refer to Appendix
A. “Schematics and Layouts” for more information.
Three-phase voltage and current signals are connected to the meter through
transformers, and connected to the MCP3909 A/D converter through a simple signal
conditioning circuit. The MCP3909 device samples the signal and performs the
analog-to-digital conversion (ADC). The MCP3909 device sends the digital conversion
results to the dsPIC device via the SPI interface.
The MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design uses a quasi-synchronous sampling algorithm via the dsPIC33F, therefore eliminating the need for
voltage zero-crossing detect and clock generating circuit, which are otherwise needed
in a synchronous sampling algorithm. The clock of the MCP3909 device is an active
external 3.2768 MHz crystal.
.
Phase A
CT
Phase C
CT
FIGURE 2-1:
© 2009 Microchip Technology Inc.
MCP3909
Phase C
PT
3.2768
MHz
ADC
UART
I/O
dsPIC33FJ64GP206
SPI
SPI
+5V
Power Supply
RS232
ICD2
Interface
Phase B
CT
CLK
Gain
Control
MCP3909
Phase B
PT
MCP3909
Phase A
PT
Op Amp
UART
Interface
Neutral
line CT
+3.3V
Power Supply
Hardware Block Diagram.
DS51723A-page 17
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
2.2
ANALOG FRONT END CIRCUITRY
For safety, current transformers are used between the voltage and current input signals
and the measurement module to isolate it from the 3-phase power supply.
The transformer used for the voltage path is a 1:1 transformer, SPT204B from Beijing
Singure Measurement & Control Technology Co., with non-linearity less than 0.1%, and
rated input/output current of 2 mA/2 mA.
The transformer used on the current side is a SPT254FK, from Yhehua Shanghai, with
rated input /output current of 20A/2 mA, non-linearity less than 0.1%, and linear range
of 0-20 A. See Appendix B. “Bill Of Materials (BOM)” for more information on the
input circuitry.
Using phase A as an example for the voltage signal path, a 150 kΩ (R1) resistor is used
before the CT to transform the signal to an appropriate current. After the CT, burden
resistors R125 and R126 are needed to transform the current signal to a differential voltage signal for the MCP3909 device to sample. Signals are coupled into the MCP3909
device's signal input port via R110 and R111. C111 and C112 are used to filter high-frequency signals.
Using phase A as an example for the current signal path, transformation of the current
signal is similar to that of the voltage signal. The burden resistors R125/R126 and
R116/R117 are chosen to be 20Ω for the current channel and 100Ω for the voltage
channel after considering the following 3 factors:
• The MCP3909 device's differential voltage input range: 1V for voltage channel
and 0.705V for current channel
• Maximum current/voltage for the meter: rated current of 5A, maximum current of
20A, and maximum voltage of 300V)
• Transformer ratio for the current and voltage transformer.
A non-isolated voltage input circuit is included. In practice, a voltage divider network of
resistors is often used for sampling AC voltage input. This measuring method is
therefore included in the hardware design. In Figure 2-2, voltage divider resistors R3,
R4, and sampling resistor R1 construct a network for sampling AC voltage.
CTA-1
1.0 kΩ
SCT220B
CH0+
R125
CURRENT
R126
R110
20 Ω
20 Ω
1 nF
C111
1.0 kΩ
CTA-2
CH0T103
150 kΩ
PA
R111
1.0 kΩ
SPT204B
R1
CH1+
R116
VOLTAGE
1 nF
C112
R117
R108
100 Ω
100 Ω
1 nF
C109
1.0 kΩ
N
CH1+
T1
R109
Jumper
499 kΩ
499 kΩ
R4
R3
1 nF
C110
MCP3909
PA
VOLTAGE (Non-Isolated Option)
R1
1 kΩ
C3
33 nF
N
FIGURE 2-2:
DS51723A-page 18
Input Signal Conditioning Circuit (Phase A).
© 2009 Microchip Technology Inc.
Hardware Description
2.2.1
Burden Resistor Temperature Coefficient
The high precision class 0.2S requirement for the energy meter makes it crucial to
select proper burden resistors for the output of the current transformers.
Metal film resistors with low inherent noise and temperature coefficient are ideal. Given
that the secondary current of the CT is I, then the input voltage of the MCP3909 device
is U = IR, where R is the resistance of R125 and R126 (Using Phase A as an example).
If the temperature varies by ΔT, and the temperature coefficient of sampling resistor R
is β ppm/°C, then the output voltage is:
EQUATION 2-1:
U' = I ( R + Δ T × β × R )
The voltage variation is:
EQUATION 2-2:
ΔU = U' – U = I × ΔT × β × R = U × ΔT × β
This relationship shows that the output voltage variation caused by temperature
variation is in proportion to the temperature coefficient of the burden resistor.
In addition, a smaller temperature coefficient benefits meter start stabilization after
startup. It takes a longer time for resistors with larger temperature coefficient to
stabilize. Therefore, accurate measurements would require a longer wait after
power-up. This affects the efficiency, or speed of meter calibration.
© 2009 Microchip Technology Inc.
DS51723A-page 19
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
2.3
ANALOG-TO-DIGITAL CONVERSION
This meter design uses Microchip's MCP3909 energy meter ICs. Small-signal current
inputs can be amplified by the programmable gain amplifier inside the MCP3909
device. The programmable range for the MCP3909's PGA is 16:1 V/V. The MCP3909
PGA gain can be configured by G0 and G1 (Pin 15 and 16 of the dsPIC33F).
The MCP3909 device's clock is provided by a 3.3768 MHz active crystal (for 50 Hz
system, see Appendix D. “50/60 Hz Meter Operation” for 60 Hz line frequency
information). The MCP3909 device's output data rate is 12.8 ksps. The clock lines and
MCLR lines of all 3 MCP3909 devices are connected together, which ensures that the
3 phases are strictly synchronous.
3.3768 MHz (50 Hz Version)
X100
MCLK input
MCP3909
SDO
SDO
MCP3909
SDO
MCP3909
To dsPIC33F
IRQ IC1
(Input Capture)
DR Pulse
fSAMPLE1 = 12.8 ksps (Active Power)
fSAMPLE2 = 6.4 ksps (Reactive Power, RMS Current / Voltage)
fSAMPLE3 = 3.2 ksps (Harmonic Analysis, Distortion)
tLINE_CYC
Phase A,B,C I & V Data
SDO DR
16 bits
x 6 ADCs
DR
tSAMPLE
FIGURE 2-3:
DS51723A-page 20
Clock Generation, Sampling Times and Calculation Frequencies.
© 2009 Microchip Technology Inc.
Hardware Description
2.3.1
Samples And Processing
Input capture IC1 on the dsPIC33F is used to detect if A/D conversion is complete.
However, not all MCP3909 device samples are stored in the MCU, depending on the
parameter being calculated. The ADC conversion rate of the MCP3909 device is
determined by the frequency of the master clock (3.378 MHz for the case of a 50 Hz
line), and the output data rate is MCLK/256 or 12.8 ksps. After each conversion is
complete, a Data Ready signal is generated by the SDO of the MCP3909 device. The
signal is fed into IC1, allowing the Interrupt Service Routine (ISR) of IC1 to read the
data. When the MCP3909 device outputs data, it first sends an ADC result of the
voltage channel, then an ADC result of the current channel, with MSB first.
As noted, not all MCP3909 device samples are used for calculating all the parameters.
In practice, 6.4 ksps sampling rate is required, which means only 1 output data is used
for every 2 data sampled. For 50 Hz input signal, 6.4 ksps sampling rate will take 128
samples for each cycle. For example, the active power metering is computed based on
this condition.
But for other parameters for which precision is not critical, such as reactive energy,
voltage, current and frequency, the sampling rate may be reduced to save data storage
space and processing time. In this design, the 3.2 ksps sampling rate is used, which
means only 1 result is stored for every 4 ADC conversions.
After each conversion, a positive pulse with the width of 4 clock cycles is output by the
SDO pin of the MCP3909 device. IC1 is used to detect the falling edge of the pulse and
generate an interrupt for every 2 falling edges, i.e., 1 data is read for every 2 conversions, thus realizing 6.4 ksps sampling rate.
© 2009 Microchip Technology Inc.
DS51723A-page 21
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
2.4
DSPIC33F HARDWARE CONNECTION AND PERIPHERAL USAGE
Table 2-1 is the pin allocation for the dsPIC33FJ64GP206 MCU.
TABLE 2-1:
dsPIC
Pin
FUNCTIONAL ALLOCATION FOR dsPIC33F PINS
Pin Function
Name
Corresponding Name
in Schematic
Diagram
Functional Description
62, 1
RG14, RG15
G0A, G1A
MCP3909's gain control for Phase A
2, 3
RC1, RC2
G0B, G1B
MCP3909's gain control for Phase B
13, 12 RB3, RB4
G0C, G1C
MCP3909's gain control for Phase C
14
RB2
CSA
MCP3909's chip-select signal corresponding to Phase A
15
RB1
CSB
MCP3909's chip-select signal corresponding to Phase B
16
RB0
CSC
MCP3909's chip-select signal corresponding to Phase C
4, 5, 6 SCK, SDI, SDO
SPI I/F
Interface signal of SPI2. SPI interface operates under master
mode, used for the MCP3909 device communications
42
INT1
SDO
SDO line of SPI interface, for detecting MCP3909's A/D
conversion complete flag
49-51
RD1-RD3
PULSE1, PULSE2,
PULSE3
PULSE1 is total phase active power pulse output, PULSE2 is
total phase reactive power pulse output. PULSE3's function is
to be determined.
61
RG0
AD_MCLR
Master clear signal of 3 MCP3909 devices (tied together)
53
RC13, RC14
LED1, LED2
LED drive pins, can be used as energy pulse output indicator,
for meter calibration. Its function is similar to those of
PULSE1 and PULSE2
36, 37 SDA1 / SCL1
SDA/SCL
I2C™ interface, used to read/write EEPROM externally
33, 34 U1TX / U1RX
RF3/RF2
UART interface
33, 34, SDO1/SDI1/SCK1
35
RF3/RF2/RF6
SPI1 interface, can be designed by customers, used for
communication with host MCU to obtain measurement results
and calibrate a meter. Its function is the same as UART
interface, but have a faster communication rate and higher
efficiency. SPI operates in slave mode. If UART interface is
used to communicate with host MCU, then this interface
cannot be used.
27
AN12
Current_N
Detect neutral current
28
AN13
Ref_V
Detect boost voltage of neutral current
17, 18 ICSPCLK, ICSPDAT
ICSP I/F
Online debugging/programming interface
7
MCLR
Master clear input
MCLR
DS51723A-page 22
© 2009 Microchip Technology Inc.
Hardware Description
2.4.1
UART and SPI1 Interface
The UART and SPI1 interfaces are multiplexed. Through the UART or SPI1 interface,
the host MCU can communicate with the metering front-end to perform calibration or
obtain metering results. The SPI interface may also be used if high-speed data transfer
is desired. In this case, the SPI interface of the dsPIC device works in the slave mode.
The UART and SPI1 share a common pin, so only one of the two interfaces can be
used at a time. Since the reference design uses a PC to simulate the host MCU, the
UART interface is chosen as the communication interface. SPI and RS232 interfaces
are not isolated from the PC. A general-purpose transceiver device, MAX232, is used
for the UART interface.
2.4.2
Energy Pulse Output Interface
Three sets of outputs for energy measurement pulses are available in this design,
corresponding to the I/O pins of RD1-RD3. Two of them, output total active energy and
total reactive energy, respectively, and the other is not yet specified. Outputs are
isolated by a photo-electronic coupling device, U3. The photo-electronic coupler is
active when corresponding I/O pin is high.
In addition, the design also provides two sets of LED outputs for energy meter calibration. The output pins for these LEDs are RC13 and RC14. The LED is on when the
output is low. Figure 2-4 is the circuit of energy pulse output interface.
1 kΩ
RD3
R301
Total Active
1 kΩ
RD2
R301
Total Reactive
1 kΩ
RD1
R301
(not used)
3.3V
470Ω
R314
D303
Total Active
RC13
3.3V
470Ω
R310
Total Reactive
D302
RC14
FIGURE 2-4:
© 2009 Microchip Technology Inc.
Energy Output Pulse Configuration.
DS51723A-page 23
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
2.4.3
Neutral Current Detection
Detection of neutral wire tampering is performed by the on-chip A/D of the dsPIC
device. The purpose of the detection is to prevent electricity theft, balance 3-phase
signals and detect electricity leakage. Since the precision is not critical, a dsPIC
on-chip A/D is sufficient.
Figure 2-5 shows the circuit for neutral current detection. The neutral input uses R128
for sampling. To bias the AC signals to the A/D measurement range, a 3.3V power
supply is divided by R130 and R129 and connected to the emitter-follower of the
MCP6002 device to output a boost voltage REV_V of 1.65V. The biased voltage is then
connected to the CT in series. The CT's sampling voltage is connected to Op-amp B of
the MCP6002 device via R127 as emitter-follower output, generating the sampling
voltage, Current_N, for the current. Both Current_N and 1.65V VREF signals are
sampled and measured by the dsPIC on-chip A/D.
.
Neutral Wire Input
T100 Secondary
3.3V
R130
4.7 kΩ
U101_A
+
470Ω
4.7 kΩ
R128
R127
U101_B
+
Current_N
R129
4.7 kΩ
1.65V VREF
FIGURE 2-5:
DS51723A-page 24
Circuit of Neutral Line Detection.
© 2009 Microchip Technology Inc.
Hardware Description
2.5
POWER SUPPLY
The power supply used in this design provides 3.3V and 5V. Since the energy meter for
a 3-phase 4-wire system is required to operate properly when any one phase is active,
a switching power supply module is used for convenience. T4 is the switching power
supply module.
Prior to the input of this module, additional protection circuitry is included with the meter
design. Figure 2-6 shows the input to the switching power supply module and the
additional filtering and protection circuitry. In Figure 2-6, R5 is the integrated ferrite
bead, C1, C4, C6, RV1, RV2 and RV3 are CBB capacitors and varistors. They are used
to improve anti-surge performance of the system.
.
1
R5
JP4
1
JP5
1
1
2
3
4
A
B
C
N
A
B
C
N
8
7
6
5
PA
PB
PC
RV1
JP6
C1
0.1U
COILS
1
RV2
C4
0.1u
RV3
C6
0.1u
N
N
12V
T4
N
PC
PB
PA
FIGURE 2-6:
1
2
3
4
N
Vo2
C
G2
B
Vo1
A
G1
8
7
6
+ C2
100uf
JP6
1
2
3
Header 3
5
Switching Power Supply Module (T4) and Additional Input Protection Circuitry.
© 2009 Microchip Technology Inc.
DS51723A-page 25
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
An LDO is connected in series at the output of the switching power supply to obtain a
more stable power supply. Figure 2-7 shows this circuit. Microchip's MCP1701 device
and MCP1700 device, low drop-out high efficiency LDOs, are selected for use.
D301 is the power LED for the meter and is active when the meter is connected to the
proper line input voltage.
J1
MCP1701(5V-SOT89)
L302
2
1
2
5V_IN
INDUCTOR
L301
INDUCTOR
R316
470
C322
CAP
D301
LED
3
DS51723A-page 26
Vout
C336
0.1uF
+ C338
100uf
U306
MCP1700(3.3V-SOT23)
2
Vin
Vout
Gnd
1
C323
CAP
FIGURE 2-7:
Vin
VSS
1
+ C337
100uf
5V
3
3.3V
C324
CAP
+ C339
47uF
5V and 3.3V LDO Modules.
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 3. Firmware
3.1
OVERVIEW
This section discusses the dsPIC firmware structure, peripheral resources, important
program flows, and explanations of project files included in the firmware zip files
included with the system. See Appendix D. “50/60 Hz Meter Operation” for
converting to 60 Hz code.
•
•
•
•
•
3.2
Calculate all electrical parameters in frequency domain
MCP3909 device communication
Detect voltage/current phase order, and determine missing phases
Generation imp/kWh power pulse
UART communication
MAIN LOOP
The main loop of the entire dsPIC33F program is shown in Figure 3-1.
Main Program
Initialize on-chip
peripherals and variables
and MCP3909 device
Process UART
comm. commands
Calculation()
Sampled 3 cycles?
Yes
Compute elec.
parameters
Yes
Compute
neutral line
current
No
Neutral data
acquisition complete?
No
Clear WDT
FIGURE 3-1:
© 2009 Microchip Technology Inc.
Main Loop Chart.
DS51723A-page 27
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
After system power-up, the MCU enters the initialization process, which includes
proper configuration of the I/O ports and on-chip peripherals (such as timer, UART, SPI
and IC). At the same time, the system control parameters can be loaded from the
external EEPROM and variables are all initialized.
Since most tasks of this system are accomplished through interrupts, only three tasks
are carried out in the system main loop, which are interpreting/processing UART
communication protocol, calculating parameters, and detecting neutral current. The
UART communication is performed in function UART_process(), the executing
frequency of which depends on the polling frequency of the upper computer.
Parameters are calculated by the function Calculation(), which is executed once every
3 cycles of the power grid. Neutral current is detected by function ComputeNeutralCurrent(), and computing is performed once every 16 cycles of the power grid.
DS51723A-page 28
© 2009 Microchip Technology Inc.
Firmware
3.3
CALCULATION() - CALCULATING ELECTRICAL PARAMETERS
All power parameters are calculated with the function Calculation(), which is executed
once every 3 cycles of the power grid. As shown in Figure 3-2, all calculations are
performed post DFT (direct fourier transform), in the frequency domain.
•
•
•
•
•
•
•
•
•
•
•
RMS Voltage/current Of Each phase
Phase Angle
Measuring Line Frequency
Active, Reactive, Apparent Power Of Each Phase
Positive and Negative Active Power
Positive/negative Reactive Power
4-quadrant Reactive Energy
Total Active Power, Total Reactive Power, and Total Apparent Power
Total Power Factor
Voltage and Current Distortion Of Each Phase
Voltage and Current Harmonic Contents Of Each Phase
Note:
Algorithms for all calculations are shown in Appendix C. “Power
Calculation Theory”.
Calculate function
Select sync. window
function and
sine/cosine table
phase sequence 1~3
loop
Voltage signal and
sync. window process
DTF transformation
Calculate voltage
RMS value
Calculate current
RMS value
Calculate current
harmonic
Active/reactive
power calculation
and compensation
phase sequence 1~3
end
Calculate combined
power
Combined energy
accumulation
Calculate voltage
harmonic
Current signal and
sync. window process
DTF transformation
Calculate frequency,
determine phase
sequence
Update loop array
pointer and data
length
End of function
calculation
FIGURE 3-2:
© 2009 Microchip Technology Inc.
Calculation() Flow Chart.
DS51723A-page 29
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
Each block can be categorized into one of three types of calculations:
1. Calculation for an individual phase
2. Calculation for all 3 phases
3. Calculation of power accumulation (Energy)
3.3.1
Process Quasi-Synchronization Window
The first blocks of the calculation flow are to determine how many samples to use for
the quasi-synchronization sampling algorithm. See Appendix C. “Power Calculation
Theory” for more information on this approach.
The firmware selects the proper quasi-synchronization window function (array of data)
and corresponding sine/cosine table according to the number of sampling points in
present current cycle.
The number of sampling points is obtained from the last calculation of frequency
function. As the line frequency fluctuates in slow motion and typically varies by a small
amount over three (3) line cycles, the period of line frequency measurements of the
previous cycle can be used to determine data length of sampling.
The quasi-synchronization window function is an array established in advance, and its
length is the same as that of the sampling data, which is obtained by the weight
coefficient multiplied by 32768. In this design, the method of quadrature by
complexification echelon is used, corresponding weight coefficient is calculated with
three (3) iterations. Three iterations implies that the length of the input original data is
equal to the number of sampling points in three cycles. The number of sampling points
in each cycle is usually different from the input signal cycle, but it will be close to an
integral multiple of the input signal cycle. For example, at a sampling rate of 3.2 ksps,
50 Hz input signal corresponds to 64 sampling points for each cycle, and 50.1 Hz input
signal would also be close to 64 sampling points for each cycle. Again, 51 Hz input
signal would be close to 63 sampling points for each cycle. Therefore, at different input
frequencies, the corresponding numbers of sampling points in each cycle are different.
Consequently, the corresponding quasi-sync window function and sine/cosine table
need to be established according to different numbers of sampling points. The
sine/cosine table is established by evenly dividing a cycle into a number of segments
equal to the number of sampling points, calculating corresponding sine/cosine values
and then multiplying the values by 32768. The purpose of the multiplication is to
change the original operation of floating point numbers into that of fixed-point numbers.
Adjustments will be performed in the final stage of calculation.
The processing of the original signal being processed by the quasi-synchronization
window function is actually a process of array multiplication, i.e. the original input signal
is multiplied with a corresponding array of window functions. It's accomplished by the
function qusi_syn_wnd(), which is written in assembly to take full advantages of DSP
features.
3.3.2
DFT Transformation
A Direct Fourier Transform (DFT) is performed on the collected sets of data. Processing of the original signal by quasi-synchronization window can effectively reduce
spectrum leakage caused by non-entire-cycle sampling during DFT transformation.
The data length is not a power of 2, therefore, the FFT algorithm cannot be used in DFT
transformation. DFT transform is accomplished by function DFT(), which is written in
assembly to take full advantages of the DSP feature of accumulated multiplication.
Since an FFT algorithm cannot be used, and it takes longer to perform a DFT
calculation, this is the most time-consuming process in the entire system.
DS51723A-page 30
© 2009 Microchip Technology Inc.
Firmware
3.3.3
Calculating RMS Voltage/Current
After the data set of either a voltage or current signal (of each phase), has been DFT
transformed, the voltage or current magnitudes of the different harmonics can then be
calculated. The total effective voltage or current (RMS) can be obtained by further
calculaton, by simply combining the results of the individual harmonics (including the
fundamental or the 1ST harmonic).
Computing the magnitude of the voltage or current is accomplished by function
ComputeMagnitude(). The result, called amplitude, is a long integer, and is the
squared magnitude of voltage or current. To speed up the computation, fixed-point
operation is used. The ComputeMagnitude() routine is written in assembly language.
After the magnitude is computed, there is an adjustment process which is based on a
floating-point operation. The limited number of computations will not affect the
operation speed, and will instead greatly improve precision.
Parameter ratio1 in the firmware is a coefficient related to the number of sampling
points (see Equation 2-2). Division is accomplished by a simple shift operation in
firmware. If the sampling cycle is not a power of 2, it cannot be accomplished by
shifting. However, division by shifting can be accomplished by multiplying a compensating coefficient Coeff.data.linear.V_channel[]. This is the calibration coefficient for
ratio errors.
Since the current signal has a wide dynamic range, for small signals, the ADC output
data range is small and is limited by DSP's bit resolution (16-bit MCU). If division by
shift is used in the same way that is used for large signals in computation, precision
may be affected. Therefore, for computing magnitudes of small signals, ComputeSmallMagnitude() function is used instead. This function is similar to ComputeMagnitude(), the only difference being that the shift length is shortened in division operation,
and will be compensated during data adjustment. The computation precision will not be
affected as the data adjustment process uses float-point operation.
3.3.4
Calculating Harmonics
Computing harmonics is accomplished in assembly language, by the function ComputeHarmonic(). The computation is based on Equation C-62, in Appendix
C. “Power Calculation Theory”. The result is the ratio of the magnitude of K-th
harmonic to fundamental magnitude, and is given in a percentage.
Note:
3.3.5
Since the output of ComputeHarmonic() is the squared harmonic
magnitude, extraction of square root is needed in computation. The
calculated harmonic content is stored as a fixed-point number, and the
actual value stored is the harmonic content multiplied by 10.
Calculating Power
Computing power is accomplished in assembly language by the function ComputePower() based on Equation C-39 and Equation C-40 in Appendix C. “Power Calculation Theory”.
After the ComputePower() function is complete, an adjusting process for computed
power is required. First, the computed result is adjusted according to the gain of current
amplifier. Then the calibrating coefficient ratio2 is determined according to present
number of sampling points.
Additional compensation to the power calculation is required, for phase compensation.
This compensation is based on the present load current. The difference between signal
frequency and the central frequency is also taken into consideration. Consequently
computed power is compensated.
© 2009 Microchip Technology Inc.
DS51723A-page 31
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
3.3.6
Active Energy Accumulation
Energy accumulation is done by calculating the total energy, which is the algebraic sum
of energy of each phase. Active energy is obtained by accumulating the multiplication
of voltage and current of each sample, which ensures the high accuracy of measurement.
3.3.7
Reactive Energy Accumulation
The required measurement accuracy of reactive energy low, so in this design, it is
obtained by accumulating the product of the present measured reactive power and the
time interval between two measurements.
3.3.8
Output Pulse Generation
Refer to Section 2.4.2 “Energy Pulse Output Interface” for pulse output. To ensure
the uniformity of output pulses, the calculation is divided in the measurement cycle into
in a number of equal sections, and accumulate them. For simplification and lowering
computation complexity, a counter is used to substitute the process of accumulation.
The counter is only enabled when accumulated energy approaches to the threshold of
the pulse output.
3.3.9
Line Frequency Calculation
Frequency calculation is based on Equation C-52 and Equation C-53, in Appendix
C. “Power Calculation Theory”. The dsPIC33F collects 3-line cycles worth of data.
The first two cycles of data of all sampled data is analyzed, and then the frequency of
two successive cycles is used.
The data of two successive cycles are transformed via DFT for the fundamental, which
is accomplished by assemble function DFT_Fundamental(). This is followed by the
computation of the initial phase angle of the first two line cycles. Then the phase lag
and frequency offset of the two line cycles of signal can be calculated.
When measuring frequency, only the first two cycles of data are used. It must be
assumed the input frequency is 50 Hz and the chosen appropriate sine/cosine table to
carry out DFT transform for fundamentals of the 1st and 2nd cycles of data. See
Appendix D. “50/60 Hz Meter Operation” for 60 Hz firmware.
Frequency offset is calculated by determining the initial phase angle for each line cycle.
The greater the frequency offset, the greater the measurement error.
Since one of the 3 phases may be missing, if the voltage magnitude for phase A is less
than the threshold, it is necessary to switch to phase B. Consequently, if sufficient voltage magnitude of phase B is not detected, it is necessary to switch to phase C.
The basic algorithm for measuring line frequency is based on the method described in
Appendix A, Section C.4 “Measuring The Voltage/current Rms Value And Power
Using Quasi-synchronous Sampling Algorithm”.
Frequency will be measured once for every 3 times the data is sampled.
DS51723A-page 32
© 2009 Microchip Technology Inc.
Firmware
3.4
ADC SAMPLING SCHEME FOR CALCULATIONS
The ADC conversion rate of the MCP3909 device is determined by the frequency of
master clock, MCLK, and the rate will be MCLK/256. After each conversion is complete,
a DataReady signal (4-CLK length) is generated by the SDO of the MCP3909 device.
The signal is fed into IC1 (Input Capture 1 on the dsPIC33F), allowing the Interrupt
Service Routine (ISR) of IC1 to invoke data-read function of the MCP3909 device.
When the MCP3909 device outputs data, it first sends the ADC result of the voltage
channel, then that of the current channel, with the MSB first.
The frequency of the master clock, MCLK, of the MCP3909 device is 3.2768 MHz, and
ADC outputs @12.8 ksps. In practice 6.4 ksps sampling rate is used in the program,
which means only 1 output data is used for every 2 data sampled. For a 50 Hz input
signal, a 6.4 ksps sampling rate will take 128 samples for each cycle. The active power
calculation is computed based on this condition.
The other parameters for which precision is not critical, such as reactive energy,
voltage, current and frequency, the sampling rate may be reduced to save data storage
space and processing time. In this design, the 3.2 ksps sampling rate is used, which
means only 1 result is stored for every 4 ADC conversions.
In the program, sampling and calculation are carried out concurrently, and data is
stored in the cyclic array in the dsPIC33F RAM. A calculation may be performed after
either 1 cycle, 2 cycles or 3 cycles of data are sampled, which can be configured in the
program. The user should note that frequent calculations will increase the measurement precision at the price of system overhead and response speed, therefore making
proper tradeoffs based on practical requirement. In this design, 3 cycles of signals are
sampled before an AC electrical parameter calculation is performed. Refer to
Figure 3-3.
Sampling
Cycle n
Sampling
Cycle n+1
Sampling
Cycle n+2
Idle
Idle
Idle
FIGURE 3-3:
3.4.1
Sampling
Cycle n+3
Calculate
n,n+1,n+2
Sampling
Cycle n+4
Idle
AC Signal Sampling alnd Computing.
Processing IC1 Interrupt
Input capture IC1 is used to detect if the A/D conversion is complete. After each
conversion, a positive pulse the width of 4 clock cycles is outputted by the SDO pin of
the MCP3909 device. IC1 is used to detect the falling edge of the pulse and generate
an interrupt for every 2 falling edges, i.e., 1 data is read for every 2 conversions, thus
realizing 6.4 ksps sampling rate.
In addition to reading the data of the MCP3909 device, the IC1 interrupt service routine
(ISR) also controls the energy pulse output generation. Energy pulse processing
consists of active/reactive energy pulse processing. For the pulses to be outputted
more uniformly, the clock resolution used to generate the pulses must be as high as
possible. The interval of the IC1 interrupt is 156.25 µs, therefore, the resolution
generated by the pulse can be up to 156.25 µs.
© 2009 Microchip Technology Inc.
DS51723A-page 33
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
The energy pulse processing program only begins when the level is close to outputting
pulse level. To simplify the process and shorten the ISR execution time, a counter is
used in place of the energy accumulation function for each pulse and to determine if a
pulse will be outputted. When the count is greater than the threshold of pulse output,
an energy pulse will be outputted, and the appropriate amount of energy will be subtracted from the energy accumulating register. Output toggling will then be processed.
Once the width of the output pulse exceeds 80 ms, the pulse output will be turned off.
The program flow chart is shown in Figure 3-4.
ICI Interrupt Service
routine
Call MCP3909 data
read program
Processing
falling edge of
active energy?
Yes
Update pulse width
counter, if pulse
width > 80 ms, toggle
pulse output level.
End pulse output
process.
No
Processing
rising edge of
active energy?
Yes
Update pulse width
counter, if counter >
flip threshold, output
pulse and update
energy accumulation
register.
No
Processing
falling edge of
reactive energy?
Yes
Update pulse width
counter, if pulse
width > 80 ms, toggle
pulse output level.
End pulse output
process.
No
Processing
rising edge of
active energy?
Yes
Update pulse width
counter, if counter >
flip threshold, output
pulse and update
energy accumulation
register.
No
Return
FIGURE 3-4:
DS51723A-page 34
IC1 Interrupt Service Routine.
© 2009 Microchip Technology Inc.
Firmware
3.5
READING A/D DATA OF THE MCP3909 DEVICE
All three MCP3909 devices use the same clock source and reset signal, so all 6 A/D
channels of the 3 MCP3909 devices are synchronous. Only a single Data Ready (SDO)
signal of any of the MCP3909 device is required to read A/D data of the 3 phases in
turn. This module is invoked by IC1 interrupt triggered by the "data ready" signal on the
SDO of the MCP3909 device. IC1 is set to generate an interrupt for every two falling
edges. Therefore, only one of the two sampling data of the MCP3909 device is
read.The flow of reading the MCP3909 device's data is as follows:
• Retrieve all values of 3-phases, both current channel and voltage channel data.
Bits 0-15 of each phase data are voltage channel data, bits 16-31 are current
channel data
• Accumulate the active power of each phase. On every other interrupt, the current
and voltage values are stored into RAM in the cyclic sampling array
• Update the pointer of sampling array and length of sampling data. If the length of
sampling data is 3-line cycles long, set the sampling complete flag, and then the
calculation function Calculate() will be called by the main flow to start computing
all corresponding parameters.
Read MCP3909 data
Select phase A of
The MCP3909 device,
clear SPI flag
Read phase A data,
No accumulate active
energy of phase A
and save data to array
Even count
data read?
Yes
Read phase A data
and accumulate active
energy of phase A
Read phase B data
and accumulate active
energy of phase B
Read phase C data
and accumulate active
energy of phase C
Read phase A data,
accumulate active
energy of phase A
and save data to array
Read phase A data,
accumulate active
energy of phase A
and save data to array
Update array pointer,
sample pass count
flag and data length
End of sampling of
this cycle?
y
No
Set data sampling
complete flag
End
FIGURE 3-5:
© 2009 Microchip Technology Inc.
Flow Chart of Read A/D Data.
DS51723A-page 35
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
3.5.1
Initialize and Configure MCP3909 Operation Mode
The task of this module is to enable the MCP3909 device to enter the "Channel Output"
mode. This design uses the "Channel Output" mode of the MCP3909 device. In this mode,
the current and voltage data channels measured by the ADC is sent through the
MCP3909 device's SPI port. To enable the MCP3909 device to enter the Channel
Output mode, certain instructions must be sent to the device via the SPI interface within
a specified time (32CLK) after resetting the MCP3909 device.
• Enable all MCP3909 devices: enable ADCS1, ADCS2 and ADCS3, and configure
MCU's SPI to 8-bit mode
• Reset the MCP3909 device through the RESET pin. The pin must be pulled low
for no less than 1 clock cycle of the MCP3909 device
• After the RESET pin is pulled high, wait for 4 clock cycles for the MCP3909 device
pin functions to reset
• Send Instruction 0x94 to the MCP3909 device through the SPI Interface
• Configure the SPI interface to 16-bit mode and strobe the MCP3909 device for
Phase A
Initialize MCP3909
Strobe all
MCP3909 devices
Reset the
MCP3909 devices
Wait for 4 CLK cycles
Send instruction 0xac
Set SPI to 16-bit mode
Strobe phase A of
the MCP3909 device
End
FIGURE 3-6:
DS51723A-page 36
Initializing the MCP3909 Device Flow Chart.
© 2009 Microchip Technology Inc.
Firmware
3.6
COMMUNICATION OF UART INTERFACE
The UART interface is used to communicate with the upper computer (MCU or PC). Via
the UART interface, the upper computer reads the measured parameters of the power
grid, and may also send system parameters and calibration parameters to the target
board as well.
The communication interface is a bidirectional interface based on UART, using
master/slave half-duplex mode. The baud rate is 19,200 bps, with 1 start bit, 8 data bits
and 1 stop bit. Communication is done by frames with non-fixed-length frame structure,
definition of which is shown in Table 3-1. The Communication protocol is specified in a
master-slave structure. The system in this design is the slave, and the upper computer
is the master. The master sends commands to the slave, and slave responds to the
master.
Each command is defined in Chapter 6. “Meter Communications Protocol”.
TABLE 3-1:
FRAME STRUCTURE OF COMMUNICATION PROTOCOL
Sync Field
Command Type
Data Length
Data Field
Checkout Byte
End Byte
1 byte
1 byte
N bytes
1 byte
1 byte
2 bytes
3.7
RESOURCE CONFIGURATION
Details of the MCU resources used in this design and their configurations are listed in
Table 3-2.
TABLE 3-2:
CONFIGURATION OF MCU RESOURCES
Resource Name
Interrupt
Priority
System Clock
Timer
Interrupt
Functional Description
Fcy = 29.4912M, provided by an external 7.3728 Mz timer through an internal PLL frequency doubler.
Timer2
1
System clock, used for timing. Its cycle is 10 ms. The interrupt flag may be
set in the IRS. Used to extend the indication of timer. Also used to deal with
UART reception overtime.
Timer3
none
Used to detect ADC's sampling synchronization of neutral current. After the
frequency of the power grid is measured, the period of TMR3 is adjusted
accordingly. 16 points are sampled by ADC for each cycle of power grid.
TMR2
1
ditto
IC1
5
Driven by a 3.2768 MHz clock. The MCP3909 device can generate
12.8 ksps of data output.
Sampling input capture. An interrupt is generated for every two MCP3909
device samplings. 6.4 ksps sampling rate is realized.
In fact, active power is cumulated at 6.4 ksps sampling rate (128 sampling
points each cycle at 50 Hz), but other parameters are cumulated at 3.2 ksps
sampling rate
UART RX
2
Receive data of UART communication
UART TX
2
Transmit data of UART communication
ADC
2
Detect current of neutral line
SPI2
none
Used in communicating with the MCP390X device - set the MCP390X
device's modes and read A/D results
SPI1
none
Unused, but the interface is reserved and may be used to communicate
with upper computer in substitution of the UART interface
© 2009 Microchip Technology Inc.
DS51723A-page 37
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
3.8
DESCRIPTION OF PROJECT FILES
TABLE 3-3:
FILE DESCRIPTION
File Name
Description
main.h
main.c
Main program
global.h
global.c
Mainly define important system macros, key data structures, and declare global variables.
MCP390x.h
Declare macros, constants, local global variables, some of the global variables and functions
used in the MCP390X device.
MCP390x.c
Functions involved with the MCP390X device, including set SPI, initialize the MCP390X device
and read data.
calcu.h
Declare macros, constants, local global variables, some of the global variables and functions
used in calcu.c.
calcu.c
Main module to calculate parameters, including calculate frequencies, current/voltage RMS,
power, power factors and energy, and analyze harmonics.
uart_comm.c
Declare macros, constants, local global variables, some of the global variables and functions
used in uart.c.
uart.c
Receive, transmit, process protocol and so forth for UART communication.
Calibrate.c
Program for Ratio error calibration, power calibration and phase lag calibration, it stores and
initializes calibration data.
Calibrate.h
Declare constants, local variables and global variables used in calibration.
Adc.c
Adc.h
On-chip ADC operation, detecting the current of neutral wire.
I2Csubs.h
I2Csubs.c
Control EEPROM of off-chip I2C interface.
interrupt.h
Declare macros, constants, local global variables, some of the global variables and functions
used in interrupt.c.
interrupt.c
Set interrupts and ISRs.
Asmcode.c
Some assemble functions used in calculation.
DS51723A-page 38
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 4. Meter Calibration
4.1
INTRODUCTION
Meter calibration consists of using standard electrical power equipment that supplies
the power to the meter and calculates the error and correction factor at each calibration
point. This equipment must be accurate in order to calibrate the energy meter. The
supplied PC software is then used to send calibration commands and correction factors
down to the dsPIC33F, completing meter calibration.
Why Is Calibration Necessary?
An energy meter usually consists of errors due to transformers, VREF tolerance, ADC
gain errors, and other passive component errors. Energy meters are factory calibrated
before shipping to eliminate the impact from such elements and reduce the error. The
non-linearity and inconsistency of signals in the path of sampling circuit and A/D
conversion circuit cannot be ignored in high-accuracy measurement. The impact needs
to be corrected to improve measurement accuracy.
The calibration described in this chapter are calibrated with the help of the PC software
PM_Viewer, described in detail in the next chapter. To summarize the process, the
measurement error is fed into the software, and the data is then sent to the meter to via
the UART. The details of this procedure are detailed in the next chapter, "PC Software".
4.2
CURRENT/VOLTAGE CALIBRATION
Current and voltage calibration is a ratio error calibration from the upper computer by
sending commands and data for correction to the MCU. The dsPIC33F will call a
firmware module after receiving the command from the host PC. The flow is as follows:
1. Determine the phase to be calibrated and the magnitude of current and voltage
being applied to the meter, and read measurement (RMS) values of that channel.
2. Calculate the calibration coefficient of the ratio error by the ratio of standard value
received to the measured value.
3. Multiply the original coefficient by calibration coefficient and obtain the calibration
coefficient after correction.
4. Store the final calibration coefficient after correction into EEPROM.
Note:
Voltage and current calibration is a two step process using 100% and
10% IB.
Since the dynamic range of the voltage channel is usually very small, single-point
calibration is enough to meet the accuracy requirements for full range. However, the
dynamic range of the current channel is larger, and the transformer has different ratio
errors at different current loads.
The MCP3909 device’s current channel, CH0, contains a PGA with gain options of 1,
2, 8, 16. For high-accuracy energy meters, current ratio error needs to be segmented
and calibrated for different current loads. The ratio error calibration of current channel
uses a two-point calibration method. One point is calibrated when the load is at the
rated current (IB) and the PGA gain is 1. The second point is calibrated under small-signal input condition (0.1 IB) and the PGA gain is 16.
© 2009 Microchip Technology Inc.
DS51723A-page 39
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
4.2.1
Current/Voltage Calibration Process
The process of calibration is as follows:
1. Supply the meter with balanced load, PF = 1.0, VCAL = 220V, IB = 5A.
2. Load the dsPIC33F with the proper correction factors. Automatically done using
the PC software. See Chapter 5. “PC Software”.
Repeat these following steps for the second point at 10% IB:
1. Set the three-phase balance input conditions PF = 1.0, VCAL = 220V, ICAL = 0.1 IB or
500 mA.
2. Load the dsPIC33F with the proper correction factors. Automatically done using
the PC software. See Chapter 5. “PC Software”.
If accuracy is not critical, the single-point calibration method can be used. The number
of calibration points can be defined in the header file of the program.
4.3
APPARENT POWER CALIBRATION
Apparent Power calibration function is implemented by the upper computer by sending
the commands. Before the power calibration process can be entered, power calibration
mode command needs to be sent first. The error data of the calibration workbench and
channel information to be calibrated are sent to the metering front-end. When the
front-end receives the command, it calls this module. The flow is as follows:
1. Determine the phase to be calibrated according to the parameters received.
2. Calculate new power calibration coefficient according to the error value received
and the measured value, together with the original power calibration coefficient.
3. Store the coefficient after correction into the EEPROM.
4.3.1
Apparent Power Calibration Process
The process of calibration is as follows:
1. Set the input condition as: Phase A PF = 1.0, VCAL = 220V, input current is the
current when region N is being calibrated, the voltage and current inputs of phase
B and C are zero.
2. Choose the energy pulse output to be the apparent power output mode Refer to
Chapter 6. “Meter Communications Protocol”. At this time, the energy pulse
is the accumulated multiplication of power and time.
3. Load the dsPIC33F with the proper correction factors. This is automatically done
using PC software. See Chapter 5. “PC Software”.
4. Repeat steps 1 - 3 for phase lag calibrations for all current regions of phase A.
Note:
At this time, the phase lag has not been calibrated, so when the input
PF = 1.0, the measured value of the reactive power isn't equal to zero.
5. Repeat the above steps for Phases B and C.
DS51723A-page 40
© 2009 Microchip Technology Inc.
Meter Calibration
4.4
PHASE LAG CALIBRATION
The phase lag calibration function is implemented by the upper computer by sending
the proper commands via the UART. When calibrating phase lag, error from the
calibration equipment and channel information to be adjusted are sent to the dsPIC33F
energy meter. When the front-end receives the command, it calls this module. The flow
is as follows:
1. Determine the phase to be calibrated according to parameters received.
2. Calculate new phase lag calibration coef. according to the error value received
and the measured value.
3. Store the coefficient after correction into the EEPROM.
This meter design supports single, two, and five point calibration for phase lag error
correction.
The purpose of phase lag calibration is to eliminate the impact of phase lag introduced
by the current transformer (CT), and voltage transformer (PT) over the power measurement range.
The voltage transformer usually has a constant load, thereby introducing a phase lag
that varies insignificantly. The dynamic range of current is larger, and under different
current loads, phase lags caused by CT vary greatly. In order to meet the requirements
of measurement accuracy in the entire range, it is usually necessary to segment the
phase lag and calibrate.
In this design, current is partitioned into 5 regions.
TABLE 4-1:
CURRENT REGIONS FOR PHASE CALIBRATION
Region
Current Range
1
0 - 0.075 IB
2
0.075 IB - 0.2 IB
3
0.2 IB - 0.75 IB
4
0.75 IB -1.5 IB
5
1.5 IB - 4.0 IB
The partition limit for the current region can be modified in the header file of the
program. If accuracy is not critical, single-point calibration and two-point calibration can
be used to improve the efficiency of meter calibration.
Single, Two, or Five Point Calibration
Single-, two- or five-point calibration method can be configured by modifying the
header file. When using the single-point calibration, the phase lag compensation
values of all regions are the same; When using two-point calibration, the compensation
values of region 1 and 2 (0-0.075 IB, 0.075 IB - 0.2 IB) are the same, and the phase lag
compensation values for region 3, 4 and 5 (0.2 IB - 0.75 IB, 0.75 IB - 1.5 IB, 1.5 IB 4.0 IB) are the same.
© 2009 Microchip Technology Inc.
DS51723A-page 41
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
4.4.1
Phase Lag Calibration Process
The process of phase calibration is as follow:
1. Setup input condition: Phase A, voltage input 220V, current input is the current
for region 1, voltage and current inputs of phase B and phase C are zero.
2. Load the dsPIC33F with the proper correction factors. This is automatically done
using the PC software. See Chapter 5. “PC Software”.
3. Repeat steps 1 and 2 for phase lag calibrations for all current regions of
phase A.
Note:
If the power metering error still can not meet the requirement, the meter can
be calibrated a few more times. When doing so, simply input a new error
value into the front-end of the meter.
4. Repeat for Phases B & C.
DS51723A-page 42
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 5. PC Software
5.1
OVERVIEW AND INSTALLATION
The PC software “PM_Viewer” or “Power Meter Viewer” has two main functions: view
the calculated parameters and calibrate the meter. The PC software has seven output
display screens, or “work modes”, selected from the toolbar pull-down menu.
•
•
•
•
•
•
•
Basic Parameters
Phase A Harmonic
Phase B Harmonic
Phase C Harmonic
Distortion Rate
Harmonic Power
Energy Accumulation
In addition, the PC software has four calibration screens, selected from the toolbar
pull-down menu.
•
•
•
•
Reset All Calibration
Linearity Calibration
Apparent Power Calibration
Phase Lag Calibration
5.1.1
System Required
• HDD space > 25 MB
• Microsoft Windows OS98 or later
• Hardware COM interface
5.1.2
1.
2.
3.
4.
Installation
Unzip PM_Viewer setup.zip.
Double click on setup.exe.
Finish the installation according the prompt.
To PM_Viewer.exe - Start -> Program -> Energy Meter ->PM_Viewer.exe.
© 2009 Microchip Technology Inc.
DS51723A-page 43
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
5.2
ESTABLISH COMMUNICATION
1. Open PM_Viewer.
2. Click on Comm Port selection, and select the com port (com1, com2, or com3)
that you will use on the menu, noting the baud rate is 19200 bps in 1-8-1 format,
and can not be changed.
3. Click on the Link to establish communication with the demo. “Communication
OK!” will be displayed on the bottom, if communication is established.
FIGURE 5-1:
DS51723A-page 44
Establising Communications.
© 2009 Microchip Technology Inc.
PC Software
5.3
BASIC PARAMETERS OUTPUT SCREEN
FIGURE 5-2:
5.4
Basic Parameters Work Mode Screen.
PHASE A/B/C HARMONIC OUTPUT SCREEN
FIGURE 5-3:
© 2009 Microchip Technology Inc.
Phase N Harmonic Work Mode Screen.
DS51723A-page 45
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
5.5
DISTORTION RATE
FIGURE 5-4:
5.6
Distortion Mode Screen.
HARMONIC POWER
FIGURE 5-5:
DS51723A-page 46
Harmonic Power Work Mode Screen.
© 2009 Microchip Technology Inc.
PC Software
5.7
ENERGY ACCUMULATION
FIGURE 5-6:
5.8
Energy Accumulation Work Mode Screen.
CALIBRATION STEP 1 - RESET ALL CALIBRATION
1. Select Reset All Calibration from the toolbar menu.
2. Meter Calibration Values are Reset.
FIGURE 5-7:
© 2009 Microchip Technology Inc.
Reset All Calibration Command.
DS51723A-page 47
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
5.9
LINEARITY CALIBRATION
1.
2.
3.
4.
5.
6.
DS51723A-page 48
Select Channel, either Voltage or Current.
Select Phase A, B or C.
Select Region, either 100% or 10%.
Using a standard meter, supply the input conditions given here.
Enter the error recorded from the standard meter here.
Click the Set button.
© 2009 Microchip Technology Inc.
PC Software
5.10
APPARENT POWER CALIBRATION
1.
2.
3.
4.
5.
6.
7.
8.
Select Phase A, B or C.
Select Region n.
Using a standard meter, supply the input conditions given here.
Click the Set Apparent button.
Enter the error recorded from the standard meter here.
Click the Set button.
Repeat steps 2-5 for the different regions.
Repeat for other 2 phases.
© 2009 Microchip Technology Inc.
DS51723A-page 49
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
5.11
PHASE LAG CALIBRATION
1.
2.
3.
4.
5.
6.
7.
DS51723A-page 50
Select Phase A, B or C.
Select Region n.
Using a standard meter, supply the input conditions given here.
Enter the error recorded from the standard meter here.
Click the Set button.
Repeat steps 2-5 for the different regions.
Repeat for other 2 phases.
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Chapter 6. Meter Communications Protocol
6.1
INTRODUCTION
The UART interface is used to communicate with the upper computer (MCU or PC). Via
the UART interface, the upper computer reads the measured parameters of the power
grid, and may also send system parameters and calibration parameters to target board
as well.
The communication interface is a bidirectional interface based on UART, using
master/slave half-duplex mode. The baud rate is 19,200 bps, with 1 start bit, 8 data bits
and 1 stop bit. Communication is done by frames with non-fixed-length frame structure,
definition of which is shonw in Table 6-1. Communication protocol is specified in
master-slave structure. The system in this design is the slave, and the upper computer
is the master. The master sends commands to the slave, and the slave responds to the
master.
UART communication uses half-duplex mode. The data format is 8-1-1 and the rate is
19,200 bps. The PC is the host computer, and the target board is the slave.
There are 14 command strings that the meter uses. These command strings are
defined in Table 6-1
TABLE 6-1:
COMMAND STRINGS
Command Description
Test Connection
Command
0x41
Total Data Request
0x42
Harmonic Content, Phase A
0x43
Harmonic Content, Phase B
0x44
Harmonic Content, Phase C
0x45
Total Harmonic Distortion
0x46
Energy
0x47
Stop Energy Measurement and Clear Energy Values
0x48
Harmonic Power
0x49
Write Calibration Values to Meter
0x62
Write Phase Lag Calibration Values to Meter
0x63
Write Power Calibration Values to Meter
0x64
Write Energy Pulse Configuration - Active/Apparent
0x65
Reset All Calibration Values
0x66
Write Energy Pulse Constant
0x67
© 2009 Microchip Technology Inc.
DS51723A-page 51
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
Variant-length frame structure is used and data communiate in bytes. The command
protocol structure is defined as following.
TABLE 6-2:
TYPICAL PROTOTOL
START
Command
Data
Length
Data Field
Check Sum
STOP
2 Bytes
1 Bytes
1 Byte
N Bytes
1 Byte
1 Byte
• The START word has 2 bytes, which are 0x00, 0xFF ( PC to target board) or 0xFF,
0x00 (target board to PC)
• Command word is 1 byte which indicates the type of the command
• Data length word is 1 byte that indicates the length of data field
• The data field word has multiple byte(s) that varies with command types
• Checksum word is a single byte, whose content equals to the XOR value of all
bytes sent before it
• Stop word is 1 byte with the content of 0xE0
6.2
TEST CONNECTION COMMAND
This command is sent from the PC to the meter to setup and test the connection.
TABLE 6-3:
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x41
0x00
0x00
XX
0xE0
TABLE 6-4:
6.3
PC TO METER (7 BYTES)
METER RESPONSE (8 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x42
0x02
0xA5, 0X5A
XX
0xE0
TOTAL DATA REQUEST
This is the main command retrieves all the calcuated data from the dsPIC33F. This
command gathers data from all 3 phases including total energy, power, and power
factor data.
TABLE 6-5:
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x42
0x00
0x00
XX
0xE0
TABLE 6-6:
DS51723A-page 52
PC TO METER (7 BYTES)
METER RESPONSE (104 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x43
0x62
98 bytes
XX
0xE0
© 2009 Microchip Technology Inc.
Meter Communications Protocol
TABLE 6-7:
METER RESPONSE, TOTAL REQUEST FRAME, DETAILED
DESCRIPTION
Data Field
Byte
Name
Value
1,2
Status
See Definition Below
3-6
Frequency
Float, 4 Bytes Total
7-10
Phase A Voltage
Float, 4 Bytes Total
Phase B Voltage
Float, 4 Bytes Total
94-98
© 2009 Microchip Technology Inc.
Phase C Voltage
Float, 4 Bytes Total
Phase A Current
Float, 4 Bytes Total
Phase B Current
Float, 4 Bytes Total
Phase C Current
Float, 4 Bytes Total
Neutral Current
Float, 4 Bytes Total
Active Power, Phase A
Float, 4 Bytes Total
Reactive Power, Phase A
Float, 4 Bytes Total
Apparent Power, Phase A
Float, 4 Bytes Total
Power Factor, Phase A
Float, 4 Bytes Total
Active Power, Phase B
Float, 4 Bytes Total
Reactive Power, Phase B
Float, 4 Bytes Total
Apparent Power, Phase B
Float, 4 Bytes Total
Power Factor, Phase B
Float, 4 Bytes Total
Active Power, Phase C
Float, 4 Bytes Total
Reactive Power, Phase C
Float, 4 Bytes Total
Apparent Power, Phase C
Float, 4 Bytes Total
Power Factor, Phase C
Float, 4 Bytes Total
Total Active Power
Float, 4 Bytes Total
Total Reactive Power
Float, 4 Bytes Total
Total Apparent Power
Float, 4 Bytes Total
Total Power Factor
Float, 4 Bytes Total
DS51723A-page 53
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
6.4
STATUS REGISTER
This register contains the gain register.
REGISTER 6-1:
STATUS REGISTER
R-0
R-0
R-0
R-0
R-0
R-0
R-0
R-0
CPO
VPO
PHC_S1
PHC_S0
PHB_S1
PHB_S0
PHA_S1
PHA_S0
bit 7
bit 0
Legend:
R = Readable bit
W = Writable bit
U = Unimplemented bit, read as ‘0’
-n = Value at POR
‘1’ = Bit is set
‘0’ = Bit is cleared
bit 7
CPO: Current Phase Order
1 = Problem Detected
0 = Normal
bit 6
VPO: Voltage Phase Order
1 = Problem Detected
0 = Normal
bit 5:4
PHC_S: Phase C Status
11 = High Votage
10 = No Input
01 = Low Voltage
00 = Normal
bit 3:2
PHB_S: Phase C Status
11 = High Votage
10 = No Input
01 = Low Voltage
00 = Normal
bit 1:0
PHA_S: Phase C Status
11 = High Votage
10 = No Input
01 = Low Voltage
00 = Normal
6.5
x = Bit is unknown
HARMONIC CONTENT COMMAND
TABLE 6-8:
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x43
0x00
0x00
XX
0xE0
TABLE 6-9:
DS51723A-page 54
PC TO METER (7 BYTES)
METER RESPONSE (134 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x44
0x80
128 bytes
XX
0xE0
© 2009 Microchip Technology Inc.
Meter Communications Protocol
TABLE 6-10:
HARMONIC ANALYSIS DETAILED DESCRIPTION
Data Field
Byte
6.6
Description
Value
1,2
Fundamential or 1st Harmonic, Voltage Content
Unsigned Int, 2 bytes
3,4
2nd Harmonic, Voltage Content
Unsigned Int, 2 bytes
5-56
3-31st Harmonic, Voltage Content
Unsigned Int, 2 bytes
57,58
Total Voltage Harmonic Content (not including
Fundamental)
Unsigned Int, 2 bytes, /
1000 * (100%)
59,60
Fundamential or 1st Harmonic, Current Content
Unsigned Int, 2 bytes
61,62
2nd Harmonic, Current Content
Unsigned Int, 2 bytes
63-126
3-31st Harmonic, Current Content
Unsigned Int, 2 bytes
127,128
Total Current Harmonic Content (not including
Fundamental)
Unsigned Int, 2 bytes, /
1000 * (100%)
TOTAL HARMONIC DISTORTION (THD) COMMAND
TABLE 6-11:
PC TO METER (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x46
0x00
0x00
XX
0xE0
TABLE 6-12:
METER RESPONSE (29 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x47
0x18
24 bytes
XX
0xE0
TABLE 6-13:
Data Field
Byte
TOTAL HARMONIC DISTORTION DESCRIPTION
Description
Value
1-4
Total Harmonic Distortion of Phase A Voltage
Float, 4 Bytes Total
5-8
Total Harmonic Distortion of Phase B Voltage
Float, 4 Bytes Total
9-12
Total Harmonic Distortion of Phase C Voltage
Float, 4 Bytes Total
13-16
Total Harmonic Distortion of Phase A Current
Float, 4 Bytes Total
17-20
Total Harmonic Distortion of Phase B Current
Float, 4 Bytes Total
21-24
Total Harmonic Distortion of Phase C Current
Float, 4 Bytes Total
© 2009 Microchip Technology Inc.
DS51723A-page 55
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
6.7
START ENERGY MEASUREMENT COMMAND
TABLE 6-14:
PC TO METER (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x47
0x00
0x00
XX
0xE0
TABLE 6-15:
METER RESPONSE (42 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x48
0x24
36 bytes
XX
0xE0
TABLE 6-16:
TOTAL HARMONIC DISTORTION DESCRIPTION
Data Field
Byte
6.8
Description
Value
1-4
First quadrant reactive energy
Float, 4 Bytes Total
5-8
Second quadrant reactive energy
Float, 4 Bytes Total
9-12
Third quadrant reactive energy
Float, 4 Bytes Total
13-16
Fouth quadrant reactive energy
Float, 4 Bytes Total
17-20
Forward Reactive energy
Float, 4 Bytes Total
21-24
Reverse Reactive Energy
Float, 4 Bytes Total
25-28
Forward Active Energy
Float, 4 Bytes Total
29-32
Reverse Active Energy
Float, 4 Bytes Total
33-36
Reverse Active Energy
Float, 4 Bytes Total
STOP ENERGY MEASUREMENT COMMAND
TABLE 6-17:
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x48
0x00
0x00
XX
0xE0
TABLE 6-18:
DS51723A-page 56
PC TO METER (7 BYTES)
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x49
0x00
0 bytes
XX
0xE0
© 2009 Microchip Technology Inc.
Meter Communications Protocol
6.9
HARMONIC POWER COMMAND
TABLE 6-19:
PC TO METER (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x49
0x00
0x00
XX
0xE0
TABLE 6-20:
METER RESPONSE (54 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x4A
0x30
48 bytes
XX
0xE0
TABLE 6-21:
Data Field
Byte
1-4
HARMONIC POWER MEASUREMENTS
Description
Fundamental Active Power Of Phase A
Value
Float, 4 Bytes Total
5-8
Fundamental Reactive Power Of Phase A
Float, 4 Bytes Total
9-12
Fundamental Active Power Of Phase B
Float, 4 Bytes Total
13-16
Fundamental Reactive Power Of Phase B
Float, 4 Bytes Total
17-20
Fundamental Active Power Of Phase C
Float, 4 Bytes Total
21-24
Fundamental Reactive Power Of Phase C
Float, 4 Bytes Total
25-28
Harmonic Active Power Of Phase A
Float, 4 Bytes Total
29-32
Harmonic Reactive Power Of Phase A
Float, 4 Bytes Total
33-36
Harmonic Active Power Of Phase B
Float, 4 Bytes Total
37-40
Harmonic Reactive Power Of Phase B
Float, 4 Bytes Total
41-44
Harmonic Active Power Of Phase C
Float, 4 Bytes Total
45-48
Harmonic Reactive Power Of Phase C
Float, 4 Bytes Total
© 2009 Microchip Technology Inc.
DS51723A-page 57
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
6.10
CALIBRATE METER VOLTAGE/CURRENT COMMAND
TABLE 6-22:
PC TO METER (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x63
7
0x07
XX
0xE0
TABLE 6-23:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x64
0
0x00
XX
0xE0
TABLE 6-24:
CALIBRATION OF GAIN AND OFFSET VALUES
Data Field
Byte
DS51723A-page 58
Description
Value
1
Phase Select
0x01 = Phase A
0x02 = Phase B
0x03 = Phase C
2
Range Select
0x01 = 10%
0x02 = 100%
3
Channel Select
0x00 = Current
0x01 = Voltage
4-7
Correction Factor (Error Being Calibrated Out)
Float, 4 Bytes Total
© 2009 Microchip Technology Inc.
Meter Communications Protocol
6.11
CALIBRATE PHASE LAG COMMAND
TABLE 6-25:
PC TO METER (12 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x63
6
0x06
XX
0xE0
TABLE 6-26:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x64
0
0x00
XX
0xE0
TABLE 6-27:
CALIBRATION OF PHASE LAG
Data Field
Byte
6.12
Description
Value
1
Phase Select
0x01 = Phase A
0x02 = Phase B
0x03 = Phase C
2
Range Select
1-7
3-6
Correction Factor (Error Being Calibrated Out)
Float, 4 Bytes Total
CALIBRATE APPARENT POWER COMMAND
TABLE 6-28:
PC TO METER (12 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x64
6
0x06
XX
0xE0
TABLE 6-29:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x65
0
0x00
XX
0xE0
TABLE 6-30:
CALIBRATION OF POWER
Data Field
Byte
Description
Value
1
Phase Select
0x01 = Phase A
0x02 = Phase B
0x03 = Phase C
2
Current Range Select
1-7
3-6
Correction Factor (Error Being Calibrated Out)
Float, 4 Bytes Total
© 2009 Microchip Technology Inc.
DS51723A-page 59
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
6.13
CALIBRATE ENERGY PULSE COMMAND
TABLE 6-31:
PC TO METER (12 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x65
2
0x02
XX
0xE0
TABLE 6-32:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x66
0
0x00
XX
0xE0
TABLE 6-33:
CALIBRATION OF POWER
Data Field
Byte
6.14
Description
Value
1
Phase Select
0x01 = Phase A
0x02 = Phase B
0x03 = Phase C
2
Energy Output Mode
0x00 = Active Power
0x01 = Apparent Power
RESET ALL METER CALIBRATION VALUES COMMAND
TABLE 6-34:
START
Command
Data
Length
Data Field
Check Sum
STOP
0x00, 0xFF
0x66
4
0x04
XX
0xE0
TABLE 6-35:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x67
0
0x00
XX
0xE0
TABLE 6-36:
Data Field
Byte
DS51723A-page 60
PC TO METER (12 BYTES)
RESET METER OPTIONS
Description
Value
1
Command Type
0x55 - Reset All
0xA1 = Reset
0xA2 - Reset Power Calibration Only
0xA3 - Reset Phase Calibration
2
Phase Select
0x01 = Phase A
0x02 = Phase B
0x03 = Phase C
3
Current Range Select
4
Reserved
© 2009 Microchip Technology Inc.
Meter Communications Protocol
6.15
CALIBRATE METER CONSTANT (ENERGY PULSE OUTPUT CONSTANT)
TABLE 6-37:
PC TO METER (9 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0x00, 0xFF
0x67
2
0x02
XX
0xE0
TABLE 6-38:
METER RESPONSE (7 BYTES)
START
Command
Data
Length
Data Field
Check Sum
STOP
0xFF, 0x00
0x68
0
0x00
XX
0xE0
TABLE 6-39:
Data Field
Byte
1-2
© 2009 Microchip Technology Inc.
ENERGY CONSTANT OPTIONS OPTIONS
Description
Energy Constant
Value
Range of 10064000 (Decimal)
DS51723A-page 61
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
NOTES:
DS51723A-page 62
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Appendix A. Schematics and Layouts
A.1
INTRODUCTION
This appendix contains the following schematics and layouts for the MCP3909 /
dsPIC33F 3-Phase Energy Meter Reference Design.
•
•
•
•
•
•
•
A.2
Power Supply Board Schematic
Main Board Schematic - Page 1
Main Board Schematic - Page 2
Power Supply Board - Assembly Drawing
Power Supply Board - Composite Drawing
Main Board - Assembly Drawing
Main Board - Composite Drawing
SCHEMATICS AND PCB LAYOUT
FIGURE A-1:
LAYER ORDER
Top Layer
Bottom Layer
© 2009 Microchip Technology Inc.
DS51723A-page 63
DS51723A-page 64
A
B
N
PC
N
PC
N
PB
N
PB
N
PA
N
1
150k
T3
499K 1%
499K 1%
R11
R13
R14
150k
T2
499K 1%
499K 1%
R7
R8
R9
150k
T1
499K 1%
499K 1%
R2
R3
R4
SPT204B
R10
1K 1%
SPT204B
R6
1K 1%
SPT204B
R1
1K 1%
C7
33N
Jumper
W3
C5
33N
Jumper
W2
C3
33N
Jumper
W1
PC-
PC+
PC-
PB-
PB+
PB-
PA-
PA+
PA-
2
2
PA
PB
PC
N
1
JP6
1
JP5
1
JP4
1
JP1
4
3
2
1
1
2
3
4
PC+
PC-
PB+
PB-
PA+
PA-
N
0
R12
DB-12CY
A
B
C
N
T4
COILS
A
B
C
N
R5
5
6
7
8
8
7
6
5
Header 10
1
2
3
4
5
6
7
8
9
10
JP3
G1
Vo1
G2
Vo2
A
B
C
N
3
3
Date:
File:
A4
Size
RV2
Header 3
1
2
3
JP6
C4
0.1u
C6
0.1u
N
RV3
N
1-1
2.0
Revision
A
B
C
D
4
30-Nov-2007
Sheet of
X:\AIPD Eval Boards\102-00011 thru 00020\102-00018\PCB
Drawn By:
Files (Protel )\PCB-R5\PM33_
Number
Power supply for 3phs power meter
RV1
Title
+ C2
100uf
12V
C1
0.1U
PA
PB
PC
three phase voltage input
2R = Vour*Rin/(Kpt*Vin)
4
FIGURE A-2:
C
D
PA
1
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
POWER SUPPLY BOARD SCHEMATIC
© 2009 Microchip Technology Inc.
© 2009 Microchip Technology Inc.
A
C104
1
R129
4.7K 1%
R130
4.7K 1%
3.3V
REF_V
SCT220B
REV_V
4
8
6
2
1
2
3
4
1K
R111
R127
4.7K
2
MCP6002
VDD
OutB
InBInB+
U101
R128
470
OutA
InAInA+
Vss
20
R126
1.65V
Neutral line current detection
T100
R110
R107
R106
R103
20
2R = Vout/(Kct*Iin)
R125
20
R124
20
R123
20
R102
R109
1nf
8
7
6
5
1nf
1K
1K
C100
1nf
3.3V
C112
1nf
C111
C108
1nf
C107
1nf
C103
1K 1nf
1K
1K 1nf
Current_N
C128
0.1uF
PAC-
PAC+
PBC-
PBC+
PCC-
PCC+
PAV+
PAV-
Current_N
3
5V
5V
5V
AD_MCLR
R118
C127
0.1
10
C125
10
C116
10
C120
R119
R120
C114
C118
+
4
8
6
2
4
8
6
2
20
R122
R121
100
R117
1nf
C110
C109
1nf
PBV+
PBV-
PCV+
PCV-
5V
5V
5V
C117
C115
C119
C124
C123
1u
PAC+
PACPAVPAV+
AD MCLR
0.1 C126
500
FB1
C122
PBC+
PBCPBVPBV+
AD_MCLR
500
FB2
PCC+
PCCPCVPCV+
AD MCLR
C121
1uf
C113
500
FB3
+
SCT220B
T103
SCT220B
T102
4
8
6
2
1K
R108
1K
C106
1nf
C105
1K 1nf
C102
C101
1nf
+
SCT220B
T101
1K
R105
100
100
R104
R101
1K
100
2R = Vour*Rin/(Kpt*Vin)
R116
R115
R114
100
3
+
B
C
3
2
1
3
2
1
100
R113
R112
R100
+
J8
J7
3
2
1
2
1
2
3
4
5
6
7
8
9
10
11
12
1
2
3
4
5
6
7
8
9
10
11
12
1
2
3
4
5
6
7
8
9
10
11
12
U104
4
DVDD
HPF
AVDD
NC
CH0+
CH0CH1_
CH1+
MCLR
REFi/o
AGND
F2
U102
DVDD
HPF
AVDD
NC
CH0+
CH0CH1_
CH1+
MCLR
REFi/o
AGND
F2
U103
3.2768M
VCC
CLKOUT
GND
X100
Fo0
Fo1
HFo
DGND
NEG
NC
OSC2
OSC1
G0
G1
F0
F1
MCP3909
Fo0
Fo1
HFo
DGND
NEG
NC
OSC2
OSC1
G0
G1
F0
F1
MCP3909
Fo0
Fo1
HFo
DGND
NEG
NC
OSC2
OSC1
G0
G1
F0
F1
MCP3909
7
5V_IN
14
DVDD
HPF
AVDD
NC
CH0+
CH0CH1_
CH1+
MCLR
REFi/o
AGND
F2
4
24
23
22
21
20
19
18
17
16 G0A
15 G1A
14
13
24
23
22
21
20
19
18
17
16
15
14
13
24
23
22
21
20
19
18
17
16
15
14
13
8
G0B
G1B
SCK
G0C
G1C
CSC
SDI
SDO
AD_CLK
5
G0A
5
CSC
CSB
CSA
SDO
SDI
SCK
R132
4.7K
GAIN
3905 CS
SPI I/F
R134
4.7K
R135
4.7K
G1C
R136
4.7K
GAIN
6
A
B
C
D
Date:
File:
B
Size
2-1
Revision
1.0
6
30-Nov-2007
Sheet of
Jemmey Huang
X:\AIPD Eval Boards\102-00011 thru 00020\102-00018\PCB
Drawn By:
Files (Protel )\PCB-R5\PM33_
Number
Front-end of three Phase Power Meter Design
Title
R131
4.7K
R133
4.7K
Gain selection
G1 G0 Gain
0 0
1
0 1
2
1 0
8
1 1
16
G1A
J6
G0B
+
G1B
FIGURE A-3:
G0C
D
1
Schematics and Layouts
MAIN BOARD SCHEMATIC - PAGE 1
DS51723A-page 65
AD_CLK
A
B
C
D
J1
2
1
6
5
4
3
2
1
CON6
J2
MCLR
1
L301
INDUCTOR
INDUCTOR
5V_IN
D301
LED
R316
470
510
R313
3.3V
S301
SW-PB
ICSPCLK
ICSPDAT
L302
C340
0.1uf
R321
10k
3.3V
F = 50Hz * 64 * 256 = 819.2 KHz
F = 50Hz * 128 * 256 = 1638.4 KHz
* 3905 clk range 1M to 4M
AD CLK = Fac * SampleRate * Kad
1
2 1
2
+ C337
100uf
GAIN
3909 CS
SPI I/F
MCLR
C328
3
2
C323
CAP
C322
CAP
G0A
G1A
G0B
G1B
G0C
G1C
CSA
CSB
CSC
SDI
SDO
SCK
3.3V
C330
100
100
100
3.3V
C331
Vout
U306
MCP1700(3.3V-SOT23)
2
Vin
Vout
Vin
3
3.3V
U305
MCP1701(5V-SOT89)
R319
R318
R317
3.3V
2
C332
C324
CAP
C336
0.1uF
G1C
G0C
CSA
CSB
CSC
GND
3.3V
+ C339
47uF
RG15
AN16/RC1
AN17/RC2
RG6
RG7
RG8
MCLR
SS2/RG9
Vss
Vdd
AN15/RB5
AN4/RB4
AN3/RB3
AN2/RB2
PGC3/RB1
PGD3/RB0
3.3V
+ C338
100uf
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
C335
5V
3.3V
AD_MCLR
C333
G1A
G0B
G1B
SCK2
SDO2
SDI2
MCLR
3.3V
3
C329
3
1
2
3
4
24LC04B
A0 VCC
A1
WP
A2 SCL
Vss SDA
U309
Current_N
REF_V
2
VSS
1
Gnd
1
3.3V
G0A
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
8
7
6
5
3.3V
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
SDA
R311
1k
4
D302
D303
SCL
R312
1k
4
C334
R309
100
SDO
GND
OSC2
OSC1
3.3V
SCL
SDA
SCK1
U1RX(SDI1)
U1TX(SDO1)
DSPIC33FJ128GP706
U307
RC14
RC13
RD0
IC4/RD11
RD10
RD9
RD8
Vss
OSC2/RC15
OSC1/RC12
Vdd
SCL1/RG2
SDA1/RG3
INT0/RF6
RF2
RF3
RG13
RG12
RG14
RG0
RG1
RF1
RF0
Vdd
Vddcore
RD7
RD6
RD5
RD4
RD3
RD2
RD1
DS51723A-page 66
PGC1/RB6
PGD1/RB7
AVdd
AVss
AN8/RB8
AN9/RB9
AN10/RB10
AN11/RB11
Vss
Vdd
AN12/RB12
AN13/RB13
AN14/RB14
AN15/RB15
RF4
RF5
R308
100
LED
LED
R307
100
X303
470
R310
470
R314
C346
C345
0.1uf
3.3V C303
C302
0.1uf
C301
0.1uf
5
3
4
1
U1RX/SDI1
12
9
11
U1TX/SDO1 10
0.1
C342
0.1
C341
100
R306
R303
1K
R302
1K
R301
1K
5
5
R1IN
R2IN
T1OUT
T2OUT
VCC
GND
V-
V+
13
8
14
7
16
15
6
2
AD_CLK
0.1
0.1
C344
3.3V
C343
U1TX/SDO1
U1RX/SDI1
SPI1_SCK
1
6
2
7
3
8
4
9
5
DB9
J5
1
2
3
4
J4
6
5
4
3
2
1
J3
Date:
File:
B
Size
2-2
Revision
1.0
A
B
C
D
6
30-Nov-2007
Sheet of
Jemmey Huang
X:\AIPD Eval Boards\102-00011 thru 00020\102-00018\PCB
Drawn By:
Files (Protel )\PCB-R5\PM33
Number
Front-end of three Phase Power Meter Design
Title
MAX232
R1OUT
R2OUT
T1IN
T2IN
C2-
C1C2+
C1+
U308
AD_CLK
U303
TLP521-1
U302
TLP521-1
U301
TLP521-1
Output Pulse
1: active power 2: active power +
3: Reactive power 4: Reactive power +
5: TBD 6: TBD +
6
FIGURE A-4:
ICSPCLK 17
ICSPDAT 18
3.3V
19
GND
20
21
22
23
24
GND
25
3.3V
26
Current_N
27
28
29
30
31
32
1
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
MAIN BOARD SCHEMATIC - PAGE 2
© 2009 Microchip Technology Inc.
Schematics and Layouts
FIGURE A-5:
POWER SUPPLY BOARD LAYOUT - ASSEMBLY DRAWING
© 2009 Microchip Technology Inc.
DS51723A-page 67
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
FIGURE A-6:
DS51723A-page 68
POWER SUPPLY BOARD LAYOUT - COMPOSITE DRAWING
© 2009 Microchip Technology Inc.
Schematics and Layouts
FIGURE A-7:
MAIN BOARD LAYOUT - ASSEMBLY DRAWING
© 2009 Microchip Technology Inc.
DS51723A-page 69
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
FIGURE A-8:
DS51723A-page 70
MAIN BOARD LAYOUT - COMPOSITE DRAWING
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Appendix B. Bill Of Materials (BOM)
TABLE B-1:
Qty
BILL OF MATERIALS - POWER SUPPLY (BOTTOM BOARD)
Reference
Description
3
1
3
3
C1, C2, C3
C4
C5, C6, C7
J4, J5, J6
CAP 0.1UF 305VAC EMI SUPPRESS
CAP 470UF 25V ALUM LYTIC RADIAL
DO NOT POPULATE
CONN HEADER 3POS .100 VERT TIN
4
HOOK-UP WIRE 18AWG STRAND RED
1
JP1, JP2, JP3,
JP4
JP1, JP2, JP3,
JP4
JP6
CONN HEADER 3POS .156 VERT TIN
1
JP6
CONN HOUSING 3POS .156 W/O RAMP
3
JP6
CONN TERM FEMALE 18-24AWG TIN
1
PCB
3
P4, P5, P6
9
P4, P5, P6
RoHS Compliant Bare PCB, dsPIC33F
and MCP3909 3-Phase Energy Meter
(Power) Bottom Bd.
CONN HOUS 3POS .100 W/RAMP/RIB
(Connects to Above)
CRIMP TERM FEMALE 22-30AWG TIN
1
1
6
R1
R2
R3, R4, R5, R6,
R7, R8
R9, R10, R11
4
CONN RING TERM #6 18-22AWG
Part of T1
DO NOT POPULATE
DO NOT POPULATE
Manufacturer
Part Number
EPCOS Inc
Panasonic® - ECG
—
Molex®/Waldom
Electronics Corp
Alpha Wire
Company
Molex/Waldom Electronics Corp
Molex/Waldom
Electronics Corp
Molex/Waldom
Electronics Corp
Molex/Waldom
Electronics Corp
Microchip
Technology Inc.
B32922A2104M
ECA-1EM471
—
22-23-2031
Molex/Waldom
Electronics Corp
Molex/Waldom
Electronics Corp
Sanki
—
—
22-01-3037
3055 RD005
19070-0040
26-48-1035
09-50-7031
08-50-0105
104-00158
08-65-0805
—
—
—
5033ED150K0F12AF5
Vishay®/Phoenix
Passive
Components
3
R12, R13, R14
DO NOT POPULATE
—
—
3
RV1, RV2, RV3
VARISTOR 300V RMS 14MM RADIAL
EPCOS Inc
S14K300E2
1
T1
3P AC-DC converter, 220V - 12V/5V
Sanki
DB-12CY220
3
T2, T3, T4
2mA/2mA Current transformer
Xinge
SPT204
3
W1, W2, W3
DO NOT POPULATE
—
—
Note 1: The components listed in this Bill of Materials are representative of the PCB assembly. The released BOM
used in manufacturing uses all RoHS-compliant components.
3
RES 150K OHM METAL FILM .50W 1%
© 2009 Microchip Technology Inc.
DS51723A-page 71
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
TABLE B-2:
Qty
13
6
26
4
BILL OF MATERIALS - TOP BOARD
Reference
C100, C101, C102,
C103, C104, C105,
C106, C107, C108,
C109, C110, C111,
C112
C113, C114, C115,
C116, C124, C125
C117, C118, C119,
C120, C126, C127,
C128, C301, C302,
C303, C322, C323,
C324, C328, C330,
C331, C332, C333,
C334, C335, C336,
C340, C341, C342,
C343, C344
C121, C122, C123,
C329
C337, C338, C339
Description
Manufacturer
Part Number
®
CAP 1000PF 50V CERAMIC X7R
0805
Kemet Electronics C0805C102K5RACTU
Corp
CAP CER 10UF 16V X7R 1206
Murata Electronics® GRM31CR71C106KAC7L
North America
Panasonic® - ECG ECJ-2VB1E104K
CAP .1UF 25V CERAMIC X7R 0805
CAP 1.0UF 10V CERAMIC X7R 0805 Kemet Electronics C0805C105K8RACTU
Corp
3
CAP 470UF 25V ALUM LYTIC
Panasonic - ECG
ECA-1EM471
RADIAL
2
C345, C346
CAP 22PF 50V CERM CHIP 0805
Panasonic - ECG
ECJ-2VC1H220J
SMD
3
6" Wire Thru CT
HOOK-UP WIRE 16AWG STRAND
Alpha Wire
3057 RD005
RED
Company
6
CT Wire terminals
CONN RING TERM #6 18-22AWG
Molex/Waldom
19070-0040
Electronics Corp
1
D301
LED 3MM ALGAAS RED CLEAR
LITE-ON INC
LTL-4266N
1
D302
LED 3MM GREEN CLEAR
LITE-ON INC
LTL-4236N
1
D303
LED 3MM YELLOW CLEAR
LITE-ON INC
LTL-4256N
3
FB1, FB2, FB3
FERRITE SMT
luying
STBL2012-121
1
J1
CONN HOUSING 3POS .156 W/O
Molex/Waldom
09-50-7031
RAMP
Electronics Corp
1
J1
CONN TERM FEMALE 18-24AWG
Molex/Waldom
08-50-0105
TIN
Electronics Corp
1
J1
HOOK-UP WIRE 22AWG STRAND
Alpha Wire
3051 RD005
RED
Company
1
J1
HOOK-UP WIRE 22AWG STRAND
Alpha Wire
3051 BK005
BLACK
Company
1
J2
CONN MOD JACK 6-6 VERT PCB
Tyco
5520258-3
50AU
Electronics/Amp
1
J3
CONN HEADER 6POS .100 VERT
Molex/Waldom
22-27-2061
TIN
Electronics Corp
1
J4
CONN HEADER 4POS .100 VERT
Molex/Waldom
22-27-2041
TIN
Electronics Corp
1
J5
DB9 Female vertical
Tyco
747091-2
Electronics/Amp
3
J6,J7 & J8
CONN HOUS 3POS .100
—
—
W/RAMP/RIB (Connects to J4,J5 & J6
of Lower Board)
9
J6,J7 & J8
CRIMP TERM FEMALE 22-30AWG
Molex/Waldom
08-65-0805
TIN
Electronics Corp
Note 1: The components listed in this Bill of Materials are representative of the PCB assembly. The released BOM
used in manufacturing uses all RoHS-compliant components.
DS51723A-page 72
© 2009 Microchip Technology Inc.
Bill Of Materials (BOM)
TABLE B-2:
Qty
3
BILL OF MATERIALS - TOP BOARD
Reference
Description
1
HOOK-UP WIRE 22AWG STRAND
GREEN
J6,J7 & J8
HOOK-UP WIRE 22AWG STRAND
RED
J6,J7 & J8
HOOK-UP WIRE 22AWG STRAND
BLACK
L301, L302
60ohm bead
PCB
RoHS Compliant Bare PCB,
dsPIC33F and MCP3909 3-Phase
Energy Meter (Power) Bottom Bd.
R100, R101, R102, RES 1.00K OHM 1/8W 1% 0805 SMD
R103, R104, R105,
R106, R107, R108,
R109, R110, R111,
R301, R302, R303,
R311, R312
R112, R113, R114, RES 100 OHM 1/8W 1% 0805 SMD
R115, R116, R117,
R307, R308, R309,
R317, R318, R319
R118, R119, R120 RES 10.0 OHM 1/8W 1% 0805 SMD
R121, R122, R123, RES 20.0 OHM 1/8W 1% 0805 SMD
R124, R125, R126
R127, R131, R132, RES 4.7K OHM 1/8W 5% 0805 SMD
R133, R134, R135,
R136
R128, R310, R314, RES 470 OHM 1/8W 5% 0805 SMD
R316
R129, R130
RES 4.70K OHM 1/8W 1% 0805 SMD
R313
RES 510 OHM 1/8W 5% 0805 SMD
R321
RES 10.0K OHM 1/8W 1% 0805 SMD
S301
SWITCH TACT 6MM MOMENTARY
250GF
X100
3.2768Mhz Crystal
X303
7.3728Mhz Crystal
U305
2uA Low Dropout Positive Voltage
Regulator
U306
Low Quiescent Current LDO
1
U307
1
U308
1
U309
3
3
2
1
17
12
3
6
7
4
2
1
1
1
1
1
1
J6,J7 & J8
High-Performance, 16-bit Digital Signal Controllers
IC DRVR/RCVR MULTCH RS232
16SOIC
4K I2C™ Serial EEPROM
Manufacturer
Part Number
Alpha Wire
Company
Alpha Wire
Company
Alpha Wire
Company
Jones
—
3051 GR005
B60
104-00159
Panasonic - ECG
ERJ-6ENF1001V
Panasonic - ECG
ERJ-6ENF1000V
Panasonic - ECG
Panasonic - ECG
ERJ-6ENF10R0V
ERJ-6ENF20R0V
Panasonic - ECG
ERJ-6GEYJ472V
Panasonic - ECG
ERJ-6GEYJ471V
Yageo Corporation
Panasonic - ECG
Panasonic - ECG
E-Switch
9C08052A4701FKHFT
ERJ-6GEYJ511V
ERJ-6ENF1002V
TL1105BF250Q
Koan
Koan
Microchip
Technology Inc
Microchip
Technology Inc
Microchip
Technology Inc
Texas Instruments
DIP-8-3.2768M
HC-49S-SMD-7.3728M
MCP1701T-5002I/MB
3051 RD005
3051 BK005
MCP1700T-3302E/TT
dsPIC33FJ128GP206
MAX3232CDR
Microchip
24LC04B-E/SN
Technology Inc
3
U102, U103 U104 Energy Metering IC with SPI Interface Microchip
MCP3909-I/SS
and Active Power Pulse Output
Technology Inc
1
U101
1MHz, Low Power Op-Amp
Microchip
MCP6002-I/SN
Technology Inc
3
T101 T102, T103
5A/5mA Current transformer
Xinge
SCT954F
1
T100
" DO NOT POPULATE
—
—
Note 1: The components listed in this Bill of Materials are representative of the PCB assembly. The released BOM
used in manufacturing uses all RoHS-compliant components.
© 2009 Microchip Technology Inc.
DS51723A-page 73
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
NOTES:
DS51723A-page 74
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Appendix C. Power Calculation Theory
C.1
OVERVIEW
This MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design is unique in that
all calculations are done in the frequency domain. This is easily realized using the DSP
engine core of the advanced 16-bit MCU, the dsPIC33F. In addition to performing direct
fourier transforms (DFTs) on all the input channel measurements, an additional
firmware function is included, quasi-synchronous sampling.
C.2
SYNCHRONOUS SAMPLING AND QUASI-SYNCHRONOUS SAMPLING
The fundamentals of quasi-synchronous sampling and corresponding methods for
measuring AC electrical parameters are discussed in this section. Typically, a
synchronous sampling method is used for measuring electrical parameters. The
method requires synchronization between sampling intervals and power grid
frequency. For these types of meter designs, an external hardware PLL circuit is used
to track power grid frequency, and the clock of the MCP3909 device is automatically
updated to change the sampling frequency. Since the PLL output frequency drops
behind the power grid frequency, a synchronous error exists in the system, and fully
synchronous sampling is hard to achieve. In addition, as non-sine waves exist in the
power grid, which may affect zero-crossing detection, when such conditions worsen, it
may cause PLL failure, preventing the system from working normally.
For the MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design presented
here, the dsPIC33F performs additionall calculations that eliminates the need for a
costly external PLL circuit. This quasi-synchronous sampling method has an
advantage in the engineering practice, which actually is periodic sampling, without
synchronizing with the power grid frequency. Therefore, the zero-crossing detection
and PLL circuit can be reduced to lower the hardware complexity. The tradeoff is the
increased software requirements of the system, easily realized using the powerful
dsPIC33F.
For DFT or FFT harmonic analysis of periodic signals, a Fourier transform may only
bring accurate spectrum analysis when the sampling points satisfy N > 2M for each line
cycle and strict synchronous sampling is realized. (Where M is the maximal harmonic
order of periodic signals, and N is the number of samples per line cycle).
Otherwise, if N ≤ 2M, it will cause spectrum aliasing. In addition, if strict synchronous
sampling cannot be realized, spectral-leakage will occur (the Hurdle Effect). However,
in the quasi-synchronous sampling method emplyed here, strict synchronization
between sampling intervals and the period of sampled signal is not guaranteed but is
overcome through post-processing and iteration of the collected data. To reduce the
error caused by this problem and to obtain better accuracy when measuring the
fundamental and harmonics of higher orders, an increase in the number of iterations
when processing data to improve accuracy is performed.
The iterations can effectively reduce the impact of synchronization error over the measurement accuracy, and is one of the methods to realize accurate measurement of the
frequency and harmonics under steady-state conditions.
© 2009 Microchip Technology Inc.
DS51723A-page 75
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
T≠N*Ts
V
t
Ts
FIGURE C-1:
C.2.1
Quasi Synchronous Sampline.
Basic Idea of Quasi-Synchronous Sampling
Assuming that the average of a periodic signal in one cycle is g(t),
EQUATION C-1:
( TO + T )
T
1
1
g ( t ) = --- ⋅ ∫ g ( t )dt = --- ⋅
T
T
∫
0
g ( t )dt
TO
Make t = x/ω, then,
EQUATION C-2:
2π
1
g ( t ) = ------ ⋅
2π
∫ f ( x ) dx
= f(x)
0
where f(x) = g(x/ω), and the period is 2π.
If the entire period sampling cannot be realized while a sampling frequency deviation
Δ exists, then:
EQUATION C-3:
(2π + Δ)
1
f ( x ) ≠ ----------------- ⋅
2π + Δ
∫
(α + 2π + Δ)
1
f ( x ) dx ≠ ----------------- ⋅
2π + Δ
0
∫
f ( x ) dx
α
We have:
EQUATION C-4:
(α + 2π + Δ)
1
1
F ( α ) = ----------------- ⋅
2π + Δ
∫
f ( x ) dx
α
The value of F1(α) is a function of α and also a function with 2π as its perod. The
non--synchronous sampling error E = f(x)- F1(α).
As F1(α) is function with 2π as its period, its value may be averaged through integration
within the range of 0-2π, and it can be deduced that f(x) = F1(α).
DS51723A-page 76
© 2009 Microchip Technology Inc.
Power Calculation Theory
Assuming that the integral starts at β, then:
EQUATION C-5:
(β + 2π)
1
f ( x ) = F ( α ) = ------ ⋅
2π
∫
1
1
F ( α ) dα
β
Likewise, as strict integration cannot be realized in the entire cycle, so:
EQUATION C-6:
(β + 2π + Δ)
1
f ( x ) = F ( α ) ≠ ----------------- ⋅
2π + Δ
∫
1
1
F ( α ) dα
β
Similarly, the integral value of above equation is related to β with 2π as its period, let’s
denote it as F2(β). If it won't confuse people, we'll write F1(α) and F2(β) as F1(x) and
F2(x), and a recurrence formula can be obtained as the following:
EQUATION C-7:
(x + 2π + Δ)
n
1
F ( α ) = ----------------- ⋅
2π + Δ
∫
F
n–1
( x ) dx
x
It can be proved that,
EQUATION C-8:
n
lim F ( α ) = f ( x )
n–∞
In practical applications, it is necessary to sample the continuous analog signals and
process the data obtained with discrete algorithms. The quasi-synchronous recursive
process mentioned above can be expressed as follows:
For Equation C-4, the integral interval [x0, x0 + n x (2π + Δ)] whose width is n x (2π + Δ)
can be divided equally into n x N sections, which results in n x N + 1 sampled data,
f(xi), (i=0,1,...,nxN), and we can iterate as follows:
© 2009 Microchip Technology Inc.
DS51723A-page 77
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
First iteration:
N
1
1 F 0 = -------------⋅
N
∑ ρi
∑ ρi ⋅ f ( xi )
i=0
i=0
N+1
1
1 ⋅
F 1 = --------------N+1
∑ ρi
∑ ρi ⋅ f ( xi )
i=1
i=1
…
N×n
F
1
(n – 1) × N
1
-⋅
= --------------------------------N×n
∑
∑
i = (n – 1) × N
ρi
ρi ⋅ f ( xi )
i = (n – 1 ) × N
Second iteration:
N
2
1 F 0 = -------------⋅
N
∑ ρi
∑ ρi ⋅ Fi
1
i=0
i=0
N+1
F1
2
1 = --------------⋅
N+1
∑ ρi
∑ ρi ⋅ Fi
1
i=1
i=1
…
N × (n – 1)
F
2
(n – 2) × N
1
-⋅
= --------------------------------N × (n – 1)
∑
∑
i = (n – 2) × N
ρi
ρi ⋅ Fi
1
i = (n – 2) × N
DS51723A-page 78
© 2009 Microchip Technology Inc.
Power Calculation Theory
Third iteration:
N
F0
3
1 = -------------⋅
N
∑ ρi
∑ ρi ⋅ Fi
2
i=0
i=0
N+1
F1
3
1 = --------------⋅
N+1
∑ ρi
∑ ρi ⋅ Fi
2
i=1
i=1
…
N × (n – 2)
F
3
(n – 3) × N
1
-⋅
= --------------------------------N × (n – 2)
∑
∑
i = (n – 3) × N
ρi
ρi ⋅ Fi
2
i = (n – 3) × N
…
© 2009 Microchip Technology Inc.
DS51723A-page 79
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
N-th iteration:
N
F0
n
1 = -------------⋅
N
∑ ρi
∑ ρi ⋅ Fi
n–1
i=0
i=0
Where ρi is the weight coefficient which is decided by the digital quadrature formula.
Complex rectangular quadrature algorithm or complex trapezoidal quadrature
algorithm is usually used in quasi-synchronous sampling.
Figure C-2 shows a 3-cycle interative process.
Original data
f0
f1
f2
......
First Iteration
F01 F11 F21
Second Iteration
F02 F12
Third Iteration
F03
FIGURE C-2:
fN
fN+1
fN+2
...... FN1 F1N+1 F1N+2
......
f2N
f2N+1
f2N+2 ......
f3N
...... F12N
F22 ...... FN2
3-Cycle Iterative Process.
In practical applicatons, a frequency offset Δ is usually small, and good results may
usually be obtained through 3-5 iterations.
As mentioned above, the iterative process will result in a group of weight coefficients
ηi, called weight coefficients of quasi-synchronous algorithm, they may be deduced
from the numeric quadrature formula. The relationship between the iterative result and
original data is shown in Equation C-9.
EQUATION C-9:
n×N
F0
n
1 = --------------⋅
n×N
∑ ηi
∑ ηi ⋅ f ( xi )
i=0
i=0
n×N
1
= -----n- ⋅
N
∑ ηi ⋅ f ( xi )
i=0
n×N
=
∑ Ri ⋅ f ( xi )
i=0
DS51723A-page 80
© 2009 Microchip Technology Inc.
Power Calculation Theory
Where:
EQUATION C-10:
1
R i = -----n- ⋅ η i
N
（i = 0 ~ n x N）
Equation C-10 is called the quasi-synchronous window function. After determining the
sampling points, the number of iterations and numerical quadrature method, the
coefficients of quasi-synchronous window function will become definite, and a
quasi-synchronous window function arrays may be established in advance.
Using the quasi-synchronous window function to carry out the weighted process of the
original data is equivalent to carrying out data synchronization one time, and the
algorithm realization is also very simple that only a multiplication of the original data and
quasi-synchronous window function arrays is required. After processing, the new
periodic signal will have the same period and frequency components as the original
signal, and the synchronization error of the new signal becomes smaller.
Figure C-3 is a schematic of the quasi-synchronous window function characteristics in
the time domain and data processing. In Figure C-3, the red curve is the characteristic
of the window function, and the blue curve is input signal, and the green curve is the
output signal.
Window function
Amplitude
Raw data
Data processed
Sampling point (1-193) Time 3.2 ksps
FIGURE C-3:
Quasi-Synchronous Window Function Characteristics Curve and Data
Processing.
© 2009 Microchip Technology Inc.
DS51723A-page 81
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.3
THE HARMONIC ANALYSIS ALGORITHM OF QUASI-SYNCHRONOUS
SAMPLING
Periodic signal can be expressed as trigonometric Fourier series or exponential Fourier
series. A periodic signal with a period of T can be expressed as :
EQUATION C-11:
α0
∞
f ( t ) = ------- +
2
∑
( α k cos ( k ⋅ ω ⋅ t ) + β cos ( k ⋅ ω ⋅ t ) )
k=1
where:
EQUATION C-12:
T
2
a k = --- ⋅ ∫ f ( t ) ⋅ cos ( k ⋅ ω ⋅ t )dt
T
0
EQUATION C-13:
T
2
b k = --- ⋅ ∫ f ( t ) ⋅ sin ( k ⋅ ω ⋅ t ) dt
T
0
or as:
EQUATION C-14:
∞
f ( t ) = a0 +
∑ ck sin ( k ⋅ ω ⋅ t + ϕk )
k=1
where the relationship between ak, bk， ck and ϕk is:
EQUATION C-15:
ck =
2
2
ak + bk
EQUATION C-16:
a
bk
ϕ k = atan ----k-
EQUATION C-17:
a k = c k sin ϕ k
DS51723A-page 82
© 2009 Microchip Technology Inc.
Power Calculation Theory
EQUATION C-18:
b k = c k cos ϕ k
Make g(t) = f(t) • cos(K•ω•t), it can be proved that g(t) is also a periondic function with T
as its period. Averaging g(t) in the range of [0 ~ T] results in:
EQUATION C-19:
T
1
g ( t ) = --- ⋅ ∫ f ( t ) ⋅ cos ( k ⋅ ω ⋅ t )dt
T
0
So ak = 2 × g(t). Therefore, ak can be obtained if only g(t) can be calculated.
EQUATION C-20:
n×N
2
a k = 2g ( t ) = -----n- ⋅
N
∑ ηi ⋅ gi
i=0
n×N
2
= ------n ⋅
N
- ⋅ i⎞
∑ ηi ⋅ fi ⋅ cos ⎛⎝ k ⋅ ----N ⎠
2π
i=0
EQUATION C-21:
n×N
2
b k = -----n- ⋅
N
2π
- ⋅ i⎞
∑ ηi ⋅ fi ⋅ sin ⎛⎝ k ⋅ ----N ⎠
i=0
Where N, n and ηi are constants. The equation may therefore be written as:
EQUATION C-22:
n×N
ak =
2π
- ⋅ i⎞
∑ Ii ⋅ fi ⋅ cos ⎛⎝ k ⋅ ----N ⎠
i=0
EQUATION C-23:
n×N
bk =
2π
∑ Ii ⋅ fi ⋅ sin ⎛⎝ k ⋅ -----N- ⋅ i⎞⎠
i=0
Where:
EQUATION C-24:
2
I i = -----n- ⋅ η 1 = R 1 × 2
N
© 2009 Microchip Technology Inc.
(i = 0 ∼ n × N)
DS51723A-page 83
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.4
MEASURING THE VOLTAGE/CURRENT RMS VALUE AND POWER USING
QUASI-SYNCHRONOUS SAMPLING ALGORITHM
From Equation C-11, a periodic voltage can be expressed as:
EQUATION C-25:
U a0
U ( t ) = ----------+
2
∞
∑ ( uak cos ( k ⋅ ω ⋅ t ) + ubk sin ( k ⋅ ω ⋅ t ) )
k=1
So the voltage fundamental and the voltage of each harmonic can be expressed as:
EQUATION C-26:
U k ( t ) = u ak cos ( k ⋅ ω ⋅ t ) + u k sin ( k ⋅ ω ⋅ t )
From Equation C-25, the fundamental voltage and voltage of each harmonic can also
be expressed as:
EQUATION C-27:
U k ( t ) = u ck sin ( k ⋅ ω ⋅ t + ϕ uk )
Where:
EQUATION C-28:
u ck =
2
u ak + u bk
2
EQUATION C-29:
u
u bk
ak
ϕ uk = atan -------
The voltage RMS value of each harmonic, then can be expressed as shown in
Equation C-31 with its initial phase angle shown in Equation C-29.
EQUATION C-30:
1
U k = ------- ⋅ u ck =
2
DS51723A-page 84
2
2
u ak + u bk
---------------------------2
© 2009 Microchip Technology Inc.
Power Calculation Theory
The relationship between uak, ubk and Uk can be expressed as:
EQUATION C-31:
u ak =
2 ⋅ U k ⋅ sin ϕ uk
u bk =
2 ⋅ U k ⋅ cos ϕ uk
EQUATION C-32:
Total effective voltage can be expressed as:
EQUATION C-33:
∞
U total =
∑ Uk
2
k=0
Similarly, the effective values and initial phase angles of fundamental current and
current of each other harmonic can be expressed as:
EQUATION C-34:
2
2
i ak + i bk
------------------------2
1
I k = ------- ⋅ i ck =
2
EQUATION C-35:
i
i bk
ak
ϕ ik = atan -----
The relationship between iak, ibk and Ik can be expressed as:
EQUATION C-36:
i ak =
2 ⋅ I k ⋅ sin ϕ ik
i bk =
2 ⋅ I k ⋅ cos ϕ ik
EQUATION C-37:
Total current RMS can be expressed as:
EQUATION C-38:
∞
I total =
∑ Ik
2
k=0
© 2009 Microchip Technology Inc.
DS51723A-page 85
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
According to the definition of power measurement, the active power and reactive power
of the fundamental and each other harmonic can be expressed as:
EQUATION C-39:
P k = U k I k cos ( ϕ uk – ϕ ik )
1
= --- ⋅ U k I k ( sin ϕ uk sin ϕ ik + cos ϕ uk cos ϕ ik )
2
EQUATION C-40:
Q k = U k I k sin ( ϕ uk – ϕ ik )
1
= --- ⋅ U k I k ( sin ϕ uk cos ϕ ik – cos ϕ uk sin ϕ ik )
2
Substituting Equation C-31, C-32, C-33 and C-37 into Equation C-39 and C-40, the
power of each harmonic component can be obtained with the following:
EQUATION C-41:
1
P k = --- ⋅ ( u ak i ak + u bk i ik )
2
EQUATION C-42:
1
Q k = --- ⋅ ( u ak i ak – u bk i ik )
2
Total active power and reactive power can be expressed as:
EQUATION C-43:
∞
P total =
∑ Pk
k=0
EQUATION C-44:
∞
Q total =
∑ Qk
k=0
DS51723A-page 86
© 2009 Microchip Technology Inc.
Power Calculation Theory
C.5
MEASURING FREQUENCY
There are many ways to measure frequency, with the most common being counting the
signal cycle. In this method, a counter increments each time a zero-crossing is
detected. Based on the counts, the width of a cycle can be measured. If the zero-crossing is accurate and the counter precision is high enough, cycle counting can be a
simple and practical method. But if the input signal has large harmonic components,
causing distortion around zero-crossing, then this approach may produce large errors.
Another method is to analyze and process the sampled data and calculate the frequencies. Analysis may be carried out in time domain, such as digital differential ND and
interpolation method; or may be carried out in frequency domain after DFT transformation, such as gravity center method, spectrum zoom method and phase difference
method, among which the phase difference method is the most common one. It is not
sensitive to signal distortion around zero-crossing points.
The basic idea of the phase difference method is: if the rough range of to-be-measured
signal frequency is known, then we may assume a frequency that is close to the actual
frequency and then acquire an array of samples based on the assumed frequency. In
the sampled data, the phase of the 1st cycle and the subsequent N-th cycle are meaured and their difference may be calculated. Then the phase difference may be used to
calculate the difference between the actual freqency and the assumed frequence, thus
figuring out the actual frequency.
If the frequency f0 to be measured is known to be a definite value f, i.e., f0 = f + Δf,
Δf << f, then from Equation C-27, the fundamental signal can be expressed as:
EQUATION C-45:
U 1 ( t ) = u c1 sin ( ω ⋅ t + ϕ u1 ) = u c1 sin ( 2 π f 0 t + ϕ u1 )
If:
EQUATION C-46:
T = 1--f
EQUATION C-47:
T
2
u a = --- ⋅ ∫ U 1 ( t ) cos ( ω t ) dt
T
0
T
2
= --- ⋅ ∫ u c1 sin ( 2 π f 0 t + ϕ u1 ) cos ( 2 π ft ) dt
T
0
EQUATION C-48:
T
2
u b = --- ⋅ ∫ U 1 ( t ) sin ( ω t ) dt
T
0
T
2
= --- ⋅ ∫ u c1 sin ( 2 π f 0 t + ϕ u1 ) sin ( 2 π ft ) dt
T
0
© 2009 Microchip Technology Inc.
DS51723A-page 87
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
We get:
EQUATION C-49:
πΔ f
( f + Δ f ) ⋅ sin ( ϕ u1 ) ⋅ sin ⎛ ---------⎞
⎝ f ⎠
2u c1
u a = ----------- ⋅ ------------------------------------------------------------------------π ⋅ ( 2f + Δ f ) ⋅ Δ f
T
EQUATION C-50:
πΔ f
f ⋅ cos ( ϕ u1 ) ⋅ sin ⎛⎝ ---------⎞⎠
2u c1
f
u b = ----------- ⋅ -------------------------------------------------------T
π ⋅ ( 2f + Δ f ) ⋅ Δ f
As Δf << f, from Equation C-49 and C-50, we have:
EQUATION C-51:
u a sin ( ϕ u1 )
----- ≈ ---------------------u b cos ( ϕ u1 )
Therefore,
EQUATION C-52:
π
⎧
, (ub = 0, u a > 0)
⎪
2
⎪
3π
⎪
, (ub = 0, u a < 0)
2
⎪
−1 u a
⎪⎪
ϕ u1 ≈ ⎨ tg ( u ), (ub > 0, u a > 0)
b
⎪
⎪ tg −1 ( u a ) + π , (ub > 0, u a < 0)
ub
⎪
⎪ −1 u a
⎪tg ( ) + 2π , (ub < 0, u a > 0)
ub
⎩⎪
Assuming that the signal's initial phase angle measured in the 1st cycle is ϕ1 and in
the N-th is ϕN, then the difference between actual frequency and the assumed
frequency is:
EQUATION C-53:
⎧ (ϕ N − ϕ1 ) ⋅ f
, ( ϕ N − ϕ1 < π )
⎪
2π ⋅ N
⎪ (ϕ − ϕ + 2π ) ⋅ f
⎪
1
Δf ≈ ⎨ N
, (ϕ N − ϕ1 < −π )
2π ⋅ N
⎪
⎪ (ϕ N − ϕ1 − 2π ) ⋅ f , (ϕ − ϕ > π )
N
1
2π ⋅ N
⎩⎪
DS51723A-page 88
© 2009 Microchip Technology Inc.
Power Calculation Theory
C.6
IMPROVING MEASUREMENT PRECISION OF QUASI-SYNCHRONOUS
SAMPLING ALGORITHM
When using the quasi-synchronous sampling method for harmonic analysis and
calculation of power as well as voltage and current, strict restrictions apply for the
algorithm and compensation, (i.e., the frequency offset must not exceed 1% of the
central frequency). Precision of the result increases as the frequency offset gets less.
Measurement accuracy is not guaranteed, if this condition can not be met. Figure C-4
shows the quasi-synchronization algorithm using 3 iterations with input signal ranging
from 47.5 Hz to 52.5 Hz. The algorithm is for calculating the active power, the reactive
power and the relative error of current and voltage. Figure C-4 shows that the algorithm
works well when the frequency falls in the range of 47.5 Hz to 52.5 Hz. As the
frequency deviates from the range, the error increases significantly. Therefore, the
algorithm needs to be improved to fit into more applications with a more relaxed
restriction.
FIGURE C-4:
Quasi-sync Algorithm Error Analysis of 3 Iterations.
The quasi-sync sampling algorithm has relative high accuracy in frequency measurement and the error can be less than 0.005 Hz. If the frequency range to be measured
can be segmented to make the frequency input closest to the multiple of cycle point,
and processed using appropriate quasi-sync window function and sine/cosine tale,
then the algorithm can be used for a much wider range of frequency .
Figure C-5 is the error analysis of the improved 3-iteration quasi-synchronous
algorithm at 3.2 ksps. It shows that the relative error for each result can be well
controlled when the frequency of the input signal falls in the range of 47.5 Hz to
52.5 Hz.
Figure C-5 clearly shows that the relative errors of the current, voltage, active power
and reactive power in the entire frequency range are less than 0.08%. Also, when the
input frequency is around the multiple of cycle frequencies (52.459 Hz, 51.613 Hz,
50.794 Hz, 50.0 Hz, 49.231 Hz, 48.485 Hz and 47.761 Hz), the calculateion error is
© 2009 Microchip Technology Inc.
DS51723A-page 89
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
minimal (<0.01%). When the input frequency deviates from the multiple of cycle
frequencies, the calculation error increases rapidly. As the calculation error is related
to the frequency offset to the multiple of freqency point, the calculation error caused by
frequency offset can be corrected. Figure C-6 is the error analysis after frequency offset correction using simple parabolic interpolation. Calcultion errors for all parameters
are shown to be less than 0.015% after the correction.
DS51723A-page 90
FIGURE C-5:
Iterations.
Error Analysis Of Improved Quasi-sync Calculation Using 3
FIGURE C-6:
Compensation.
Calculation Error Analysis After The Frequency Offset
© 2009 Microchip Technology Inc.
Power Calculation Theory
C.7
MEASURING SECONDARY PARAMETERS
The methods of measuring parameters such as RMS values of voltage and current,
active power, reactive power and frequency have been discussed in previous sections.
These are primary parameters that need to be calculated from the original data. There
are some other parameters called secondary parameter, such as power factor of each
phase, total reactive power, total active power, total power factor, harmonic
components and cumulative energy. They are obtained indirectly from primary parameters.
The measurement of secondary parameters is discussed in this section.
C.7.0.1
TOTAL ACTIVE POWER AND TOTAL REACTIVE POWER
For 3-phase 4-wire systems, 3-phase total active power and recative power are the
sum of power of each phase, respectively, which can be expressed as:
EQUATION C-54:
P = PA + P B + PC
EQUATION C-55:
Q = QA + QB + QC
C.8
APPARENT POWER OF EACH PHASE AND TOTAL APPARENT POWER
Apparent power is defined as:
EQUATION C-56:
S =
C.9
2
Q +p
2
POWER FACTOR OF EACH PHASE AND TOTAL POWER FACTOR
Power factor is defined as the ratio of active power to apparent power. The definition
can be represented as shown in Equation C-57:
EQUATION C-57:
P PF = ----------------------2
2
P +Q
© 2009 Microchip Technology Inc.
DS51723A-page 91
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.10 Active Energy AND REACTIVE ENERGY
Active energy is defined as the integral of active power over time, which is:
EQUATION C-58:
N
T
W =
∫0 P ( t ) dt
∑ u(k ) ⋅ i( k) ⋅ Δt
=
k=0
In this design, active energy is obtained from multiplying the voltage by the current
sampled each time. The phase angle difference is compensated after each power
measurement is completed.
For reactive power, the cumulative reactive energy over a time period can be calculated
by measuring the average power and calculating the time interval between 2 measurements.
EQUATION C-59:
T
Vr =
∫0 Q ( t )dt
C.11 POSITIVE/NEGATIVE ACTIVE ENERGY, POSITIVE/NEGATIVE REACTIVE
ENERGY AND FOUR-QUADRANT REACTIVE ENERGY
In the measurement plane, the horizontal axis denotes voltage vector U (fixed on the
horizontal axis). The instantaneous current vector is used to represent the power
transfer, and has a phase angle φ against vector U. φ is positive in counter-clockwise
direction. Power exchange can be defined in 4 scenarios:
• Quadrant I (P>0, Q>0): active energy and reactive energy are sent out at the
same time;
• Quadrant II (P<0, Q>0): active energy is sent in while reactive energy is sent out;
• Quadrant III (P<0, Q<0): active energy is sent in while reactive energy is
absorbed;
• Quadrant IV (P>0, Q<0): active energy is sent out while the reactive energy is
absorbed.
1. Positive active energy and negative active energy: accumulated active
energy can be defined as positive and negative depending on the direction of
active current. When the direction of active current is positive (from power grid to
loads), active energy is positive (where active power P>0, corresponding to
quadrants I and IV, which means that loads are drawing energy from grid). When
current moves from loads to power grid, it is defined as negative active energy
(where active power P < 0, corresponding to quadrants II and III, which means
energy is provided to grid). Usually only positive active energy is taken into
account in active energy, but in practice negative active energy may be taken into
account as well, if necessary.
2. Positive reactive energy and negative reactive energy: If reactive power
Q > 0 (corresponding to quadrants I and II), it means power grid is providing
reactive energy to loads, so the energy is defined as positive reactive energy.
When reactive power Q < 0 (corresponding to quadrants III and IV), it means
that loads are providing reactive energy to power grid, so the energy is defined
as negative reactive energy.
DS51723A-page 92
© 2009 Microchip Technology Inc.
Power Calculation Theory
3. Four-quadrant reactive energy: metering reactive energy in positive/negative
reactive energy cannot truly reflect the status of reactive energy, whereas
4-quandrant reactive energy measuring gives a true picture of energy exchange.
Reactive energy in four quadrants represents four different reactive energy (see
Figure C-7). And the reactive energy is accumulated depending on which
quadrant it is located.
Reactive in (+R)
电流向量
I (RL)
II (RC)
Current vector
Φ
Active in (+A)
Active out (-A)
Voltage
vector
电压向量
III (-RL)
IV (-RC)
Reactive out (-R)
Where:
A－有功电能；R—无功电能；R —感性无功电能；R —容性无功电能
L
=
is active energy,
R
=
is reactive energy
RL
=
is inductive reactive energy
RC
=
s capacitive reactive energy
图2.4
FIGURE C-7:
© 2009 Microchip Technology Inc.
C
A
电能测量四象限定义
Definition Of 4 Quadrants To Measure Electrical Energy.
DS51723A-page 93
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.12 HARMONIC COMPONENTS OF CURRENT, VOLTAGE AND TOTAL
HARMONIC DISTORTION
In Section C.3 “The Harmonic Analysis Algorithm Of Quasi-synchronous Sampling”, we discussed the measuring of current and voltage signals for each order of
harmonics. 3 parameters are used to show to what extent a distorted wave deviates
from a sine wave. They are: harmonic content, total distortion and harmonic ratio of the
k-th harmonic. The term harmonic content means the root of square of effective values
for all harmonics, which is defined as:
EQUATION C-60:
N
UH =
∑ Uk 2
k=2
The total voltage distortion ratio of harmonics is the ratio of harmonic content to the
fundamental in percentage, which is defined as:
EQUATION C-61:
THD U
UH
= ------- × 100%
UI
N
∑ ( THDUk )
2
k=2
The k-th harmonic ratio for voltage is the ratio of the k-th harmonic to the fundamental
in percentage, defined as:
EQUATION C-62:
Uk
THD Uk = ------ × 100%
U1
Similarly, harmonics of each order for the current and total distortion ratio can be sorted
out.
DS51723A-page 94
© 2009 Microchip Technology Inc.
Power Calculation Theory
C.13 COMPENSATION FOR RATIO ERROR AND PHASE LAG
The error of the current transformer (CT) is a complex, which can usually be expressed
by two orthogonal parts, current error (f) and phase lag (δ).
EQUATION C-63:
ε = f+j⋅δ
The current error, also known as ratio error, can be written in percentage as:
EQUATION C-64:
( kn I2 – I1 )
f = 100 ------------------------I1
Where:
Kn
=
I1
=
I2
=
the rated current ratio
the primary current
the secondary current that passes I1
under the test condition
Phase lag, also known as angle error, is the phase difference between primary and
secondary current vector, in 'minute'.
Different current transformers have different errors. Current transformers are classified
into different accuracy classes depending on their error magnitudes. The accuracy
class of a transformer is nominated by the percentage of the maximal ratio error
allowed for a given rated current.
Accuracy classes and corresponding error limits for a current transformer are listed in
Table C-1.
Accurate
Class
TABLE C-1:
ACCURACY CLASS AND ERROR LIMIT FOR THE CURRENT TRANSFORMER
Ratio Error± (%)
Phase lag
Rated Current (%)
1
± (Grad)
Rated Current (%)
5
20
100
120
0.1
0.4
0.2
0.1
0.1
0.2
0.75
0.35
0.2
0.2
30
15
10
10
0.9
0.45
0.3
0.3
0.5
1.5
0.75
0.5
0.5
90
45
30
30
2.7
1.35
0.9
0.9
3
1.5
1.0
1.0
180
90
60
60
5.4
2.7
1.8
1.8
1
Rated Current (%)
50
120
3
3
3
5
5
5
© 2009 Microchip Technology Inc.
1
± (%)
Rated Current (%)
5
20
100
120
15
8
5
5
1
5
20
100
120
0.45
0.24
0.15
0.15
Phase difference
Phase difference of class 3 and class 5 are not specified.
DS51723A-page 95
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.14 RELATIONSHIP BETWEEN ERROR AND CURRENT
For a given load and frequency, the absolute ratio error and angle error increase when
the primary current decreases from the rated value for un-compensated current transformer. The reason is that with the decrease of the secondary current, the magnetic
permeability µ of the iron core varies non-linearly, resulting in less reduction in the field
ampere turns.
Figure C-8 is a typical curve for load current and the phase lag of a CT. Generally,
phase lag of the output signal is great when the load current is small, and also
increases at a fast rate.
Phase Lag
Current Load
FIGURE C-8:
DS51723A-page 96
Current Load Versus CT Phase Lag.
© 2009 Microchip Technology Inc.
Power Calculation Theory
C.15 RATIO ERROR COMPENSATION
As non-linearity and inconsistency exist in the sampling circuit (including CT and
back-end shunt resistors) and the ADC circuit, it is necessary to compensate for ratio
error to the system. Figure C-9 is the transfer link of the voltage and current channels.
The compensation for ratio error is quite simple. It compares the measured value
against the actual input value under certain input conditions and obtains a correction
coefficient.
EQUATION C-65:
Calibration Value
Correction coefficient = Coefficient before correction × ------------------------------------------Measured Value
Voltage secondary
Sampling resistor
Input
Voltage
R0
R0
Voltage primary
Sampling resistor
CT
1:1
ADC
R0
R2
Input
Current
CT
2000:1
ADC
R2
Current secondary
Sampling resistor
FIGURE C-9:
Error Caused By Sampling Circuit And ADC.
Since current has a large dynamic range, for a meter which requires high accuracy
(0.2s and 0.5s), the multi-point calibration method is needed to meet input requirement
for full range. The MCP3909 device's current channel includes an adjustable gain
amplifier. The ratio error must be recalibrated for different amplification, but only needs
to be calibrated once under the same amplification conditions.
© 2009 Microchip Technology Inc.
DS51723A-page 97
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
C.16 PHASE LAG COMPENSATION
Phase lag of a CT has no effect on the metering of RMS current/voltage and apparent
power, but will affect the metering of power, since the phase lag will change the phase
relationship between the input current and the voltage. This will result in a deviation of
the calculated active power from the calculated reactive power.
Figure C-10 shows how a transformer's phase lag affects the measured results under
both inductive and capacitive loads. Let's assume that the output of CT has no phase
lag from the input voltage, while the CT has a phase lag from the input signal. With
inductive loads, the phase angle increases between the volatge and the current
because of the phase lag induced by the CT, resulting in a decrease of the measured
active power and an increase of the reactive power. While with capacitive loads, the
phase angle between the voltage and the current decreases because of the phase lag
induced by the CT, resulting in a decrease of the measured reactive power and an
increase of active power.
There are many methods for phase lag compensation. In this design the result correction method is used. It compensates with a coefficient after the active power and
reactive power are figured out, which has a small amount of calculation.
Inductive
Load
Inductive
Load
CT Output
current
Input
voltage
Input
current
Δθ
θ
CT Output
current
θ
Δθ
Input
voltage
Input
current
Capacitive
Load
Capacitive
Load
FIGURE C-10:
Measurement Change Caused By Transformer Phase Lag.
Assuming that the phase lag of CT is ϕi, of PT is ϕu, after PT and CT, the variation of
phase lag between current and voltage is:Δϕ = ϕu – ϕi.
S
Q'
Q
Δf
F
P'
P
FIGURE C-11:
DS51723A-page 98
Principle Of Phase Lag Correction.
© 2009 Microchip Technology Inc.
Power Calculation Theory
Given that the phase lag between the input voltage and current is φ, after PT and CT,
the actual measured active power is P', reactive power is Q'. RMS current is I, RMS
voltage is V, input apparent power is S, actual input active power is P and reactive
power is Q, then Figure C-11 can be drawn based on the principles of power triangle.
EQUATION C-66:
P' = V ⋅ I ⋅ cos ( φ + Δϕ ) = P cos Δϕ – Q sin Δϕ
EQUATION C-67:
Q' = V ⋅ I ⋅ sin ( φ + Δϕ ) = Q cos Δϕ – P sin Δϕ
From the above 2 equations, we have:
EQUATION C-68:
P = k 1 P' + k 2 Q'
EQUATION C-69:
Q = k 1 Q' – k 2 P'
Where:
EQUATION C-70:
k 1 = cos Δϕ
EQUATION C-71:
k 1 = sin Δ ϕ
By setting up certain input conditions, Δϕ can be measured, and then k1 and k2 can be
calculated.
In this design, we use an input of 0.5L for calibration. With this condition, Δϕ can be
calculated using the difference between the meaured and the actual input value of
active power. For accurate calculations, the user may also use the difference between
the measured cumulative energy and the actual cumulative energy to calculate the
difference.
EQUATION C-72:
P'
Δϕ = a cos ⎛⎝ -----------⎞⎠ – --π- = a cos ⎛⎝ ( 0.5 ⋅ ( 1 + err ) ) – --π-⎞⎠
2⋅P
3
3
Where err is the error rate of the energy measurement, which results from calculating
the error between the actual energy measured by a standard meter and the energy
measured by the dsPIC devices. The error can be obtained from the output of the meter
calibration workbench.
EQUATION C-73:
– P- = Δ
------P- × 100
err = P'
------------P
P
© 2009 Microchip Technology Inc.
DS51723A-page 99
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
Since the phase lag of a CT's output signal is related to the magnitude of current,
different correction coefficients, K, can be set according to different RMS current
values. In this design, 5 calibration points can be set. If it does not require high-precision, fewer points can be set to simplify calibration.
If one-time calibration cannot meet the precision requirement, more calibrations can be
done. The new angle error may still be calculated using Equation C-72. The new
correction coefficient is:
EQUATION C-74:
k' 1 = cos ( Δϕ 1 + Δϕ 2 ) = k 1 ⋅ cos Δϕ 2 – k 2 ⋅ sin Δϕ 2
EQUATION C-75:
k' 2 = sin ( Δϕ 1 + Δϕ 2 ) = k 2 ⋅ cos Δϕ 2 – k 1 ⋅ sin Δϕ 2
C.16.0.1 PHASE LAG COMPENSATION WHEN FREQUENCY VARIES
For the same current intensity, the signal delay caused by the CT is the same. When
the frequency of the input signal varies, the phase lag will be different. Normally,
calibration is done at 50 Hz. When the frequency varies, if the same phase lag
compensation coefficient for 50 Hz is still used, it will cause an error in the power
measurement. In most cases, the frequency varies in a small range (test specification
requires ±2.5%), so it has little effect on the measurement accuracy. For meters with
an accuracy of 0.5s or above, this measurement error can be ignored. But for 0.2s
meters, the error cannot be ignored and the frequency variation needs to be corrected
during calculation.
The phase lag compensation coefficient k1 and k2 are corrected during calculation.
Assuming that the freqnency is 50 Hz, the signal delay caused by CT is t, then after
correction, the compensation coefficient k1 and k2 will be:
EQUATION C-76:
k 1 = cos Δϕ = cos ( t ⋅ 50 ⋅ 2 π )
EQUATION C-77:
k 2 = sin Δ ϕ = sin ( t ⋅ 50 ⋅ 2 π )
When frequency varies, assuming that the frequency offset is Δf, i.e. the input signal
frequency is 50 + Δf, then the compensation coefficient will be:
EQUATION C-78:
k' 1 = cos Δϕ = cos ( t ⋅ ( 50 + Δ f ) ⋅ 2 π )
EQUATION C-79:
k' 2 = sin Δ ϕ = sin ( t ⋅ ( 50 + Δ f ) ⋅ 2 π )
DS51723A-page 100
© 2009 Microchip Technology Inc.
Power Calculation Theory
To avoid complexity in calculation and to maximize the correction precision, the
following equations may be used to approximate K1 and K2 when the phase angle is
small.
EQUATION C-80:
Δf
2
k' 1 = cos Δϕ ≈ cos Δϕ – ------ ⋅ sin Δϕ
50
Δf
= k 1 – ------ ⋅ k 2 ⋅ k 2
50
EQUATION C-81:
Δf
k' 2 = sin Δ ϕ ≈ sin Δ ϕ – ------ ⋅ sin Δϕ ⋅ cos Δϕ
50
Δf
= k 2 – ------ ⋅ k 1 ⋅ k 2
50
© 2009 Microchip Technology Inc.
DS51723A-page 101
MCP3909 / dsPIC33F 3-Phase Energy Meter Reference Design
NOTES:
DS51723A-page 102
© 2009 Microchip Technology Inc.
MCP3909 / DSPIC33F 3-PHASE
ENERGY METER REFERENCE DESIGN
Appendix D. 50/60 Hz Meter Operation
D.1
FIRMWARE VERSIONS
There are two versions of firmware for the meter due to the quasi-synchronous
sampling scheme employed by the dsPIC33F firmware. The design covers the
frequency of rated frequency ±2.5 Hz.
To change the rated frequency, just change the definition in the beginning of
cacul.h - #define STD_FREQ 50.0.
This is provided for download from Microchip’s website, file names and checksums
below.
At the same time you need to change the crystal to provide clock for metering IC.
TABLE D-1:
FIRMWARE FILES
Line Frequency
Firmware Name
Hex File Checksum
YXX VALUE
50 Hz
60 Hz
PM_1_50.ZIP
PM_1_60.ZIP
TBD
TBD
3.857 MHz
TBD
© 2009 Microchip Technology Inc.
DS51723A-page 103
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DS51723A-page 104
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