PD-97392 IR3870MBF SupIRBuck TM 10A HIGHLY INTEGRATED WIDE-INPUT VOLTAGE, SYNCHRONOUS BUCK REGULATOR Features Description • • • • • • The IR3870M SupIRBuckTM is an easy-to-use, fully integrated and highly efficient DC/DC voltage regulator. The onboard constant on time hysteretic controller and MOSFETs make IR3870 a space-efficient solution that delivers up to 10A of precisely controlled output voltage in 60°C ambient temperature applications without airflow. • • • • • • • • • Input Voltage Range: 3V to 26V Output Voltage Range: 0.5V to 12V Continuous 10A Load Capability Constant On-Time control Excellent Efficiency at very low output current levels Gate drive charge pump option to maximize efficiency at higher output current levels Compensation Loop not Required Programmable switching frequency, soft start, and over current protection Power Good Output Precision Voltage Reference (0.5V, +/-1%) Enable Input with Voltage Monitoring Capability Pre-bias Start Up Under/Over Voltage Fault Protection Ultra small, low profile 5 x 6mm QFN Package Lead-free, halogen-free and RoHS compliant Applications • • • • Notebook and desktop computers Game consoles Consumer electronics - STB, LCD TV, Printers General purpose POL DC-DC Converters Programmable switching frequency, soft start, and over current protection allows for a very flexible solution suitable for many different applications. The combination of the gate drive charge pump option and constant on time control allow efficiency optimization in the whole output current range, making this device an ideal choice for battery powered applications. Additional features include pre-bias startup, very precise 0.5V reference, over/under voltage shut down, power good output, and enable input with voltage monitoring capability. Typical Notebook Application Circuit Diagram 1/27/2010 1 IR3870MBF ABSOLUTE MAXIMUM RATINGS (Voltages referenced to GND unless otherwise specified) • VIN ……………………………………………………. 30V • VCC ……………….….………………………………. 3.9V • PVCC ……………….….…………….……..………… 7.5V • Boot • PHASE …………………………………………..…….. -0.3V to 30V(DC), -5V(100ns) • Boot to PHASE…..…………………………….…..…… 7.5V • FF • PGND to GND ……………...…………………………. -0.3V to +0.3V • All other pins ……………...…………………………… 3.9V • Storage Temperature Range ..................................... -65°C To 150°C • Junction Temperature Range .................................... -10°C To 150°C • ESD Classification …………………………….……… JEDEC Class 1C • Moisture sensitivity level………………...…………….. JEDEC Level 3@260 °C ……………………………………..……….….. 40V ………………………………………….………… 30V Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications are not implied. PACKAGE INFORMATION 5mm x 6mm POWER QFN θJA = 35 o C / W θJ -PCB = 2 o C / W ORDERING INFORMATION PKG DESIG PACKAGE DESCRIPTION PIN COUNT PARTS PER REEL M IR3870MTRPbF 23 4000 M IR3870MTR1PbF 23 750 2 IR3870MBF Block Diagram 3 IR3870MBF Pin Description NAME NUMBER I/O LEVEL DESCRIPTION NC 1 No connection ISET 2 Connecting resistor to PHASE pin sets over current trip point PGOOD 3 GND 4,17 5V Power good – pull up to 5V Reference Bias return and signal reference FB 5 3.3V Inverting pin of PWM comparator and OVP/PGood sense SS 6 3.3V Set soft start slew-rate with a capacitor to GND NC 7 VCC 8 3.3V Internal bias supply CPO 9 0.75V Charge pump output PVCC 10 5V Gate drive supply PGND 11 Reference Power return PHASE 12 VIN Phase node (or switching node) of MOSFET half bridge VIN 13 VIN Input voltage for the system. BOOT 14 VIN +PVCC Bootstrapped gate drive supply – connect a capacitor to PHASE FF 15 VIN Input voltage feed forward – sets on-time with a resistor to VIN EN 16 5V Enable – turns device on or off No connection 4 IR3870MBF Recommended Operating Conditions Symbol VIN VCC Boot to PHASE VOUT IOUT Fs Definition Input Voltage Supply Voltage Supply Voltage Output Voltage Output Current Switching Frequency Min Max Units 3 3 26* 3.6 7.0 V 0.5 0 12 10 1000 A kHz * Note: PHASE pin must not exceed 30V. Electrical Specifications Unless otherwise specified, these specification apply over VIN = 12V, PVCC = 7VDC, 0oC ≤ TJ ≤ 125oC. PARAMETER BIAS SUPPLIES VCC Turn-on Threshold NOTE TEST CONDITION MIN TYP UNIT 3 V V 2.65 VCC Turn-off Threshold 60 VCC Threshold Hysteresis mV 3.05 PVCC Turn-on Threshold V 2.65 PVCC Turn-off Threshold V 60 PVCC Threshold Hysteresis VCC Shutdown Current VCC Operating Current PVCC Operating Current PVCC Shutdown Current FF Shutdown Current VIN Shutdown Current PHASE Shutdown Current CONTROL LOOP Reference Accuracy, VREF On-Time Accuracy EN=LOW 30 RFF = 180kΩ, EN = HIGH RFF = 180kΩ, EN = HIGH, Fs = 500kHz EN=LOW EN=LOW EN=LOW EN=LOW 1.2 8 0 RFF = 180 kΩ, TJ = 65 C mV μA 60 mA mA 20 0.1 0 0.25 60 2 1 2 μA μA μA μA 0.495 0.5 0.505 V 280 300 320 ns Min Off Time 300 Soft-Start Current FAULT PROTECTION ISET pin output current Under Voltage Threshold Under Voltage Hysteresis Over Voltage Threshold Over Voltage Hysteresis PGOOD Delay Threshold (VSS) MAX EN = HIGH 1 Falling VFB & monitor PGOOD Rising VFB Rising VFB & monitor PGOOD Rising VFB ns 8 10 12 μA 18 0.37 20 0.4 7.5 0.6 7.5 0.6 22 0.43 μA V mV V mV V 0.56 0.64 5 IR3870MBF Electrical Specifications (continued) Unless otherwise specified, these specification apply over VIN = 12V, PVCC = 7VDC, 0oC ≤ TJ ≤ 125oC. PARAMETER GATE DRIVE Dead time NOTE 1 UPPER MOSFET Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Diode Forward Voltage LOWER MOSFET Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Static Drain-to-Source OnResistance Diode Forward Voltage CHARGE PUMP OUTPUT Source Resistance Sink Resistance Charge Pump Disable Threshold, VCP TH LOGIC INPUT AND OUTPUT EN Rising Threshold EN Hysteresis EN Input Current PGOOD pull down resistance TEST CONDITION MIN Monitor body diode conduction on PHASE pin TYP 5 o PVCC =5V, ID =10A, TJ =25 C UNIT 30 ns 14.7 o PVCC =5V, ID =10A, TJ =125 C mΩ 26 o PVCC =7V, ID =10A, TJ =25 C mΩ 11.9 o PVCC =7V, ID =10A, TJ =125 C o VGS =0V, IS =10A, TJ =25 C o PVCC =5V, ID =10A, TJ =25 C mΩ 22 mΩ 1.0 V 6.2 o PVCC =5V, ID =10A, TJ =125 C mΩ 11 o PVCC =7V, ID =10A, TJ =25 C mΩ 5.1 o PVCC =7V, ID =10A, TJ =125 C mΩ 9 mΩ 1.0 V 5 2.1 6.7 3.3 1 7.2 Ω Ω V 1.13 40 1.21 100 1.29 160 1 100 V mV o VGS =0V, IS =10A, TJ =25 C ICPO =15mA ICPO =15mA IPGOOD =2mA MAX 50 μA Ω Notoe1: Guaranteed by design, not tested in production 6 IR3870MBF TYPICAL OPERATING DATA (25oC) (Circuit of Figure 15, VCC = 3.3V, V5 = 5V, VIN = 12.6V, F = 500kHz Unless otherwise noted) 336 1000000 1000 334 Rff( 0.5, fsw) 332 Feedforward Resistance (KΩ) Rff( 1.5, fsw) On-Tim e (nS) Feedforward Resistance Rff( 1, fsw) Rff( 2, fsw) Rff( 2.5, fsw) VOUT = 5V Rff( 3, fsw) Rff( 3.5, fsw) Rff( 4, fsw) 330 328 326 Rff( 4.5, fsw) Rff( 5, fsw) 324 V 000000 OUT 200000 0 200 322 = 0.5V fsw Switching Frequency 1000000 0 1000 50 75 100 125 Temperature (C) Switching Frequency (KHz) Figure 3. On-Time Variation in DCM: VIN = 12V, VOUT =1.1V, RFF = 180kOhm Figure 1. Feedforward Resistance Vs Switching Frequency: 0.5V VOUT step 500 315 450 314.5 400 314 350 On-Tim e (nS) S w itc h in g F re q u e n c y (k H z ) 25 300 250 200 150 313.5 313 312.5 100 312 50 311.5 0 0 2 4 6 8 10 12 Output Current (A) Figure 2. Switching Frequency Vs Output Current 311 0 25 50 75 100 125 Temperature (C) Figure 4. On-Time Variation in CCM: VIN = 12V, VOUT =1.1V, RFF = 180kOhm 7 IR3870MBF TYPICAL OPERATING DATA (25oC) (Circuit of Figure 15, VCC = 3.3V, V5 = 5V, VIN = 12.6V, F = 500kHz Unless otherwise noted) Eff: 12.6Vin 1.11400 Eff: 19Vin Ploss: 12.6Vin Ploss: 19Vin 88 0.5 0.5 87 0.4 1.10400 1.09900 1.09400 0.4 85 0.3 84 0.3 0.2 83 0.2 19Vin 82 12.6Vin 81 0.1 1.08900 2 4 6 8 10 0.1 80 12 0.0 0.2 0.4 Output Current (A) Eff: 19Vin Ploss: 12.6Vin 89 3.0 86 2.0 85 1.5 84 83 1.0 82 0.5 81 80 0.0 8 10 Output Current(A) Figure 6. System Efficiency: VOUT = 1.1V 12 VIN Power Loss (W) 2.5 87 Pow er Loss (W ) Efficiency (%) 88 6 1.2 1.4 0.0 1.8 1.6 Ploss: 19Vin 3.5 4 1.0 Figure 7. Light load Efficiency: VOUT = 1.1V 90 2 0.8 Output Current(A) Figure 5. Output Voltage Regulation Vs Output Current Eff: 12.6Vin 0.6 16 0.080 14 0.070 12 0.060 10 0.050 8 0.040 6 0.030 4 0.020 Vin Ploss 5V PVcc Ploss 3.3V Vcc Ploss 2 0.010 0 0.000 0 2 4 6 8 10 12 Output Current(A) Figure 8. Input Power Vs Output Current 8 PVCC and VCC Power Loss (W) 0 0 P o w e r L o s s (W ) 86 E fficiency (% ) O u tp u t Vo ltag e (V) 1.10900 IR3870MBF TYPICAL OPERATING WAVEFORM (25oC) (Circuit of Figure 15, VCC = 3.3V, V5 = 5V, VIN = 12.6V, F = 500kHz, Unless otherwise noted) CH1: Vout (500mV/div); 500uS/div CH2: PHASE (10V/div) CH3: EN (2V/div) CH4: PGOOD (5V/div) Figure 9: Shutdown/EN at IOUT = 500mA CH1: Vout (50mV/div); 20uS/div CH2: PHASE (10V/div) Figure 11: FCCM/CCM transition from 0.5A to 5A at 12.6VIN: 30mV overshoot, 15mV undershoot CH1: Vout (50mV/div); 20uS/div CH2: Phase (10V/div) CH1: Vout (50mV/div); CH2: Phase (10V/div) CH4: CPO (2V/div); Time: 2uS/div) Figure 10: Load Step (2A to 10A) Transient (5A/us) at 12.6VIN: 50mV overshoot, 25mV undershoot Figure 12: Charge Pump Off at IOUT = 1A 9 IR3870MBF TYPICAL OPERATING WAVEFORM con’t (25oC) (Circuit of Figure 15, VCC = 3.3V, V5 = 5V, VIN = 12.6V, F = 500kHz, Unless otherwise noted) CH1: Vout (50mV/div); CH2: Phase (10V/div) CH4: CPO (2V/div); Time: 2uS/div) Figure 13: Charge Pump On at IOUT = 3A IC: 750C, Inductor: 530C, PCB: 540C Figure 14: Thermal Image: VIN = 19V, IOUT = 10A, Ta= 250C, no air flow 10 IR3870MBF TYPICAL APPLICATION CIRCUIT WITH CHARGE PUMP OPTION Figure 15. Application Circuit with Charge Pump used to boost Sync FET gate drive voltage from 5V to ~7V, resulting in improved efficiency at higher output current levels F = 500kHz, VOUT = 1.1V 11 IR3870MBF Circuit Description PWM COMPARATOR The PWM comparator initiates a SET signal (PWM pulse) when the FB pin falls below the reference (VREF) or the soft start (SS) voltage. ON-TIME GENERATOR The PWM comparator initiates a SET signal (PWM pulse) when the FB pin falls below the reference (VREF) or the soft start (SS) voltage. The PWM on-time duration is programmed with an external resistor (RFF) from the input supply (VIN) to the FF pin. The simplified equation for RFF is shown in equation 1. The FF pin is held to an internal reference after EN goes HIGH. A copy of the current in RFF charges a timing capacitor, which sets the on-time duration, as shown in equation 2. VOUT RFF = (1) 1V ⋅ 20 pF ⋅ FSW TON = RFF ⋅ 1V ⋅ 20 pF (2) VIN SOFT START An internal 10uA current source charges the external capacitor on the SS pin to set the output voltage slew rate during the soft start interval. The output voltage reaches regulation when the FB pin is above the under voltage threshold and the UV# = HIGH. Once the voltage on the SS pin is above the PGOOD delay threshold, the combination of the SSDelay and UV# signals release the PGOOD pin. With EN = LOW, the capacitor voltage and SS pin is held to the FB pin voltage. OVER CURRENT MONITOR The over current circuitry monitors the output current during each switching cycle. The voltage across the synchronous MOSFET, VPHASE, is monitored for over current and zero crossing. The minimum LGATE interval allows time to sample VPHASE. The over current trip point is programmed with a resistor from the ISET pin to PHASE pin, as shown in equation 3. When over current is detected, the output gates are tri-state and SS voltage is pulled to 0V. This initiates a new soft start cycle. If there are three consecutive OC events the IR3870 will disable switching. Toggling VCC or EN will allow the next start up. RSET = RDSON ⋅ IOC 20 μA (3) GATE DRIVE LOGIC The gate drive logic features adaptive dead time, diode emulation, and a minimum lower gate interval. An adaptive dead time prevents the simultaneous conduction of the upper and lower MOSFETs. The lower gate voltage must be below approximately 1V after PWM goes HIGH before the upper MOSFET can be gated on. Also, the upper gate voltage, the difference voltage between UGATE and PHASE, must be below approximately 1V after PWM goes LOW and before the lower MOSFET can be gated on. The control MOSFET is gated on after the adaptive delay for PWM = HIGH and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. The lower MOSFET is driven ‘off’ when the signal ZCROSS indicates that the inductor current has reversed as detected by the PHASE voltage crossing the zero current threshold. The synchronous MOSFET stays ‘off’ until the next PWM falling edge. When the lower peak of inductor current is above zero, a forced continuous current condition is selected set. The control MOSFET is gated on after the adaptive delay for PWM = HIGH, and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. The synchronous MOSFET gate is driven on for a minimum duration. This minimum duration allows time to recharge the bootstrap capacitor and allows the current monitor to sample the phase voltage. CONTROL LOGIC The control logic monitors input power sources, sequences the converter through the soft-start and protective modes and indicates output voltage status on the PGOOD pin. PVCC and VCC pins are continuously monitored and will disable the IR3870 if the voltage of either pin drops below the falling thresholds. The IR3870 will initiate a soft start when the PVCC is in the normal range and the EN pin = HIGH. In the event of a sustained overload a counter keeps track of 4 consecutive soft-start cycles and then disables the IR3870. If the overload is momentary and the output 12 IR3870MBF Circuit Description voltage is within regulation before four consecutive soft-start cycles, PGOOD transitions HIGH to reset the counter. OVER VOLTAGE PROTECTION The IR3870 monitors the voltage at the FB node. If the FB voltage is above the over voltage threshold, the gates are turned off and the PGOOD signal is pulled low. Toggling VCC will allow the next start up. CHARGE PUMP The purpose of the charge pump is to improve the system efficiency. A combination of VCC, V5 and three external components are used to boost PVCC up to VCPTH. PVCC drives the synchronous MOSFET and reduces the RDSON when compared to a regular 5V rail driver. The lower FET RDSON reduces the conduction power loss as discussed in the Power Loss section. The charge pump is continuously enabled for FCCM = HIGH. The charge pump circuit is disabled when FCCM = LOW and the output loading is less than half of inductor current ripple. In this case, PVCC is two diode voltages away from the V5 rail. Therefore, the power loss for driver is reduced. The charge pump circuit stops switching the CPO pin for PVCC above VCPTH. COMPONENT SELECTION Selection of components for the converter is an iterative process which involves meeting the specifications and trade-offs between performance and cost. The following sections will guide one through the process. Inductor Selection Inductor selection involves meeting the steady state output ripple requirement, minimizing the switching loss of upper MOSFETs, meeting transient response specifications and minimizing the output capacitance. The output voltage includes a DC voltage and a small AC ripple component due to the low pass filter which has incomplete attenuation of the switching harmonics. Neglecting the inductance in series with the output capacitor, the magnitude of the AC voltage ripple is determined by the total inductor ripple current flow through the total equivalent series resistance (ESR) of the output capacitor bank. ΔI = One can use equation 4 to find the required inductance. The main advantage of small inductance is increased inductor current slew rate during a load transient, which leads to a smaller output capacitance requirement as discussed in the Output Capacitor Selection section. The draw back of using smaller inductances is increased switching power loss in upper MOSFET, which reduces the system efficiency and increases the thermal dissipation as discussed in the Power Loss section. Input Capacitor Selection The main function of the input capacitor bank is to provide the input ripple current and fast slew rate current during the load current step up. The input capacitor bank must have adequate ripple current carrying capability to handle the total RMS current. Figure 16 shows a typical input current. Equation 5 shows the RMS input current. The RMS input current contains the DC load current and the inductor ripple current. As shown in equation 4, the inductor ripple current is unrelated to the load current. The maximum RMS input current occurs at the maximum output current. The maximum power dissipation in the input capacitor equals the square of the maximum RMS input current times the input capacitor’s total ESR. Figure 16. Typical Input Current Waveform. IIN_RMS = 1 ⋅ Ts Ts ∫f 2 (t )⋅ dt 0 2 = IOUT ⋅ D ⋅ 1 + 1 ⎛ ΔI ⎞ ⋅⎜ ⎟ (5) 3 ⎝ IOUT ⎠ The voltage rating of the input capacitor needs to be greater than the maximum input voltage because of high frequency ringing at the phase node. The typical percentage is 25%. VOUT VOUT ⋅ (VIN − VOUT ) ⋅ (1 − D) ⋅ Ts = (4) VIN ⋅ L ⋅ Fs L 13 IR3870MBF Output Capacitor Selection Selection of the output capacitor requires meeting voltage overshoot requirements during load removal, and meeting steady state output ripple voltage requirements. The output capacitor is the most expensive converter component and increases the overall system cost. The output capacitor decoupling in the converter typically includes the low frequency capacitor, such as Specialty Polymer Aluminum, and mid frequency ceramic capacitors. The first purpose of output capacitors is to provide current when the load demand exceeds the inductor current, as shown in Figure 17. Equation 6 shows the charge requirement for a certain load. The advantage provided by the IR3870 at a load step is to reduce the delay compared to a fixed frequency control method (in microseconds or (1-D)*Ts). If the load increases right after the PWM signal goes low, the longest delay will be equal to the minimum lower gate on as shown in the Electrical Specification table. The IR3870 also reduces the inductor current slew time, the time it takes for the inductor current to reach equality with the output current, by increasing the switching frequency up to 2.5MHz. The result reduces the recovery time. VESR is usually much greater than VESL. The IR3870 requires a total ESR such that the ripple voltage at the FB pin is greater than 7mV. The second purpose of the output capacitor is to minimize the overshoot of the output voltage when the load decreases as shown in Figure 18. By using the law of energy before and after the load removal, equation 7 shows the output capacitance requirement for a load step. C OUT = L ⋅ I STEP 2 V OS 2 − V OUT 2 (7) Figure 18. Typical Output Voltage Response Waveform. Figure 17. Charge Requirement during Load Step Q = C ⋅ V = 0.5 ⋅ Istep ⋅ Δt COUT = (6a) ⎡ 1 L ⋅ ΔIstep 2 ⎤ ⎢ ⋅ ⎥ (6b) VDROP ⎢⎣ 2 (VIN − VOUT ) ⎥⎦ 1 The output voltage drops, VDROP, initially depending on the characteristic of the output capacitor. VDROP is the sum of the equivalent series inductance (ESL) of the output capacitor times the rate of change of the output current and the ESR times the change of the output current. 14 IR3870MBF Boot Capacitor Selection The boot capacitor starts the cycle fully charged to a voltage of VB(0). Cg equals 1.16nF in IR3870. Choose a sufficiently small ΔV such that VB(0)-ΔV exceeds the maximum gate threshold voltage to turn on the high side MOSFET. ⎛ V (0) ⎞ − 1⎟ (8) CBOOT = Cg ⋅ ⎜ B ⎝ ΔV ⎠ Choose a boot capacitor value larger than the calculated CBOOT in equation 8. A typically value of 0.1uF uses. The voltage rating of this part needs to be larger than VB(0) plus the desired derating voltage. Its ESR and ESL needs to be low in order to allow it to deliver the large current and di/dt’s which drive MOSFETs most efficiently. In support of these requirements a ceramic capacitor should be chosen. DESIGN EXAMPLE Design Criteria: Input Voltage, VIN, = 6V to 21V Output Voltage, VOUT = 1.1V Switching Frequency, FS = 500KHz Inductor Ripple Current, ΔI = 4A Maximum Output Current, IOUT = 10A Over Current Trip, IOC = 14A Overshoot Allowance, VOS = VOUT + 100mV Undershoot Allowance, VDROP = 100mV Find RFF : RFF = 1.1V = 110 KΩ 1V ⋅ 20 pF ⋅ 500KHz Pick a standard value 110 kΩ, 1% resistor. Find RSET : 1.4 ⋅ 6.8mΩ ⋅ 14 A = 6.7KΩ 20 μA The RDSON of the lower MOSFET could be expected to increase by a factor of 1.4 over temperature. Therefore, pick a 6.81 kΩ, 1% standard resistor. RISET = Find a resistive voltage divider for VOUT = 1.1V: VFB = R2 ⋅ VOUT = 0.5V R2 + R1 R2 = 1.65kΩ, R1 = 1.96 kΩ, both 1% standard resistors. Choose the soft start capacitor: Once the soft start time has chosen, such as 1000us to reach to the reference voltage, a 22nF for CSS is used to meet 1000us. Choose an inductor to meet the design specification: L= VOUT ⋅ (VIN − VOUT ) 1.1V ⋅ (21V - 1.1V ) = = 0.52u H VIN ⋅ ΔI ⋅ Fs 21V ⋅ 4 A ⋅ 500K Hz ΔI = 1.1V ⋅ (21V - 1.1V ) = 3.7 A 21V ⋅ 0.56u H ⋅ 500K Hz Choose an inductor with the lowest DCR and AC power loss as possible to increase the overall system efficiency. For instance, choose an FDU0650-R56M manufactured by TOKO. The inductance of this part is 820nH and has 3.2mΩ DCR. Ripple current needs to be recalculated using the chosen inductor. Choose an input capacitor: 2 IIN_RMS = 10 A ⋅ 1.1V 1 ⎛ 3.7 A ⎞ ⋅ 1+ ⋅ ⎜ ⎟ = 2.4 A 21V 3 ⎝ 10 A ⎠ A Panasonic 10uF (ECJ3YB1E106M) accommodates 6 Arms of ripple current at 300KHz. Due to the chemistry of multilayer ceramic capacitors, the capacitance varies over temperature and operating voltage, both AC and DC. One 10uF capacitor is recommended. In a practical solution, one 1uF capacitor is required along with 10uF. The purpose of the 1uF capacitor is to suppress the switching noise and deliver high frequency current. Choose an output capacitor: To meet the undershoot specification, select a set of output capacitors which has an equivalent ESR of 10mΩ (100mV/10A). To meet the overshoot specification, equation 7 will be used to calculate the minimum output capacitance. As a result, 243uF will be needed for 10A load remover. Combine those two requirements, one can choose a set of output capacitors from manufactures such as SP-Cap (Specialty Polymer Capacitor) from Panasonic or POSCAP from Sanyo. A 270uF (EEFSX0D271) from Panasonic is recommended. 15 IR3870MBF This capacitor has 9mΩ ESR which leaves margin for the voltage drop of the ESL during load step up. The typical ESL for this capacitor is around 2nH. Refer to Output Capacitor Selection section for all ceramic capacitor solution. LAYOUT RECOMMENDATION Bypass Capacitor: One 1uF high quality ceramic capacitor should be placed as near VCC pin as possible. The other end of capacitor can be connected to a via or connected directly to GND plane. Use a GND plane instead of thin trace to the GND pin because this thin traces have too much higher impedance. A 1uF is recommended for both V5 and PVCC and repeat the layout procedure above for those signals. The purpose of the tight loop from the input ceramic capacitor is to suppress the high frequency (10MHz range) switching noise and reduce Electromagnetic Interference (EMI). If this path has high inductance, the circuit will cause voltage spikes and ringing, and increase the switching loss. The off time path has low AC and high average DC current. Therefore, it should be laid out with a tight loop and wide trace at both ends of the inductor. Lowering the loop resistance reduces the power loss. The typical resistance value of 1-ounce copper thickness is 0.5mΩ per square inch. Charge Pump: We recommend that D1, D2 and CCPO be placed as close to the CPO and PVCC pins as possible. If those components can not be placed on the same layer as IR3870, a minimum of two vias are needed for the connection of CCPO and CPO pin and the connection of D2 and PVCC. Boot Circuit: CBOOT should be placed near the BOOT and PHASE pins to reduce the impedance when the upper MOSFET turns on. DBOOT does not need to be close to CBOOT because the average current to charge CBOOT is small during the on time of lower MOSFET. Figure 19. Current Path of Power Stage Power Stage: Figure 19 shows the flowing current path for the on and off periods. The on time path has low average DC current with high AC current. Therefore, it is recommended to place the input ceramic capacitor, upper, and lower MOSFET in a tight loop as shown in Figure 19. 16 IR3870MBF PCB Metal and Components Placement Lead lands (the 13 IC pins) width should be equal to nominal part lead width. The minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting. Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension. The outboard extension ensures a large toe fillet that can be easily inspected. Pad lands (the 4 big pads) length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be no less than; 0.17mm for 2 oz. Copper or no less than 0.1mm for 1 oz. Copper or no less than 0.23mm for 3 oz. Copper. 17 IR3870MBF Solder Resist It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder resist should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure NSMD pads. The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto the copper of 0.05mm to accommodate solder resist misalignment. Ensure that the solder resist in between the lead lands and the pad land is ≥ 0.15mm due to the high aspect ratio of the solder resist strip separating the lead lands from the pad land. 18 IR3870MBF Stencil Design The Stencil apertures for the lead lands should be approximately 80% of the area of the lead lads. Reducing the amount of solder deposited will minimize the occurrences of lead shorts. If too much solder is deposited on the center pad the part will float and the lead lands will open. The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back in order to decrease the risk of shorting the center land to the lead lands when the part is pushed into the solder paste. 19 IR3870MBF DIM A A1 b b1 c D E e e1 e2 MILIMITERS MIN MAX 0.8 1 0 0.05 0.375 0.475 0.25 0.35 0.203 REF. 5.000 BASIC 6.000 BASIC 1.033 BASIC 0.650 BASIC 0.852 BASIC INCHES MIN MAX 0.0315 0.0394 0 0.002 0.1477 0.1871 0.0098 0.1379 0.008 REF. 1.970 BASIC 2.364 BASIC 0.0407 BASIC 0.0256 BASIC 0.0259 BASIC DIM L M N O P Q R S t1, t2, t3 t4 t5 MILIMITERS MIN MAX 0.35 0.45 2.441 2.541 0.703 0.803 2.079 2.179 3.242 3.342 1.265 1.365 2.644 2.744 1.5 1.6 0.401 BASIC 1.153 BASIC 0.727 BASIC INCHES MIN MAX 0.0138 0.0177 0.0962 0.1001 0.0277 0.0314 0.0819 0.0858 0.1276 0.1316 0.0498 0.05374 0.1042 0.1081 0.0591 0.063 0.016 BACIS 0.045 BASIC 0.0286 BASIC IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 05/09 20